Datasheet ML4894ES Datasheet (Micro Linear Corporation)

September 1996

FEATURING

Extended Commercial Temperature Range

-20˚C to 70˚C

for Portable Handheld Equipment

PRELIMINARY

ML4894

Synchronous Buck Controller

GENERAL DESCRIPTION

The ML4894 synchronous buck controller has been
designed to provide high efficiency DC/DC conversion for
portable products. The ML4894 can deliver a fixed 5V
output from input voltages of 5.9V to 15V.

The ML4894 drives external P- and N-channel MOSFETs
in a synchronous buck topology, allowing an overall
conversion efficiency of greater than 90% over an output
current range exceeding three decades, with an output
current capability of up to 5A.

The regulator can be disabled via the SHDN pin. While
disabled, the output of the regulator is completely
isolated from the circuit’s input supply, and the supply
current is reduced to less than 5µA to help extend battery
life.

FEATURES

■ Regulation to ±3% maximum

■ Fixed 5V output synchronous buck

■ Wide input voltage range (5.9V to 15V)

■ Power conversion efficiencies of >90% over 3 decades

of output current

■ Integrated antishoot-through logic

■ Shutdown control provides load isolation and

minimum sleep mode power consumption

■ Low shutdown current

BLOCK DIAGRAM

V

IN

5.9V - 15V

TO SYSTEM

POWER

MANAGEMENT

8

1

5

4

V

I

N

V

REG

GND

SHDN

BIAS

CIRCUITS

SLEEP

LOGIC

BUCK REGULATOR

P DRV

N DRV

I

SENSE

V

7

+

6

V

3

FB

2



OUT

5V

–

1

ML4894

PIN CONFIGURATION

ML4894

8-Pin SOIC (S08)

PIN DESCRIPTION

PIN# NAME FUNCTION

1V

2V

3I

REG

FB

SENSE

Connection point for internal linear
regulator bypass capacitor

Voltage feedback pin for regulating
the 5V output

Current sense input

V

REG

VFB

I

SENSE
SHDN





1

2

3

4



TOP VIEW

8

7

6

5

PIN# NAME FUNCTION

VIN

P DRV

N DRV

GND

5 GND Analog signal ground

6 N DRV NMOS driver output

7 P DRV PMOS driver output

8V

IN

Battery input voltage

4 SHDN a logic low on this pin shuts down the

regulator and all internal bias
circuitry for minimum power
consumption

2

ABSOLUTE MAXIMUM RATINGS

ML4894

Absolute maximum ratings are those values beyond
which the device could be permanently damaged.
Absolute maximum ratings are stress ratings only and
functional device operation is not implied.

V

............................................................................................ 16.5V

IN

Peak Driver Output Current....................................... ±2A

Junction Temperature............................................. 150°C

Storage Temperature Range .....................–65°C to 150°C

Lead Temperature (Soldering, 10 sec) .................... 150°C

Thermal Resistance (θ

) ....................................160°C/W

JA

OPERATING CONDITIONS

VFB Voltage .........................................GND - 0.3V to 6V

I

Voltage..................................................... +500mV

SENSE

All Other Inputs ..................... GND - 0.3V to VIN + 0.3V

VIN Range .................................................... 5.9V to 15V

Temperature Range ................................... -20°C to 70°C

SHDN Input Current .............................................100µA

ELECTRICAL CHARACTERISTICS

Unless otherwise specified, VIN = 10V, TA = Operating Temperature Range (Note 1)

PARAMETER CONDITIONS MIN TYP MAX UNITS

LINEAR REGULATOR

Output Voltage TA = 25°C 3.29 3.33 3.37 V

Line Regulation 5.9V < VIN < 15V 1.7 4 mV/V

Total Variation Line, Temp 3.24 3.42 V

SHUTDOWN

Input Low Voltage 1.0 V

Input High Voltage 3.0 V

Input Low Current VIL = 0V 100 nA

Input High Current VIH = V

BUCK REGULATOR

Duty Cycle Ratio VIN = 5.9V, I

VFB Threshold Voltage 5.9V < VIN < 15V 4.85 5 5.15 V

I

Threshold Voltage -60 -80 -100 mV

SENSE

Transition Time CL = 1000 pF, GND to V

SUPPLY

VIN Current SHDN = 0V 2 5 µA

SHDN = 5V 300 750 µA

Note 1: Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.

IN

= VFB = 0V 90 95 %

SENSE

IN

50 100 ns

50 µA

3

ML4894

FUNCTIONAL DESCRIPTION

The ML4894 converts a 5.9V to 15V input to a fixed 5V
output using a unique current mode PFM synchronous
buck control architecture. The output current is set by
external components, and can exceed 2A. Even at light
loads, the PFM architecture maintains high conversion
efficiencies over a wide range of input voltages. If it is
necessary to further extend battery life, the user can
shutdown and fully disconnect the load from the input
when the supply is not in use.

BIAS CIRCUITS

The bias circuits are comprised of a linear regulator and a
precision 2.5V reference. The V
bypassed to GND with a 1µF capacitor. The 2.5V
reference is used by the feedback circuit of the controller
to maintain an accurate output voltage.

SHUTDOWN LOGIC

The controller has a shutdown function which prevents
switching from occurring and disconnects the load from
the input. The ML4894 is shut down by applying a logic
low to the SHDN pin. The supply current in shutdown
ranges from 0.5µA at VIN = 5.9V to 3µA at VIN = 15V

BUCK CONTROLLER

A block diagram of the buck controller is shown in Figure

1. The circuit utilizes a constant ON-time PFM control
architecture. The circuit determines the OFF-time by
waiting for the inductor current to drop to a level set by
the feedback voltage (VFB).

The oscillator/one shot block generates a constant ON­time and a minimum OFF-time. The OFF-time is extended
for as long as the output of the current comparator stays
low. Note that the inductor current flows in the current
sense resistor during the OFF-time. Therefore, a minimum
OFF-time is required to allow for the finite circuit delays
in sensing the inductor current. The ON-time is triggered
when the current comparator’s output goes high.
However, unlike conventional fixed ON-time controllers,
this one shot has an inverse relationship with the input
voltage as shown in Figure 2. Figure 3 plots the inductor
voltage-ON-time product. Note that the volt-second
product is nearly constant over the entire input voltage
range. The inductor current is given by:

TVV

×−

bg

∆I

This means that the ripple current also remains nearly
constant over the entire input voltage range.

The transconductance amplifier generates a current from
the voltage difference between the reference and the
feedback voltage, VFB. This current produces a voltage
across Rgm that adds to the negative voltage that is
developed across the current sense resistor. When the
current level in the inductor drops low enough (a less

ON IN OUT

=

L

L

pin should be

REG

(1)

negative sense voltage) to cause the voltage at the non­inverting input of the current comparator to go positive,
the comparator trips and starts a new ON cycle. In other
words, the current programming comparator controls the
length of the OFF-time by waiting until the inductor
current decreases to a value determined by the
transconductance amplifier.

This technique allows the feedback transconductance
amplifier’s output current to steer the current level in the
inductor. The higher the transconductance amplifier’s
output current, the higher the inductor current. For
example, when the output voltage drops due to a load
increase, the transconductance amplifier will increase its
output current and generate a larger voltage across Rgm,
which in turn raises the inductor current trip level,
shortening the OFF-time. At some level of increasing the
output load, the transconductance amplifier can no
longer continue to increase its output current. When this
occurs, the voltage across Rgm reaches a maximum and
the inductor current cannot increase. If the inductor
current tries to increase, the voltage developed across the
current sense resistor would become more negative,
causing the non-inverting input of the current comparator
to be negative, which extends the OFF-time and reduces
the inductor current.

If the output voltage is too high, the transconductance
amplifier’s output current will eventually become
negative. However, since the inductor current flows in
only one direction (assuming no shoot-through current)
the non-inverting input of the current comparator will
also stay negative. This extends the OFF-time allowing the
inductor current to decrease to zero, causing the
converter to stop operation until the output voltage drops
enough to increase the output current of the
transconductance amp above zero.

In summary, the three operation modes can be defined by
the voltage at the I

V

> 0V - Discontinuous current mode

SENSE

0V > V

-60mV > V

The synchronous rectifier comparator, flip-flop, and NOR
gate make up the synchronous rectifier control circuit.
The synchronous control does not influence the operation
of the main control loop, and operation with a Schottky
diode in place of the synchronous rectifier is possible, but
at a lower conversion efficiency. The synchronous rectifier
(N DRV) is turned on during the minimum OFF-time. N
DRV will remain on until a new ON-time is started or
until the I
goes above -7mV, the current in the inductor has gone to
zero or the buck regulator is operating in discontinuous
current mode (DCM). Therefore, the synchronous rectifier
comparator is used only for DCM operation. A timing
diagram is shown in Figure 4.

> -60mV - Continuous current mode

SENSE

SENSE

pin goes above -7mV. When the I

SENSE

pin at the end of the OFF-time:

SENSE

> -100mV - Current limit

SENSE

pin

4

OSCILLATOR

ONE SHOT

t

ON

P DRV

ML4894

V

IN

C

IN

7

CURRENT

COMPARATOR

SRQ

Q

+

-

TRANSCONDUCTANCE

AMPLIFIER

Figure 1. ML4894 Functional Block Diagram

SHOOT-THRU

PROTECTION

SYNCHRONOUS

RECTIFIER

COMPARATOR

V

SR

L1

+

C

I

R

SENSE

OUT

L

N DRV

6

-

-7mV
I

+

R

gm

­+

2.5V

SENSE

V

FB

3

2

V

OUT

–

25

20

15

(µs)

ON

t

10

5

0

5 7.5 10 15

VIN (V)

12.5

60

50

40

30

VOLT-SECONDS

20

10

0

5 7.5 10 15

VIN (V)

12.5

Figure 2. ON-Time vs. Input Voltage Figure 3. Volt-seconds vs. Input Voltage

5

ML4894

V

C

t

ON

V

SR

Q (ONE SHOT)

INDUCTOR CURRENT (IL)



Figure 4. One Shot and Synchronous Rectifier Timing

DESIGN CONSIDERATIONS

A typical design can be implemented by using the
following design procedure. Note that this procedure is
not intended to give final values, but to give a good
starting point, and provide the relationships necessary to
make trade-off decisions. Some experimentation will be
necessary to optimize values and to verify that the design
operates over worst case conditions.

DESIGN SPECIFICATIONS

It is important to start with a clear definition of the design
specifications. Make sure the specifications reflect worst
case conditions. Key specifications include the minimum
and maximum input voltage and the output voltage and
load current.

INDUCTOR AND SENSE RESISTOR SELECTION

Figure 5 shows the inductor current of the buck regulator.
The inductor current is made up of two components: the
DC current level set by the transconductance amplifier,
I

, and the inductor ripple current, ∆I

SENSE

shows that I

II I

OUT SENSE L

is the summation of I

OUT

V

=+= +

1
2

SENS E

R

SENS E

. The figure also

L

and ½∆I

SENSE

TVV

×−

bg

ON IN

L

×

2

.

L

5

(3)

Therefore, the selection of the inductance value
determines how much of the output current is made up of
the ripple current. Higher inductor ripple current allows
smaller inductor values, but results in higher peak
currents, lower efficiency, and higher output voltage
ripple.

Inductor ripple currents in the range of 30% to 70% of the
maximum output current are typical. As a good starting
point, set the inductor ripple current to 50% of the
maximum output current:

TVV

∆I

=

L

×−

bg

ON IN

L

5

FI

=×

IRC OUT MAX

()

(4)

where FIRC = ratio of inductor ripple current to the
maximum output current, or:

TVV

L

=

×−

bg

ON IN

I

×

05

.

()

OUT MAX

5

(5)

Calculate the inductance using the volt-seconds value
given in Figure 3 at the maximum input voltage. Choose
the nearest standard value, realizing the trade-offs
mentioned before. Then, using the inductance value
chosen, determine the actual inductor ripple current at
the maximum and minimum input voltage using Equation
4 and Figure 3.

The sense resistor value can be determined using the
inductor ripple current value calculated above and
Equation 3 rearranged as follows:

V

SENSE MIN

R

SENSE

=

II

OUT MAX L MIN

()

1

−

() ()

∆

2

(6)

Having determined the values for the inductor and sense
resistor, we can now specify the inductor peak current
rating. This value is calculated at current limit and at the
maximum input voltage, and is given by:

II I

L PEAK MAX SENSE MAX L MAX(()) () ()

I

LPEAKMAX

af

ch

I

LPEAKMAX

ch

=+∆

V

SENSE MAX

af

=+∆

R

SENSE

=+

af

R

I

V

01.

SENSE

LMAX

af

I

∆

LMAX

af

(7)

6

I

L

V

IN – VOUT

L

ML4894

V

OUT

L

I

LPK

I

= I

L

I

SENSE

OUT

V

SENSE

=

R

SENSE

∆I

SENSE

+ 1/2 ∆I

L

T

ON

T

OFF

Figure 5. Buck Regulator Inductor Current

For reliable operation, the inductor current rating should
exceed the value calculated by 10%-20%.

For future reference, determine the peak inductor current
at the minimum input voltage:

II I

L PEAK MIN SENSE MIN L MIN(()) () ()

I

LPEAKMIN

ch

I

LPEAKMIN

ch

=+∆

V

SENSE MIN

=+∆

af

=+

af

af

R

SENSE

V

006.

R

SENSE

I

∆

LMIN

af

I

LMIN

af

(8)

Now the sense resistor’s power rating can be determined.
The sense resistor must be able to carry the peak current
in the inductor during the OFF-time:

PI R

=×

R RMS OFF SENSE

SENSE

2

()

(9)

where:

22

SENSE(MAX SENSE(MAX L PEAK MAX L P EAK MAX

+× +

) ) (()) (())

3

I

RMS OFF

V

F

2

=− ×

()

OUT

1

G

V

H

IN MAX

()

IIII

I
J

K

The final parameter that should be specified is the
winding resistance of the inductor. In general, the
winding resistance should be as low as possible,
preferably in the low mΩ range. Since the inductor is in
series with the load at all times, the copper losses can be
approximated by:

PI R

Cu OUT L

2

=×

(10)

A good rule of thumb is to allow 2 mΩ of winding
resistance per µH of inductance.

MOSFET SELECTION

The switching MOSFETs must be logic level types with
the ON resistance specified at VGS = 4.5V. In general, the

t

ON resistance - gate charge product provides a good
figure of merit by which to compare various MOSFETs,
the lower the figure the better. The internal gate drivers of
the ML4894 can drive over 100nC of total gate charge,
but 60nC to 70nC is a more practical limit to ensure good
switching times.

The drain-source breakdown voltage rating is determined
by the input voltage. For input voltages up to 10V, a drain
to source rating of 20V is acceptable. For input voltages
up to 15V, a drain to source rating of 30V is
recommended. For a more reliable design, look for
MOSFETs that are avalanche rated.

In high current applications, the MOSFET’s power
dissipation often becomes a major design factor. The I2R
losses generate the largest portion of heat in the MOSFET
package. Make sure that the MOSFETs are within their
rated junction temperature at the maximum ambient
temperature by calculating the temperature rise using the
thermal resistance specifications.

The worst case power dissipation for the P-MOS switch
occurs at the minimum input voltage and is determined
as follows:

PI R

=×

PMOS RMSON DSON

−

where:

F

5

2

I

RMS O N

=×

G

()

V

H

IN MAX

2

() ()

IIII

I

V

J

K

()

22

SENSE( MAX SENSE( MAX L PEAK MAX L PEA K MAX

+× +

) ) (()) (())

(11)

3

The worst case power dissipation for the N-MOS switch
occurs at the maximum input voltage and is determined
using:

PI R

=×

N MOS RMS OFF DS ON

−

2

() ()

(12)

7

ML4894

DESIGN CONSIDERATIONS (cont.)

INPUT CAPACITOR SELECTION

The choice of the input capacitor is based on its ripple
current and voltage ratings rather than its capacitance
value. The input capacitor should be a low ESR type and
located as close to the source of the P-MOS switch as
possible. The input capacitor’s ripple current is
determined by the load current and input voltage, with
the worst case condition occurring at VIN = 2 x V

VV V

×−

II I

RMSC SENSEMAX LMAX

()≈+×

() ( ) ( )

IN

1

∆

2

55

:

OUT

bg

IN

V

IN

The capacitor’s voltage rating is based on the maximum
input voltage, V

IN(MAX)

. Capacitor manufacturers typically
recommend derating the capacitor voltage rating by 20%
to 50% for aluminum electrolytic types and 50% to 70%
for tantalum types.

In high current applications it may necessary to add a
small 0.1µF ceramic capacitor to bypass V

(pin 8) of the

IN

ML4894.

OUTPUT CAPACITOR SELECTION

The output capacitors determine the loop stability and the
output ripple voltage. Use only low ESR capacitors
intended for switching power supply applications, such as
AVX TPS, Sprague 593D, Sanyo OS-CON, or Nichicon PL
series. To ensure stability, the minimum capacitance value
is given by:

OUT

T

ON MAX

R

SENSE

()

(14)

C

≥×

OUT

V

43.

The maximum ESR value can be estimated using:

V

∆

≤

OUT

I

∆

()

LMAX

(15)

ESR

The selected capacitor must meet both the capacitance
and ESR requirements. As a final check, make sure the
output capacitor can handle the ripple current, I

I

∆

LMAX

≈

()

12

I

RMS

RMS

:

(16)

8

LAYOUT

ML4894

A typical application circuit is shown in Figure 6.
Proximity of passive devices and adequate power and
ground planes are critical for reliable operation of the
circuit. In general, use the top layer for the high current
connections and the bottom layer for the quiet
connections such as GND, feedback and current sense.
Some more specific guidelines follow.

1. The connection from the current sense resistor to the

I

pin should be made by a separate trace and

SENSE

located as close to the lead of the resistor as possible.
The trace length from the sense resistor to the ML4894
should be kept as short as possible and away from
switching components and their traces.

2. The trace lengths from the buck regulator’s input

capacitor to the switching MOSFET, from the MOSFETs
to the inductor, from the synchronous rectifier MOSFET
to the sense resistor, and from the inductor to the
output capacitor should all be as short as possible.

3. The high current ground paths need to be kept separate

from the signal ground paths. The GND connection
should be made at a single-point star ground. It is very
important that the ground for the ML4894 ground pin
(pin 5) be made using a separate trace.

4. Concentrating on keeping the current sense and high
current connections short as well as keeping the
switching components and traces away from the
sensitive analog components and traces during layout
will eliminate the majority of problems created by a
poor layout.

5. The V

and bypass capacitor needs to be located

REG

close to the ML4894 for adequate filtering of the IC’s
internal bias voltage.

6. Remote sensing the output for improved load
regulation can be implemented with the ML4894. The
output can be remote sensed by using the top of the
external resistor divider as the remote sense point.

C1

22µF

V

IN

C4

1µF

C2

22µF

C3

22µF

1

2

3

4

ML4894

V

REG

VFB

I

SENSE

SHDN

Q1

IRF7406

R1

25mΩ

L1

39µH

C6

100µF

C7

100µF

C8

100µF

V

+

OUT

5V

–



8

VIN



Q2

IRF7403



P DRV

N DRV

GND

7

6

5

Figure 6. 5V, 3A DC/DC Converter Circuit

9

ML4894

PHYSICAL DIMENSIONS inches (millimeters)

Package: S08

8-Pin SOIC

.187/.198

(4.75/5.03)

85

1

4

.050 ± .008

(1.27 ± 0.20)

.018 MIN (.475)
(4 PLCS)

.011/.021

(.280/.533)

.059/.070

(1.50/1.79)

TYP.

SEATING
PLANE

.007/.010

(.177/.254)

.148/.159

(3.76/4.04)

.228/.246

(5.79/6.25)

.014/.037

(.355/.940)

ORDERING INFORMATION

PART NUMBER TEMPERATURE RANGE PACKAGE

ML4894ES -20°C to 70°C 8-Pin SOIC (S08)

© Micro Linear 1996 is a registered trademark of Micro Linear Corporation.
Products described herein may be covered by one or more of the following patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940; 5,546,017;
5,559,470. Other patents are pending

Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design.
Micro Linear does not assume any liability arising out of the application or use of any product described herein,
neither does it convey any license under its patent right nor the rights of others. The circuits contained in this
data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to
whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility
or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel
before deciding on a particular application.

10

2092 Concourse Drive

San Jose, CA 95131

Tel: [phone]

Fax: [phone]

DS4894-01