Datasheet ML4880CS, ML4880ES Datasheet (Micro Linear Corporation)

FEATURING

Extended Commercial Temperature Range

–20°C to 70°C

for Portable Handheld Equipment

July 2000

ML4880*

Portable PC/PCMCIA Power Controller

GENERAL DESCRIPTION

The ML4880 Portable PC and PCMCIA Power Controller
is a complete solution for DC/DC power conversion for
portable computing systems with single or multiple
PCMCIA slots.

The device provides two synchronous buck controllers to
implement mixed voltage systems and a flyback controller
for 12V VPP generation for PCMCIA slots. The flyback
architecture enables generation of high currents (150mA
or more per slot) on the 12V bus for multiple slot PCMCIA
applications.

Each regulator can be independently switched off to fully
isolate the load from the power supply. The PFM

FEATURES

■Regulation to ±3% maximum: provides 2% PCMCIA

switch matrix margin

■ Two synchronous buck controllers for 3.3/3V, 5V

generation, and a flyback for high current, 12V
generation from 5.5V to 18V input

■ Regulator power conversion efficiencies > 90%

■ Pulse frequency modulation for high efficiency

operation

■ Independent regulator shutdown for full load isolation

■ Adjustable current limit

■ Wide input voltage range (5.5V to 18V)

architecture will automatically adjust switching frequency
at light loads in order to maintain power conversion
efficiencies in excess of 90% over a wide output power
range. (* Indicates Part is End Of Life as of July 1, 2000)

BLOCK DIAGRAM

V

IN

V

IN

5.5 – 18V

+

V

C

OUT

6V – 15V

–

TO SYSTEM

POWER

MANAGEMENT

V

IN

16

6

GND

7

GND

SHDN A

11

SHDN B

18

SHDN C

10

N DRV C

17

I

SENSE

3

V

FB

12

21

PWR GND C

C

C

15

V

BIAS

CIRCUITS

SHUTDOWN

LOGIC

REG

GATE

CLAMP

FLYBACK REGULATOR C

24

CLAMP

BUCK REGULATOR ABUCK REGULATOR B

CLAMP

GATE

CLAMP

CLAMP A

P DRV A

N DRV A
I

SENSE

PWR GND A

VFB A

CLAMP B

P DRV B

N DRV B
I

SENSE

PWR GND B

VFB B

2

13

14

A

4

23

9

1

19

20

B

5

22

8

V

IN

+

V

A

OUT

2.5 – 3.5V

–

+

V

B

OUT

4.5 – 5.0V

–

1

ML4880

PIN CONFIGURATION

ML4880

24-Pin SOIC (S24)

CLAMP B
CLAMP A

I

SENSE

I

SENSE

I

SENSE

GND
GND

V

FB

V

FB

SHDN C
SHDN A

V

FB

1
2

C
A
B

B
A

C

3
4
5
6
7
8
9
10
11
12

TOP VIEW

24
23
22
21
20
19
18
17
16
15
14
13

CLAMP
PWR GND A
PWR GND B
PWR GND C
NDRV B
PDRV B
SHDN B
NDRV C
V

IN

V

REG

NDRV A
PDRV A

PIN DESCRIPTION

PIN# NAME FUNCTION PIN# NAME FUNCTION

1 CLAMP B Gate clamp, regulator B
2 CLAMP A Gate clamp, regulator A
3I
4I
5I

C Current sense, regulator C

SENSE

A Current sense, regulator A

SENSE

B Current sense, regulator B

SENSE

6 GND Ground
7 GND Ground
8VFB B Feedback node, buck regulator B

9VFB A Feedback node, buck regulator A
10 SHDN C Shutdown input, regulator C
11 SHDN A Shutdown input, regulator A
12 VFB C Feedback node, regulator C

13 PDRV A P-channel drive, regulator A
14 NDRV A N-channel drive, regulator A
15 V
16 V

REG

IN

Linear regulator output

Power supply input voltage
17 NDRV C N-channel drive, regulator C
18 SHDN B Shutdown input, regulator B
19 PDRV B P-channel drive, regulator B
20 NDRV B N-channel drive, regulator B
21 PWR GND C Power Ground, regulator C
22 PWR GND B Power Ground, regulator B
23 PWR GND A Power Ground, regulator A
24 CLAMP Charge pump capacitor input

2

ML4880

ABSOLUTE MAXIMUM RATINGS

Absolute maximum ratings are those values beyond
which the device could be permanently damaged.
Absolute maximum ratings are stress ratings only and
functional device operation is not implied.

V

............................................................................................... 20V

IN

OPERATING CONDITIONS

VIN Range ..................................................... 5.5V to 18V

Temperature Range

ML4880CS ................................................ 0°C to 70°C

ML4880ES ............................................. –20°C to 70°C

Peak Driver Output Current ........................................ 2A

VFB Voltage .........................................GND – 0.3V to 6V

I

Voltage..................................................... ±500mV

SENSE

All Other Analog Inputs ......... GND – 0.3V to VIN + 0.3V

All Digital Inputs .................. GND – 0.3V to V

REG

+ 0.3V

Junction Temperature ............................................. 150°C

Storage Temperature Range .................... –65°C to 150°C

Lead Temperature (Soldering, 10 sec) ..................... 150°C

Thermal Resistance (θJA)...................................... 80°C/W

ELECTRICAL CHARACTERISTICS

Unless otherwise specified, V

PARAMETER CONDITIONS MIN TYP MAX UNITS

Shutdown Inputs

Input Low Voltage 0.8 V

Input High Voltage 2.0 V
Input Bias Current –1 1 µA

Buck Regulator A

Duty Cycle Ratio VIN = 5.5V, I
VFB A Threshold Voltage 1.21 1.25 1.29 V
I

A Threshold Voltage –140 –200 –250 mV

SENSE

Transition Time CL = 1000pF, 0 to V

Buck Regulator B

Duty Cycle Ratio VIN = 5.5V, I
VFB B Threshold Voltage 1.21 1.25 1.29 V
I

B Threshold Voltage –140 –200 –250 mV

SENSE

Transition Time CL = 1000pF, 0 to V

Flyback Regulator C

Duty Cycle Ratio VIN = 5.5V, I

VFB C Threshold Voltage 1.21 1.25 1.29 V

I

C Threshold Voltage –140 –200 –250 mV

SENSE

Transition Time CL = 1000pF, 0 to V

Supply

Linear Regulator Output Voltage 7V ≤ VIN ≤ 18V 6.25 6.99 V
Linear Regulator Load Regulation I(V
VIN Current SHDN A/B/C = 5V 300 350 µA

= 14V, C(V

IN

) = 10µF, T

REG

VIN = 18V, I

) = 0 to 10mA 6.10 6.99 V

REG

SHDN A/B/C = 0V 350 400 µA

= Operating Temperature Range (Note 1)

A

A = VFB A = 0V 86 96 %

SENSE

REG

B = VFB B = 0V 92 98 %

SENSE

REG

C = VFBC = 0V 86 94 %

SENSE

C = VFBC = 0V 61 77 %

SENSE

REG

80 100 ns

80 100 ns

80 100 ns

Note 1: Limits are guaranteed by 100% testing, sampling or correlation with worst case test conditions.

3

ML4880

FUNCTIONAL DESCRIPTION

The ML4880 converts a 5.5V to 18V input to three outputs
via two synchronous buck controllers and a flyback
controller. The two buck controllers utilize a unique
current mode PFM control architecture that generate
output voltages in the range of 2.5V to 3.5V (output A),
and 4.5V to 5V (output B). The output current is set by
external components, and can exceed 2A on each supply.
The flyback controller also uses a current mode PFM
control scheme and can be used to generate a 6V to 15V
output. Again, the output current of the flyback is
dependent on external components, and output currents
of 500mA are obtainable. Even at light loads, the PFM
architecture maintains high conversion efficiencies over a
wide range of input voltages. If it is necessary to further
extend battery life, the user can shutdown and fully
disconnect each load from the input independently.

BIAS CIRCUITS

The bias circuits are comprised of a linear regulator and a
precision voltage reference. The linear regulator produces
a supply voltage (V
V

pin should be bypassed to GND with a 4.7µF to

REG

10µF capacitor. The precision voltage reference is used by
the feedback circuit of each controller to maintain an
accurate output voltage.

) used by the control circuits. The

REG

GATE CLAMP

The gate clamp circuit provides a method of preventing
the buck regulator P-channel MOSFET switches from
accidentally turning on when the input is suddenly
switched to a higher voltage. This condition can occur
during start-up or when an adapter voltage is applied. As
shown in the block diagram, the P DRV drivers are
capacitively coupled to the gates.

Assume that P DRV is in the OFF state, or at V

REG

, and the
gate voltage is held at VIN through the resistor. The gate­source voltage of the MOSFET is 0V and the switch stays
off. If a higher input voltage (VIN + 5V for example) is
suddenly applied with P DRV still in the OFF state, the
source voltage of the MOSFET would jump up to VIN +
5V, but the gate would still be at VIN. The gate-source
voltage becomes –5V and the P-channel MOSFET would
turn on even though P DRV is still in the OFF state. In
order to prevent the MOSFET from turning on, the gate
clamp circuit senses the increase in the input voltage and
pumps charge into the gate side of the coupling capacitor,
quickly charging the capacitor to the new input voltage
level.

The CLAMP pin requires a capacitor to GND in order for
the gate clamp circuit to function properly. The capacitor
value should be 1.5 to 2 times the value used for the
coupling capacitor.

SHUTDOWN LOGIC

Each controller has a separate shutdown pin. By applying
a logic high to the appropriate pin, the transconductance
amplifier and current comparator of each controller
(shown in Figures 1 and 3) can be disabled. This prevents
switching from occurring and disconnects the load from
the input. All other circuitry within the ML4880 remains
active during shutdown.

T

OSCILLATOR/

ONE SHOT

V

CURRENT

COMPARATOR

ON

SHOOT THRU
PROTECTION

SYNCHRONOUS

RECTIFIER

COMPARATOR

V

SR

Rgm

C

+

–

TRANSCONDUCTANCE

AMPLIFIER

BUCK CONTROLLERS

A block diagram of the buck controllers is shown in Figure

1. The circuit utilizes a constant ON-time PFM control
architecture. The circuit determines the OFF-time by
waiting for the inductor current to drop to a level set by
the feedback voltage (VFB).

V

IN

C

R

SENSE

IN

L

V

OUT

C

OUT

R1

I

L

P DRV

N DRV

–18mV

+

I

–

–

SENSE

V

FB

+

1.25V

R2

Figure 1. Buck Controller Functional Diagram

4

ML4880

The oscillator/one shot block generates a constant ON­time and a minimum OFF-time. The OFF-time is extended
for as long as the output of the current comparator stays
low. Note that the inductor current flows in the current
sense resistor during the OFF-time. Therefore, a minimum
OFF-time is required to allow for the finite circuit delays
in sensing the inductor current. The ON-time is triggered
when the current comparator’s output goes high.
However, unlike conventional fixed ON-time controllers,
the ML4880’s one shot has an inverse relationship with
the input voltage as shown in Figure 4. Figure 5 plots the
inductor voltage-ON-time product. Note that the volt­second product is nearly constant at voltages above 7V
input. This results in an inductor current ripple of:

TVV

×−()

ON IN OUT

∆I

=

L

L

(1)

It is important to note that the ripple current does not vary
in proportion with VIN, but remains nearly constant over a
wide input voltage range.

The transconductance amplifier generates a current from
the voltage difference between the reference and the
feedback voltage, VFB. This current produces a voltage
across Rgm that adds to the negative voltage that is
developed across the current sense resistor. When the
current level in the inductor drops low enough (a less
negative sense voltage) to cause the voltage at the non­inverting input of the current comparator to go positive,
the comparator trips and starts a new ON cycle. In other
words, the current programming comparator controls the
length of the OFF-time by waiting until the inductor
current decreases to a value determined by the
transconductance amplifier.

This technique allows the feedback transconductance
amplifier’s output current to steer the current level in the
inductor. The higher the transconductance amplifier’s
output current, the higher the inductor current. For
example, when the output voltage drops due to a load
increase, the transconductance amplifier will increase its
output current and generate a larger voltage across Rgm,

which in turn raises the inductor current trip level,
shortening the OFF-time. At some level of increasing the
output load, the transconductance amplifier can no longer
continue to increase its output current. When this occurs,
the voltage across Rgm reaches a maximum and the
inductor current cannot increase. If the inductor current
tries to increase, the voltage developed across the current
sense resistor would become more negative, causing the
non-inverting input of the current comparator to be
negative, which extends the OFF-time and reduces the
inductor current.

When the output voltage is too high, the
transconductance amplifier’s output current will
eventually become negative. However, since the inductor
current flows in only one direction (assuming no shoot
through current) the non-inverting input of the current
comparator will also stay negative. This extends the OFF­time allowing the inductor current to decrease to zero and
causing the converter to stop operation until the output
voltage drops enough to increase the output current of the
transconductance amp above zero.

In summary, the three operation modes can be defined by
the voltage at the I

V

≥ 0V Discontinuous

SENSE

pin at the end of the OFF-time:

SENSE

current mode

0V > V

> –140mV Continuous

SENSE

current mode

–140mV > V

> –250mV Current limit

SENSE

The synchronous rectifier comparator and the two NOR
gates make up the synchronous rectifier control circuit.
The synchronous control does not influence the operation
of the main control loop, and operation with a Schottky
diode in place of the synchronous rectifier is possible, but
at a lower conversion efficiency. The synchronous rectifier
(N DRV) is turned on during the minimum OFF-time or
whenever the I

pin goes below –18mV. N DRV will

SENSE

remain on until a new ON-time is started or until the
I

pin goes above –18mV. When the ISENSE pin goes

SENSE

above –18mV, the current in the inductor has gone to zero

T

OFF(MIN)

V

C

T

ON

V

SR

I

L

Figure 2. One Shot and Synchronous Rectifier Timing Diagram

5

ML4880

or the buck regulator is operating in a discontinuous
current mode (DCM). Therefore, the synchronous rectifier
comparator is used only for DCM operation. A timing
diagram is shown in Figure 2.

FLYBACK CONTROLLER

A block diagram of the flyback regulator is shown in
Figure 3. The circuit also utilizes a constant ON-time PFM
control architecture. As with the buck controllers, the
flyback controller determines the OFF-time by waiting for
the inductor current to drop to a level set by the feedback
voltage VFB. As a result of the control scheme, the valley
of the inductor current ripple is programmed instead of
the peak, and features an automatic transition into and out
of burst mode which does not require additional control
circuitry.

The flyback controller’s oscillator/one shot block has a
constant ON-time and a minimum OFF-time similar to
that of the buck controllers. The OFF-time is extended for
as long as the current comparator’s output stays low.
Again, since the inductor current flows in the current
sense resistor during the OFF-time, a minimum OFF-time
is required to allow for the finite circuit delays in sensing
the inductor current. The ON-time is triggered when the
current comparator’s output goes high. However, unlike
conventional fixed ON-time controllers, the ML4880’s
one shot has an inverse relationship with the input voltage
as shown in Figure 4. Figure 5 plots the inductor voltage­ON-time product. Note that the volt-second product is
nearly constant over the input voltage range. This results
in an inductor current ripple of:

TV

×

ON IN

∆I

=

L

L

(2)

It is important to note that the ripple current does not vary
in proportion with VIN, but remains nearly constant over a
wide input voltage range.

The transconductance amplifier and current comparator
operate in the same manner as the buck regulator. The
current programming comparator controls the length of
the OFF-time by waiting until the inductor current
decreases to a value determined by the transconductance
amplifier before starting a new ON-time.

Like the buck controllers, when the flyback’s output
voltage drops due to a load increase, the
transconductance amplifier will increase its output current
and generate a larger voltage across Rgm, which in turn
causes the current comparator’s output to go positive
earlier or shortens the OFF-time. Conversely, when the
output voltage is too high, the transconductance
amplifier’s output current will eventually become
negative. This holds the current comparator’s output low
and extends the OFF-time causing the converter to stop
operation until the output voltage drops enough to
increase the output current of the transconductance amp
above zero.

In summary, the flyback has three operation modes that
can be defined by the voltage at the SENSE pin at the end
of the OFF-time:

V

≥ 0V Discontinuous

SENSE

current mode

0V > V

> –140mV Continuous

SENSE

current mode

–140mV > VSENSE > –250mV Current limit

OSCILLATOR/

ONE SHOT

V

CURRENT

COMPARATOR

V

IN

T

ON

Rgm

C

+

–

TRANSCONDUCTANCE

AMPLIFIER

–

+

1.25V

N DRV

I

SENSE

V

FB

Figure 3. Flyback Controller Functional Diagram

6

ML4880

DESIGN CONSIDERATIONS

A typical design can be implemented by using the
following design procedure. Note that this procedure is
not intended to give final values, but to give a good
starting point, and provide the relationships necessary to
make trade-off decisions. Some experimentation will be
necessary to optimize values and to verify that the design
operates over worst case conditions.

DESIGN SPECIFICATIONS

It is important to start with a clear definition of the design
specifications. Make sure the specifications reflect worst
case conditions. Key specifications include the minimum
and maximum input voltage and the output voltage and
load current for each output.

BUCK REGULATORS - INDUCTOR AND SENSE
RESISTOR SELECTION

Figure 6 shows the inductor current of the step down
regulators. The inductor current is made up of two
components: the DC current level set by the
transconductance amplifier, I
current, ∆I
summation of I

II I

OUT SENSE L

. The figure also shows that I

L

SENSE

=+= +

1

∆

22

and

V

R

1

/2 ∆I

SENSE
SENSE

, and the inductor ripple

SENSE

:

L

TVV

×−

ON IN OUT

is the

OUT

()

L

×

(3)

Inductor ripple currents in the range of 30% to 70% of the
maximum output current are typical. As a good starting
point set the inductor ripple current to 50% of the
maximum output current:

TVV

×−

()

ON IN OUT

where F

∆I

=

L

= ratio of inductor ripple current to the

IRC

L

FI

=×

IRC OUT MAX

()

(4)

maximum output current, or:

TVV

×−

()

ON IN OUT

L

=

05

I

×

.

OUT MAX

()

(5)

Calculate the inductance using the volt-seconds value
given in Figure 5 at the maximum input voltage. Choose
the nearest standard value, realizing the trade-offs
mentioned before. Then, using the inductance value
chosen, determine the actual inductor ripple current at the
maximum and minimum input voltage using Equation 4
and Figure 5.

The sense resistor value can be determined using the
inductor ripple current value calculated above and
Equation 3 rearranged as follows:

R

SENSE

V

SENSE MIN

=

IIII

OUT MAX L MIN OUT MAX L MIN

()

1

−

() () () ()

∆∆

2

=

.

014

−

1

(6)

2

Therefore, the selection of the inductance value
determines how much of the output current is made up of
the ripple current. Higher inductor ripple current allows
smaller inductor values, but results in higher peak
currents, lower efficiency, and higher output voltage
ripple.

(µs)
T

ON

18

16

14

12

10

3.3V BUCK

8

12V FLYBACK

6

4

5V BUCK

45

40

35

30

25

20

VOLT-SECONDS (V-µs)

15

10

12V FLYBACK

3.3V BUCK

5V BUCK

2

0 5 10 15 20

VIN (V)

Figure 4. TON vs. V

IN

5

0510

V

(V)

IN

Figure 5. Volt-seconds vs V

15 20

IN

7

ML4880

Having determined the values for the inductor and sense
resistor, we can now specify the inductor peak current
rating. This value is calculated at current limit and at the
maximum input voltage, and is given by:

II I

L PK MAX SENSE MAX L MAX

(@ ) ( ) ( )

V

SENSE MAX

R

=+=

()

+=+

∆∆

I

L MAX

SENSE

() ()

∆

025

V

.

R

SENSE

I

L MAX

(7)

For reliable operation, the inductor current rating should
exceed the value calculated by 10%-20%.

For future reference, determine the peak inductor current
at the minimum input voltage:

II I

L PK MIN SENSE MAX L MIN

(@ ) ( ) ( )

V

SENSE MAX

R

=+=

()

SENSE

I

+=+

∆∆

L MIN

() ()

∆

V

.

025

R

SENSE

I

L MIN

(8)

Now the sense resistor’s power rating can be determined.
The sense resistor must be able to carry the peak current
in the inductor during the OFF-time:

PI R

R RMS OFF SENSE

SENSE

where

2

I

RMS OFF

2

=×

()

,



()()()()

IIII

V

OUT

V

IN MAX

()



SENSE MAX










1

=−

()




22

++

()

SENSE MAX L PK MAX

(9)

()(@)

3

L PK MAX

(@ )

One final parameter should be specified: the winding
resistance of the inductor. In general the winding
resistance should be as low as possible, preferably in the
low mΩ range. Since the inductor is in series with the
load at all times, the copper losses can be approximated
by:

BUCK REGULATORS — MOSFET SELECTION

The switching MOSFETs must be logic level types with the
ON resistance specified at VGS = 4.5V. In general, the ON
resistance - gate charge product provides a good figure of
merit by which to compare various MOSFETs, the lower
the figure the better. The internal gate drivers of the
ML4880 can drive over 100nC of total gate charge, but
60nC to 70nC is a more practical limit to ensure good
switching times.

The drain-source breakdown voltage rating is determined
by the input voltage. For input voltages up to 12V, a drain
to source rating of 20V is acceptable. For input voltages
up to 18V, a drain to source rating of 30V is
recommended. For a more reliable design, look for
MOSFETs that are avalanche rated.

In high current applications, the MOSFET’s power
dissipation often becomes a major design factor. The I2R
losses generate the largest portion of heat in the MOSFET
package. Make sure that the MOSFETs are within their
rated junction temperature at the maximum ambient
temperature by calculating the temperature rise using the
thermal resistance specifications.

The worst case power dissipation for the P-MOS switch
occurs at the minimum input voltage and is determined as
follows:

PIR

P MOS RMS ON DS ON( ) () ()−






where

2

I

RMS ON

()

2

=×



()()()()

IIII

SENSE MAX SENSE MAX L PK MIN L PK MIN





V





OUT

=





V





IN MIN

()

22

++

() ()(@) (@)

3

The worst case power dissipation for the N-MOS switch
occurs at the maximum input voltage and is determined
using:

(11)






2

PI R

=×

CU OUT L

A good rule of thumb is to allow 2mΩ of winding
resistance per µH of inductance.

I

L

V

IN – VOUT

L

T

ON

V

T

OFF

Figure 6. Buck Regulator Inductor Current

8

OUT

L

(10)

∆I

I

LPK

L

I

SENSE

PIR

N MOS RMS OFF DS ON()()()−

I

= I

OUT

SENSE

V

SENSE

=

R

SENSE

2

=×

+ 1/2 ∆I

L

t

(12)

ML4880

BUCK REGULATORS — INPUT CAPACITOR SELECTION

The choice of the input capacitor is based on it’s ripple
current and voltage ratings rather than its capacitance
value. The input capacitor should be a low ESR type and
located as close to the source of the P-MOS switch as
possible. The input capacitor’s ripple current is
determined by the load current and the input voltage, with
the worst case condition occurring at V





II I

RMS C SENSE MAX L MAX

≈+



() () ()

IN



1

∆

2










= 2 × V

IN

()

VVV

OUT IN OUT

V

OUT

−

IN

:




(13)



The capacitor’s voltage rating is based on the maximum
input voltage, V

. Capacitor manufacturers typically

IN(MAX)

recommend derating the capacitor voltage rating by 20%
to 50% for aluminum electrolytic types and 50% to 70%
for tantalum types.

In high current applications it may necessary to add a
10µF bulk capacitor and a small 0.1µF ceramic capacitor
to bypass VIN (pin 16) right at the ML4880.

BUCK REGULATORS — OUTPUT CAPACITOR
SELECTION

The output capacitors determine the loop stability and the
output ripple voltage. Use only low ESR capacitors
intended for switching power supply applications, such as
AVX TPS, Sprague 593D Sanyo OS-CON, or Nichicon PL
series. To ensure stability, the minimum capacitance value
is given by:

C

OUT



V

43.

OUT










T

ON MAX

R



≥





()

SENSE






(14)

The maximum ESR value can be estimated using:

V

∆

≤

∆

OUT

I

L MAX

()

(15)

ESR

The selected capacitor must meet both the capacitance
and ESR requirements. As a final check, make sure the
output capacitor can handle the ripple current,
I

RMS(C

OUT

:

)

I

∆

L MAX

I

RMS C

()

OUT

≈

()

12

(16)

BUCK REGULATORS - OUTPUT VOLTAGE

The output of buck regulator A is adjustable and can be
set to any voltage between 2.5V and 3.5V by connecting a
resistor divider to the feedback pin as shown in Figure 1.
Similarly, the output voltage of buck regulator B is
adjustable and has a usable range of 4.5V to 5.0V. The
resistor values R1 and R2 can be calculated using the
following equation:



.

VV

125

=×

OUT

RR







12

+





2

R

(17)

The value of R2 should be 125kΩ or less to minimize bias
current errors.

It is important to note that the accuracy of these resistors
directly affects the accuracy of the output. Use precision
resistors and set the nominal voltage approximately 1% to
2% high with no load in order to make up for the drops
that normally occur when supplying heavy loads. This
offset results in the best overall output accuracy over line
and load.

FLYBACK REGULATOR — INDUCTOR AND SENSE
RESISTOR SELECTION

Figure 7 shows the inductor current of the flyback
regulator. The inductor current is made up of two
components: the DC current level set by the
transconductance amplifier, I
current, ∆I

. Therefore, the selection of the inductance

L

, and the inductor ripple

SENSE

value determines how much of the output current is made
up of the inductor ripple current. Higher inductor ripple
current allows smaller inductor values, but results in
higher peak currents, lower efficiency, and higher output
voltage ripple.

I

L

V

IN

L

T

ON

V

OUT

L

T

OFF

I

OUT

∆I

∆IL =

I

SENSE

= (I

L(PEAK)

VIN T

SENSE

L

V

=

R

ON

SENSE

SENSE

+ 1/2 ∆IL)

V

VIN + V

t

IN

OUT

Figure 7. Flyback Regulator Inductor Current

9

ML4880

The relationship of the inductor current to the output
current is given by the following equation:

+

IN OUT



(18)









II I

=+



OUT SENSE L





1

∆







22

VVVR



V

IN

+

IN OUT





=+









TVLV

SENSE

ON IN IN

SENSE





×





×





VV

The inductance value can be determined by setting the
ripple current portion of Equation 18 to some percentage
of the output current. Inductor ripple currents in the range
of 50% to 80% of the maximum output current are
typical. As a good starting point set the inductor ripple
current to 65% of the maximum output current:

IN






(19)

TV

∆I

=

L

where F



×

ON IN

L

= ratio of inductor ripple current to the

IRC

FI

=× ×

IRC OUT MAX

()

VV

IN OUT





+

V

maximum output current, or:

TV

ON IN MIN

L

=

.065



×

()

I

×+

OUT MAX

()

V

IN MIN




()

VV

IN MIN OUT

()



(20)




Calculate the inductance using the volt-seconds value
given in Figure 5 at the minimum input voltage. Choose
the nearest standard value, and determine the actual
inductor ripple current at the maximum and minimum
input voltage using Equation 19 and Figure 5.

Next, determine the sense resistor value using the inductor
ripple current value calculated above using:

The winding resistance and the turns ratio of the coupled
inductor need to be specified. In general the winding
resistance should be as low as possible, preferably in the
low milliohm range. A good rule of thumb is to allow 2
mW of winding resistance per mH of inductance. The
turns ratio is specified at 1:1 for simplicity and availability.

FLYBACK REGULATOR - MOSFET SELECTION

The internal gate drive of the ML4880 limits the selection
of the switching FET to logic level types with the ON
resistance specified at VGS = 4.5V. Also, look for a FET
with a maximum total gate charge of around 70nC.

The drain to source breakdown voltage rating is
determined as follows:

VV V F

=+×()

DS IN MAX OUT DS

()

(24)

A safety derating factor, FDS, in the range of 1.5 to 2.0 is
recommended due to the voltage spikes caused by the
leakage inductance of a transformer. For a more reliable
design, look for MOSFETs that are avalanche rated.

The maximum ON resistance of the MOSFET can be
determined using the worst case power dissipation for the
package type chosen. This assumes that the major portion
of heat generated in the package is due to I2R losses.

P

R

DS ON

()

=

2

I

RMS ON

PKG

(25)

()

V

()

R

SENSE

=

I

OUT MAX

=

I

()

OUT MAX



VV

()

IN MIN OUT

()




SENSE MIN



VV

+

()

IN MIN OUT



V



V

IN MIN

014

.

+

()

()

IN MIN






−

1

∆

I

2



1

−



2



()

L MIN

∆

I

()

L MIN

(21)

Having determined the values for the inductor and sense
resistor, the inductor peak current rating can now be
specified. This value is calculated at current limit and at
the maximum input voltage, and is given by:

II I

L PK MAX SENSE MAX L MAX

(@ ) ( ) ( )

V

=+=+

=+

SENSE MAX

()

R

SENSE

∆∆

∆

025

V

I

L MAX

() ()

.

R

SENSE

I

L MAX

(22)

For reliable operation, the inductor current rating should
exceed the value calculated by 10%-20%.

Now the sense resistor’s power rating can be determined.
The sense resistor must be able to carry the peak current
in the inductor, but only during the OFF-time:

PI R

R RMS OFF SENSE

2

=×

SENSE

()

(23)

where,



()()()()

IIII



2

I

()

RMS OFF

V

IN MAX

=



VV



()

IN MAX OUT



SENSE MAX



()





+



22

++

()(@) (@)

SENSE MAX L PK MAX L PK MAX

()

3

where,

2

I

RMS ON

()



V

OUT

=



+

VV



IN MIN OUT

()



()()()()

IIII



SENSE MAX








2

++

SENSE MAX L PK MAX

()

()(@)

3

FLYBACK REGULATOR — RECTIFIER SELECTION

The output diode must be able to carry the worst case
output current as well as withstand the worst case reverse
voltage. The worst case output current occurs at current
limit and at the maximum input voltage. Therefore, the
average forward current rating of the output diode can be
determined by the following:



I

I

D AVG

OUT MAX

==+

()

FR



025

() () ()

ID SENSE

.





TV

×

ON IN MAX IN MAX

2







L

×





VVF

IN MAX OUT ID

()

where,

F derating factor

=

ID

The derating factor is typically in the range of 0.5 to 0.9
depending on reliability goals. Typically a value of 0.8 is
used for most applications.

The reverse voltage rating is determined by the sum of the
output voltage and the maximum input voltage:






V

+

L PK MAX

(@ )












(26)

2








1





10

ML4880

VV

+

()

F

VR

(27)

where,

OUT IN MAX

V

=

R

F derating factor

=

VR

Again a derating factor of 0.8 is typical for most
applications.

FLYBACK REGULATOR - INPUT CAPACITOR
SELECTION

The input capacitor should be a low ESR type and located
as close to the primary winding of the transformer as
possible. The capacitor’s ability to handle ripple current is
the selection criteria used. The input capacitor’s ripple
current is determined by the load current and the input
voltage, with the worst case condition occurring at
VIN = V

II I

RMS C SENSE MAX L MAX

() () ()

.

OUT



≈+



IN



1

∆

2












×

VV

OUT IN

+

VV

OUT IN




(28)



The capacitor must also be able to handle the maximum
input voltage. For reliable operation derate the capacitor
voltage rating by 20% to 50% for aluminum electrolytic
types and 50% to 70% for tantalum types.

FLYBACK REGULATOR - OUTPUT CAPACITOR
SELECTION

The output capacitors determine the flyback regulator’s
loop stability and the output ripple voltage. Use only low
ESR capacitors intended for switching power supply
applications. To ensure stability, the minimum capacitance
value is given by:

L MIN

()



(29)




C

OUT



V

86.

OUT










T

ON MAX

R

SENSE



≥









I

() ()

OUT MAX









∆

I

The maximum ESR value can be estimated using:

V

∆

≤

∆

OUT

I

L MAX

()

(30)

ESR

The selected capacitor must meet both the capacitance
and ESR requirements. As a final check, make sure the
output capacitor can handle the ripple current,
I

RMS(C

OUT

.

)

II

RMS C RMS C

≈

() ()

OUT IN

(31)

FLYBACK REGULATOR - OUTPUT VOLTAGE

The output of the flyback regulator is adjustable and can
be set to any voltage between 6.0V and 15.0V by
connecting a resistor divider to the feedback pin as shown
in Figure 3. The resistor values R1 and R2 can be
calculated using the following equation:



125

VV

OUT

.

=×

RR







12

+





2

R

(32)

The value of R2 should be 125kΩ or less to minimize bias
current errors.

It is important to note that the accuracy of these resistors
directly affects the accuracy of the output. Use precision
resistors and set the nominal voltage approximately 1% to
2% high in order to make up for the ML4880’s load
regulation. This offset results in the best overall output
accuracy over line and load.

LAYOUT

Proximity of passive devices and adequate power and
ground planes are critical for reliable operation of the
circuit. In general, use the top layer for the high current
connections and the bottom layer for the quiet
connections such as GND, feedback and current sense.
Some more specific guidelines follow.

1. The connection from the current sense resistor to the
I

pin should be made by a separate trace and

SENSE

located as close to the lead of the resistor as
possible. The trace length from the sense resistor to
the ML4880 should be kept as short as possible and
away from switching components and their traces.

2. The trace lengths from the buck regulator’s input
capacitor to the switching MOSFET, from the
MOSFETs to the inductor, from the synchronous
rectifier MOSFET to the SENSE resistor, and from the
inductor to the output capacitor should all be as
short as possible. The flyback regulator’s trace length
from the input capacitor to the inductor, from the
inductor to the MOSFET, from the sense resistor to
the inductor, and from the diode to the output
capacitor should also be kept short to minimize
noise and ground bounce.

3. The high current ground paths need to be kept
separate from the signal ground paths. The GND
connection should be made at a single-point star
ground. It is very important that the ground for the
ML4880 ground pins (pins 6 and 7) be made using a
separate trace.

4. Note: Concentrating on keeping the current sense
and high current connections short as well as
keeping the switching components and traces away
from the sensitive analog components and traces
during layout will eliminate the majority of problems
created by a poor layout.

5. The VIN and V
located close to the ML4880 for adequate filtering of
the IC’s internal bias voltage.

6. Remote sensing the output for improved load
regulation can be implemented with the ML4880.
Any output can be remote sensed by using the top of
the external resistor divider as the remote sense
point.

bypass capacitors need to be

REG

11

ML4880

ML4480

16

V

IN

11

SHDN A

18

SHDN B

10

SHDN C

17

N DRV C

3

I

SENSE

C

21

PWR GND C

12

V

FB

C

GND

GND

V

REG

CLAMP

5.5V to 18V

C1

33µF

25V

C2

33µF

25V

C3

33µF

25V

C4

10µF

35V

C5

0.1µF

C8

68µF

20V

C9

68µF

20V

C10

10nF

R4

887k

1%

R5

100k

1%

D1

MBRS340T3

T1

39µH

V

IN

Q1 A&B

Si9955

R6

0.12

C6

0.1µF

C7

10µF

35V

15 24

67

R12

80m

R14

124k

1%

D5*

L2

33µH

C17

10nF

R13

383k

1%

C18

100µF

10V

C19

100µF

10V

*Optional

Q5

IRF7403

Q4

IRF7406

R11

100k

D4

IN4148

C16

47nF

R8

80m

C12

47nF

Q3

IRF7403

Q2

IRF7406

R7

100k

D2

IN4148

C11

68nF

D3*

C13

10nF

R9

205k

1%

R10

124k

1%

L1

33µH

C14

100µF

10V

C15

100µF

10V

3.3V, 2A

5.0V, 2A

12V, 0.5A

3.3 SHDN

5.0 SHDN

12 SHDN

V

IN

V

IN

V

IN

CLAMP A

P DRV A

N DRV A

I

SENSE

A

PWR GND A

V

FB

A

CLAMP B

P DRV B

N DRV B

I

SENSE

B

PWR GND B

V

FB

B

21314423

9

11920522

8

12

Figure 8. Typical Application Circuit

SUGGESTED VENDORS

See Table 1 for a list of suggested component manufacturers.

TABLE 1: COMPONENT SUPPLIERS

COMPONENT MANUFACTURER PART NUMBER PHONE

SENSE RESISTORS Dale WSL Series [phone]

IRC LRC Series [phone]

INDUCTORS Coilcraft DO/DT Series [phone]

Coiltronics OCTA-PAC Series [phone]
Dale LPE-6562 Series [phone]

LPT-4545 Series
Premier Magnetics VTK & HTK Series [phone]
Pulse Engineering LCI & HCI Series [phone]
Sumida CD & CDR Series [phone]
XFMRS, Inc. Surface Mt Series [phone]

CAPACITORS AVX TPS Series [phone]

Sprague 593D Series [phone]
Sanyo SC & SA Series [phone]
Nichicon PL Series [phone]
Panasonic HFQ Series [phone]

ML4880

SEMICONDUCTORS Diodes, Inc. SK Series [phone]

International Rectifier IRF7XXX Series [phone]
Motorola MMSF Series [phone]

MMDF Series
National Semiconductor NDS94XX [phone]

NDS99XX
Siliconix Littlefoot Series [phone]

Litefoot Series

13

ML4880

PHYSICAL DIMENSIONS

0.600 - 0.614

(15.24 - 15.60)

0.050 BSC
(1.27 BSC)

0.024 - 0.034
(0.61 - 0.86)

(4 PLACES)

24

PIN 1 ID

1

inches (millimeters)

Package: S24

24-Pin SOIC

0.291 - 0.301
(7.39 - 7.65)

0.095 - 0.107
(2.41 - 2.72)

0.398 - 0.412

(10.11 - 10.47)

0º - 8º

0.090 - 0.094
(2.28 - 2.39)

0.012 - 0.020
(0.30 - 0.51)

SEATING PLANE

0.005 - 0.013
(0.13 - 0.33)

0.022 - 0.042
(0.56 - 1.07)

0.009 - 0.013
(0.22 - 0.33)

14

ML4880

ORDERING INFORMATION

PART NUMBER TEMPERATURE RANGE PACKAGE

ML4880CS0°C to 70°C24-PIN SOIC (S24) (End Of Life)

ML4880ES–20°C to 70°C24-PIN SOIC (S24) (Obsolete)

© Micro Linear 1997 is a registered trademark of Micro Linear Corporation
Products described in this document may be covered by one or more of the following patents, U.S.: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502; 5,508,570; 5,510,727; 5,523,940;
5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; Japan: 2598946; 2619299. Other patents are pending.

Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design.
Micro Linear does not assume any liability arising out of the application or use of any product described herein,
neither does it convey any license under its patent right nor the rights of others. The circuits contained in this
data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to
whether the illustrated circuits infringe any intellectual property rights of others, and will accept no responsibility
or liability for use of any application herein. The customer is urged to consult with appropriate legal counsel
before deciding on a particular application. DS4880-01

16

2092 Concourse Drive

San Jose, CA 95131

Tel: [phone]

Fax: [phone]

04/28/97 Printed in U.S.A.