Datasheet ML4863IS, ML4863CS, ML4863ES Datasheet (Micro Linear Corporation)

July 2000

FEATURING

Extended Commercial Temperature Range

-20˚C to 70˚C

for Portable Handheld Equipment

ML4863*

High Efficiency Flyback Controller

GENERAL DESCRIPTION

The ML4863 is a flyback controller designed for use in
multi-cell battery powered systems such as PDAs and

FEATURES

■ Variable frequency current mode control and

synchronous rectification for high efficiency
notebook computers. The flyback topology is ideal for
systems where the battery voltage can be either above or

■ Minimum external components

below the output voltage, and where multiple output
voltages are required.

■ Guaranteed start-up and operation over a wide input

voltage range (3.15V to 15V)
The ML4863 uses the output voltage as the feedback
control signal to the current mode variable frequency
flyback controller. In addition, a synchronous rectifier

■ High frequency operation (>200kHz) minimizes the

size of the magnetics
control output is supplied to provide the highest possible
conversion efficiency (greater than 85% efficiency over a
1mA to 1A load range).

The ML4863 has been designed to operate with a

■ Flyback topology allows multiple outputs in addition to

the regulated 5V

■ Built-in overvoltage and current limit protection

minimum number of external components to optimize
space and cost. *Some Packages Are Obsolete

BLOCK DIAGRAM

3

1

4

SHDN

V

IN

V

FB

R

gm

4.5V
LDO

I

BIAS & UVLO

18mV

18mV

CURRENT

COMPARATOR

+

COMP

–

RECTIFIER

COMPARATOR

–

COMP

+

V

FB

V

CC

SWITCHING

CONTROL

BLANKING

V

CC

5

–
+

V

REF

V

CC

A1

CROSS-CONDUCTION

PROTECTION

A2

V

CC

V

FB

GND

8

OUT 1

6

OUT 2

7

SENSE

2

1

ML4863

PIN CONFIGURATION

PIN DESCRIPTION

8-Pin SOIC (S08)

V

IN

SENSE

SHDN

V

FB

TOP VIEW

ML4863

1

2

3

4

8

GND

7

OUT 2

6

OUT 1

5

V

CC

PIN NAME FUNCTION

1V

IN

Battery input voltage

2 SENSE Secondary side current sense

3 SHDN Pulling this pin high initiates a

shutdown mode to minimize battery
drain

4V

FB

Feedback input from transformer
secondary, and supply voltage when
V

> 4.5V

OUT

PIN NAME FUNCTION

5V

CC

Internal power supply node for
connection of a bypass capacitor

6 OUT 1 Flyback primary switch MOSFET driver

output

7 OUT 2 Flyback synchronous rectifier MOSFET

driver output

8 GND Analog signal ground

2

ABSOLUTE MAXIMUM RATINGS

ML4863

Absolute maximum ratings are those values beyond which
the device could be permanently damaged. Absolute

Lead Temperature (Soldering 10 Sec.) ..................... 260ºC

Thermal Resistance (qJA) .................................... 160ºC/W

maximum ratings are stress ratings only and functional
device operation is not implied.

V

................................................................. GND – 0.3V to 18V

IN

Voltage on any other pin ........................... GND – 0.3V to 7V

Source or Sink Current (OUT1 & OUT2)...................... 1A

Junction Temperature ..............................................150ºC

Storage Temperature Range...................... –65ºC to 150ºC

OPERATING CONDITIONS

Temperature Range

ML4863CS................................................. 0ºC to 70ºC

ML4863ES ............................................. –20ºC to 70ºC

ML4863IS .............................................. –40ºC to 85ºC

VIN Operating Range ...................................3.15V to 15V

ELECTRICAL CHARACTERISTICS

Unless otherwise specified, VIN = 12V, TA = Operating Temperature Range (Note 1)

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

OSCILLATOR

t

ON

ON Time C Suffix 2.1 2.5 2.8 µs

E/I Suffix 2.1 2.5 2.95 µs

Minimum Off Time VFB = 0V 450 650 850 ns

VFB REGULATION

Total Variation Line, Load, & Temp 4.85 5 5.15 V

OUTPUT DRIVERS

OUT1 Rise Time C

OUT1 Fall Time C

OUT2 Rise Time C

OUT2 Fall Time Continuous Mode, C

SHDN

Input High Voltage 2.0 V

Input Low Voltage 0.8 V

Input Bias Current SHDN = 5V 5 10 µA

SENSE

SENSE Threshold – Full Load VIN = 5V, VFB = V

SENSE Threshold – Short Circuit VFB = 0V 235 mV

CIRCUIT PROTECTION

= 3nF, 20% to 90% of V

LOAD

= 3nF, 90% to 20% of V

LOAD

= 3nF, 20% to 90% of V

LOAD

90% to 20% of V

Discontinuous Mode, C
90% to 20% of V

LOAD

CC

CC

FB (No Load)

CC

CC

CC

= 3nF,

= 3nF,

LOAD

– 100mV 130 150 160 mV

60 70 ns

60 70 ns

60 70 ns

60 70 ns

125 150 ns

Undervoltage Lockout Start-up Threshold 3.0 3.15 V

Undervoltage Lockout Hysteresis 0.5 0.6 V

3

ML4863

ELECTRICAL CHARACTERISTICS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

SUPPLY

I

FB

I

V

Note 1: Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.

VFB Quiescent Current 100 150 µA

VIN Shutdown Current SHDN = 5V 20 25 µA

IN

VCC Output Voltage VFB = 0V, VIN = 15V, C

CC

(Continued)

SHDN = 5V, VIN < 6V 5 10 µA

= 0.1µF 4.5 5.5 V

VCC

VFB = 0V, VIN = 6V, C

VFB = 0V, VIN = 3.15V, C

VFB = 5V 4.5 5 5.15 V

= 0.1µF 4.0 5.0 V

VCC

= 0.1µF 2.8 V

VCC

4

FUNCTIONAL DESCRIPTION

ML4863

The ML4863 utilizes a flyback topology with constant on­time control. The circuit determines the length of the off­time by waiting for the inductor current to drop to a level
determined by the feedback voltage (VFB). Consequently,
the current programming is somewhat unconventional
because the valley of the current ripple is programmed
instead of the peak. The controller automatically enters
burst mode when the programmed current falls below
zero. Constant on-time control therefore features a
transition into and out of burst mode which does not
require additional control circuitry.

The control circuit is made up of four distinctive blocks;
the constant on-time oscillator, the current programming
comparator, the feedback transconductance amplifier, and
the synchronous rectifier controller. A simplified circuit
diagram is shown in Figure 1.

OSCILLATOR & COMPARATOR

The oscillator has a constant on-time and a minimum off­time. The off-time is extended as long as the output of the
current programming comparator is low. Note that in
constant on-time control, a discharge (off-time) cycle is
needed for the inductor current to be sensed. The
minimum off-time is required to account for the finite
circuit delays in sensing the inductor output current.

TRANSCONDUCTANCE AMPLIFIER

The feedback transconductance amplifier generates a
current from the voltage difference between the output
and the reference. This current produces a voltage across
Rgm that adds to the negative voltage on the current sense
resistor, R

. When the current level in the inductor

SENSE

drops low enough to cause the voltage at the non-inverting
input of the current programming comparator to go
positive, the comparator trips and the converter starts a
new on-cycle. The current programming comparator
controls the length of the off-time by waiting until the
current in the secondary decreases to the value specified
by the feedback transconductance amplifier.

In this way, the feedback transconductance amplifier‘s
output current steers the current level in the inductor.
When the output voltage drops due to a load increase, it
will increase the output current of the feedback amplifier
and generate a larger voltage across Rgm which in turn
raises the secondary current trip level. However, when the
output voltage is too high, the feedback amplifier’s output
current will eventually become negative. Because the
output current of the inductor can never go negative by
virtue of the diode, the non-inverting input of the
comparator will also stay negative. This causes the
converter to stop operation until the output voltage drops
enough to increase the output current of the feedback
transconductance amplifier above zero.

4

V

FB

R

P

C

P

ML4863

FEEDBACK

TRANSCONDUCTANCE

AMPLIFIER

+

REF

–

V

PROGRAMMING

COMPARATOR

R

gm

COMPARATOR

CURRENT

+

COMP

–

RECTIFIER

–

COMP

+

CONSTANT ON-TIME
MINIMUM OFF-TIME

OSCILLATOR

ONE SHOT

t

OFF

500ns

BLANKING

ONE SHOT

t

ON

2.5µs

A2

OUT 2

SENSE

OUT 1

6

7

2

V

OUT

I

V

IN

L

S

1:1

P

R

SENSE

R

ESR

LOAD

C

Figure 1. Schematic of the ML4863 Controller and Power Stage

5

ML4863

FUNCTIONAL DESCRIPTION (Continued)

SYNCHRONOUS RECTIFIER CONTROL

The control circuitry for the synchronous rectifier does not
influence the operation of the main controller. The
synchronous rectifier is turned on during the minimum off
time, or whenever the SENSE pin is less than –18mV.
During transitions where the primary switch is turned on
before the voltage on the SENSE pin goes above –18mV,
the gate of the synchronous rectifier is discharged softly to
avoid accidently triggering the current-mode comparator
with the gate discharge spike on the sense resistor.

The part will also operate with a Schottky diode in place
of the synchronous rectifier, but the conversion efficiency
will suffer.

CURRENT LIMIT AND MODES OF OPERATION

The normal operating range and current limit point are
determined by the current programming comparator. They
are dependent on the value of the synchronous rectifier
current sense resistor (R
primary inductance (LP), and the input voltage.

R

can be calculated by:

SENSE

R

SENSE

V

IN MIN

=

VVmVI

OUT IN



05

´+



+



), the nominal transformer

SENSE

150

OUT MAX

05

V

IN MIN

05

VI

20

´´

IN MAX OUT MAX

05 05



´

h



(1)



where h = converter efficiency.

Once R

LVR

P

Three operational modes are defined by the voltage at the
SENSE pin at the end of the off-time: discontinuous mode,
continuous mode, and current limit. The following values
can be used to determine the current levels of each mode:

V

SENSE

0V < V

160mV < V

Inserting the maximum value of V
operational mode into the following equation will
determine the maximum current levels for each
operational mode:

I

OUT

has been determined, LP can be found:

SENSE

−

=× × ×

()25 10

< 0V: discontinuous mode

SENSE

=

6

IN MAX

05

< 160mV: continuous mode

< 235mV: current limit

SENSE

V

IN

VV

+

OUT IN



V

×+



R



SENSE

SENSE

SENSE

SENSE

tV

ON IN

×

2

for each



×

×

η



L



P

(2)

(3)

6

DESIGN CONSIDERATIONS

ML4863

DESIGN PROCEDURE

A typical design can be implemented by using the
following procedure.

1. Specify the application by defining:

The maximum input voltage (V
The mainimum input voltage (V
The maximum output current (I
The maximum output ripple (DV

As a general design rule, the output ripple should be kept
below 100mV to ensure stability.

2. Select a sense resistor, R

3a. Determine the inductance required for the

optimum output ripple using equation 2.

3b. Determine the minimum inductor current rating

required. The peak inductor current is calculated
using the following formula:

I

LPEAK

3c. Specify the inductor's DC winding resistance. A

3d. Specify the coupled inductor's turns ratio:

235

=+

R

SENSE

good rule of thumb is to allow 5mW, or less, of
resistance per µH of inductance. For minimum
core loss, choose a high frequency core material
such as Kool-Mu, ferrite, or MPP.

IN MAX

Np : Ns = 1:1

V

mV



()

L

IN(MAX)

, using equation 1.

SENSE

25 10

(. )

P

)

)

IN(MIN)

OUT(MAX)

)

OUT

-

6

)

(4)

See Table 1 for suggested component manufacturers.

Part

Component Manufacturer Number Phone

Sense Dale LRC Series [phone]
Resistors IRC WSL Series [phone]

Inductors Coilcraft R4999 [phone]

Coiltronics OCTA-PAC Series [phone]
Dale LPE-6562 Series [phone]

LPT-4545 series

Capacitors AVX TPS series [phone]

Sprague 593D Series [phone]

MOSFETs National NDS94XX [phone]

NDS99XX

Motorola MMDF Series [phone]

MMSF Series

Siliconix Littlefoot Series [phone]

Table 1. Component Suppliers

DESIGN EXAMPLE

1. Specify the application by defining:
V

IN(MAX)

V

IN(MIN)

I

OUT(MAX)

DV

2. Select the sense resistor, R

R

SENSE

OUT

=

54

= 6V

= 4V

= 500mA

= 100mV

4

+

mV

150



×+





mA

500

, using Equation 1:

SENSE

V

4

××

20605






×

085..

(1a)

4a. Calculate the minimum output capacitance

required using:

-

VV

CI

=

OUT MAX

4b. Establish the maximum allowable ESR for the

ouput capacitor:

R

5. As a final design check, evaluate the system

D

<

ESR

stability (no compensation, single pole response)
by using the following equation:

∆V

≤× ×

OUT

where R
used.



()




VR



OUT SENSE

mV

150

()

610

SENSE

+

OUT IN MAX

VV

OUT OUT

RVV



−

SENSE OUT IN MIN

6

!

and LP are the actual values to be



()

×+

.25 10






()

L

P

6



D

()

3a. Determine the inductance required using

(5)

3b. Determine the minimum inductor current rating

(6)

"
#

(7)

$

= 138mW @ 120mW

R

SENSE

equation 2.

−

235

120

6

mV
m

Ω

L

=× ××

().25 10 6 0 12

P

required.

I

=+

LPEAK

= 18µH (2a)

−

××

62510

(. )

×

18 10

6

=

279

–6

.

(4a)

A

7

ML4863

DESIGN CONSIDERATIONS (Continued)

3c. Specify the inductor’s DC winding resistance:

L

= 90mW

DCR

3d. Specify the coupled inductor's turn ratio:

Np : Ns = 1:1

4a. Calculate the minimum output capacitance

required using equation 5.

−

050

.





C =×

56525 10

+



×




.

6

×

= 55µF (5a)

01

.

4b. Establish the maximum ESR for the output

capacitor using equation 6.

ESR

..

<

150

= 80mW (6a)

mV

R

×01 012

Based on these calculations, the design should use two
100µF capacitors, with an ESR of 100mW each, in parallel
to meet the capacitance and ESR requirements.

5. As a final design check, evaluate the system
stability using equation 7.

LAYOUT

Good PC board layout practices will ensure the proper
operation of the ML4863. Important layout considerations
follow:

• The connection from the current sense resistor to the
SENSE pin of the ML4863 should be made by a
separate trace and connected right at the sense resistor
lead.

• The VCC bypass capacitor needs to be located close to
the ML4863 for adequate filtering of the IC's internal
bias voltage.

• Trace lengths from the capacitors to the inductor, and
from the inductor to the FET should be as short as
possible to minimize noise and ground bounce.

• Power and ground planes must be large enough to
handle the current the converter has been designed for.

See Figure 5 for a sample PC board layout.

100 6 10

mV ≤× ×

()

−

6



18 10

!

×

"

= 360mV (7a)

–6

#

$

012 5 4

×+

.()

Since the inequality is met, the circuit should be stable.

Some typical application circuits are shown in Figures 2, 3,
and 4.

V

OUT

5V, 1A

400µF

V

IN

47µF

ML4863

V

IN

SENSE

SHDN

V

FB

GND

OUT 2

OUT 1

V

CC

Coiltronics

CTX20-4

NDS9955

1µF

V

IN

100µF

ML4863

V

IN

SENSE

SHDN

V

FB

GND

OUT 2

OUT 1

V

CC

1µF

NDS9410

Dale

LPE6562

NDS9410

V
5V, 2A

800µF

OUT

100mΩ

50mΩ

Figure 3. 5V, 2A CircuitFigure 2. 5V, 1A Circuit

8

ML4863

V

SHDN

C4

33µF

20V

T1

DALE

LPE-6562-A145

6,10

NDS9955

Q1A Q1B

ML4863

V

GND

IN

C1

33µF

20V

C2

33µF

20V

IN

SENSE

SHDN

V

FB

OUT 2

OUT 1

V

CC

C3

1µF

50V

798

1,5

423

Q2A Q2B

R1

120mΩ

R2

30mΩ

R3

60mΩ

C5

33µF

20V

C6

100µF

6.3V

12V

C7

100µF

6.3V

C10

100µF

6.3V

MMDF3N03

C8

100µF

6.3V

C11

100µF

6.3V

C9

100µF

6.3V

C12

100µF

6.3V

5V

C13

100µF

6.3V

3.3V

Figure 4. 5W Multiple Output DC-DC Converter

Figure 5. Typical PC Board Layout

9

ML4863

PHYSICAL DIMENSIONS inches (millimeters)

0.189 - 0.199
(4.80 - 5.06)

8

Package: S08

8-Pin SOIC

0.017 - 0.027
(0.43 - 0.69)

(4 PLACES)

0.055 - 0.061

(1.40 - 1.55)

PIN 1 ID

1

0.050 BSC
(1.27 BSC)

0.012 - 0.020
(0.30 - 0.51)

SEATING PLANE

0.148 - 0.158
(3.76 - 4.01)

0.059 - 0.069
(1.49 - 1.75)

0.228 - 0.244
(5.79 - 6.20)

0.004 - 0.010
(0.10 - 0.26)

0º - 8º

0.015 - 0.035
(0.38 - 0.89)

0.006 - 0.010
(0.15 - 0.26)

ORDERING INFORMATION

PART NUMBER TEMPERATURE RANGE PACKAGE

ML4863CS 0ºC to 70ºC 8-Pin SOIC (S08)
ML4863ES –20ºC to 70ºC 8-Pin SOIC (S08)

ML4863IS (Obsolete) –40ºC to 85ºC 8-Pin SOIC (S08)

© Micro Linear 1997. is a registered trademark of Micro Linear Corporation. All other trademarks are the property of their respective owners.

Products described herein may be covered by one or more of the following U.S. patents: 4,897,611; 4,964,026; 5,027,116; 5,281,862; 5,283,483; 5,418,502;
5,508,570; 5,510,727; 5,523,940; 5,546,017; 5,559,470; 5,565,761; 5,592,128; 5,594,376; 5,652,479; 5,661,427; 5,663,874; 5,672,959; 5,689,167. Japan: 2,598,946;
2,619,299; 2,704,176. Other patents are pending.

Micro Linear reserves the right to make changes to any product herein to improve reliability, function or design. Micro Linear does not assume any liability
arising out of the application or use of any product described herein, neither does it convey any license under its patent right nor the rights of others. The circuits
contained in this data sheet are offered as possible applications only. Micro Linear makes no warranties or representations as to whether the illustrated circuits
infringe any intellectual property rights of others, and will accept no responsibility or liability for use of any application herein. The customer is urged to consult
with appropriate legal counsel before deciding on a particular application.

10

2092 Concourse Drive

San Jose, CA 95131

Tel: [phone]

Fax: [phone]

www.microlinear.com

DS4863-01