Datasheet MIC2168 Datasheet (Micrel)

MIC2168 Micrel, Inc.

1.2µH

3.3V

VIN= 5V

VDD

COMP/EN

VIN

CS

FB

GND

LSD

BST

1kΩ

10kΩ

4kΩ

3.24kΩ

4.7µF

100µF

0.1µF

100nF

IRF7821

SD103BWS

IRF7821

100pF

HSD

VSW

MIC2168

150µF x 2

MIC2168

1MHz PWM Synchronous Buck Control IC

General Description

The MIC2168 is a high-efficiency, simple to use 1MHz PWM
synchronous buck control IC housed in a small MSOP-10
package. The MIC2168 allows compact DC/DC solutions
with a minimal external component count and cost.

The MIC2168 operates from a 3V to 14.5V input, without
the need of any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range.

The MIC2168 senses current across the high-side N-Chan­nel MOSFET, eliminating the need for an expensive and
lossy current-sense resistor. Current limit accuracy is main­tained by a positive temperature coefficient that tracks the
increasing R

DS(ON)

and space are saved by the internal in-rush-current limiting
digital soft-start.

The MIC2168 is available in a 10-pin MSOP package, with a
wide junction operating range of –40°C to +125°C.

All support documentation can be found on Micrel’s web site
at www.micrel.com.

of the external MOSFET. Further cost

Features

• 3V to 14.5V input voltage range

• Adjustable output voltage down to 0.8V

• Up to 95% efficiency

• 1MHz PWM operation

• Adjustable current-limit senses high-side N-Channel
MOSFET current

• No external current sense resistor

• Adaptive gate drive increases efficiency

• Ultra-fast response with hysteretic transient recovery
mode

• Overvoltage protection protects the load in fault
conditions

• Dual mode current limit speeds up recovery time

• Hiccup mode short-circuit protection

• Internal soft-start

• Dual function COMP and EN pin allows low-power shut­down

• Small size MSOP 10-lead package

Applications

• Point-of-load DC/DC conversion

• Set-top boxes

• Graphic cards

• LCD power supplies

• Telecom power supplies

• Networking power supplies

• Cable modems and routers

Typical Application

Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 [phone] • fax + 1 [phone] • http://www.micrel.com

April 2005 1 M9999-040805

MIC2168 Adjustable Output 1MHz Converter

MIC2168 Micrel, Inc.

FB GND65

1VIN

VDD

CS

COMP/EN

10 BST

HSD

VSW

LSD

9

8

7

2

3

4

Ordering Information

Part Number Frequency Junction Temp. Range Package

Standard Pb-Free

MIC2168BMM MIC2168YMM 1MHz -40°C to +125°C 10-Lead MSOP

Pin Configuration

10-Pin MSOP (MM)

Pin Description

Pin Number Pin Name Pin Function

1 VIN Supply Voltage (Input): 3V to 14.5V.

2 VDD 5V Internal Linear Regulator (Output): V

drive supply voltage and an internal supply bus for the IC. When VIN is <5V,
this regulator operates in dropout mode.

3 CS Current Sense / Enable (Input): Current-limit comparator noninverting input.

The current limit is sensed across the MOSFET during the ON time. The cur
rent can be set by the resistor in series with the CS pin.

4 COMP/EN Compensation (Input): Dual function pin. Pin for external compensation.

If this pin is pulled below 0.2V, with the reference fully up the device shuts
down (50µA typical current draw).

5 FB Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.

6 GND Ground (Return).

7 LSD Low-Side Drive (Output): High-current driver output for external synchro

nous MOSFET.

8 VSW Switch (Return): High-side MOSFET driver return.

9 HSD High-Side Drive (Output): High-current output-driver for the high-side

MOSFET. When V
should be used. At V

10 BST Boost (Input): Provides the drive voltage for the high-side MOSFET driver.

The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.

is between 3.0V to 5V, 2.5V threshold-rated MOSFETs

IN

> 5V, 5V threshold MOSFETs should be used.

IN

is the external MOSFET gate

DD

-

-

M9999-040805 2 April 2005

MIC2168 Micrel, Inc.

Absolute Maximum Ratings

(1)

Supply Voltage (VIN) ................................................... 15.5V

Booststrapped Voltage (V
Junction Temperature (T
Storage Temperature (T

) .................................VIN +5V

BST

) ...................–40°C ≤ TJ ≤+125°C

J

) ........................ –65°C to +150°C

S

Operating Ratings

Supply Voltage (VIN) ..................................... +3V to +14.5V

Output Voltage Range ..........................

Package Thermal Resistance

θJA 10-lead MSOP ............................................. 180°C/W

(2)

0.8V to VIN × D

MAX

Electrical Characteristics

(3)

TJ = 25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C

Parameter Condition Min Typ Max Units

Feedback Voltage Reference (± 1%) 0.792 0.8 0.808 V

Feedback Voltage Reference (± 2% over temp)

Feedback Bias Current 30

0.784 0.8 0.816 V

100 nA

Output Voltage Line Regulation 0.03 % / V

Output Voltage Load Regulation 0.5 %

Output Voltage Total Regulation 3V ≤ V

≤ 14.5V; 1A ≤ I

IN

≤ 10A; (V

OUT

OUT

= 2.5V)

(4)

0.6 %

Oscillator Section

Oscillator Frequency 900 1000 1100 kHz

Maximum Duty Cycle 90 %

(4)

Minimum On-Time

30 60 ns

Input and VDD Supply

PWM Mode Supply Current V

= VIN –0.25V; VFB = 0.7V (output switching but excluding 1.6 3 mA

CS

external MOSFET gate current.)

Shutdown Quiescent Current V

V

Shutdown Threshold 0.1 0.25 0.4 V

COMP

V

Shutdown Blanking C

COMP

COMP/EN

COMP

= 0V 50 150 µA

= 100nF 4 ms

Period

Digital Supply Voltage (V

) VIN ≥ 6V 4.7 5 5.3 V

DD

Notes:

1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, T
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature, and the regulator will go into thermal shutdown.

2. Devices are ESD sensitive, handling precautions required.

3. Specification for packaged product only.

4. Guaranteed by design.

(max),

J

April 2005 3 M9999-040805

MIC2168 Micrel, Inc.

Electrical Characteristics

Parameter Condition Min Typ Max Units

Error Amplifier

DC Gain 70 dB

Transconductance 1 ms

Soft-Start

Soft-Start Current After timeout of internal timer. See

Current Sense

CS Over Current Trip Point V

Temperature Coefficient

Output Fault Correction Thresholds

Upper Threshold, V

Lower Threshold, V

Gate Drivers

Rise/Fall Time Into 3000pF at V

Output Driver Impedance Source, V

Sink, V

Source, V

Sink, V

Driver Non-Overlap Time

Notes:

5. Specification for packaged product only.

6. Guaranteed by design.

(relative to VFB) +3 %

FB_OVT

(relative to VFB) –3 %

FB_UVT

(5)

“Soft-Start” section. 8.5 µA

= VIN –0.25V 160 200 240 µA

CS

+1800 ppm/°C

> 5V 30 ns

IN

= 5V 6 Ω

IN

= 5V 6 Ω

IN

= 3V 10 Ω

IN

= 3V 10 Ω

IN

Note 6 10 20 ns

M9999-040805 4 April 2005

MIC2168 Micrel, Inc.

Typical Characteristics

VIN = 5V

April 2005 5 M9999-040805

MIC2168 Micrel, Inc.

Current Limit

Reference

Current Limit

Comparator

Error

Amp

Low-Side

Driver

High-Side

Driver

PWM

Comparator

FB

COMP

GND

LSD

V

REF

+3%

V

REF

3%

HSD

V

DD

C

BST

CS

V

DD

5V

5V

5V

C2

C1

R1

5V

0.8V

V

IN

SW

Q2

Q1

L1

Driver

Logic

0.8V

BG Valid

Clamp &
Startup
Current

Enable
Error
Loop

Hys

Comparator

5V LDO

Bandgap

Reference

Soft-Start &

Digital Delay

Counter

MIC2168

Ramp
Clock

BOOST

R2

R3

4W

RSW

RCS

C

OUT

V

OUT

C

IN

D1

Functional Diagram

MIC2168 Block Diagram

Functional Description

The MIC2168 is a voltage mode, synchronous step-down
switching regulator controller designed for high output power
without the use of an external sense resistor. It includes an
internal soft-start function which reduces the power supply
input surge current at start-up by controlling the output volt­age rise time, a PWM generator, a reference voltage, two
MOSFET drivers, and short-circuit current limiting circuitry
to form a complete 1MHz switching regulator.

Theory of Operation

The MIC2168 is a voltage mode step-down regulator. The
figure above illustrates the block diagram for the voltage
control loop. The output voltage variation due to load or line
changes will be sensed by the inverting input of the transcon­ductance error amplifier via the feedback resistors R3, and
R2 and compared to a reference voltage at the non-invert­ing input. This will cause a small change in the DC voltage
level at the output of the error amplifier which is the input to
the PWM comparator. The other input to the comparator is
a 0 to 1V triangular waveform. The comparator generates
a rectangular waveform whose width tON is equal to the
time from the start of the clock cycle t0 until t1, the time the
triangle crosses the output waveform of the error amplifier.
To illustrate the control loop, let us assume the output volt­age drops due to sudden load turn-on, this would cause the
inverting input of the error amplifier which is divided down

M9999-040805 6 April 2005

version of V
causing the output voltage of the error amplifier to go high.
This will cause the PWM comparator to increase tON time of
the top side MOSFET, causing the output voltage to go up
and bringing V

Soft-Start

The COMP/EN pin on the MIC2168 is used for the following
three functions:

1. Disables the part by grounding this pin

2. External compensation to stabilize the voltage
control loop

3. Soft-start

For better understanding of the soft-start feature, let’s as­sume V
by un-grounding the COMP/EN pin. The COMP pin has an
internal 8.5µA current source that charges the external com-

IN

pensation capacitor. As soon as this voltage rises to 180mV
(t = Cap_COMP × 0.18V/8.5µA), the MIC2168 allows the
internal VDD linear regulator to power up and as soon as it
crosses the undervoltage lockout of 2.6V, the chip’s internal
oscillator starts switching. At this point in time, the COMP
pin current source increases to 40µA and an internal 11-bit
counter starts counting which takes approximately 2ms to
complete. During counting, the COMP voltage is clamped
at 0.65V. After this counting cycle the COMP current source

to be slightly less than the reference voltage

OUT

back in regulation.

OUT

= 12V, and the MIC2168 is allowed to power-up

MIC2168 Micrel, Inc.

V

V

OUT

IN

t4











x

0.5

- x

Cap_COMP

8.5µA

L1 Inductor

V

IN

HSD

LSD

RCS

CS

200µA

0

C2
C

IN

C1
C

OUT

Q1
MOSFET N

Q2
MOSFET N

V

OUT

R

CS

=

200µA

R

DS(ON) Q1

× L

L

I

L

=

2(Inductor Ripple Current)

1

I

LOAD

=

(V

IN

V

OUT

=

- V

OUT

)

V

IN

F

SWITCHING

L

× ×

is reduced to 8.5µA and the COMP pin voltage rises from

0.65V to 0.95V, the bottom edge of the saw-tooth oscillator.
This is the beginning of 0% duty cycle and it increases slowly
causing the output voltage to rise slowly. The MIC2168 has
two hysteretic comparators that are enabled when V
within ±3% of steady state. When the output voltage reaches
97% of programmed output voltage, then the gm error amplifier
is enabled along with the hysteretic comparator. This point
onwards, the voltage control loop (gm error amplifier) is fully
in control and will regulate the output voltage.

Soft-start time can be calculated approximately by adding
the following four time frames:

t1 = Cap_COMP × 0.18V/8.5µA

t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5µA

Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 +
t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms

Current Limit

The MIC2168 uses the R
to measure output current. Since it uses the drain to source
resistance of the power MOSFET, it is not very accurate. This
scheme is adequate to protect the power supply and external
components during a fault condition by cutting back the time
the top MOSFET is on if the feedback voltage is greater than

0.67V. In case of a hard short when feedback voltage is less
than 0.67V, the MIC2168 discharges the COMP capacitor to

0.65V, resets the digital counter and automatically shuts off
the top gate drive, and the gm error amplifier and the –3%
hysteretic comparators are completely disabled and the
soft-start cycles restarts. This mode of operation is called the
“hiccup mode” and its purpose is to protect the down stream
load in case of a hard short. The circuit in Figure 1 illustrates
the MIC2168 current limiting circuit.

Figure 1. The MIC2168 Current Limiting Circuit

April 2005 7 M9999-040805

of the top power MOSFET

DS(ON)

The current limiting resistor RCS is calculated by the follow­ing equation:

is

OUT

Equation (1)

where:
Inductor Ripple Current =

F

SWITCHING

= 1MHz

200µA is the internal sink current to program the MIC2168
current limit.

The MOSFET R

varies 30% to 40% with temperature;

DS(ON)

therefore, it is recommended to add a 50% margin to the load
current (I

) in the above equation to avoid false current

LOAD

limiting due to increased MOSFET junction temperature rise.
It is also recommended to connect RCS resistor directly to
the drain of the top MOSFET Q1, and the RSW resistor to the
source of Q1 to accurately sense the MOSFETs R

DS(ON)

. A

0.1µF capacitor in parallel with RCS should be connected to
filter some of the switching noise.

Internal VDD Supply

The MIC2168 controller internally generates VDD for self bias­ing and to provide power to the gate drives. This VDD supply
is generated through a low-dropout regulator and generates
5V from VIN supply greater than 5V. For supply voltage less
than 5V, the VDD linear regulator is approximately 200mV
in dropout. Therefore, it is recommended to short the VDD
supply to the input supply through a 10Ω resistor for input
supplies between 2.9V to 5V.

MOSFET Gate Drive

The MIC2168 high-side drive circuit is designed to switch an
N-Channel MOSFET. The block diagram in Figure 2 shows a
bootstrap circuit, consisting of D2 and CBST, supplies energy
to the high-side drive circuit. Capacitor CBST is charged while
the low-side MOSFET is on and the voltage on the VSW pin
is approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the MOSFET turns on, the voltage on the VSW pin
increases to approximately VIN. Diode D2 is reversed biased
and CBST floats high while continuing to keep the high-side
MOSFET on. When the low-side switch is turned back on,
CBST is recharged through D2. The drive voltage is derived
from the internal 5V VDD bias supply. The nominal low-side
gate drive voltage is 5V and the nominal high-side gate drive
voltage is approximately 4.5V due the voltage drop across D2.
An approximate 20ns delay between the high- and low-side
driver transitions is used to prevent current from simultane­ously flowing unimpeded through both MOSFETs.

MOSFET Selection

The MIC2168 controller works from input voltages of 3V to

13.2V and has an internal 5V regulator to provide power to
turn the external N-Channel power MOSFETs for high- and
low-side switches. For applications where V

< 5V, the internal

IN

MIC2168 Micrel, Inc.

I Q f

G[high-side](avg) G S

= ×

I C V f

G[low -side](avg) ISS GS S

= × ×

P V I I

GATEDRIVE IN G[high -side](avg) G[low -side](avg)

= +

( )

P P P

SW CONDUCTION AC

= +

P I R

CONDUCTION

SW(rms)

SW

2

= ×

P P P

AC AC(off) AC(on)

= +

D

=

V

V

O

IN











duty cycle

t

C V C V

I

T

ISS GS OSS IN

G

=

× + ×

P (V V ) I t f

AC IN D PK T S

= + × × ×

L

V (V m

ax V )

V m

ax f 0.2 I max

OUT IN OUT

IN S OUT

=

× −

× ×

×

( )

( ) ( )

VDD regulator operates in dropout mode, and it is necessary
that the power MOSFETs used are low threshold and are in
full conduction mode for VGS of 2.5V. For applications when
V

> 5V; logic-level MOSFETs, whose operation is specified

IN

at VGS = 4.5V must be used.

It is important to note the on-resistance of a MOSFET increases
with increasing temperature. A 75°C rise in junction tempera­ture will increase the channel resistance of the MOSFET by
50% to 75% of the resistance specified at 25°C. This change
in resistance must be accounted for when calculating MOSFET
power dissipation and in calculating the value of current-sense
(CS) resistor. Total gate charge is the charge required to turn
the MOSFET on and off under specified operating conditions
(VDS and VGS). The gate charge is supplied by the MIC2168
gate drive circuit. At 1MHz switching frequency and above, the
gate charge can be a significant source of power dissipation
in the MIC2168. At low output load, this power dissipation is
noticeable as a reduction in efficiency. The average current
required to drive the high-side MOSFET is:

where:

I

G[high-side](avg)

= average high-side MOSFET gate

current.

QG = total gate charge for the high-side MOSFET taken from
manufacturer’s data sheet for VGS = 5V.

The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during this
time. The switching loss for the low-side MOSFET is usu­ally negligible. Also, the gate-drive current for the low-side
MOSFET is more accurately calculated using CISS at
VDS = 0 instead of gate charge.

For the low-side MOSFET:

Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2168 due to gate
drive is:

A convenient figure of merit for switching MOSFETs is the on
resistance times the total gate charge R

DS(ON)

× QG. Lower

numbers translate into higher efficiency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2168.

Parameters that are important to MOSFET switch selection
are:

• Voltage rating

• On-resistance

• Total gate charge

The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
should be added to the VDS(max) of the MOSFETs to account
for voltage spikes due to circuit parasitics.

The power dissipated in the switching transistor is the sum
of the conduction losses during the on-time (P

CONDUCTION

and the switching losses that occur during the period of time
when the MOSFETs turn on and off (P

AC

).

where:

RSW = on-resistance of the MOSFET switch

Making the assumption the turn-on and turn-off transition times
are equal; the transition times can be approximated by:

where:

C

ISS

and C

are measured at VDS = 0

OSS

IG = gate-drive current (1A for the MIC2168)

The total high-side MOSFET switching loss is:

where:

tT = switching transition time (typically 20ns to 50ns)

VD = freewheeling diode drop, typically 0.5V

fS it the switching frequency, nominally 1MHz

The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.

Inductor Selection

Values for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak induc­tor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFETs. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple cur­rent. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.

where:

fS = switching frequency, 1MHz

0.2 = ratio of AC ripple current to DC output current

VIN(max) = maximum input voltage

The peak-to-peak inductor current (AC ripple current) is:

)

M9999-040805 8 April 2005

MIC2168 Micrel, Inc.

I

V (V m

ax V )

V m

ax f L

PP

OUT IN OUT

IN S

=

× −

× ×

( )

( )

I I max 0.5 I

PK OUT PP

= + ×( )

I I max 1

1

3

I

I max

INDUCTOR(rms) OUT

P

OUT

2

= × +











( )

( )

P I R

INDUCTOR Cu

INDUCTOR(rms)

WINDING

2

= ×

R R 1 0.0042 (T T )

WINDING(hot) WINDING(20 C) HOT 20 C

= × + × −

( )

° °

R

V

I

ESR

OUT

PP

≤

∆

∆V

I (1 D

)

C f

I R

OUT

PP

OUT S

2

PP ESR

2

=

× −

×













+ ×

( )

I

I

12

C

PP

OUT(rms)

=

P I R

DISS(C C ESR(C )

OUT

OUT(rms)

2

OUT

)

= ×

∆V I R

IN INDUCTOR(peak) ESR(C )

IN

= ×

The output capacitor ESR also affects the overall voltage
feedback loop from stability point of view. See “Feedback Loop

The peak inductor current is equal to the average output current

Compensation” section for more information. The maximum
value of ESR is calculated:

plus one half of the peak-to-peak inductor ripple current.

The RMS inductor current is used to calculate the I2 × R
losses in the inductor.

Maximizing efficiency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2168 requires the use of fer­rite materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the efficiency of the power supply.
This is especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At lower
output currents, the core losses can be a significant con­tributor. Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is calculated
by the equation below:

The resistance of the copper wire, R

WINDING

, increases with
temperature. The value of the winding resistance used should
be at the operating temperature.

where:

T

T

R

WINDING(20°C)

= temperature of the wire under operating load

HOT

= ambient temperature

20°C

is room temperature winding resistance (usu-

ally specified by the manufacturer)

Output Capacitor Selection

The output capacitor values are usually determined capaci­tors ESR (equivalent series resistance). Voltage and RMS
current capability are two other important factors selecting
the output capacitor. Recommended capacitors tantalum,
low-ESR aluminum electrolytics, and POSCAPS. The output
capacitor’s ESR is usually the main cause of output ripple.

where:

V

= peak-to-peak output voltage ripple

OUT

IPP = peak-to-peak inductor ripple current

The total output ripple is a combination of the ESR output
capacitance. The total ripple is calculated below:

where:

D = duty cycle

C

= output capacitance value

OUT

fS = switching frequency

The voltage rating of capacitor should be twice the voltage for
a tantalum and 20% greater for an aluminum electrolytic.

The output capacitor RMS current is calculated below:

The power dissipated in the output capacitor is:

Input Capacitor Selection

The input capacitor should be selected for ripple current rating
and voltage rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the input
supply on. Tantalum input capacitor voltage rating should
be at least 2 times the maximum input voltage to maximize
reliability. Aluminum electrolytic, OS-CON, and multilayer
polymer film capacitors can handle the higher inrush currents
without voltage derating. The input voltage ripple will primarily
depend on the input capacitor’s ESR. The peak input current
is equal to the peak inductor current, so:

The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at
the maximum output current. Assuming the peak-to-peak
inductor ripple current is low:

April 2005 9 M9999-040805

MIC2168 Micrel, Inc.

I I max D (1 D)

C (rms) OUT

IN

≈ × × −( )

P I R

DISS(C )

C (rms)

ESR(C )

IN

IN

2

IN

= ×

Error
Amp

7

MIC2168 [adj.]

FB

V

REF

0.8V

R2

R1

V V 1

R1

R2

O REF

= × +











R2

V R

1

V V

REF

O REF

=

×

−

V V

DIODE(rrm)

IN

=

P I V

DIODE D(avg) F

= ×

ESR

C

OUT

V

O

DCRL

(1 + ESR × s × C)











-

G

(S)

DCR × s × C + s2× L × C + 1 + ESR × s × C

The power dissipated in the input capacitor is:

Voltage Setting Components

The MIC2168 requires two resistors to set the output voltage
as shown in Figure 2.

Figure 2. Voltage-Divider Configuration

Where:

V

for the MIC2168 is typically 0.8V

REF

The output voltage is determined by the equation:

A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is
too large, it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small, in value, it will decrease the
efficiency of the power supply, especially at light loads. Once
R1 is selected, R2 can be calculated using:

External Schottky Diode

An external freewheeling diode is used to keep the inductor
current flow continuous while both MOSFETs are turned off.
This dead time prevents current from flowing unimpeded
through both MOSFETs and is typically 15ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must
be able to handle the peak current.

V

= forward voltage at the peak diode current

F

The external Schottky diode, D1, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve efficiency and
decrease high frequency noise. If the MOSFET body diode
is used, it must be rated to handle the peak and average cur­rent. The body diode has a relatively slow reverse recovery
time and a relatively high forward voltage drop. The power
lost in the diode is proportional to the forward voltage drop
of the diode. As the high-side MOSFET starts to turn on, the
body diode becomes a short circuit for the reverse recovery
period, dissipating additional power. The diode recovery and
the circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts at
a lower forward voltage preventing the body diode in the
MOSFET from turning on. The lower forward voltage drop
dissipates less power than the body diode. The lack of a
reverse recovery mechanism in a Schottky diode causes
less ringing and less power loss. Depending on the circuit
components and operating conditions, an external Schottky
diode will give a 1/2% to 1% improvement in efficiency.

Feedback Loop Compensation

The MIC2168 controller comes with an internal transcon­ductance error amplifier used for compensating the voltage
feedback loop by placing a capacitor (C1) in series with a
resistor (R1) and another capacitor C2 in parallel from the
COMP pin to ground. See “Functional Block Diagram.”

Power Stage

The power stage of a voltage mode controller has an induc­tor, L1, with its winding resistance (DCR) connected to the
output capacitor, C

, with its electrical series resistance

OUT

(ESR) as shown in Figure 3. The transfer function G(s), for
such a system is:

Figure 3. The Output LC Filter in a Voltage Mode

Buck Converter

The reverse voltage requirement of the diode is:

The power dissipated by the Schottky diode is:

where:

M9999-040805 10 April 2005

Plotting this transfer function with the following assumed values
(L=2 µH, DCR=0.009Ω, C

=1000µF, ESR=0.050Ω) gives

OUT

lot of insight as to why one needs to compensate the loop by
adding resistor and capacitors on the COMP pin. Figures 4
and 5 show the gain curve and phase curve for the above
transfer function.

MIC2168 Micrel, Inc.

100 1

.

1

0

3

1

.

1

0

4

1

.

1

0

5

1

.

1

0

6

60

37.5

15

7.5

30

30

60

GAIN

1000000100

f

100 1

.

10

3

1

.

10

4

1.10

5

1

.

10

6

150

100

50

0

0

180

P

1000000100 f

1

=

f

C

2 × π √ L × C

OUT

1

=

f

ZERO

2 × π × ESR × C

OUT

100 1

.

10

3

1

.

10

4

1

.

10

5

1

.

10

6

150

100

50

0

0

180

P

1000000100 f

1+ R1 × S × C1

×

g

m

s × (C1 + C2)

Error Amplifier (z)

-











1 + R1 ×

C1 × C2 × S

C1 + C2

1+ R1 × S × C1

×

g

m

s × (C1)(1+ R1 × C2 × S)

Error Amplifier (z)

-











It can be seen from the transfer function G(s) and the gain
curve that the output inductor and capacitor create a two pole
system with a break frequency at:

Therefore, fLC = 3.6kHz

By looking at the phase curve, it can be seen that the output
capacitor ESR (0.050Ω) cancels one of the two poles (LC
system by introducing a zero at:

Therefore, F

From the point of view of compensating the voltage loop, it is
recommended to use higher ESR output capacitors since they
provide a 90° phase gain in the power path. For comparison
purposes, Figure 6, shows the same phase curve with an
ESR value of 0.002Ω.

April 2005 11 M9999-040805

Figure 4. The Gain Curve for G(s)

Figure 5. Phase Curve for G(s)

= 6.36kHz.

ZERO

Figure 6. The Phase Curve with ESR = 0.002Ω

It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number is
–150°. This means that the transconductance error ampli­fier has to provide a phase boost of about 45° to achieve a
closed loop phase margin of 45° at a crossover frequency
of 50kHz for Figure 4, versus 105° for Figure 6. The simple
RC and C2 compensation scheme allows a maximum error
amplifier phase boost of about 90°. Therefore, it is easier to
stabilize the MIC2168 voltage control loop by using high ESR
value output capacitors.

gm Error Amplifier

It is undesirable to have high error amplifier gain at high
frequencies because high frequency noise spikes would be
picked up and transmitted at large amplitude to the output,
thus, gain should be permitted to fall off at high frequencies.
At low frequency, it is desired to have high open-loop gain to
attenuate the power line ripple. Thus, the error amplifier gain
should be allowed to increase rapidly at low frequencies.

The transfer function with R1, C1, and C2 for the internal
gm error amplifier can be approximated by the following
equation:

)

OUT

The above equation can be simplified by assuming
C2<<C1,

From the above transfer function, one can see that R1 and
C1 introduce a zero and R1 and C2 a pole at the following
frequencies:

Fzero= 1/2 π × R1 × C1

Fpole = 1/2 π × C2 × R1

Fpole@origin = 1/2 π × C1

Figures 7 and 8 show the gain and phase curves for the above
transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF,
and gm = .005Ω–1. It can be seen that at 50kHz, the error
amplifier exhibits approximately 45° of phase margin.

MIC2168 Micrel, Inc.

1.10

3

1

.

10

4

1

.

10

5

1

.

10

6

1

.

10

7

20

40

60

60

.0

01

E

100000001000 f

10 100 1

.

1

031.1041

.

10

5

1

.

1

0

6

260

240

220

200

215.856

270

E

100000010 f

100 1

.

10

3

1

.

10

4

1

.

10

5

1

.

10

6

50

0

50

100

71.607

42.933

OPEN LOOP GAIN MARGIN

1000000100 f

10 100 1

.

1

0

3

1

.

1

0

4

1.10

5

1

.

1

0

6

350

300

250

269.097

360

100000010 f

OPEN LOOP PHASE MARGIN

Figure 7. Error Amplifier Gain Curve

Figure 8. Error Amplifier Phase Curve

Total Open-Loop Response

The open-loop response for the MIC2168 controller is easily
obtained by adding the power path and the error amplifier
gains together, since they already are in Log scale. It is
desirable to have the gain curve intersect zero dB at tens of
kilohertz, this is commonly called crossover frequency; the
phase margin at crossover frequency should be at least 45°.
Phase margins of 30° or less cause the power supply to have
substantial ringing when subjected to transients, and have
little tolerance for component or environmental variations.
Figures 9 and 10 show the open-loop gain and phase margin.
It can be seen from Figure 9 that the gain curve intersects
the 0dB at approximately 50kHz, and from Figure 10 that at
50kHz, the phase shows approximately 50° of margin.

Figure 9. Open-Loop Gain Margin

Figure 10. Open-Loop Phase Margin

M9999-040805 12 April 2005

MIC2168 Micrel, Inc.

Design Example

Layout and Checklist:

1. Connect the current limiting (CS) resistor directly
to the drain of top MOSFET Q1.

2. Connect the VSW pin directly to the source of top
MOSFET Q1 thru a 4Ω to 10Ω resistor. The purpose
of this resistor is to filter the switch node.

3. The feedback resistors R1 and R2 should be placed
close to the FB pin. The top side of R1 should
connect directly to the output node. Run this trace
away from the switch node (junction of Q1, Q2, and
L1). The bottom side of R1 should connect to the
GND pin on the MIC2168.

4. The compensation resistor and capacitors should
be placed right next to the COMP/EN pin and the
other side should connect directly to the GND pin
on the MIC2168 rather than going to the plane.

5. The input bulk capacitors should be placed close
to the drain of the top MOSFET.

6. The 1µF ceramic capacitor should be placed right
on the VIN pin of the MIC2168.

7. The 4.7µF to 10µF ceramic capacitor should be
placed right on the VDD pin.

8. The source of the bottom MOSFET should connect
directly to the input capacitor GND with a thick
trace. The output capacitor and the input capacitor
should connect directly to the GND plane.

9. Place a 0.1µF ceramic capacitor in parallel with
the CS resistor to filter any switching noise.

April 2005 13 M9999-040805

MIC2168 Micrel, Inc.

Rev. 00

Package Information

10-Pin MSOP (MM)

MICREL INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA

TEL + 1 [phone] FAX + 1 [phone] WEB http://www.micrel.com

This information furnished by Micrel in this data sheet is believed to be accurate and reliable. However no responsibility is assumed by Micrel for its use.

Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can

reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into

the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's

use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify

M9999-040805 14 April 2005

Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.

Micrel for any damages resulting from such use or sale.

© 2003 Micrel Incorporated