Datasheet MC44602P Specification

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The MC44602 is an enhanced high performance fixed frequency current
mode controller that is specifically designed for off–line and high voltage
dc–to–dc converter applications. This device has the unique ability of
changing operating modes if the converter output is overloaded or shorted,
offering the designer additional protection for increased system reliability.
The MC44602 has several distinguishing features when compared to
conventional current mode controllers. These features consist of a foldback
amplifier for overload detection, valid load and demag comparators with a
fault latch for short circuit detection, thermal shutdown, and separate high
current source and sink outputs that are ideally suited for driving a high
voltage bipolar power transistor, such as the MJE18002, MJE18004, or
MJE18006.

Standard features include an oscillator with a sync input, a temperature
compensated reference, high gain error amplifier, and a current sensing
comparator. Protective features consist of input and reference undervoltage
lockouts each with hysteresis, cycle–by–cycle current limiting, a latch for
single pulse metering, and a flip–flop which blanks the output off every other
oscillator cycle, allowing output deadtimes to be programmed from 50% to
70%. This device is manufactured in a 16 pin dual–in–line heat tab package
for improved thermal conduction.

• Separate High Current Source and Sink Outputs Ideally Suited for

Driving Bipolar Power Transistors: 1.0 A Source, 1.5 A Sink

• Unique Overload and Short Circuit Protection

• Thermal Protection

• Oscillator with Sync Input

• Current Mode Operation to 500 kHz Output Switching Frequency

• Output Deadtime Adjustable from 50% to 70%

• Automatic Feed Forward Compensation

• Latching PWM for Cycle–By–Cycle Current Limiting

• Input and Reference Undervoltage Lockouts with Hysteresis

• Low Startup and Operating Current

Simplified Block Diagram

V

V

ref

16

Sync Input

RT/C

T

Compensation

Voltage Feedback–Input

7

8

1

3

V

ref

Undervoltage

Lockout

Oscillator

Error

Amplifier

Foldback
Amplifier

5.0V

Reference

Flip Flop

and

Latching

PWM

Gnd 9

CC

Undervoltage

Lockout

Short Circuit

Detection

Thermal

V

CC

15

Load Detect Input

2

V

C

14

Source Output

11

Sink Output

10

Sink Ground

4, 5, 12, 13

Current Sense Input

6

Order this document by MC44602/D



HIGH PERFORMANCE

CURRENT MODE

CONTROLLER

SEMICONDUCTOR

TECHNICAL DATA

16

1

P2 SUFFIX

PLASTIC PACKAGE

CASE 648C

DIP (12 + 2 + 2)

PIN CONNECTIONS

RT/C

T

1

2

3

4

5

6

7

8

(Top View)

Compensation

Load Detect Input

Voltage Feedback Input

Sink Gnd

Current Sense Input

Sync Input

ORDERING INFORMATION

Operating

Device

MC44602 TA = –25 to 85°C DIP (12 + 2 + 2)

Temperature Range

16

V

ref

15

V

CC

14

V

C

13

Sink Gnd

12

11

Source Output

10

Sink Output

9

Gnd

Package

MOTOROLA ANALOG IC DEVICE DATA

 Motorola, Inc. 1996 Rev 0

1

MC44602

MAXIMUM RATINGS

Rating Symbol Value Unit

Total Power Supply and Zener Current (ICC + IZ) 30 mA
Sink Ground Voltage

with Respect to Gnd (Pin 9)

Output Supply Voltage

with Respect to Sink Gnd (Pins 4, 5, 12, 13)

Output Current (Note 1)

Source

Sink
Output Energy (Capacitive Load per Cycle) W 5.0 µJ
Current Sense and Voltage Feedback Inputs V
Sync Input

High State Voltage

Low State Reverse Current
Load Detect Input Current I
Error Amplifier Output Sink Current IEA
Power Dissipation and Thermal Characteristics

Maximum Power Dissipation at TA = 25°C

Thermal Resistance, Junction–to–Air

Thermal Resistance, Junction–to–Case
Operating Junction Temperature T
Operating Ambient Temperature T

NOTE: 1.Maximum package power dissipation limits must be observed.

V

Sink(neg)

V

C

I

O(Source)

I

O(Sink)

in

V

IH

I

IL
in

(Sink)

P

D

R

θJA

R

θJC

J

A

–5.0 V

20 V

1.0

1.5

–0.3 to 5.5 V

5.5

–20

–20 to +10 mA

10 mA

2.5
80
15

150 °C

–25 to +85 °C

A

V

mA

W

°C/W
°C/W

ELECTRICAL CHARACTERISTICS (V

values TA = –25°C to +85°C [Note 3] unless otherwise noted.)

Characteristic

ERROR AMPLIFIER SECTION

Voltage Feedback Input (VO = 2.5V) V
Input Bias Current (VFB = 2.5 V) I
Open Loop Voltage Gain (VO = 2.0 V to 4.0 V) A
Unity Gain Bandwidth

TJ = 25°C

TA = –25 to +85°C
Power Supply Rejection Ratio (VCC = 10 V to 16 V) PSRR 65 70 – dB
Output Current

Sink (VO = 1.5 V, VFB = 2.7 V)

Sink TJ = 25°C

Sink TA = –25 to +85°C

Source (VO = 5.0 V, VFB = 2.3 V)

Source TJ = 25°C

Source TA = –25 to +85°C

Output Voltage Swing

High State (I

Low State (I

NOTES: 2. Adjust VCC above the startup threshold before setting to 12V.

O(Source)

O(Sink)

3.Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.

= 0.5 mA, VFB = 2.3 V)

= 0.33 mA, VFB = 2.7 V)

and VC = 12 V [Note 2], RT = 10k, CT = 1.0 nF, for typical values TA = 25°C, for min/max

CC

Symbol Min Typ Max Unit

FB
IB

VOL

BW

I

Sink

I

Source

V

OH

V

OL

2.45 2.5 2.65 V
– –0.6 –2.0 µA

65 90 – dB

1.0

0.8

–

1.5

–

–2.0

6.0
–

1.4
–

5.0
–

–1.1

–

7.0

1.0

1.8

2.0

–

10

–

–0.2

–

1.1

MHz

mA

V

2

MOTOROLA ANALOG IC DEVICE DATA

MC44602

ELECTRICAL CHARACTERISTICS (V

and VC = 12 V [Note 2], RT = 10k, CT = 1.0 nF, for typical values TA = 25°C, for min/max

CC

values TA = –25°C to +85°C [Note 3] unless otherwise noted.)

Characteristic

Symbol Min Typ Max Unit

OSCILLATOR SECTION

Frequency

TJ = 25°C

TA = –25°C to +85°C
Frequency Change with Voltage (VCC = 12 V to 18 V) ∆f
Frequency Change with Temperature ∆f
Oscillator Voltage Swing (Peak–to–Peak) V
Discharge Current (V

TJ = 25°C

OSC

= 3.0 V)

TA = –25°C to +85°C

f

OSC

/∆V – 0.1 0.2 %/V

OSC

/∆T – 0.05 – %/°C

OSC

OSC(pp)
I

dischg

168
160

180

–

192
200

1.3 1.6 – V

6.5

6.0

10

–

14

13.5

Sync Input Threshold Voltage

High State

Low State
Sync Input Resistance

TJ = 25°C

TA = –25°C to +85°C

V

IH

V

IL

R

in

2.5

1.0

6.5

6.0

2.8

1.3

10

3.2

1.7

13.5

–

18

REFERENCE SECTION

Reference Output Voltage (IO = 1.0 mA) V
Line Regulation (VCC = 12 V to 18 V) Reg
Load Regulation (IO = 1.0 mA to 20 mA) Reg
T emperature Stability T
Total Output Variation over Line, Load and Temperature V
Output Noise Voltage (f = 10 Hz to 10 kHz, TJ = 25°C) V

ref

line
load
S

ref

n

4.7 5.0 5.3 V
– 1.0 10 mV
– 3.0 15 mV
– 0.2 – mV/°C

4.65 – 5.35 V
– 50 – µV

Long Term Stability (TA = 125°C for 1000 Hours) S – 5.0 – mV
Output Short Circuit Current

TJ = 25°C
TA = –25°C to +85°C

I

SC

–

–70

–130

–

–

–180

CURRENT SENSE SECTION

Current Sense Input Voltage Gain (Notes 4 & 5)

TJ = 25°C

TA = –25°C to +85°C
Maximum Current Sense Input Threshold (Note 4) V
Input Bias Current I
Propagation Delay (Current Sense Input to Sink Output) t

A

V

th

IB

PLH(in/out)

2.85

2.7

3.0
–

3.2

3.15

0.9 1.0 1.1 V
– –4.0 –10 µA
– 100 150 ns

UNDERVOLTAGE LOCKOUT SECTIONS

Startup Threshold (VCC Increasing) V
Minimum Operating Voltage After Turn–On (VCC Decreasing) V
Reference Undervoltage Threshold (V

NOTES: 2.Adjust VCC above the startup threshold before setting to 12V.

3.Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.

4.This parameter is measured at the latch trip point with IFB = –5.0

5.Comparator gain is defined as AV =

Decreasing) V

ref

µA, refer to Figure 9.

∆V Compensation

∆V Current Sense Input

th

CC(min)

(UVLO) 3.0 3.35 3.7 V

ref

13 14.1 15 V

9.0 10.2 11 V

kHz

mA

V

kΩ

mA

V/V

MOTOROLA ANALOG IC DEVICE DATA

3

MC44602

Ä

ÄÄÄÄ

ÄÄÄÄ

ÄÄÄÄ

ELECTRICAL CHARACTERISTICS

(V

and VC = 12 V [Note 2], RT = 10k, CT = 1.0 nF, for typical values TA = 25°C, for min/max

CC

values TA = –25°C to +85°C [Note 3] unless otherwise noted.)

Characteristic

Symbol Min Typ Max Unit

OUTPUT SECTION

Output Voltage (TA = 25°C)

Low State (I

Low State (I
Low State (I

High State (I

High State (I

High State (I

Output Voltage with UVLO Activated (VCC = 6.0 V, I
Output Voltage Rise T ime (CL = 1.0 nF, TJ = 25°C) t
Output Voltage Fall T ime (CL = 1.0 nF, TJ = 25°C) t

Sink

= 1.0A)

Sink

= 1.5 A)

Sink

Source
Source
Source

= 100 mA)

= 50 mA)
= 0.5 A)
= 0.75 A)

= 1.0 mA) V

Sink

V

OL

(VCC–VOH)

OL(UVLO)

r

f

PWM SECTION

Duty Cycle

Maximum
Minimum

DC

DC

(max)

(min)

TOTAL DEVICE

Power Supply Current

Startup (VCC = 5 V)

I

CC

Operating (Note 2)

TJ = 25° C
TA = –25°C to +85° C

Power Supply Zener Voltage (ICC = 25 mA) V

Z

OVERLOAD AND SHORT CIRCUIT PROTECTION

Foldback Amplifier Threshold (Figures 9,10) ∆V
Load Detect Input

Valid Load Comparator Threshold (V
Demag Comparator Threshold (V
Propagation Delay (Input to Sink or Source Output)

Pin 2

Decreasing)

Pin 2

Increasing)

Input Resistance

NOTES: 2. Adjust VCC above the startup threshold before setting to 12V.

3.Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.

FB

V

th(VL)

V

th(Demag)

t

PLH(in/out)

R

in

(VFB–100) (VFB–200) (VFB–300) mV

–
–
–

–
–
–

0.6

1.8

2.1

1.4

1.7

1.8

0.3

2.0

2.6

1.7

2.0

2.2
– 0.1 1.1 V
– 50 150 ns
– 50 150 ns

46

–

48

–

0

50

–

–

10

0.2

17

0.5

20

–

22

18 20 23 V

2.0
50

–

12

2.5
88

1.1
18

3.0

120

1.6
30

V

%

mA

V

mV

µS
kΩ

80
50

30

Ω

20

CT=5.0 nF

10

8.0
CT=10 nF

5.0

3.0

, TIMING RESISTOR (k )

VCC = 12 V

T

R

2.0

TA = 25

°

C

Note: Output switches at
one–half the oscillator frequency.

1.0

0.8

4

Figure 1. Timing Resistor

versus Oscillator Frequency

CT=500 pF

CT=2.0 nF

f

, OSCILLAT OR FREQUENCY (Hz)

OSC

CT=100 pF

CT=200 pF

CT=1.0 nF

1.0 M500 k200 k100 k50 k20 k10 k

75

1. CT = 10 nF

2. CT = 5.0 nF

3. CT = 2.0 nF

70

4. CT = 1.0 nF

5. CT = 500 pF

6. CT = 100 pF

65

60

55

% DT, PERCENT OUTPUT DEADTIME

50

Figure 2. Output Deadtime

versus Oscillator Frequency

Note: Output switches at
one–half the oscillator
frequency .

5

4

3

2

1

VCC = 12 V
TA = 25

20 k 50 k 200 k 500 k

f

, OSCILLAT OR FREQUENCY (Hz)

OSC

MOTOROLA ANALOG IC DEVICE DATA

6

°

C

1.0 M100 k10 k

MC44602

12

11

10

9.0

, DISCHARGE CURRENT (mA)

8.0

dischg

I

7.0

2.55 V

–55

Figure 3. Oscillator Discharge Current

versus T emperature

VCC = 12 V
V

= 3.0 V

OSC

–25 0 25 50 75 100 125

°

TA, AMBIENT TEMPERATURE (

C)

Figure 5. Error Amp Small Signal

Transient Response

VCC = 12 V
AV = –1.0

°

C

TA = 25

5.0

4.0

3.0

2.0

, OSCILLAT OR VOLTAGE SWING (V)V

1.0

OSC

3.0 V

0

–55

Figure 4. Oscillator V oltage Swing

versus T emperature

VCC = 12 V
RT = 10 k
CT = 1.0 nF

Peak Voltage

Valley Voltage

–25 0 25 50 75 100 125

TA, AMBIENT TEMPERATURE (

°

C)

Figure 6. Error Amp Large Signal

Transient Response

VCC = 12 V
AV = –1.0

°

C

TA = 25

2.5 V

2.45 V

Figure 7. Error Amp Open Loop Gain and

100

80

60

40

20

, OPEN LOOP VOL TAGE GAIN (dB)

0

VOL

A

–20

µ

t, TIME (0.5

s/DIV) t, TIME (1.0 µs/DIV)

Phase versus Frequency

VCC = 12 V
VO = 2.0 V to 4.0 V
RL = 100 k

°

C

Gain

1.0 k 10 k 100 k 1.0 M
f, FREQUENCY (Hz)

TA = 25

Phase

2.5 V

20 mV/DIV

2.0 V

200 mV/DIV

Figure 8. Current Sense Input Threshold versus

Error Amp Output Voltage

0

30

60

90

120

150

180

10 M0.1 k

1.2

1.0
TA = 125°C

0.8

0.6

TA = 25°C

0.4

EXCESS PHASSE (DEGREES)

0.2

, CURRENT SENSE INPUT THRESHOLD (V)V

th

0

1.0 3.0 5.0 7.0

0

2.0 4.0 6.0

VO, ERROR AMP OUTPUT VOLTAGE (V)

TA = –40°C

VCC = 12 V

MOTOROLA ANALOG IC DEVICE DATA

5

MC44602

2.6

2.2

1.8

, INPUT VOLTAGE (V)

in

V

1.4

1.0
–500

200

160

120

Figure 9. V oltage Feedback Input,

V oltage versus Current

V

= 1.0 V

Clamp

VCC = 12 V

°

C

TA = 25

V

= 0.3 V

Clamp

V

Clamp

–400 –300 –200 –100 0

Iin, INPUT CURRENT (

V

= 0.1 V

Clamp

= 0.7 V

µ

A)

V

Clamp

= 0.5 V

Figure 11. Reference Short Circuit Current

versus T emperature

VCC = 12 V

≤

0.1

RL

Figure 10. V oltage Feedback Input

versus Current Sense Clamp Level

2.6
VCC = 12 V

2.2

TA = 125°C

1.8

, INPUT VOLTAGE (V)

in

1.4

V

1.0

TA = 25°C

TA = –55°C

0

0.2 0.4 0.6 0.8 1.0
V

, CURRENT SENSE CLAMP LEVEL (V)

Clamp

Figure 12. Reference Line and Load

Regulation versus T emperature

3.0

Ω

2.0

1.0

–1.0

Line Regulation

0

VCC = 12 V to 18 V
I

= 0 mA

ref

80

, REFERENCE SHORT CIRCUIT CURRENT (mA)

SC

40

I

–55

–25 0 25 50 75 100 125

Figure 13. Reference V oltage Change

0

–5.0

–10

–15

–20

, REFERENCE VOLTAGE CHANGE (mV)

–25

ref

V

∆

–30

0

VCC = 12 V

30 60 90 120 150 180

I

ref

°

TA, AMBIENT TEMPERATURE (

C)

versus Source Current

TA = –55°C

TA = 25°C

TA = 125°C

, REFERENCE SOURCE CURRENT (mA)

–2.0

Load Regulation

–3.0

VCC = 12 V
I

, REFERENCE VOLTAGE CHANGE (mA)

–4.0

ref

V

∆

–5.0

= 1.0 mA to 20 mA

ref

–55

–25 0 25 50 75 100 125

TA, AMBIENT TEMPERATURE (

Figure 14. Thermal Resistance and Maximum

Power Dissipation versus P.C.B. Copper Length

°

100

80

60

40

20

, THERMAL RESISTANCE JUNCTION T O AIR ( C/W)

JA

0

θ

0

R

R

θ

JA

P

for TA = 70°C

D(max)

10 20 30 40 50

L, LENGTH OF COPPER (mm)

Printed circuit board heatsink example

2.0 oz

L

Copper

L

Graphs represent symmetrical layout

°

C)

5.0

4.0

3.0 mm

3.0

2.0

1.0

0

, MAXIMUM POWER DISSIPATION (W)

D

P

6

MOTOROLA ANALOG IC DEVICE DATA

MC44602

Figure 15. Output Waveform Figure 16. Output Cross Conduction

90%

10%

3.0
Sink Saturation

(Load to VCC)

2.5

2.0

1.5

Voltage

Current

t, TIME (100 ns/DIV)

VCC = 12 V
CL = 2.0 nF

°

C

TA = 25

Figure 17. Sink Output Saturation Voltage

versus Sink Current

TJ = –55°C

TJ = 25°C

1.0 A

0

–1.0 A

O

V

, SUPPLY CURRENT

CC

I , OUTPUT VOLTAGE

–0.5

–1.0

–1.5

t, TIME (50 ns/DIV)

Figure 18. Source Output Saturation Voltage

versus Load Current

0

V

CC

VCC = 12 V
80
120 Hz Rate

TJ = 125°C

VCC = 12 V
CL = 15 pF

°

C

TA = 25

µ

s Pulsed Load

–90%

–10%

20 mA/DIV

1.0

0.5

, SINK OUTPUT SA TURATION VOLTAGE (V)

sat

V

0

0

Gnd

250 500 750 1000 1250 1500 1750

I

sink

TJ = 125°C

VCC = 12 V
80
120 Hz Rate

, SINK OUTPUT CURRENT (mA)

Figure 19. Supply Current versus Supply V oltage

32

RT = 10 k
CT = 1.0 nF
VFB = 0 V
Current Sense = 0 V

24

16

, SUPPLY CURRENT (mA)

8.0

CC

I

0

TA = 25

0

°

C

4.0 8.0 12 16 20 24
VCC, SUPPLY VOLTAGE (V)

µ

s Pulsed Load

–2.0

–2.5

, SINK OUTPUT SA TURATION VOLTAGE (V)

sat

V

–3.0

0

23

22

21

, ZENER VOLTAGE (V)

CC

20

V

19

–55

TJ = –55°C

Source Saturation
(Load to Ground)

150 300 450 600 750 900

I

, OUTPUT SOURCE CURRENT (mA)

source

TJ = 25°C

Figure 20. Power Supply Zener V oltage

versus T emperature

ICC = 25 mA

–25 0 25 50 75 125

TA, AMBIENT TEMPERATURE (°C)

100

MOTOROLA ANALOG IC DEVICE DATA

7

MC44602

3.2

2.8

2.4

, VALID LOAD COMPARATOR THRESHOLD (V)

2.0

th(VL)

V

µ

1.4

1.2

Figure 21. Valid Load Comparator Threshold

versus T emperature

–55

–25 0 25 50 75 125

°

TA, AMBIENT TEMPERATURE (

C)

Figure 23. Load Detect Input

Propagation Delay versus T emperature

VCC = 12 V
RT = 10 k
CT = 1.0 nF

VCC = 12 V

100

120

100

80

, DEMAG COMPARATOR THRESHOLD (mV)

60

th(Demag)

–55

V

14.5

14.3

Figure 22. Demag Comparator Threshold

versus T emperature

VCC = 12 V

–25 0 25 50 75 125

TA, AMBIENT TEMPERATURE (

°

100

C)

Figure 24. Startup Threshold V oltage

versus T emperature

VCC Increasing

1.0

, LOAD DETECT PROPAGATION DELAY ( s)

0.8
–55

PLH(IN/OUT)

t

10.35

10.25

10.15

10.05

, MINIMUM OPERATING VOLTAGE (V)

CC(min)

V

9.95
–55

–25 0 25 50 75 125

TA, AMBIENT TEMPERATURE (

°

100

C)

Figure 25. Minimum Operating V oltage

After Turn–On versus Temperature

VCC Decreasing

–25 0 25 50 75 125100

TA, AMBIENT TEMPERATURE (

°

C)

14.1

13.9

, STARTUP THRESHOLD VOLTAGE (V)

th

V

13.7
–55

3.42

3.38

3.34

, REFERENCE UNDERVOL TAGE THRESHOLD (V)

3.30
–55

ref(UVLO)

V

–25 0 25 50 75 125

TA, AMBIENT TEMPERATURE (

°

100

C)

Figure 26. Reference Undervoltage Threshold

versus T emperature

V

Decreasing

ref

–25 0 25 50 75 125100

TA, AMBIENT TEMPERATURE (

°

C)

8

MOTOROLA ANALOG IC DEVICE DATA

V

ref

16

Sync Input

R

T

C

T

Compensation

Voltage Feedback

Input

MC44602

Figure 27. Representative Block Diagram

V

CC

V

CC

+

R
R

2.5V

7

10k

Oscillator

8

1

2.5V

3

Amplifier

2.5V

Internal

Bias

Error

I

3.6V

Fault Latch

+

1.0 mA

Foldback

Amplifier

Demag

Comparator

Valid Load

Comparator

R

Q

S

2R

1.0V

Reference

Regulator

Reference

UVLO

Thermal RSQ

R

Current Sense

Comparator

V
UVLO

85mV

2.5V

TQ

R

PWM
Latch

CC

18k

20V

14V

+

Substrate

15

Load
Detect
Input

2

V

C

14
Source Output
11

Sink Output
10
Sink Ground

4, 5, 12, 13
Current Sense Input

6

V

in

V

out

Q1

R

S

Capacitor C

T

PWM Latch
“Set” Input

Toggle Flip Flop

Output

Q

Current Sense
Input

PWM Latch
“Set” Input

Source Output

Load Detect

85mV

Input

2.8V

1.2V
0V

2.5V

0V

C

R

V

Clamp

Gnd 9

1

R

2

Sink Only

=

Positive True Logic

C

O

Figure 28. Timing Diagram

C

*

C

NC

NC

NC

NC

C

C

NC

Demag Output

Fault Latch Q

Sync Input

2.5V
0V

Startup With Foldback

Startup Without Foldback

*C = Comparison of Current Sense Input With V

MOTOROLA ANALOG IC DEVICE DATA

Clamp

Normal Operation Output Overload

NC = No Comparison of Current Sense Input With V

Clamp

9

MC44602

OPERA TING DESCRIPTION

The MC44602 is a high performance, fixed frequency,
current mode controller specifically designed to directly drive
a bipolar power switch in off–line and high voltage dc–to–dc
converter applications. This device offers the designer a cost
effective solution with minimal external components. The
representative block and timing diagrams are shown in
Figures 27 and 28.

Oscillator

The oscillator frequency is programmed by the values
selected for the timing components RT and CT. Capacitor C
is charged from the 5.0 V reference through resistor RT to
approximately 2.8 V and discharged to 1.2 V by an internal
current sink. During the discharge of CT, the oscillator
generates an internal blanking pulse that holds one of the
inputs of the NOR gate high. This causes the Source and
Sink outputs to be in a low state, thus producing a controlled
amount of output deadtime. An internal toggle flip–flop has
been incorporated in the MC44602 which blanks the output
off every other clock cycle by holding one of the inputs of the
NOR gate high. This in combination with the CT discharge
period yields output deadtimes programmable from 50% to
70%. Figure 1 shows RT versus Oscillator Frequency and
Figure 2, Output Deadtime versus Frequency, both for a
given value of CT. Note that many values of RT and CT will
give the same oscillator frequency but only one combination
will yield a specific output deadtime at a given frequency .

In many noise sensitive applications it may be desirable to
frequency–lock the converter to an external system clock.
This can be accomplished by applying a narrow rectangular
clock signal with an amplitude of 3.2 V to 5.5 V to the Sync
Input (Pin 7). For reliable locking, the free–running oscillator
frequency should be set about 10% less than the clock
frequency. If the clock signal is ac coupled through a
capacitor, an external clamp diode may be required if the
negative sync input current is greater than –5.0 mA.
Connecting Pin 7 to V

will cause CT to discharge to 0 V,

ref

inhibiting the Oscillator and conduction of the Source Output.
Multi–unit synchronization can be accomplished by
connecting the CT pin of each IC to a single MC1455 timer.

Error Amplifier

A fully compensated Error Amplifier with access to the
inverting input and output is provided. It features a typical dc
voltage gain of 90 dB, and a unity gain bandwith of

1.0 MHz with 57 degrees of phase margin (Figure 7). The
noninverting input is internally biased at 2.5 V and is not
pinned out. The converter output voltage is typically divided
down and monitored by the inverting input. The maximum
input bias current with the inverting input at 2.5 V is –2.0 µA.
This can cause an output voltage error that is equal to the
product of the input bias current and the equivalent input
divider source resistance.

The Error Amp Output (Pin 1) is provided for external loop
compensation (Figure 29). The output voltage is offset by two
diodes drops (≈1.4 V) and divided by three before it connects
to the inverting input of the Current Sense Comparator. This

guarantees that no drive pulses appear at the Source Output
(Pin 11) when Pin 1 is at its lowest state (VOL). This occurs
when the power supply is operating and the load is removed,
or at the beginning of a soft–start interval. The Error Amp
minimum feedback resistance is limited by the amplifier’s
minimum source current (0.5 mA) and the required output
voltage (VOH) to reach the comparator’s 1.0 V clamp level:

R

f(min)

3.0 (1.0 V))1.4 V

[

0.5mA

+

T

Figure 29. Error Amplifier Compensation

+

Compensation

R

FB

R

C

f

Voltage

Feedback

R

1

f

Input

R

1

2.5V

3

2

Amplifier

2.5V

From Power Supply Output

Error

I

1.0 mA

2R

Foldback
Amplifier

1.0V

Gnd

Current Sense Comparator and PWM Latch

The MC44602 operates as a current mode controller,
where output switch conduction is initiated by the oscillator
and terminated when the peak inductor current reaches the
threshold level established by the Error Amplifier output (Pin

1). Thus the error signal controls the peak inductor current on
a cycle–by–cycle basis. The Current Sense Comparator
PWM Latch configuration used ensures that only a single
pulse appears at the Source Output during the appropriate
oscillator cycle. The inductor current is converted to a voltage
by inserting the ground referenced sense resistor RS in series
with the emitter of output switch Q1. This voltage is monitored
by the Current Sense Input (Pin 6) and compared to a level
derived from the Error Amp output. The peak inductor current
under normal operating conditions is controlled by the
voltage at Pin 1 where:

lpk[

V

(Pin1)

3R

*

S

1.4V

Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:

l

pk(max)

[

1.0 V
R

S

8800

R

Current Sense

Comparator

9

W

10

MOTOROLA ANALOG IC DEVICE DATA

MC44602

A narrow spike on the leading edge of the current
waveform can usually be observed and may cause the power
supply to exhibit an instability when the output is lightly
loaded. This spike is due to the power transformer
interwinding capacitance and the output rectifier recovery
time. The addition of an RC filter on the Current Sense Input
with a time constant that approximates the spike duration will
usually eliminate the instability; refer to Figure 30.

Undervoltage Lockout

Two undervoltage lockout comparators have been
incorporated to guarantee that the IC is fully functional before
the output stage is enabled. The positive power supply
terminal (VCC) and the reference output (V

) are each

ref

monitored by separate comparators. Each has built–in
hysteresis to prevent erratic output behavior as their
respective thresholds are crossed. The VCC comparator
upper and lower thresholds are 14.1 V/10.2 V. The V

ref

comparator upper and lower thresholds are 3.6 V/3.3 V. The
large hysteresis and low startup current of the MC44602
make it ideally suited for off–line converter applications
(Figures 33, 34) where efficient bootstrap startup techniques
are required.

A 20 V zener is connected as a shunt regulator from VCC to
ground. Its purpose is to protect the IC from excessive
voltage that can occur during system startup. The upper limit
for the minimum operating voltage of the MC44602 is 1 1V.

Outputs

The MC44602 contains a high current split totem pole
output that was specifically designed for direct drive of
Bipolar Power Transistors. By splitting the totem pole into
separate source and sink outputs, the power supply designer
has the ability to independently adjust the turn–on and
turn–off base drive to the external power transistor for optimal
switching. The Source and Sink outputs are capable of up to

1.0 A and 1.5 A respectively and feature 50 ns switching
times with a 1.0 nF load. Additional internal circuitry has been
added to keep the Source Output “Off” and the Sink Output
“On” whenever an undervoltage lockout is active. This
feature eliminates the need for an external pull–down resistor
and guarantees that the power transistor will be held in the
“Off” state.

Separate output stage power and ground pins are
provided to give the designer added flexibility in tailoring the
base drive circuitry for a specific application. The Source
Output high–state is controlled by applying a positive voltage
to VC (Pin 14) and is independent of VCC. A zener clamp is
typically connected to this input when driving power
MOSFETs in systems where VCC is greater than 20V. The
Sink Output low–state is controlled by applying a negative
voltage to the Sink Ground (Pins 4, 5, 12, 13). The Sink
Ground can be biased as much as 5.0 V negative with
respect to Ground (Pin 7). Proper implementation of the V
and Sink Ground pins will significantly reduce the level of
switching transient noise imposed on the control circuitry.

This becomes particularly useful when reducing the I
clamp level.

Reference

The 5.0 V bandgap reference has a tolerance of ±6.0%
over a junction temperature range of –25°C to 85°C. Its
primary purpose is to supply charging current to the oscillator
timing capacitor. The reference has short circuit protection
and is capable of providing in excess of 20 mA for powering
additional control system circuitry.

Figure 30. Bipolar Transistor Drive

and Current Spike Suppression

I

B

+

0

Base Charge

Removal

RB1R

C

B

B2

L

B

TQ

RSQ
R

PWM
Latch

Current Sense

Comparator

Substrate

–

V

C

14

Source

11

Sink

10

Sink Gnd

4, 5, 12, 13

Current Sense

6

Thermal Protection and Package

Internal Thermal Shutdown circuitry is provided to protect

the integrated circuit in the event that the maximum junction
temperature is exceeded. When activated, typically at 160°C,
the PWM Latch is held in the “reset” state, forcing the Source
Output “Off” and the Sink Output “On”. This feature is
provided to prevent catastrophic failures from accidental
device overheating. It is not intended to be used as a
substitute for proper heatsinking.

The MC44602 is contained in a heatsinkable 16–lead

plastic dual–in–line package in which the die is mounted on a
special heat tab copper alloy lead frame. This tab consists of
the four center Sink Ground pins that are specifically
designed to improve the thermal conduction from the die to
the circuit board. Figure 14 shows a simple and effective
method of utilizing the printed circuit medium as a heat
dissipater by soldering these pins to an adequate area of
copper foil. This permits the use of standard layout and
mounting practices while having the ability to halve the

C

junction to air thermal resistance. This example is for a
symmetrical layout on a single–sided board with two ounce
per square foot of copper.

pk(max)

V

in

Q1

R

C

R

S

MOTOROLA ANALOG IC DEVICE DATA

11

MC44602

Design Considerations

Do not attempt to construct the converter on

wire–wrap or plug–in prototype boards. High frequency

circuit layout techniques are imperative to prevent
pulse–width jitter. This is usually caused by excessive noise
pick–up imposed on the Current Sense or Voltage Feedback
inputs. Noise immunity can be improved by lowering circuit
impedances at these points. The printed circuit layout should
contain a ground plane with low–current signal, and high
current switch and output grounds returning on separate

PROTECTION MODES

The MC44602 operates as a conventional fixed frequency
current mode controller when the power supply output load is
less than the design limit. For enhanced system reliability , this
device has the unique ability of changing operating modes if
the power supply output is overloaded or shorted.

Overload Protection

Power supply overload protection is provided by the
Foldback Amplifier. As the output load gradually increases,
the Error Amplifier senses that the voltage at Pin 3 is less than
the 2.5 V threshold. This causes the voltage at Pin 1 to rise,
increasing the Current Sense Comparator threshold in order
to maintain output regulation. As the load further increases,
the inverting input of the Current Sense Comparator reaches
the internal 1.0 V clamp level, limiting the switch current to the
calculated I

pk(max)

will cause the power supply output to fall out of regulation. As
the voltage at Pin 3 falls below 2.5 V, current will flow out of
the Foldback Amplifier input, and the internal clamp level will
be proportionally reduced (Figures 9, 10). The increase in
current flowing out of the Foldback Amplifier input in
conjunction with the reduced clamp level, causes the power
supply output voltage to fall at a faster rate than the voltage at
Pin 3. This results in the output foldback characteristic shown
in Figure 31. The shape of the current limit “knee” can be
modified by the value of resistor R1 in the feedback divider.
Lower values of R1 will reduce the I
faster rate.

Improper operation of the Foldback Amp can be
encountered when the Error Amp compensation capacitor C
exceeds 2.0 nF. The problem appears at Startup when the
output voltage of the power supply is below nominal, causing
the Error Amp output to rise quickly. The rapid change in
output voltage will be coupled through Cf to the Inverting Input
(Pin 3), keeping it at its 2.5 V threshold as the 1.0 mA Error
Amp current source charges Cf. This has the effect of
disabling the Foldback Amp by preventing Pin 3 and the
clamp level at the inverting input of the Current Sense
Comparator, from rising in proportion to the power supply
output voltage. By adding resistor RFB in series with Cf, the
voltage at Pin 3 can be held to 1.0 V, corresponding to a
Current Sense clamp level of 0.08 V (Figure 10), while
allowing the Error Amp output to reach its high state VOH of

7.0 V. The required resistor to keep Pin 3 below 1.0 V during
initial Startup is:

. At this point any further increase in load

clamp level at a

2

2

RFB R

RFB + R

pk(max)

f

≥ 6

f

R1 R

R1 + R

paths back to the input filter capacitor. Ceramic bypass
capacitors (0.1 µF) connected directly to VCC, VC, and
V

may be required depending upon circuit layout. This

ref

provides a low impedance path for filtering the high frequency
noise. All high current loops should be kept as short as
possible using heavy copper runs to minimize radiated EMI.
The Error Amp compensation circuitry and the converter
output voltage divider should be located close to the IC and
as far as possible from the power switch and other noise
generating components.

Figure 31. Output Foldback Characteristic

V

VO Nominal

VCC UVLO

Threshold

out

Low Value R

New Startup

Sequence Initiated

Nominal Load

Range

l

pk(max)

1

Short Circuit Protection

Short circuit protection for the power supply is provided by
the Valid Load Comparator, Fault Latch, and Demag
Comparator. Figure 32 shows the logic truth table of the
functional blocks. When operating the power supply with
nominal output loading, the Fault Latch is “Set” by the NOR
gate driver during the Power Transistor “On” time and “Reset”
by the Fault Comparator during the “Off” time. When a severe
overload or short circuit occurs on any output, the voltage
during the “Off” time (flyback voltage) at the Load Detect
Input, is unable to reach the 2.5 V threshold of the V alid Load
Comparator. This causes the Fault Latch to remain in the

f

“Set” state with output Q

“Low”. During the “Off” time the
Demag Comparator output will also be “Low”. This causes
the NOR gate to internally hold the Sync Input “High”,
inhibiting the next fixed frequency Oscillator cycle and
switching of the Power Transistor. As the load dissipates the
stored transformer energy, the voltage at the Load Detect
Input will fall. When this voltage reaches 85 mV, the Demag
Comparator output goes “High”, allowing the Sync Input to go
“Low”, and the Power Transistor to turn “On”.

Note that as long as there is an output short, the switching
frequency will shift to a much lower frequency than that set by
RT/CT. The frequency shift has the effect of lowering the duty
cycle, resulting in a significant reduction in Power Transistor
and Output Rectifier heating when compared to conventional
current mode controllers. The extended “On” time is the result
of CT charging from 0 V to 2.8 V instead of 1.2 V to 2.8 V . The
extended “Off” time is the result of the output short time
constant. The time constant consists of the output filter
capacitance, and the equivalent series resistance (ESR) of
the capacitor plus the associated wire resistance.

High Value R

Overload

I

out

1

12

MOTOROLA ANALOG IC DEVICE DATA

MC44602

Output

Power

Figure 32. Logic Truth Table of Functional Blocks

Output Power

Load

Nominal On <85mV 1 1 0 0 0 NOR gate driver sets Fault Latch.

Transistor

Demag Fault Latch Sync

Input Out S R

Q

Input

Operating Comments

At Turn–Off >85 mV, <2.5 V 0 0 0 0

Off >2.5 V 0 0 1 1 0 Valid Load Comparator resets Fault Latch.

Short On <85 mV 1 1 0 0 0 Short is not detected until transistor turn–off.

At Turn–Off >85 mV, <2.5 V 0 0 0 0 1

Off <85 mV 1 0 0 0 0 Load dissipates transformer energy, Oscillator enabled.

During the initial power supply startup the controller
sequences through the Short Circuit and Overload Protection
modes as the output filter capacitors charge–up. If an output
is shorted and the auxiliary feedback winding is used to
power the control IC as in Figure 33, the VCC UVLO lower
threshold level will be reached after several cycles, disabling
the IC and initiating a new startup sequence. The Short
Circuit Protection mode can be disabled by grounding the
Sync Input. Narrow switching spikes are present on this pin
during normal operation. These spikes are caused by the rise
time of the flyback voltage from the 85 mV Demag
Comparator threshold to the 2.5 V Valid Load Comparator
threshold. In high power applications, the increased negative
current at the Load Detect Input can extend the switching
spikes to the point where they exceed the Sync Input
threshold. This problem can be eliminated by placing an
external small signal clamp diode at the Load Detect Input.
The diode is connected with the cathode at Pin 2 and the
anode at ground.

The divide–by–two toggle flip–flop will appear not to
function properly during power supply startup without
foldback, or operation with an overloaded output. This
phenomena appears at the end of the oscillator cycle if there
was not a current sense comparison, and after the flyback
voltage at the Load Detect Input failed to exceed 2.5 V . Under
these conditions, the Sync input will go high approximately

1.0 µs after the Load Detect Input exceeds the 85 mV Demag

Narrow spike at Sync Input (<2.5 V) as transformer voltage
rises quickly, Oscillator is not affected.

Valid Load Comparator fails to reset Fault Latch, Pulse at
Sync Input exceeds 2.5 V , Oscillator is disabled.

Comparator threshold. This causes CT to discharge down
towards ground, generating a second negative going edge
on the oscillator waveform. This second edge results in the
divide–by–two flip–flop being clocked twice for each “On”
time of the switch transistor. During initial startup, this effect
can be eliminated by insuring that the Foldback Amplifier is
fully active with the addition of resistor RFB. With the Foldback
Amplifier active, the clamp level at the inverting input of the
Current Sense Comparator will be low, allowing a comparison
to take place during the switch transistor “On” time. When the
Load Detect Input exceeds 85 mV, the Sync Input will go
high, discharging CT to ground after 1.0 µs, thus eliminating
the second negative edge. Operation with the output
overloaded will cause the toggle flip–flop to be clocked twice
for each “On” time. This should not be a problem since the
next “On” time is delayed by the Demag Comparator until the
load dissipates the transformers energy.

The point where the IC detects that there is a severe
output overload, or that the transformer has reached zero
current, is controlled by the voltage of the auxiliary winding
and a resistor divider. The divider consists of an external
series resistor and an internal shunt resistor. The shunt
resistor is nominally 18 kΩ but can range from 12 kΩ to 30 kΩ
due to process variations. If more precise overload and zero
current detection is required, the internal resistor variations
can be swamped out by connecting a low value external
resistor (≤2.7 kΩ) from Pin 2 to ground.

MOTOROLA ANALOG IC DEVICE DATA

13

MC44602

PIN FUNCTION DESCRIPTION

Pin Function Description

1 Compensation This pin is the Error Amplifier output and is made available for loop compensation.
2 Load Detect Input A voltage indicating a severe overload or short circuit condition at any output of the

3 Voltage Feedback Input This is the inverting input of the Error Amplifier and the noninverting input of the

4, 5, 12, 13 Sink Ground The Sink Ground pins form a single power return that is typically connected back to the

6 Current Sense Input A voltage proportional to inductor current is connected to this input. The PWM uses this

7 Sync Input A narrow rectangular waveform applied to this input will synchronize the Oscillator. A dc

8 RT/C

9 Ground This pin is the control circuitry ground and is typically connected back to the power

10 Sink Output Peak currents up to 1.5 A are sunk by this output suiting it ideally for turning–off a bipolar

11 Source Output Peak currents up to 1.0 A are sourced by this output suiting it ideally for turning–on a

14 V

15 V

16 V

T

C

CC

ref

switching power supply is connected to this input. The Oscillator is controlled by this
information making the power supply short circuit proof.

Foldback Amplifier. It is normally connected to the switching power supply output
through a resistor divider.

power source on a separate path from Pin 9 Ground, to reduce the effects of switching
transient noise on the control circuitry . These pins can be used to enhance the package
power capabilities (Figure 14). The Sink Output low state (VOL) can be modified by
applying a negative voltage to these pins with respect to Ground (Pin 9) to optimize
turn–off of a bipolar junction transistor .

information to terminate conduction of the output switch transistor.

voltage within the range of 3.2 V to 5.5 V will inhibit the Oscillator.
The Oscillator frequency and maximum Output duty cycle are programmed at this pin by

connecting resistor RT to V

source on a separate path from the Sink Ground (Pins 4, 5, 12, 13).

junction transistor. The output switches at one–half the oscillator frequency.

bipolar junction transistor. The output switches at one–half the oscillator frequency.
The Output high state (VOH) is set by the voltage applied to this pin. With a separate

connection to the power source, it can reduce the effects of switching transient noise on
the control circuitry.

This pin is the positive supply of the control IC. The minimum operating voltage range
after startup is 11 V to 18 V.

This is the 5.0 V reference output. It provides charging current for capacitor CT through
resistor RT and can be used to bias any additional system circuitry.

and capacitor CT to ground.

ref

14

MOTOROLA ANALOG IC DEVICE DATA

MC44602

Figure 33. 60 Watt Off–Line Flyback Regulator

85 to 265

2.2

Vac

1N4148

0.1 470pF

0.1µF

24k

47k

10k

470k

1.0k

10k

2.2nF

1.0nF

1N5404

15k

1N4148

1
2
3
4
5

6
7
8

470

270

15
14

12

MC44602

11
10

16

13

220pF

390

T1

47k

2.0W

MUR
4100

220

0.1

85V/0.5A

220pF

1N4934

µ

H

1.0

MUR

415

470

0.1 20V/0.6A

220pF

8.2k

2.0W

220

3.3nF

1.0

22

0.33

µ

H

9

47

MUR

460

47nF

MBR

340

470pF

MJE18006

470

0.1

6.8V/0.8A

1.0nF/1.0kV

1.0k

0.82

4.7M

Test Conditions Results

Line Regulation

85V
20V

6.8V

Load Regulation

85V
20V

6.8V

Vin = 85 Vac to 265 Vac
IO = 0.5 A
IO = 0.5 A
IO = 0.8 A

Vin = 220 Vac
IO = 0.1 A to 0.5 A
IO = 0.1 A to 0.5 A
IO = 0.1 A to 0.8 A

∆ = 1.0 V or ± 0.6%
∆ = 0.04 V or ± 0.1%
∆ = 0.07 V or ± 0.5%

∆ = 1.0 V or ± 0.6%
∆ = 0.4 V or ± 1.0%
∆ = 0.2 V or ± 1.5%

Efficiency Vin = 110 Vac, PO = 58 W 81%
Standby Power Vin = 110 Vac, PO = 0 W 2.0 W

T1

–

Orega SMT2 (G4787–01)
Primary: 41 Turns, #25AWG
Auxiliary Feedback: 12 Turns, #25AWG
Secondary: 85 V – 60 Turns, #25A WG

Secondary: 20 V – 15 Turns, #25AWG (2 Strands) Bifiliar Wound
Secondary: 6.8 V – 5 Turns, #25AWG (2 Strands) Bifiliar Wound

Core

–

ETD39 34x17x11 B52

Gap

–

≈ 0.020″ for a primary inductance of 750 µH, AL = 500 nH/Turn

2

MOTOROLA ANALOG IC DEVICE DATA

15

MC44602

Figure 34. 150 Watt Off–Line Flyback Regulator

4.7

220 Vac

1N4148

0.1 470pF

0.1µF

24k

47k

10k

470k

1.0k

10k

2.2nF

1.0nF

1N5404

15k

1N4148

1
2
3
4
5
6
7
8

100

270

16
15
14
13
12

MC44602

11
10

9

47k

2.0W

1N4934

220

3.3nF

22

2.2

1.0k

220pF

390

T1

MUR
4100

220

155V/0.5A

0.1

220pF

µ

H

1.0

MUR

415

470

0.1 24.5V/1.8A

220pF

8.2k

2.0W

47nF

MUR

460

MUR

415

470pF

470

15.5V/1.8A

0.1

1.0

µ

H

47

MJE18006

1.0nF/1.0kV

0.47

4.7M

Test Conditions Results

Line Regulation

155V

24.5V

15.5V

Load Regulation

155V

24.5V

15.5V

Vin = 185 Vac to 265 Vac
IO = 0.5 A
IO = 1.0. A
IO = 1.0 A

Vin = 220 Vac
IO = 0.1 A to 0.5 A
IO = 0.1 A to 1.0 A
IO = 0.1 A to 1.0 A

∆ = 1.0 V or ± 0.3%
∆ = 0.4 V or ± 0.8%
∆ = 0.3 V or ± 1.0%

∆ = 2.0 V or ± 0.7%
∆ = 0.4 V or ± 0.8%
∆ = 0.2 V or ± 0.7%

Efficiency Vin = 220 Vac, PO = 117.5 W 83%
Standby Power Vin = 220 Vac, PO = 0 W 5.0 W

T1

–

Orega SMT2 (G4717–01)
Primary: 55 Turns, #25AWG
Auxiliary Feedback: 6 Turns, #25AWG
Secondary: 155 V – 52 Turns, #25A WG

Secondary: 24.5 V – 9 Turns, #25AWG (2 Strands) Bifiliar Wound
Secondary: 15.5 V – 6 Turns, #25AWG (2 Strands) Bifiliar Wound

Core

–

GETV 53x18x18 B52

Gap

–

≈ 0.020″ for a primary inductance of 1.35 µH, AL = 450 nH/Turn

2

16

MOTOROLA ANALOG IC DEVICE DATA

–T–

SEATING
PLANE

MC44602

OUTLINE DIMENSIONS

P2 SUFFIX

PLASTIC PACKAGE

CASE 648C–03

ISSUE C

–A–

16 9

–B–

18

NOTE 5

C

N

F

D

0.13 (0.005) T

E

G

16 PL

M

S

A

K

L

J 16 PL

0.13 (0.005) T

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.

2. CONTROLLING DIMENSION: INCH.

3. DIMENSION L TO CENTER OF LEADS WHEN
FORMED PARALLEL.

4. DIMENSION B DOES NOT INCLUDE MOLD
FLASH.

5. INTERNAL LEAD CONNECTION BETWEEN 4 AND
5, 12 AND 13.

DIM MIN MAX MIN MAX

A 0.740 0.840 18.80 21.34
B 0.240 0.260 6.10 6.60
C 0.145 0.185 3.69 4.69
D 0.015 0.021 0.38 0.53
E 0.050 BSC 1.27 BSC
F 0.040 0.70 1.02 1.78
G 0.100 BSC 2.54 BSC

M

M

S

B

J 0.008 0.015 0.20 0.38
K 0.115 0.135 2.92 3.43
L 0.300 BSC 7.62 BSC
M 0 10 0 10
N 0.015 0.040 0.39 1.01

MILLIMETERSINCHES

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MOTOROLA ANALOG IC DEVICE DATA

17

MC44602

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty , representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “T ypical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola
was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.

How to reach us:
USA/EUROPE/Locations Not Listed: Motorola Literature Distribution; JAPAN: Nippon Motorola Ltd.; Tatsumi–SPD–JLDC, 6F Seibu–Butsuryu–Center,

P.O. Box 20912; Phoenix, Arizona 85036. 1–800–441–2447 or 602–303–5454 3–14–2 Tatsumi Koto–Ku, Tokyo 135, Japan. 03–81–3521–8315

MFAX: RMF [e-mail] – TOUCHT ONE 602–244–6609 ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park,
INTERNET: http://Design–NET.com 51 Ting Kok Road, Tai Po, N.T ., Hong Kong. 852–26629298

18

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MOTOROLA ANALOG IC DEVICE DATA

MC44602/D

*MC44602/D*