Datasheet MC34262D, MC34262P, MC33262D, MC33262P Datasheet (Motorola)

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The MC34262/MC33262 are active power factor controllers specifically
designed for use as a preconverter in electronic ballast and in off–line power
converter applications. These integrated circuits feature an internal startup
timer for stand–alone applications, a one quadrant multiplier for near unity
power factor, zero current detector to ensure critical conduction operation,
transconductance error amplifier, quickstart circuit for enhanced startup,
trimmed internal bandgap reference, current sensing comparator, and a
totem pole output ideally suited for driving a power MOSFET.

Also included are protective features consisting of an overvoltage
comparator to eliminate runaway output voltage due to load removal, input
undervoltage lockout with hysteresis, cycle–by–cycle current limiting,
multiplier output clamp that limits maximum peak switch current, an RS latch
for single pulse metering, and a drive output high state clamp for MOSFET
gate protection. These devices are available in dual–in–line and surface
mount plastic packages.

• Overvoltage Comparator Eliminates Runaway Output Voltage

• Internal Startup Timer

• One Quadrant Multiplier

• Zero Current Detector

• Trimmed 2% Internal Bandgap Reference

• Totem Pole Output with High State Clamp

• Undervoltage Lockout with 6.0 V of Hysteresis

• Low Startup and Operating Current

• Supersedes Functionality of SG3561 and TDA4817

Order this document by MC34262/D

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POWER FACTOR

CONTROLLERS

SEMICONDUCTOR

TECHNICAL DATA

PLASTIC PACKAGE

8

1

8

1

PLASTIC PACKAGE

P SUFFIX

CASE 626

D SUFFIX

CASE 751

(SO–8)

Multiplier

Input

PIN CONNECTIONS

Simplified Block Diagram

Zero Current Detector

2.5V

Reference

Multiplier,

Latch,
PWM,
Timer,

Logic

3

6

Gnd

Multiplier

Undervoltage

Lockout

&

Compensation

Error Amp

2

Overvoltage
Comparator

+

1.08 V

+

Quickstart

ref

V

ref

Zero Current
Detect Input

5

V

CC

8

Drive Output

7

Current Sense
Input

4

Voltage
Feedback

1

Input

Voltage Feedback

Compensation
Multiplier Input

Current Sense

ORDERING INFORMATION

Device

MC34262D
MC34262P
MC33262D
MC33262P

1

Input

2
3
4

Input

(Top View)

Operating

Temperature Range

TA = 0° to +85°C

TA = –40° to +105°C

V

8

CC

7

Drive Output
Gnd

6

Zero Current

5

Detect Input

Package

Plastic DIP

Plastic DIP

SO–8

SO–8

MOTOROLA ANALOG IC DEVICE DATA

 Motorola, Inc. 1996 Rev 1

1

MC34262 MC33262

MAXIMUM RATINGS

Rating Symbol Value Unit

Total Power Supply and Zener Current (ICC + IZ) 30 mA
Output Current, Source or Sink (Note 1) I
Current Sense, Multiplier, and V oltage Feedback Inputs V
Zero Current Detect Input

High State Forward Current
Low State Reverse Current

Power Dissipation and Thermal Characteristics

P Suffix, Plastic Package, Case 626

Maximum Power Dissipation @ TA = 70°C
Thermal Resistance, Junction–to–Air

D Suffix, Plastic Package, Case 751

Maximum Power Dissipation @ TA = 70°C

Thermal Resistance, Junction–to–Air
Operating Junction Temperature T
Operating Ambient Temperature (Note 3)

MC34262
MC33262

Storage Temperature T

R

R

O

I

in

P

θJA

P

θJA

T

stg

in

D

D

J

A

500 mA

–1.0 to +10 V

50
–10

800
100

450
178

+150 °C

0 to + 85

– 40 to +105
– 65 to +150 °C

mA

mW

°C/W

mW

°C/W

°C

ELECTRICAL CHARACTERISTICS (V

temperature range that applies [Note 3], unless otherwise noted.)

Characteristic

ERROR AMPLIFIER

Voltage Feedback Input Threshold

TA = 25°C

TA = T
Line Regulation (VCC = 12 V to 28 V, TA = 25°C) Reg
Input Bias Current (VFB = 0 V) I
Transconductance (TA = 25°C) g
Output Current

Source (VFB = 2.3 V)

Sink (VFB = 2.7 V)
Output Voltage Swing

High State (VFB = 2.3 V)

Low State (VFB = 2.7 V)

OVERVOLTAGE COMPARATOR

Voltage Feedback Input Threshold V

MULTIPLIER

Input Bias Current, Pin 3 (VFB = 0 V) I
Input Threshold, Pin 2 V
Dynamic Input Voltage Range

Multiplier Input (Pin 3)

Compensation (Pin 2)

Multiplier Gain (V

ZERO CURRENT DETECTOR

Input Threshold Voltage (Vin Increasing) V
Hysteresis (Vin Decreasing) V
Input Clamp Voltage

High State (I

Low State (I

low

to T

DET

DET

(VCC = 12 V to 28 V)

high

= 0.5 V, V

Pin 3

= + 3.0 mA)

= – 3.0 mA)

Pin 2

= 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient

CC

V

V

= V

+ 1.0 V) (Note 4) K 0.43 0.65 0.87 1/V

th(M)

Symbol Min Typ Max Unit

V

FB

line

IB

m

I

O

OH(ea)

OL(ea)

FB(OV)

IB

th(M)

V

Pin 3

V

Pin 2

th

H

V

IH

V

IL

2.465

2.44
— 1.0 10 mV
— –0.1 – 0.5 µA
80 100 130 µmho

—
—

5.8
—

1.065 V

FB

— –0.1 – 0.5 µA

1.05 V

OL(EA)

0 to 2.5

V

to

th(M)

(V

+ 1.0)

th(M)

1.33 1.6 1.87 V
100 200 300 mV

6.1

0.3

1.08 V

1.2 V

V

(V

th(M)

2.5
—

10
10

6.4

1.7

OL(EA)

0 to 3.5

th(M)

+ 1.5)

6.7

0.7

FB

to

2.535

2.54

—
—

—

2.4

1.095 V

— V

—
—

—

1.0

FB

V

µA

V

V

V

V

2

MOTOROLA ANALOG IC DEVICE DATA

MC34262 MC33262

.

ELECTRICAL CHARACTERISTICS (V

= 12 V (Note 2), for typical values TA = 25°C, for min/max values TA is the operating ambient

CC

temperature range that applies (Note 3), unless otherwise noted.)

Characteristic

CURRENT SENSE COMPARATOR

Input Bias Current (V
Input Offset Voltage (V

= 0 V) I

Pin 4

Pin 2

= 1.1 V, V

= 0 V) V

Pin 3

Maximum Current Sense Input Threshold (Note 5) V
Delay to Output t

DRIVE OUTPUT

Output Voltage (VCC = 12 V)

Low State (I

Low State (I

High State (I

High State (I

= 20 mA)

Sink

= 200 mA)

Sink
Source
Source

= 20 mA)
= 200 mA)

Output Voltage (VCC = 30 V)
High State (I

= 20 mA, CL = 15 pF)

Source

Output Voltage Rise T ime (CL = 1.0 nF) t
Output Voltage Fall T ime (CL = 1.0 nF) t
Output Voltage with UVLO Activated

(VCC = 7.0 V, I

Sink

= 1.0 mA)

RESTART TIMER

Restart Time Delay t

UNDERVOLTAGE LOCKOUT

Startup Threshold (VCC Increasing) V
Minimum Operating Voltage After Turn–On (VCC Decreasing) V
Hysteresis V

TOTAL DEVICE

Power Supply Current

Startup (VCC = 7.0 V)
Operating
Dynamic Operating (50 kHz, CL = 1.0 nF)

Power Supply Zener Voltage (ICC = 25 mA) V

NOTES: 1. Maximum package power dissipation limits must be observed.

2.Adjust VCC above the startup threshold before setting to 12 V.

3. T

=0°C for MC34262

low

3. T

=–40°C for MC33262

low

T

=+85°C for MC34262

high

T

= +105°C for MC33262

high

Symbol Min Typ Max Unit

IB

IO

th(max)

PHL(in/out)

V

OL

V

OH

V

O(max)

r
f

V

O(UVLO)

DLY

th(on)

Shutdown

H

I

CC

Z

— – 0.15 –1.0 µA
— 9.0 25 mV

1.3 1.5 1.8 V
— 200 400 ns

—
—

9.8

7.8

0.3

2.4

10.3

8.4

0.8

3.3
—
—

14 16 18
— 50 120 ns
— 50 120 ns
— 0.1 0.5 V

200 620 — µs

11.5 13 14.5 V

7.0 8.0 9.0 V

3.8 5.0 6.2 V

—
—
—

0.25

6.5

9.0

0.4
12
20

30 36 — V

4. K =

5.This parameter is measured with VFB = 0 V, and V

Pin 4 Threshold

V

(V

Pin 3

Pin 2

– V

th(M)

)

Pin 3

V

V

mA

= 3.0 V

Figure 1. Current Sense Input Threshold

versus Multiplier Input

1.6
VCC = 12 V

1.4

TA = 25

°

C

1.2

1.0

0.8

0.6

0.4

0.2

, CURRENT SENSE PIN 4 THRESHOLD (V)

0

CS

V

1.4– 0.2 3.80.6 2.2 3.0

VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V)

MOTOROLA ANALOG IC DEVICE DATA

V

Pin 2

= 2.0 V

, CURRENT SENSE PIN 4 THRESHOLD (V)
V

CS

0.08

0.07

0.06

0.05

0.04

0.03

0.02

0.01

– 0.12

Figure 2. Current Sense Input Threshold

versus Multiplier Input, Expanded View

V

= 3.5 V

Pin 2

V

= 3.25 V

Pin 2

0

– 0.06 0.06 0.12 0.18 0.240

VM, MULTIPLIER PIN 3 INPUT VOLTAGE (V)

VCC = 12 V
TA = 25

V

Pin 2

°

C

= 2.0 V

3

MC34262 MC33262

Figure 3. V oltage Feedback Input Threshold

Change versus T emperature

4.0
VCC = 12 V
Pins 1 to 2

0

– 4.0

– 8.0

–12

, VOLTAGE FEEDBACK THRESHOLD CHANGE (mV)

–16

FB

–55

V

∆

– 25 0 25 50 75 100 125

TA, AMBIENT TEMPERATURE (

Figure 5. Error Amp Transconductance and

Phase versus Frequency

120

100

µ

Phase

Transconductance

80

°

C)

VCC = 12 V
VO = 2.5 V to 3.5 V
RL = 100 k to 3.0 V
CL = 2.0 pF

°

C

TA = 25

0

30

60

Figure 4. Overvoltage Comparator Input

FB

110

109

108

107

, OVERVOLTAGE INPUT THRESHOLD (%V )

106

–55

FB(OV)

V

Threshold versus T emperature

– 25 0 25 50 75 100

Figure 6. Error Amp Transient Response

4.00 V

TA, AMBIENT TEMPERATURE (

VCC = 12 V

°

C)

VCC = 12 V
RL = 100 k
CL = 2.0 pF
TA = 25

125

°

C

60

40

20

, TRANSCONDUCTANCE ( mho)

m

g

0

3.0 k 10 k 30 k 100 k 300 k 1.0 M 3.0 M
f, FREQUENCY (Hz)

Figure 7. Quickstart Charge Current

versus T emperature

1.80
VCC = 12 V

1.76

1.72

Voltage

1.68

, QUICKSTAR T CHARGE VOLTAGE (V)

chg

V

1.64

– 55 – 25 0 25 50 75 100 125

TA, AMBIENT TEMPERATURE (°C)

Current

90

120

150

180

900

800

700

600

500

3.25 V

, EXCESS PHASE (DEGREES)

O

2.50 V

800

µ

µ

700

600

, RESTART TIME DELAY ( s)

500

DLY

t

, QUICKSTART CHARGE CURRENT ( A)

chg

I

400

–55

5.0 µs/DIV

Figure 8. Restart Timer Delay

versus T emperature

VCC = 12 V

– 25 0 25 50 75 100 125

TA, AMBIENT TEMPERATURE (

°

C)

0.75 V/DIV

4

MOTOROLA ANALOG IC DEVICE DATA

MC34262 MC33262

1.7

1.6

1.5

, THRESHOLD VOLTAGE (V)

1.4

th

V

1.3

90%

Figure 9. Zero Current Detector Input

Threshold V oltage versus Temperature

Upper Threshold
(Vin, Increasing)

Lower Threshold

(Vin, Decreasing)

– 25 0 25 50 75 100 125

–55

TA, AMBIENT TEMPERATURE (

°

Figure 11. Drive Output Waveform

VCC = 12 V

C)

VCC = 12 V
CL = 1.0 nF

TA = 25

Figure 10. Output Saturation Voltage

versus Load Current

0

– 2.0

– 4.0

– 6.0

4.0

2.0

, OUTPUT SA TURATION VOLTAGE (V)

sat

V

0

0 80 160 240 320

Source Saturation

(Load to Ground)

Sink Saturation

(Load to VCC)

V

CC

Gnd

IO, OUTPUT LOAD CURRENT (mA)

VCC = 12 V

µ

s Pulsed Load

80
120 Hz Rate

Figure 12. Drive Output Cross Conduction

VCC = 12 V

CL = 15 pF

°

C

°

C

, OUTPUT VOL TAGE

O

TA = 25

5.0 V/DIV100 mA/DIV

10%

100 ns/DIV 100 ns/DIV

Figure 13. Supply Current

versus Supply V oltage

16

12

8.0
VFB = 0 V

, SUPPLY CURRENT (mA)

4.0

CC

I

0

0 10203040

V

, SUPPLY VOLTAGE (V)

CC

Current Sense = 0 V
Multiplier = 0 V
CL = 1.0 nF
f = 50 kHz

°

C

TA = 25

, SUPPLY CURRENT V

CC

I

Figure 14. Undervoltage Lockout Thresholds

versus T emperature

14
13
12
11
10

, SUPPLY VOLTAGE (V)

9.0

CC

V

8.0

7.0
– 55 – 25 0 25 50 75 100 125

Startup Threshold

(VCC Increasing)

Minimum Operating Threshold

(VCC Decreasing)

TA, AMBIENT TEMPERATURE (°C)

MOTOROLA ANALOG IC DEVICE DATA

5

MC34262 MC33262

FUNCTIONAL DESCRIPTION

Introduction

With the goal of exceeding the requirements of legislation
on line–current harmonic content, there is an ever increasing
demand for an economical method of obtaining a unity power
factor. This data sheet describes a monolithic control IC that
was specifically designed for power factor control with
minimal external components. It offers the designer a simple,
cost–effective solution to obtain the benefits of active power
factor correction.

Most electronic ballasts and switching power supplies use
a bridge rectifier and a bulk storage capacitor to derive raw dc
voltage from the utility ac line, Figure 15.

Figure 15. Uncorrected Power Factor Circuit

Rectifiers Converter

AC

Line

+

Bulk
Storage
Capacitor

Load

This simple rectifying circuit draws power from the line
when the instantaneous ac voltage exceeds the capacitor
voltage. This occurs near the line voltage peak and results in
a high charge current spike, Figure 16. Since power is only
taken near the line voltage peaks, the resulting spikes of
current are extremely nonsinusoidal with a high content of
harmonics. This results in a poor power factor condition
where the apparent input power is much higher than the real
power. Power factor ratios of 0.5 to 0.7 are common.

Power factor correction can be achieved with the use of
either a passive or an active input circuit. Passive circuits
usually contain a combination of large capacitors, inductors,
and rectifiers that operate at the ac line frequency. Active
circuits incorporate some form of a high frequency switching
converter for the power processing, with the boost converter
being the most popular topology, Figure 17. Since active
input circuits operate at a frequency much higher than that of
the ac line, they are smaller, lighter in weight, and more
efficient than a passive circuit that yields similar results. With
proper control of the preconverter, almost any complex load

can be made to appear resistive to the ac line, thus
significantly reducing the harmonic current content.

Figure 16. Uncorrected Power Factor

Input Waveforms

V

pk

Rectified

DC

0

AC Line

Voltage

0

AC Line

Current

Line Sag

The MC34262, MC33262 are high performance, critical
conduction, current–mode power factor controllers
specifically designed for use in off–line active preconverters.
These devices provide the necessary features required to
significantly enhance poor power factor loads by keeping the
ac line current sinusoidal and in phase with the line voltage.

Operating Description

The MC34262, MC33262 contain many of the building
blocks and protection features that are employed in modern
high performance current mode power supply controllers.
There are, however, two areas where there is a major
difference when compared to popular devices such as the
UC3842 series. Referring to the block diagram in Figure 19,
note that a multiplier has been added to the current sense
loop and that this device does not contain an oscillator. The
reasons for these differences will become apparent in the
following discussion. A description of each of the functional
blocks is given below.

Figure 17. Active Power Factor Correction Preconverter

Rectifiers PFC Preconverter

AC

Line

6

+

High
Frequency
Bypass
Capacitor

MC34362

Bulk

+

Storage
Capacitor

MOTOROLA ANALOG IC DEVICE DATA

Converter

Load

MC34262 MC33262

Error Amplifier

An Error Amplifier with access to the inverting input and
output is provided. The amplifier is a transconductance type,
meaning that it has high output impedance with controlled
voltage–to–current gain. The amplifier features a typical gm
of 100 µmhos (Figure 5). The noninverting input is internally
biased at 2.5 V ± 2.0% and is not pinned out. The output
voltage of the power factor converter is typically divided down
and monitored by the inverting input. The maximum input
bias current is – 0.5 µA, which can cause an output voltage
error that is equal to the product of the input bias current and
the value of the upper divider resistor R2. The Error Amp
output is internally connected to the Multiplier and is pinned
out (Pin 2) for external loop compensation. Typically, the
bandwidth is set below 20 Hz, so that the amplifier’s output
voltage is relatively constant over a given ac line cycle. In
effect, the error amp monitors the average output voltage of
the converter over several line cycles. The Error Amp output
stage was designed to have a relatively constant
transconductance over temperature. This allows the
designer to define the compensated bandwidth over the
intended operating temperature range. The output stage can
sink and source 10 µA of current and is capable of swinging
from 1.7 V to 6.4 V, assuring that the Multiplier can be driven
over its entire dynamic range.

A key feature to using a transconductance type amplifier,
is that the input is allowed to move independently with
respect to the output, since the compensation capacitor is
connected to ground. This allows dual usage of of the Voltage
Feedback Input pin by the Error Amplifier and by the
Overvoltage Comparator.

Overvoltage Comparator

An Overvoltage Comparator is incorporated to eliminate
the possibility of runaway output voltage. This condition can
occur during initial startup, sudden load removal, or during
output arcing and is the result of the low bandwidth that must
be used in the Error Amplifier control loop. The Overvoltage
Comparator monitors the peak output voltage of the
converter, and when exceeded, immediately terminates
MOSFET switching. The comparator threshold is internally
set to 1.08 V
normal operation, the value of the output filter capacitor C
must be large enough to keep the peak–to–peak ripple less
than 16% of the average dc output. The Overvoltage
Comparator input to Drive Output turn–off propagation delay
is typically 400 ns. A comparison of startup overshoot without
and with the Overvoltage Comparator circuit is shown in
Figure 23.

Multiplier

A single quadrant, two input multiplier is the critical
element that enables this device to control power factor. The
ac full wave rectified haversines are monitored at Pin 3

. In order to prevent false tripping during

ref

with respect to ground while the Error Amp output at Pin 2 is
monitored with respect to the Voltage Feedback Input
threshold. The Multiplier is designed to have an extremely
linear transfer curve over a wide dynamic range, 0 V to 3.2 V
for Pin 3, and 2.0 V to 3.75 V for Pin 2, Figure 1. The Multiplier
output controls the Current Sense Comparator threshold as
the ac voltage traverses sinusoidally from zero to peak line,
Figure 18. This has the effect of forcing the MOSFET on–time
to track the input line voltage, resulting in a fixed Drive Output
on–time, thus making the preconverter load appear to be
resistive to the ac line. An approximation of the Current
Sense Comparator threshold can be calculated from the
following equation. This equation is accurate only under the
given test condition stated in the electrical table.

VCS, Pin 4 Threshold ≈ 0.65 (V

A significant reduction in line current distortion can be
attained by forcing the preconverter to switch as the ac line
voltage crosses through zero. The forced switching is
achieved by adding a controlled amount of offset to the
Multiplier and Current Sense Comparator circuits. The
equation shown below accounts for the built–in offsets and is
accurate to within ten percent. Let V

VCS, Pin 4 Threshold = 0.544 (V

+ 0.0417 (V

Zero Current Detector

The MC34262 operates as a critical conduction current
mode controller, whereby output switch conduction is initiated
by the Zero Current Detector and terminated when the peak
inductor current reaches the threshold level established by
the Multiplier output. The Zero Current Detector initiates the
next on–time by setting the RS Latch at the instant the
inductor current reaches zero. This critical conduction mode
of operation has two significant benefits. First, since the
MOSFET cannot turn–on until the inductor current reaches
zero, the output rectifier reverse recovery time becomes less
critical, allowing the use of an inexpensive rectifier. Second,
since there are no deadtime gaps between cycles, the ac line
current is continuous, thus limiting the peak switch to twice
the average input current.

3

The Zero Current Detector indirectly senses the inductor
current by monitoring when the auxiliary winding voltage falls
below 1.4 V . To prevent false tripping, 200 mV of hysteresis is
provided. Figure 9 shows that the thresholds are
well–defined over temperature. The Zero Current Detector
input is internally protected by two clamps. The upper 6.7 V
clamp prevents input overvoltage breakdown while the lower

0.7 V clamp prevents substrate injection. Current limit
protection of the lower clamp transistor is provided in the
event that the input pin is accidentally shorted to ground. The
Zero Current Detector input to Drive Output turn–on
propagation delay is typically 320 ns.

Pin 2

Pin 2

th(M)

Pin 2

– V

– V

= 1.991 V

– V

th(M)

th(M)

)

th(M)

) V

) V

Pin 3

Pin 3

MOTOROLA ANALOG IC DEVICE DATA

7

MC34262 MC33262

Figure 18. Inductor Current and MOSFET

Gate Voltage Waveforms

Peak

Inductor Current

0

On

MOSFET

Q1

Off

Current Sense Comparator and RS Latch

The Current Sense Comparator RS Latch configuration
used ensures that only a single pulse appears at the Drive
Output during a given cycle. The inductor current is
converted to a voltage by inserting a ground–referenced
sense resistor R7 in series with the source of output switch
Q1. This voltage is monitored by the Current Sense Input and
compared to a level derived from the Multiplier output. The
peak inductor current under normal operating conditions is
controlled by the threshold voltage of Pin 4 where:

I

L(pk

Pin 4 Threshold

) =

R

7

Abnormal operating conditions occur during preconverter
startup at extremely high line or if output voltage sensing is
lost. Under these conditions, the Multiplier output and Current
Sense threshold will be internally clamped to 1.5 V.
Therefore, the maximum peak switch current is limited to:

I

pk(max)

1.5 V

=

R

7

An internal RC filter has been included to attenuate any
high frequency noise that may be present on the current
waveform. This filter helps reduce the ac line current
distortion especially near the zero crossings. With the
component values shown in Figure 20, the Current Sense
Comparator threshold, at the peak of the haversine varies
from 1.1 V at 90 Vac to 100 mV at 268 Vac. The Current
Sense Input to Drive Output turn–off propagation delay is
typically less than 200 ns.

Average

Timer

A watchdog timer function was added to the IC to eliminate
the need for an external oscillator when used in stand–alone
applications. The Timer provides a means to automatically
start or restart the preconverter if the Drive Output has been
off for more than 620 µs after the inductor current reaches
zero. The restart time delay versus temperature is shown in
Figure 8.

Undervoltage Lockout and Quickstart

An Undervoltage Lockout comparator has been
incorporated to guarantee that the IC is fully functional before
enabling the output stage. The positive power supply terminal
(VCC) is monitored by the UVLO comparator with the upper
threshold set at 13 V and the lower threshold at 8.0 V. In the
stand–by mode, with VCC at 7.0 V , the required supply current
is less than 0.4 mA. This large hysteresis and low startup
current allow the implementation of efficient bootstrap startup
techniques, making these devices ideally suited for wide
input range off–line preconverter applications. An internal
36 V clamp has been added from VCC to ground to protect
the IC and capacitor C4 from an overvoltage condition. This
feature is desirable if external circuitry is used to delay the
startup of the preconverter. The supply current, startup, and
operating voltage characteristics are shown in Figures 13
and 14.

A Quickstart circuit has been incorporated to optimize
converter startup. During initial startup, compensation
capacitor C1 will be discharged, holding the error amp output
below the Multiplier threshold. This will prevent Drive Output
switching and delay bootstrapping of capacitor C4 by diode
D6. If Pin 2 does not reach the multiplier threshold before C
discharges below the lower UVLO threshold, the converter
will “hiccup” and experience a significant startup delay. The
Quickstart circuit is designed to precharge C1 to 1.7 V , Figure

7. This level is slightly below the Pin 2 Multiplier threshold,
allowing immediate Drive Output switching and bootstrap
operation when C4 crosses the upper UVLO threshold.

Drive Output

The MC34262/MC33262 contain a single totem–pole
output stage specifically designed for direct drive of power
MOSFETs. The Drive Output is capable of up to ± 500 mA
peak current with a typical rise and fall time of 50 ns with a

1.0 nF load. Additional internal circuitry has been added to
keep the Drive Output in a sinking mode whenever the
Undervoltage Lockout is active. This characteristic eliminates
the need for an external gate pull–down resistor. The
totem–pole output has been optimized to minimize
cross–conduction current during high speed operation. The
addition of two 10 Ω resistors, one in series with the source
output transistor and one in series with the sink output
transistor, helps to reduce the cross–conduction current and
radiated noise by limiting the output rise and fall time. A 16 V
clamp has been incorporated into the output stage to limit the
high state VOH. This prevents rupture of the MOSFET gate
when V

exceeds 20 V.

CC

4

8

MOTOROLA ANALOG IC DEVICE DATA

MC34262 MC33262

APPLICATIONS INFORMATION

The application circuits shown in Figures 19, 20 and 21
reveal that few external components are required for a
complete power factor preconverter. Each circuit is a peak
detecting current–mode boost converter that operates in
critical conduction mode with a fixed on–time and variable
off–time. A major benefit of critical conduction operation is
that the current loop is inherently stable, thus eliminating the
need for ramp compensation. The application in Figure 19
operates over an input voltage range of 90 Vac to 138 Vac
and provides an output power of 80 W (230 V at 350 mA) with

nominal line. Figures 20 and 21 are universal input
preconverter examples that operate over a continuous input
voltage range of 90 Vac to 268 Vac. Figure 20 provides an
output power of 175 W (400 V at 440 mA) while Figure 21
provides 450 W (400 V at 1.125 A). Both circuits have an
observed worst–case power factor of approximately 0.989.
The input current and voltage waveforms of Figure 20 are
shown in Figure 22 with operation at 115 Vac and 230 Vac.
The data for each of the applications was generated with the
test set–up shown in Figure 24.

an associated power factor of approximately 0.998 at

T able 1. Design Equations

Notes Calculation Formula

Calculate the maximum required output power. Required Converter Output Power PO = VO I

Calculated at the minimum required ac line voltage
for output regulation. Let the efficiency η = 0.92 for
low line operation.

Let the switching cycle t = 40 µs for universal input
(85 to 265 Vac) operation and 20 µs for fixed input
(92 to 138 Vac, or 184 to 276 Vac) operation.

In theory the on–time ton is constant. In practice t
tends to increase at the ac line zero crossings due
to the charge on capacitor C5. Let Vac = V ac
ton and t

The off–time t
voltage and approaches zero at the ac line zero
crossings. Theta (θ) represents the angle of the ac
line voltage.

The minimum switching frequency occurs at the peak
of the ac line voltage. As the ac line voltage traverses
from peak to zero, t
increase in switching frequency.

Set the current sense threshold VCS to 1.0 V for
universal input (85 Vac to 265 Vac) operation and
to 0.5 V for fixed input (92 Vac to 138 Vac, or
184 Vac to 276 Vac) operation. Note that VCS must
be <1.4 V.

Set the multiplier input voltage VM to 3.0 V at high
line. Empirically adjust VM for the lowest distortion
over the ac line voltage range while guaranteeing
startup at minimum line.

The IIB R1 error term can be minimized with a divider
current in excess of 50 µA.

The calculated peak–to–peak ripple must be less than
16% of the average dc output voltage to prevent false
tripping of the Overvoltage Comparator. Refer to the
Overvoltage Comparator text. ESR is the equivalent
series resistance of C

The bandwidth is typically set to 20 Hz. When operating
at high ac line, the value of C1 may need to be
increased. (See Figure 25)

The following converter characteristics must be chosen:

calculations.

off

is greatest at the peak of the ac line

off

approaches zero producing an

off

3

— Desired output voltage — AC RMS line voltage

V

O

I

— Desired output current

O

— Converter output peak–to–peak ripple voltage

∆V

O

Vac

on

for initial

(LL)

Vac

— AC RMS low line voltage

(LL)

Peak Inductor Current

Inductance

Switch On–Time

Switch Off–Time

Switching Frequency

Peak Switch Current

Multiplier Input Voltage

Converter Output Voltage

Converter Output
Peak to Peak
Ripple Voltage

Error Amplifier Bandwidth

I

=

L(pk)

V

O

ǒǓ

– Vac

t

LP =

∆V

VO = V

O(pp)

2

ton =

t

=

off

2 Vac

f =

R7 =

VM =

R

ref

R

= I

O

BW =

22 P

ηVac

(LL)

2 VO P

2 PO L

η Vac

t

on

V

O

Sin θ

1

ton + t

V

CS

I

L(pk)

Vac

R

5

R

3

2

+ 1

1

2πfac C

gm

2 π C

off

+ 1

O

(LL)

O

2

2

Ǔǒ

1

1

O

η Vac

P

– 1

Ǔǒ

– IIB R

Ǔǒ

3

(LL)

2

2
+ ESR

2

2

MOTOROLA ANALOG IC DEVICE DATA

9

92 to 138

Vac

RFI

Filter

MC34262 MC33262

Figure 19. 80 W Power Factor Controller

C

1

5

D

D

4

2

Zero Current

D

D

3

1

Detector

+

1.2V

+

1.6V/

36V

6.7V

1.4V
UVLO

+

16V

Drive

Output

Delay

2.5V

Reference

R

Timer

RS

Latch

13V/

8.0V

10

10

100k

R

6

8

5

7

1N4934

+

100

C

4

22k

R

D

6

T

4

MUR130

MTP

8N50E

Q

1

V

D

O

5

+

220

C

3

230V/0.35A

0.01
C

2

2.2M
R

5

7.5k

R

3

Current Sense

Comparator

3

4

V

ref

1

Multiplier

1.5V

Overvoltage
Comparator

+

1.08 V

ref

Error Amp

10µA

20k
10pF

+

Quickstart

6

2

0.68
C

1

0.1
R

1.0M
R

2

7

11k

R

1

Power Factor Controller Test Data

AC Line Input DC Output

Current Harmonic Distortion (% I

V

rms

P

PF I

in

fund

THD 2357V

90 85.9 0.999 0.93 2.6 0.08 1.6 0.84 0.95 4.0 230.7 0.350 80.8 94.0
100 85.3 0.999 0.85 2.3 0.13 1.0 1.2 0.73 4.0 230.7 0.350 80.8 94.7
110 85.1 0.998 0.77 2.2 0.10 0.58 1.5 0.59 4.0 230.7 0.350 80.8 94.9
120 84.7 0.998 0.71 3.0 0.09 0.73 1.9 0.58 4.1 230.7 0.350 80.8 95.3
130 84.4 0.997 0.65 3.9 0.12 1.7 2.2 0.61 4.1 230.7 0.350 80.8 95.7
138 84.1 0.996 0.62 4.6 0.16 2.4 2.3 0.60 4.1 230.7 0.350 80.8 96.0

This data was taken with the test set–up shown in Figure 24.

T

= Coilcraft N2881–A

Primary: 62 turns of # 22 AWG
Secondary: 5 turns of # 22 AWG
Core: Coilcraft PT2510, EE 25

Heatsink

Gap: 0.072″ total for a primary inductance (LP) of 320 µH

= AAVID Engineering Inc. 590302B03600, or 593002B03400

fund

)

O(pp)

V

I

O

O

P

O

η(%)

10

MOTOROLA ANALOG IC DEVICE DATA

90 to 268

Vac

RFI

Filter

MC34262 MC33262

Figure 20. 175 W Universal Input Power Factor Controller

C

1

5

D

D

4

2

36V

6.7V

+

1.2V

1.6V/

+

Zero Current

D

D

3

1

Detector

1.4V

2.5V

UVLO

Reference

Timer R

Delay

16V

Drive

Output

RS

Latch

100k

1N4934

R

6

D

+

100

C

4
22k

R

6

T

4

MUR460

MTP

14N50E

Q

1

V

D

O

5

+

330

C

3

400V/0.44A

8

5

13V/

+

8.0V

10

7

10

0.01
C

2

1.3M
R

5

12k

R

Current Sense

Comparator

Multiplier

1.5V

Overvoltage
Comparator

+

1.08 V

ref

Error Amp

10µA

20k
10pF

+

4

V

ref

1

3

3

Quickstart

6

2

0.68
C

1

0.1
R

1.6M
R

2

7

10k

R

1

Power Factor Controller Test Data

AC Line Input DC Output

Current Harmonic Distortion (% I

V

rms

P

PF I

in

fund

THD 2357V

90 193.3 0.991 2.15 2.8 0.18 2.6 0.55 1.0 3.3 402.1 0.44 176.9 91.5
120 190.1 0.998 1.59 1.6 0.10 1.4 0.23 0.72 3.3 402.1 0.44 176.9 93.1
138 188.2 0.999 1.36 1.2 0.12 1.3 0.65 0.80 3.3 402.1 0.44 176.9 94.0
180 184.9 0.998 1.03 2.0 0.10 0.49 1.2 0.82 3.4 402.1 0.44 176.9 95.7
240 182.0 0.993 0.76 4.4 0.09 1.6 2.3 0.51 3.4 402.1 0.44 176.9 97.2
268 180.9 0.989 0.69 5.9 0.10 2.3 2.9 0.46 3.4 402.1 0.44 176.9 97.8

This data was taken with the test set–up shown in Figure 24.

T

= Coilcraft N2880–A

Primary: 78 turns of # 16 AWG
Secondary: 6 turns of # 18 AWG
Core: Coilcraft PT4215, EE 42–15

Heatsink

Gap: 0.104″ total for a primary inductance (LP) of 870 µH

= AAVID Engineering Inc. 590302B03600

fund

)

O(pp)

V

O

I

O

P

O

η(%)

MOTOROLA ANALOG IC DEVICE DATA

11

90 to 268

Vac

RFI

Filter

MC34262 MC33262

Figure 21. 450 W Universal Input Power Factor Controller

C

2

5

D

D

4

2

36V

6.7V

+

1.2V

1.6V/

+

Zero Current

D

D

3

1

Detector

1.4V

2.5V

Reference

Timer R

Delay

UVLO

+

16V

Drive

Output

RS

Latch

13V/

8.0V

10

10

100k

R

6

8

5

7

1N4934

+

100

C

4

22k

R

D

6

T

4

MUR460

MTW

20N50E

Q

1

V

D

O

5

+

330

C

3

400V/1.125A

0.01
C

2

1.3M
R

5

12k

R

ref

4

330

0.001

0.05

1

Current Sense

Comparator

3

Multiplier

1.5V

Overvoltage
Comparator

+

1.08 V

ref

Error Amp

10µA

20k
10pF

+

V

3

Quickstart

6

2

0.68
C

1

1.6M
R

2

R

7

10k

R

1

Power Factor Controller Test Data

AC Line Input DC Output

Current Harmonic Distortion (% I

V

rms

P

PF I

in

fund

THD 2357V

90 489.5 0.990 5.53 2.2 0.10 1.5 0.25 0.83 8.8 395.5 1.14 450.9 92.1
120 475.1 0.998 3.94 2.5 0.12 0.29 0.62 0.52 8.8 395.5 1.14 450.9 94.9
138 470.6 0.998 3.38 2.1 0.06 0.70 1.1 0.41 8.8 395.5 1.14 450.9 95.8
180 463.4 0.998 2.57 4.1 0.21 2.0 1.6 0.71 8.9 395.5 1.14 450.9 97.3
240 460.1 0.996 1.91 4.8 0.14 4.3 2.2 0.63 8.9 395.5 1.14 450.9 98.0
268 459.1 0.995 1.72 5.8 0.10 5.0 2.5 0.61 8.9 395.5 1.14 450.9 98.2

This data was taken with the test set–up shown in Figure 24.

T

= Coilcraft P3657–A

Primary: 38 turns Litz wire, 1300 strands of #48 AWG, Kerrigan–Lewis, Chicago, IL
Secondary: 3 turns of # 20 AWG
Core: Coilcraft PT4220, EE 42–20

Heatsink

Gap: 0.180″ total for a primary inductance (LP) of 190 µH

= AAVID Engineering Inc. 604953B04000 Extrusion

fund

)

O(pp)

V

I

O

O

P

O

η(%)

12

MOTOROLA ANALOG IC DEVICE DATA

MC34262 MC33262

Figure 22. Power Factor Corrected Input Waveforms

(Figure 20 Circuit)

Input = 115 VAC, Output = 175 W Input = 230 VAC, Output = 175 W

Voltage = 100 V/DIV

500 V

26%

400 V

0 V

Current = 1.0 A/DIV

Voltage = 100 V/DIV

2.0 ms/DIV 2.0 ms/DIV

Figure 23. Output Voltage Startup Overshoot

(Figure 20 Circuit)

Without Overvoltage Comparator

432 V
400 V

80 V/DIV

0 V

Current = 1.0 A/DIV

With Overvoltage Comparator

8%

80 V/DIV

200 ms/DIV200 ms/DIV

Figure 24. Power Factor Test Set–Up

2X Step–Up

Line

115 Vac

Input

Neutral

Earth

Isolation

Transformer

Autoformer

AC POWER ANALYZER
PM 1000

OI

HIHI

ArmsVrmsPFVAW

53210

HARMFREQ

AinstAcfVcf

1397

11

LO

LO

Voltech

VA

0.1

RFI Test Filter

T

0.005

0.005

1.0

0 to 270 Vac

Output to Power

Factor

Controller Circuit

An RFI filter is required for best performance when connecting the preconverter directly to the ac line. The filter attenuates the level of high
frequency switching that appears on the ac line current waveform. Figures 19 and 20 work well with commercially available two stage filters such
as the Delta Electronics 03DPCG5. Shown above is a single stage test filter that can easily be constructed with four ac line rated capacitors and a
common–mode transformer . Coilcraft CMT3–28–2 was used to test Figures 19 and 20. It has a minimum inductance of 28 mH and a maximum
current rating of 2.0 A. Coilcraft CMT4–17–9 was used to test Figure 21. It has a minimum inductance of 17 mH and a maximum current rating of

η

9.0 A. Circuit conversion efficiency

(%) was calculated without the power loss of the RFI filter.

MOTOROLA ANALOG IC DEVICE DATA

13

MC34262 MC33262

Figure 25. Error Amp Compensation

Error Amp

10µA

6

The Error Amp output is a high impedance node and is susceptible to noise pickup. To minimize pickup, compensation capacitor C1 must be
connected as close to Pin 2 as possible with a short, heavy ground returning directly to Pin 6. When operating at high ac line, the voltage at Pin 2
may approach the lower threshold of the Multiplier,
circuit instability, high distortion and poor power factor. This problem can be eliminated by increasing the value of C

≈

2.0 V . If there is excessive ripple on Pin 2, the Multiplier will be driven into cut–off causing

2

C

1

+

Figure 26. Current Waveform Spike Suppression

R

2

1

R

1

.

1

Figure 27. Negative Current Waveform

Spike Suppression

7

22k

10pF

Current

Sense

Comparator

A narrow turn–on spike is usually present on the leading edge of the current
waveform and can cause circuit instability. The MC34262 provides an
internal RC filter with a time constant of 220 ns. An additional external RC
filter may be required in universal input applications that are above 200 W. It
is suggested that the external filter be placed directly at the Current Sense
Input and have a time constant that approximates the spike duration.

4

R

C

R

7

7

22k

10pF

Current

Sense

Comparator

A negative turn–off spike can be observed on the trailing edge of the current
waveform. This spike is due to the parasitic inductance of resistor R
is excessive, it can cause circuit instability. The addition of Shottky diode D
can effectively clamp the negative spike. The addition of the external RC filter
shown in Figure 26 may provide sufficient spike attenuation.

4

D

1

, and if it

7

R

7

1

14

MOTOROLA ANALOG IC DEVICE DATA

MC34262 MC33262

Figure 28. Printed Circuit Board and Component Layout

(Circuits of Figures 15 and 16)

(Top View)

3.0

″

NOTE: Use 2 oz. copper laminate for optimum circuit performance.

MOTOROLA ANALOG IC DEVICE DATA

4.5

″

(Bottom View)

15

NOTE 2

–T–

–T–

SEATING
PLANE

H

58

14

F

–A–

N

D

G

0.13 (0.005) B

–A–

58

4X P

–B–

14

G

C

SEATING
PLANE

8X D

K

–B–

C

K

M

A

T

0.25 (0.010)MB

SS

A0.25 (0.010)MTB

MC34262 MC33262

OUTLINE DIMENSIONS

P SUFFIX

PLASTIC PACKAGE

CASE 626–05

ISSUE K

L

J

M

M

M

D SUFFIX

PLASTIC PACKAGE

CASE 751–05

(SO–8)

ISSUE N

M

R

X 45

_

_

M

J

NOTES:

1. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.

2. PACKAGE CONTOUR OPTIONAL (ROUND OR
SQUARE CORNERS).

3. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.

DIM MIN MAX MIN MAX

A 9.40 10.16 0.370 0.400
B 6.10 6.60 0.240 0.260
C 3.94 4.45 0.155 0.175
D 0.38 0.51 0.015 0.020
F 1.02 1.78 0.040 0.070
G 2.54 BSC 0.100 BSC
H 0.76 1.27 0.030 0.050
J 0.20 0.30 0.008 0.012
K 2.92 3.43 0.115 0.135
L 7.62 BSC 0.300 BSC

M ––– 10 ––– 10

N 0.76 1.01 0.030 0.040

NOTES:

1. DIMENSIONING AND TOLERANCING PER
ANSI Y14.5M, 1982.

2. CONTROLLING DIMENSION: MILLIMETER.

3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.

4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.

5. DIMENSION D DOES NOT INCLUDE
DAMBAR PROTRUSION. ALLOWABLE
DAMBAR PROTRUSION SHALL BE 0.127
(0.005) TOTAL IN EXCESS OF THE D
DIMENSION AT MAXIMUM MATERIAL
CONDITION.

DIM MIN MAX MIN MAX

F

A 4.80 5.00 0.189 0.196
B 3.80 4.00 0.150 0.157
C 1.35 1.75 0.054 0.068
D 0.35 0.49 0.014 0.019

F 0.40 1.25 0.016 0.049

G 1.27 BSC 0.050 BSC

J 0.18 0.25 0.007 0.009
K 0.10 0.25 0.004 0.009
M 0 7 0 7
P 5.80 6.20 0.229 0.244
R 0.25 0.50 0.010 0.019

__

____

INCHESMILLIMETERS

INCHESMILLIMETERS

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty , representation or guarantee regarding
the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “T ypical” parameters which may be provided in Motorola
data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals”
must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of
others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other
applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury
or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola
and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees
arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola
was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal
Opportunity/Affirmative Action Employer.

How to reach us:
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INTERNET: http://Design–NET.com 51 Ting Kok Road, Tai Po, N.T ., Hong Kong. 852–26629298

16

◊

MOTOROLA ANALOG IC DEVICE DATA

MC34262/D

*MC34262/D*