Datasheet MC13158FTB Datasheet (Motorola)

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The MC13158 is a wideband IF subsystem that is designed for high performance data and analog applications. Excellent high frequency performance is achieved, with low cost, through the use of Motorola’s MOSAIC 1.5 RF bipolar process. The MC13158 has an on–board grounded collector VCO transistor that may be used with a fundamental or overtone crystal in single channel operation or with a PLL in multi–channel operation. The mixer is useful to 500 MHz and may be used in a balanced differential or single ended configuration. The IF amplifier is split to accommodate two low cost cascaded filters. RSSI output is derived by summing the output of both IF sections. A precision data shaper has an Off function to shut the output off to save current. An enable control is provided to power down the IC for power management in battery operated applications.

Applications include DECT , wideband wireless data links for personal and portable laptop computers and other battery operated radio systems which utilize GFSK, FSK or FM modulation.

• Designed for DECT Applications

• 1.8 to 6.0 Vdc Operating Voltage

• Low Power Consumption in Active and Standby Mode

• Greater than 600 kHz Detector Bandwidth

• Data Slicer with Special Off Function

• Enable Function for Power Down of Battery Operated Systems

• RSSI Dynamic Range of 80 dB Minimum

• Low External Component Count

WIDEBAND FM IF

SUBSYSTEM FOR DECT

AND DIGITAL APPLICATIONS

SEMICONDUCTOR

TECHNICAL DATA

32

1

FTB SUFFIX

PLASTIC PACKAGE

CASE 873

(Thin QFP)

ORDERING INFORMATION

Operating

Device

MC13158FTB TA = – 40 to +85°C TQFP–32

Temperature Range

Package

Mix Out

V

CC1

IF In IF Dec1

IF Dec2

IF Out

V

CC2

Lim In

Representative Block Diagram

Osc

N/C

11

N/C

Osc

Emit

MC13158

Data

Slicer

5.0 p

12

Lim Out

Base

13

Quad

Mix

In1

In2

1

2

3

4

5

6

7

8

IF Amp

LIM Amp

10

9

Lim

Dec2

Dec1

This device contains 234 active transistors.

N/C

14

N/C

EE1

15

Det

Gain

EnableV

Bias

V

2532 31 30 29 28 27 26

16 EE2

24

23

22

21

20

19

18

17

RSSI

RSSI Buf

DS Gnd

DS Out

DS In2

DS “off”

DS In1

Det Out

MOTOROLA ANALOG IC DEVICE DATA

 Motorola, Inc. 1996 Rev 1

1

MC13158

MAXIMUM RATINGS

Rating Pin Symbol Value Unit

Power Supply Voltage 16, 26 V Junction Temperature T Storage Temperature Range T

NOTE: 1.Devices should not be operated at or outside these values. The “Recommended Operating

Conditions” provide for actual device operation.

S(max)

JMAX

stg

6.5 Vdc

+150 °C

–65 to +150 °C

RECOMMENDED OPERATING CONDITIONS (V

Rating

Power Supply Voltage 2, 7 V

TA = 25°C –40°C ≤ TA≤ 85°C 16, 26

Input Frequency 31, 32 F Ambient Temperature Range T Input Signal Level 31, 32 V

DC ELECTRICAL CHARACTERISTICS (T

Characteristic

Total Drain Current VS = 2.0 Vdc 16, 26 I

DATA SLICER (Input Voltage Referenced to VEE; VS = 3.0 Vdc; No Input Signal)

Output Current; V18 LO; V19 = V

Data Slicer Enabled (DS “on”) V18 < V

Output Current; V18 HI; V19 = V

Data Slicer Enabled (DS “on”) V18 > V

Output Current; V19 = V

Data Slicer Disabled (DS “off”) V20 = VS/2

= 25°C; VS = 3.0 Vdc; No Input Signal; See Figure 1.)

A

Condition Pin Symbol Min Typ Max Unit

VS = 3.0 Vdc 3.5 5.7 8.5 VS = 6.0 Vdc 3.5 6.0 9.5 See Figure 2

V20 = VS/2

See Figure 3

V20 = VS/2

See Figure 4

= V2 = V7; VEE = V16 = V22 = V26; VS = VCC – VEE)

CC

Pin Symbol Value Unit

TOTAL

EE

20

EE

20

CC

21 I

21

S

in A in

2.5 5.5 8.5 mA

2.0 5.9 – mA

– 0.1 1.0 µA

2.0 to 6.0 Vdc

10 to 500 MHz

–40 to +85 °C

200 mVrms

AC ELECTRICAL CHARACTERISTICS (T

Characteristic

MIXER

Mixer Conversion Gain Vin = 1.0 mVrms 31, 32, 1 – – 22 – dB

Noise Figure Input Matched 31, 32, 1 NF – 14 – dB Mixer Input Impedance Single–Ended 31, 32 Rp – 865 – Ω

Mixer Output Impedance 1 – – 330 – Ω

2

= 25°C; VS = 3.0 Vdc; fRF = 110.7 MHz; fLO = 100 MHz; See Figure 1.)

A

Condition Pin Symbol Min Typ Max Unit

See Figure 5

See Figure 15 Cp – 1.6 – pF

MOTOROLA ANALOG IC DEVICE DATA

MC13158

AC ELECTRICAL CHARACTERISTICS (continued) (T

Characteristic

IF AMPLIFIER SECTION

IF RSSI Slope See Figure 8 23 – 0.15 0.3 0.4 µA/dB IF Gain f = 10.7 MHz 3, 6 – – 36 – dB

Input Impedance 3 – – 330 – Ω Output Impedance 6 – – 330 – Ω

LIMITING AMPLIFIER SECTION

Limiter RSSI Slope See Figure 9 23 – 0.15 0.3 0.4 µA/dB Limiter Gain f = 10.7 MHz 8, 12 – – 70 – dB Input Impedance 8 – – 330 – Ω

Condition Pin Symbol Min Typ Max Unit

See Figure 7

= 25°C; VS = 3.0 Vdc; fRF = 110.7 MHz; fLO = 100 MHz; See Figure 1.)

A

Figure 1. T est Circuit

Osc Emit

MC13158

Lim Out

12

LO Input

5.0 p

Base

13

100 MHz 200 mVrms

N/COsc

Data

Slicer

N/CQuad

14

100 k

1.0

µ

H

–3.0 Vdc

50

A

Det

Gain

15

V

EE1

Bias

2532 31 30 29 28 27 26

Enable

RSSI RSSI

Buf

DS

Gnd

DS Out

DS

In2

DS

“off”

DS

In1

Det

Out

V

EE2

16

A

51 k

24

23

22

21

20

19

18

17

–2.3 Vdc

A

0 to –3.0 Vdc

100

µ

A

–1.5 Vdc

0 to –3.0 Vdc

V

–3.0 Vdc

Mixer

Output

IF

Input

IF

Output

Limiter

Input

110.7 MHz

330

50

330

50

RF Input

1.0 n

100 n

1.0 n

100 n

1.0 n

100 n

1

2

3

4

5

6

7

8

1:4

Mix Out

V

CC1

IF In

IF Dec1

IF Dec2

IF Out

V

CC2

Lim In

Mix

In2

Lim

Dec1

9

200

Lim

Dec2

100 n

In1

Lim Amp

N/C

10

1.0 n1.0 n

N/CMix

11

MOTOROLA ANALOG IC DEVICE DATA

200 pF

6.8 k

–3.0 Vdc

3

MC13158

T ypical Performance Over Temperature

(per Figure 1)

6.4

6.2

6.0

5.8

5.6

5.4

5.2

, TOTAL SUPPLY CURRENT (mA)

5.0

TOTAL

I

4.8

0.12 Data Slicer “On”

µ

V19 = V

0.10

V20 = VS/2

0.08

0.06

0.04

DATA SLICER OUTPUT CURRENT ( A)

0.02

–40

Figure 2. T otal Supply Current versus

Ambient T emperature, Supply Voltage

VS = 6.0 V

3.0 V

2.0 V

DATA SLICER OUTPUT CURRENT (mA)

–20

0 20 40 60 80 100 120 0 20 40 60 80 100 120

TA, AMBIENT TEMPERATURE (°C)

Figure 4. Data Slicer On Output Current

versus Ambient T emperature

V18 > V

20

CC

– 0.1 – 0.2 – 0.3 – 0.4

NORMALIZED MIXER GAIN (dB)

– 0.5

– 20 0 20 40 60 80 100 120 – 20 0 20 40 60 80 100 120

TA, AMBIENT TEMPERATURE (°C)

– 0.6

Figure 3. Data Slicer On Output Current

versus Ambient T emperature

8.5 Data Slicer “On” V19 = V

8.0

7.5

7.0

6.5

6.0

5.5

5.0

EE

V20 = VS/2

–20

TA, AMBIENT TEMPERATURE (°C)

Figure 5. Normalized Mixer Gain

versus Ambient T emperature

0.2

0.1

0

–40

TA, AMBIENT TEMPERATURE (°C)

V18 < V

20

Vin = 1.0 mVrms VS = 3.0 Vdc fc = 110.7 MHz fLO = 100 MHz

Figure 6. Mixer RSSI Output Current versus

Ambient T emperature, Mixer Input Level

7.0

µ

6.0

5.0 VS = 3.0 Vdc

fc = 110.7 MHz fLO = 100 MHz

4.0

3.0

MIXER RSSI OUTPUT CURRENT ( A)

2.0

– 20 0 20 40 60 80 – 20 0 20 40 60 80 100 120100 120

–40

TA, AMBIENT TEMPERATURE (°C)

4

Vin = 10 mVrms

Vin = 1.0 mVrms

0.6 VS = 3.0 Vdc

0.4

f = 10.7 MHz Vin = 1.0 mVrms

0.2

0 – 0.2 – 0.4

NORMALIZED IF AMP GAIN (dB)

– 0.6 – 0.8

–40

Figure 7. Normalized IF Amp Gain

versus Ambient T emperature

TA, AMBIENT TEMPERATURE (°C)

MOTOROLA ANALOG IC DEVICE DATA

MC13158

tT ypical Performance Over Temperature

(per Figure 1)

Figure 8. IF Amp RSSI Output Current versus

Ambient T emperature, IF Input Level

10

µ

9.0

8.0

7.0 VS = 3.0 Vdc

6.0

f = 10.7 MHz

5.0

4.0

3.0

IF AMP RSSI OUTPUT CURRENT ( A)

2.0

– 20 0 20 40 60 80 – 20 0 20 40 60 80 100 120100 120

–40

TA, AMBIENT TEMPERATURE (°C)

Figure 10. Total RSSI Output Current versus

Ambient T emperature (No Signal)

0.60 VS = 3.0 Vdc

µ

No Input Signal

0.55

0.50

0.45

Vin = 10 mVrms

Vin = 1.0 mVrms

Figure 9. Limiter Amp RSSI Output Current

versus Ambient T emperature, Input Signal Level

8.0

µ

Vin = 100 mVrms

6.0

LIMITER AMP RSSI OUTPUT CURRENT ( A)

4.0

2.0

– 2.0

VS = 3.0 Vdc f = 10.7 MHz

0

–40

TA, AMBIENT TEMPERATURE (°C)

Vin = 10 mVrms

Vin = 1.0 mVrms

Vin = 100

Figure 11. Demodulator DC Voltage versus

Ambient Temperature

1.20

1.15

1.10

1.05

1.00

µ

Vrms

VS = 3.0 Vdc R17 = 51 k R15 = 100 k

0.40

TOTAL RSSI OUTPUT CURRENT ( A)

0.35

SYSTEM LEVEL AC ELECTRICAL CHARACTERISTICS (T

12 dB SINAD Sensitivity: fRF = 112 MHz 1 – dBm Narrowband Application f

Without Preamp Figure 25 –101 With Preamp Figure 26 –113

Third Order Intercept Point f

1.0 dB Comp. Point VS = 3.5 Vdc 1.0 dB C.Pt. –39

NOTES: 1. Test Circuit & Test Set per Figure 24.

– 20 0 20 40 60 80 100 120

–40

TA, AMBIENT TEMPERATURE (°C)

Characteristic

f

RF2

2.Test Circuit & Test Set per Figure 27.

A

Condition Notes Symbol Typ Unit

= 1.0 kHz

mod

= ±125 kHz

dev

SINAD Curve

= 112 MHz 2 IIP3 –32 dBm

RF1

= 112.1 MHz

Figure 28

0.95

DEMODULATOR OUTPUT DC VOLTAGE (Vdc)

0.90 –40

– 20 0 20 40 60 80 100 120

TA, AMBIENT TEMPERATURE (°C)

= 25°C; VS = 3.0 Vdc; fRF = 112 MHz; fLO = 122.7 MHz)

MOTOROLA ANALOG IC DEVICE DATA

5

MC13158

CIRCUIT DESCRIPTION

General

The MC13158 is a low power single conversion wideband FM receiver incorporating a split IF . This device is designated for use as the backend in digital FM systems such as Digital European Cordless Telephone (DECT) and wideband data links with data rates up to 2.0 Mbps. It contains a mixer, oscillator, Received Signal Strength Indicator (RSSI), IF amplifier, limiting IF, quadrature detector, power down or enable function, and a data slicer with output off function. Further details are covered in the Pin Function Description which shows the equivalent internal circuit and external circuit requirements.

Current Regulation/Enable

Temperature compensating voltage independent current regulators which are controlled by the enable pin (Pin 25) where “low” powers up and “high” powers down the entire circuit.

Mixer

The mixer is a double–balanced four quadrant multiplier and is designed to work up to 500 MHz. It can be used in differential or in single ended mode by connecting the other input to the positive supply rail. The linear gain of the mixer is approximately 22 dB at 100 mVrms LO drive level. The mixer gain and noise figure have been emphasized at the expense of intermodulation performance. RSSI measurements are added in the mixer to extend the range to higher signal levels. The single–ended parallel equivalent input impedance of the mixer is Rp ~ 1.0 kΩ and Cp ~ 2.0 pF. The buffered output of the mixer is internally loaded resulting in an output impedance of 330 Ω.

Local Oscillator

The on–chip transistor operates with crystal and LC resonant elements up to 220 MHz. Series resonant, overtone crystals are used to achieve excellent local oscillator stability . Third overtone crystals are used through about 65 to 70 MHz. Operation from 70 MHz up to 180 MHz is feasible using the on–chip transistor with a 5th or 7th overtone crystal. To enhance operation using an overtone crystal, the internal transistor bias is increased by adding an external resistor from Pin 29 to VEE; however, with an external resistor the oscillator stays on during power down. Typically, –10 dBm of local oscillator drive is needed to adequately drive the mixer. With an external oscillator source, the IC can be operated up to 500 MHz.

RSSI

The received signal strength indicator (RSSI) output is a current proportional to the log of the received signal amplitude. The RSSI current output is derived by summing the currents from the mixer, IF and limiting amplifier stages. An increase in RSSI dynamic range, particularly at higher input signal levels is achieved. The RSSI circuit is designed to provide typically 85 dB of dynamic range with temperature compensation.

Linearity of the RSSI is optimized by using external ceramic bandpass filters which have an insertion loss of

4.0 dB and 330 Ω source and load impedance. For higher data rates used in DECT and related applications, LC bandpass filtering is necessary to acquire the desired

bandpass response; however, the RSSI linearity will require the same insertion loss.

RSSI Buffer

The RSSI output current creates a voltage across an external resistor. A unity voltage–gain amplifier is used to buffer this voltage. The output of this buffer has an active pull–up but no pull–down, so it can also be used as a peak detector. The negative slew rate is determined by external capacitance and resistance to the negative supply .

IF Amplifier

The first IF amplifier section is composed of three differential stages with the second and third stages contributing to the RSSI. This section has internal DC feedback and external input decoupling for improved symmetry and stability. The total gain of the IF amplifier block is approximately 40 dB at 10.7 MHz.

The fixed internal input impedance is 330 Ω. When using ceramic filters requiring source and loss impedances of 330 Ω, no external matching is necessary. Overall RSSI linearity is dependent on having total midband attenuation of 10 dB (4.0 dB insertion loss plus 6.0 dB impedance matching loss) for the filter. The output of the IF amplifier is buffered and the impedance is 330 Ω.

Limiter

The limiter section is similar to the IF amplifier section except that five differential stages are used. The fixed internal input impedance is 330 Ω. The total gain of the limiting amplifier section is approximately 70 dB. This IF limiting amplifier section internally drives the quadrature detector section and it is also brought out on Pin 12.

Quadrature Detector

The quadrature detector is a doubly balanced four quadrant multiplier with an internal 5.0 pF quadrature capacitor between Pins 12 and 13. An external capacitor may be added between these pins to increase the IF signal to the external parallel RLC resonant circuit that provides the 90 degree phase shift and drives the quadrature detector. A single pin (Pin 13) provides for the external LC parallel resonant network and the internal connection to the quadrature detector.

Internal low pass filter capacitors have been selected to control the bandwidth of the detector. The recovered signal is brought out by the inverting amplifier buffer. An external feedback resistor from the output (Pin 17) to the input of the inverting amplifier (Pin 15) controls the output amplitude; it is combined with another external resistor from the input to the negative supply (Pin 16) to set the output dc level. For a resistor ratio of 1, the DC level at the detector output is

2.0 VBE (see Figure 12). A small capacitor C17 across the first resistor (from Pin 17 to 15) can be used to reduce the bandwidth.

Data Slicer

The data slicer is a comparator that is designed to square up the data signal. Across the data slicer inputs (Pins 18 and 20) are back to back diodes.

6

MOTOROLA ANALOG IC DEVICE DATA

MC13158

Á

Out

Á

Oscillator, and IF Amplifer. The operating

Á

The recovered data signal from the quadrature detector can be DC coupled to the data slicer DS IN1 (Pin 18). In the application circuit shown in Figure 1 it will be centered at

2.0 VBE and allowed to swing ± VBE. A capacitor is placed from DS IN2 (Pin 20) to VEE. The size of this capacitor and the nature of the data signal determine how faithfully the data slicer shapes up the recovered signal. The time constant is short for large peak to peak voltage swings or when there is a change in DC level at the detector output. For small signal or for continuous bits of the same polarity which drift close to the threshold voltage, the time constant is longer.

PIN FUNCTION DESCRIPTION

1

ÁÁ

2

ÁÁ

3

ÁÁ

4

Symbol

Mix Out

ÁÁÁ

V

CC1

ÁÁÁ

IF In

ÁÁÁ

IF

Internal Equivalent Circuit

26

2

V

CC1

EE1

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2

V

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CC1

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5

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IF Dec2

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330

Dec1

ÁÁ

5

ÁÁ

ÁÁÁ

IF

ÁÁÁ

Dec2

ÁÁÁ

БББББББББББББ

26

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V

EE1

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IF In

3

IF Dec1

A unique feature of the data slicer is that the inverting switching stages in the comparator are supplied through the emitter pin of the output transistor (Pin 22 – DS Gnd) to V rather than internally to VEE. This is provided in order to reduce switching feedback to the front end. A control pin is provided to shut the data slicer output off (DS “off” – Pin 19). With DS “off” pin at VCC the data slicer output is shut off by shutting down the base drive to the output transistor. When a channel is being monitored to make an RSSI measurement, but not to collect data, the data output may be shut off to save current.

Description/External Circuit Requirements

Mixer Output

The mixer output impedance is 330 Ω; it

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matches to 10.7 MHz ceramic filters with 330 Ω input impedance.

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Mix

1

Supply V oltage (V

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This pin is the VCC pin for the Mixer, Local

supply voltage range is from 1.8 Vdc to

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5.0 Vdc. In the PCB layout, the VCC trace

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must be kept as wide as possible to minimize inductive reactances along the trace; it is best

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to have it completely fill around the surface

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mount components and traces on the circuit side of the PCB.

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CC1

)

IF Input

The input impedance at Pin 3 is 330 Ω. It

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64 k 64 k

matches the 330 Ω load impedance of a

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10.7 MHz ceramic filter. Thus, no external matching is required.

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IF DEC1 & DEC2

IF decoupling pins. Decoupling capacitors should be placed directly at the pins to enhance

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stability . Two capacitors are decoupled to the

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RF ground V

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4

& DEC2.

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; one is placed between DEC1

CC1

EE

6

ÁÁ

MOTOROLA ANALOG IC DEVICE DATA

IF

ÁÁÁ

Out

ÁÁÁ

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2

V

CC1

26

V

EE1

IF

Out

IF Output

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The output impedance is 330 Ω; it matches

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the 330 input resistance of a 10.7 MHz ceramic filter.

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5

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7

Á

Limiter Decoupling

Á

Out

Á

7

ÁÁ

8

ÁÁ

9

ÁÁ

10

ÁÁ

1 1,14,

27 & 28

ÁÁ

12

ÁÁ

13

ÁÁ

15

ÁÁ

17

ÁÁ

16

ÁÁ

Symbol

V

CC2

ÁÁÁ

Lim

ÁÁÁ

In

ÁÁÁ

Lim

ÁÁÁ

Dec1

ÁÁÁ

Lim

ÁÁÁ

Dec2

ÁÁÁ

N/C

ÁÁÁ

Lim Out

ÁÁÁ

Quad

ÁÁÁ

Det

ÁÁÁ

Gain

ÁÁÁ

Det

ÁÁÁ

Out

ÁÁÁ

V

EE2

ÁÁÁ

MC13158

PIN FUNCTION DESCRIPTION (continued)

Internal Equivalent Circuit

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7 V

16

7 V

V

CC2

EE2

10

Dec2

16

V

CC2

Lim

EE2

V

15

16

7

CC2

Det

Gain

V

EE2

330 64 k

Lim In

8

Lim Dec1

Lim

Quad

Out

12

13

5.0 p

64 k

9

17

Det

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Description/External Circuit Requirements

Supply Voltage (V

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This pin is VCC supply for the Limiter, Quadrature Detector, data slicer and RSSI

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buffer circuits. In the application PC board this

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pin is tied to a common VCC trace with V

Limiter Input

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The limiter input impedance is 330 Ω.

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Limiter Decou

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Decoupling capacitors are placed directly at

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these pins and to VCC (RF ground). Use the

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same procedure as in the IF decoupling.

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lin

CC2

)

CC1

No Connects

There is no internal connection to these pins;

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however it is recommended that these pins be

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connected externally to VCC (RF ground).

Limiter Output

The output impedance is low. The limiter

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drives a quadrature detector circuit with in–

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phase and quadrature phase signals.

Quadrature Detector Circuit

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The quadrature detector is a doubly balanced

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four–quadrant multiplier with an internal 5.0 pF capacitor between Pins 12 and 13. An external

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capacitor may be added to increase the IF

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signal to Pin 13. The quadrature detector pin is provided to connect the external RLC parallel

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resonant network which provides the 90 degree

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phase shift and drives the quadrature detector.

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Detector Buffer Amplifier

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This is an inverting amplifier. An external feed-

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back resistor from Pin 17 to 15, (the inverting input) controls the output amplitude; another

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resistor from Pin 15 to the negative supply

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(Pin 16) sets the DC output level. A 1:1 resistor

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ratio sets the output DC level at two VBE with respect to VEE. A small capacitor from Pin 17 to

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15 can be used to set the bandwidth.

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Supply Ground (V

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In the PCB layout, the ground pins (also applies

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to Pin 26) should be connected directly to

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chassis ground. Decoupling capacitors to V should be placed directly at the ground pins.

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EE2

)

CC

.

8

MOTOROLA ANALOG IC DEVICE DATA

Á

CC2

Á

19

ÁÁ

21

ÁÁ

22

ÁÁ

18

ÁÁ

20

ÁÁ

23

ÁÁ

24

ÁÁ

Symbol

DS

ÁÁÁ

“off”

ÁÁÁ

DS Out

ÁÁÁ

DS

ÁÁÁ

Gnd

ÁÁÁ

DS

ÁÁÁ

In1

ÁÁÁ

DS

ÁÁÁ

In2

ÁÁÁ

RSSI

ÁÁÁ

Buf

ÁÁÁ

RSSI

ÁÁÁ

БББББББББББББ

V

БББББББББББББ

16

БББББББББББББ

7

БББББББББББББ

V

CC2

БББББББББББББ

DS In1

18

БББББББББББББ

16

V

EE2

БББББББББББББ

V

CC1

БББББББББББББ

MOTOROLA ANALOG IC DEVICE DATA

Internal Equivalent Circuit

7

CC2

V

EE2

V

2

24

RSSI

16

V

MC13158

PIN FUNCTION DESCRIPTION (continued)

Description/External Circuit Requirements

Data Slicer Off

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The data output may be shut off to save current by placing DS “off” (Pin 19) at VCC.

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Data Slicer Output

In the application example a 10 kΩ pull–up

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resistor is connected to the collector of the

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output transistor at Pin 21.

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Data Slicer Ground

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All the inverting switching stages in the

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comparator are supplied through the emitter pin of the output transistor (Pin 22) to ground

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rather than internally to VEE in order to reduce

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switching feedback to the front end.

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Data Slicer Inputs

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The data slicer has differential inputs with

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back to back diodes across them. The recovered signal is DC coupled to DS IN1

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(Pin 18) at nominally V18 with respect to VEE;

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thus, it will maintain V18 ± VBE at Pin 18. DS

ББББББББББББ

IN2 (Pin 20) is AC coupled to VEE. The choice of coupling capacitor is dependent on the

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nature of the data signal. For small signal or

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continuous bits of the same polarity , the response time is relatively large. On the other

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hand, for large peak to peak voltage swings or

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when the DC level at the detector output

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changes, the response time is short. See the discussion in the application section for

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external circuit design details.

RSSI Buffer

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A unity gain amplifier is used to buffer the

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voltage at Pin 24 to 23.The output of the unity gain buffer (Pin 23) has an active pull up but no

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pull down. An external resistor is placed from

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Pin 23 to VEE to provide the pull down.

ББББББББББББ

RSSI

ББББББББББББ

The RSSI output current creates a voltage

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drop across an external resistor from Pin 24 to VEE. The maximum RSSI current is 26 µA;

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thus, the maximum RSSI voltage using a

ББББББББББББ

100 kΩ resistor is approximately 2.6 Vdc. Fig-

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ure 22 shows the RSSI Output Voltage versus Input Signal Level in the application circuit.

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The negative slew rate is determined by an external capacitor and resistor to V

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(negative supply). The RSSI rise and fall times

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for various RF input signal levels and R

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values without the capacitor, C24 are displayed in Figure 24. This is the maximum response

ББББББББББББ

time of the RSSI.

CC2

EE2

DS Out

21

7

64 k

22

DS Gnd

19

DS “off”

DS In2

20

RSSI

Buf

23

EE

24

9

Á

p

Á

25

ÁÁ

Symbol

Enable

ÁÁÁ

MC13158

PIN FUNCTION DESCRIPTION (continued)

Internal Equivalent Circuit

БББББББББББББ

2 V

CC1

Description/External Circuit Requirements

Enable

ББББББББББББ

The IC regulators are enabled by placing this pin at VEE.

ББББББББББББ

ÁÁ

26

ÁÁ

28

ÁÁ

29

ÁÁ

31

ÁÁ

32

ÁÁ

10

ÁÁÁ

V

EE1

ÁÁÁ

Osc

Base

ÁÁÁ

Osc

Emitter

ÁÁÁ

Mix

In1

ÁÁÁ

Mix

In2

ÁÁÁ

2 V

CC1

28

Osc

Base

25

Enable

26

V

EE1

V

7

CC2

16

EE2

БББББББББББББ

2

V

CC1

БББББББББББББ

26

V

EE1

БББББББББББББ

29

26

V

26 V

EE1

Osc

Emitter

RF In2

32

БББББББББББББ

2

БББББББББББББ

V

CC1

БББББББББББББ

31

БББББББББББББ

RF

БББББББББББББ

In1

БББББББББББББ

ББББББББББББ

VCC and VEE ESD Protection

ESD protection diodes exist between the V

ББББББББББББ

and VEE pins. It is important to note that

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significant differences in potential (> 0.5 VBE) between the two VCC pins or between the V

ББББББББББББ

pins can cause these structures to start to

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conduct, thus compromising isolation between the supply busses. V

ББББББББББББ

maintained at the same DC potential, as

ББББББББББББ

should V

ББББББББББББ

EE1

& V

EE2

CC1

.

& V

CC2

should be

CC

Oscillator Base

This pin is connected to the base lead of the

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common collector transistor. Since there is no

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internal bias resistor to the base, VCC is applied through an external choke or coil.

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Oscillator Emitter

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This pin is connected to the emitter lead; the emitter is connected internally to a current source of about 200 µA. Additional emitter

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current may be obtained by connecting an

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external resistor to VEE; IE = V29/R29.

ББББББББББББ

Details of circuits using overtone crystal and

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LC varactor controlled oscillators are discussed in the application section.

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Mixer Inputs

The parallel equivalent differential input

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impedance of the mixer is approximately 2.0

ББББББББББББ

kΩ in parallel with 1.0 pF. This equates to a single ended input impedance of 1.0 kΩ in

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parallel with 2.0 pF.

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The application circuit utilizes a SAW filter having a differential output that requires a

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2.0 kΩ II 2.0 pF load. Therefore, little matching

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is required between the SAW filter and the

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mixer inputs. This and alternative circuits are discussed in more detail in the application

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section.

ББББББББББББ

MOTOROLA ANALOG IC DEVICE DATA

EE

MC13158

APPLICATIONS INFORMATION

Evaluation PC Board

The evaluation PCB is very versatile and is intended to be used across the entire useful frequency range of this device. The center section of the board provides an area for attaching all SMT components to the circuit side and radial leaded components to the component ground side (see Figures 29 and 30). Additionally, the peripheral area surrounding the RF core provides pads to add supporting and interface circuitry as a particular application dictates. This evaluation board will be discussed and referenced in this section.

Component Selection

The evaluation PC board is designed to accommodate specific components, while also being versatile enough to use components from various manufacturers and coil types. Figures 13 and 14 show the placement for the components specified in the application circuit (Figure 12). The application circuit schematic specifies particular components that were used to achieve the results shown in the typical curves and tables but alternate components should give similar results.

MOTOROLA ANALOG IC DEVICE DATA

11

MC13158

Figure 12. Application Circuit

(4) 122.7 MHz 5th OT Crystal

33 p

150

100 n

150

VCC =

2.0 to 5.0 Vdc

RF Input

112 MHz

(2) LCR Filter

680 p

330 nH

1.0 n

330 nH

680 p(2)

SMA

100 n

(1)

1

2

3

4

5

6

7

8

Saw

Filter

Mixer

V

CC1

V

CC2

IF Amp

100 n

(6) 0.68

Lim Amp

109

1.0 n1.0 n

µ

H

N/C

27 p

33

MC13158

Quad

Detector

5.0 p

1211

39 p

13

(5) 95 nH

4.7 k

N/CN/C

N/C

100 p

EE1

Bias

1514

EnableV

V

2.2 k

10 n

2532 31 30 29 28 27 26

EE2 16

(7)

Enable

24

23

22

21

20

19

18

17

82 k

R

15

C

RSSI

Out

17

10 k

1.0 k

DS In1

100 k

10 n

C

20

R 82 k

1.0 n

17

1.0

DS Out DS In2

DS “off”

µ

NOTES: 1. Saw Filter – Siemens part number Y6970M(5 pin SIP plastic package).

(3) LCR Quad Tank

2.An LCR filter reduces the broadband noise in the IF; ceramic filters may be used for data rates under 500 kHz. 4.0 dB insertion loss filters optimize the linearity of RSSI.

3.The quadrature tank components are chosen to optimize linearity of the recovered signal while maintaining adequate recovered signal level. 1.5 µH 7.0 mm variable shielded inductor: T oko part # 292SNS–T1373Z. The shunt resistor is approximately equal to Q(2πfL), where Q∼ 18 (3.0 dB BW = 600 kHz).

4.The local oscillator circuit utilizes a 122.7 MHz, 5th overtone, series resonant crystal specified with a frequency tolerance of 25 PPM, ESR of 120 Ω max. The oscillator configuration is an emitter coupled butler.

5.The 95 NH (Nominal) inductor is a 7.0 mm variable shielded inductor: Coilcraft part # 150–04J08S or equivalent.

6.0.68 µH axial lead chokes (molded inductor ): Coilcraft part # 90–11.

7.To enable the IC, Pin 25 is taken to VEE. The external pull down resistor at Pin 29 could be linked to the enable function; otherwise if it is taken to VEE as shown, it will keep the oscillator biased at about 500 µA depending on the VCC level.

8.The other resistors and capacitors are surface mount components.

12

1.5

µ

H

MOTOROLA ANALOG IC DEVICE DATA

MC13158

Figure 13. Circuit Side Component Placement

100n

1n

C17

82k

10n

10k

27p

33p

1.0k

C20

100n

150

330nH

100n

330nH

680p

1n 1n

680p

100n

1n

MC13158FB

100n

1n

100p

39p

33

47k

82k

2.2k

+

1µ

100n

V

CC

MOTOROLA ANALOG IC DEVICE DATA

13

MC13158

Figure 14. Ground Side Component Placement

V

EE

10.7 P

CERAMIC

FILTER

10.7 P

CERAMIC

FILTER

10.7 S CERAMIC FILTER

SAW

FILTER

QUAD

1.5 µH

COIL

XTAL

122.7 MHz

0.68 µH

LO

95 pH

V

CC

DS OFF

DS OPEN/ IN2

DS OUT

RSSI OUT

14

RF INPUT

SMA

MC13158

MOTOROLA ANALOG IC DEVICE DATA

MC13158

Á

ББББББ

Á

Input Matching/Components

It is desirable to use a SAW filter before the mixer to provide additional selectivity and adjacent channel rejection. In a wideband system the primary sensitivity of the receiver backend may be achieved before the last mixer. Bandpass filtering in the limiting IF is costly and difficult to achieve for bandwidths greater than 280 kHz.

The SAW filter should be selected to easily interface with the mixer differential input impedance of approximately

2.0 kΩ in parallel with 1.0 pF. The PC board is dedicated to the Siemens SAW filter (part number Y6970M); the part is designed for DECT at 112 MHz 1st IF frequency. It is designed for a load impedance of 2.0 kΩ in parallel with

Figure 15. Mixer Input Impedance

(Single–ended)

2.0 pF; thus, no or little input matching is required between the SAW filter and the mixer.

The Siemens SAW filter has an insertion loss of typically 10 dB and a 3.0 dB bandwidth of 1.0 MHz. The relatively high insertion loss significantly contributes to the system noise and a filter having lower insertion loss would be desirable. In existing low loss SAW filters, the required load impedance is 50 Ω; thus, interface matching between the filter and the mixer will be required. Figure 15 is a table of the single–ended mixer input impedance. A careful noise analysis is necessary to determine the secondary contribution to system noise.

БББББ

f

(MHz)

50 100 150 200

БББББ

250 300 400 500 600

Rs

ÁÁÁ

(Ω)

930 480 270 170 130

ÁÁÁ

110

ÁÁÁ

71

ÁÁÁ

63 49

Xs

БББББ

(Ω)

–350 –430 –400 –320 –270

БББББ

–250

БББББ

–190

БББББ

–140 –1 10

System Noise Considerations

The system block diagram in Figure 16 shows the cascaded noise stages contributing to the system noise; it represents the application circuit in Figure 12 and a low noise preamp using a MRF941 transistor (see Figure 17). The preamp is designed for a conjugately matched input and output at 2.0 Vdc VCE and 3.0 mAdc Ic. S–parameters at

2.0 V, 3.0 mA and 100 MHz are:

S1 1 = 0.86, –20 S21 = 9.0, 164 S12 = 0.02, 79 S22 = 0.96, –12

The bias network sets VCE at 2.0 V and Ic at 3.0 mA for VCC = 3.0 to 3.5 Vdc. The preamp operates with 18 dB gain and 2.7 dB noise figure.

In the cascaded noise analysis the system noise equation is:

Fsystem+F1

)

(

[

F2–1)ń

G1])

(

[

F3–1

)

]

(

[

ń

G1)(G2

where:

F1 = the Noise Factor of the Preamp G1 = the Gain of the Preamp F2 = the Noise factor of the SAW Filter G2 = the Gain of the SAW Filter F3 = the Noise factor of the Mixer

Rp

БББББ

(Ω)

1060

865 860 770 690

БББББ

680

БББББ

580

БББББ

370 300

БББББ

Xp (Ω)

–2820

–966 –580 –410 –330 –300 –220 –170 –130

БББББ

Note: the proceeding terms are defined as linear relationships and are related to the log form for gain and noise figure by the following:

F+log–1[(NF in dB)ń10]

and similarly

G+log–1[(Gain in dB)ń10]

The noise figure and gain measured in dB are shown in the system block diagram. The mixer noise figure is typically 14 dB and the SAW filter adds typically 10 dB insertion loss. Addition of a low noise preamp having a 18 dB gain and

2.7 dB noise figure not only improves the system noise figure but it increases the reverse isolation from the local oscillator to the antenna input at the receiver. Calculating in terms of gain and noise factor yields the following:

F1+1.86; G1+63.1 F2+10; G2+0.1

)

]

F3+25.12

Thus, substituting in the equation for system noise factor:

Fsystem+5.82; NFsystem+7.7 dB

Cp

(pF)

1.1

1.6

1.8

1.9

1.85

1.8

1.9

2.0

MOTOROLA ANALOG IC DEVICE DATA

15

MC13158

Figure 16. System Block Diagram for Noise Analysis

Noise

Source

fRF = 112 MHz

LNA

G1 = 18 dB NF1 = 2.7 dB

100 n

15 k

8.2 k

RF

Input

100 p

SAWF

G2 = 10 dB NF2 = 10 dB

Local Oscillator

fLO = 122.7 MHz

Figure 17. 112 MHz LNA

3.5 Vdc

510

MPS3906

1.0 k

100 nH

1.0 k

100 p

Mixer

fIF = 10.7 MHz

G3 = 18 dB

NF3 = 14 dB

680 nH

FB

MRF941

100 nH

270

100 p

47

330 nH

NF

Meter

150 p

RF Output

LOCAL OSCILLATORS

VHF Applications

The on–chip grounded collector transistor may be used for HF and VHF local oscillator with higher order overtone crystals. The local oscillator in the application circuit (Figure 12) shows a 5th overtone oscillator at 122.7 MHz. This circuit uses a Butler overtone oscillator configuration. The amplifier is an emitter follower. The crystal is driven from the emitter and is coupled to the high impedance base through a capacitive tap network. Operation at the desired overtone frequency is ensured by the parallel resonant circuit formed by the variable inductor and the tap capacitors and parasitic capacitances of the on–chip transistor and PC board. The variable inductor specified in the schematic could be replaced with a high tolerance, high Q ceramic or air wound surface mount component if the other components have tight enough tolerances. A variable inductor provides an adjustment for gain and frequency of the resonant tank ensuring lock up and start–up of the crystal oscillator. The overtone crystal is chosen with ESR of typically 80 Ω and 120 Ω maximum; if the resistive loss in the crystal is too high the performance of oscillator may be impacted by lower gain margins.

A series LC network to ground (which is VCC) is comprised of the inductance of the base lead of the on–chip transistor and PC board traces and tap capacitors. Parasitic oscillations often occur in the 200 to 800 MHz range. A small resistor is placed in series with the base (Pin 28) to cancel the

negative resistance associated with this undesired mode of oscillation. Since the base input impedance is so large a small resistor in the range of 27 to 68 Ω has very little effect on the desired Butler mode of oscillation.

The crystal parallel capacitance, Co, provides a feedback path that is low enough in reactance at frequencies of 5th overtones or higher to cause trouble. Co has little effect near resonance because of the low impedance of the crystal motional arm (Rm–Lm–Cm). As the tunable inductor which forms the resonant tank with the tap capacitors is tuned “off” the crystal resonant frequency it may be difficult to tell if the oscillation is under crystal control. Frequency jumps may occur as the inductor is tuned. In order to eliminate this behavior an inductor, Lo, is placed in parallel with the crystal. Lo is chosen to be resonant with the crystal parallel capacitance, Co, at the desired operation frequency. The inductor provides a feedback path at frequencies well below resonance; however, the parallel tank network of the tap capacitors and tunable inductor prevent oscillation at these frequencies.

IF Filtering/Matching

In wideband data systems the IF bandpass needed is greater than can be found in low cost ceramic filters operating at 10.7 MHz. It is necessary to bandpass limit with LC networks or series–parallel ceramic filter networks. Murata offers a series–parallel resonator pair (part number

16

MOTOROLA ANALOG IC DEVICE DATA

MC13158

KMFC545) with a 3.0 dB bandwidth of ± 325 kHz and a maximum insertion loss of 5.0 dB. The application PC board is laid out to accommodate this filter pair (a filter pair is used at both locations of the split IF). However, even using a series parallel ceramic filter network yields only a maximum bandpass of 650 kHz. In some applications a wider band IF bandpass is necessary.

A simple LC network yields a bandpass wider than the SAW filter but it does reduce an appreciable amount of wideband IF noise. In the application circuit an LC network is specified using surface mount components. The parallel LC components are placed from the outputs of the mixer and IF amplifier to the VCC trace; internal 330 loads are connected from the mixer and IF amplifier outputs to DEC2 (Pin 5 and 10 respectively).This loads the outputs with the optimal load impedance but creates a low insertion loss filter. An external shunt resistor may be used to widen the bandpass and to acquire the 10 dB composite loss necessary to linearize the RSSI output. The equivalent circuit is shown in Figure 18.

Figure 18. IF LCR Filter

R

out

1, 6

150

V

330 nH

CC

680 p

DEC1

DEC2

2, 7

3, 8

4, 9

5, 10

The following equations satisfy the 12 dB loss (1:4 resistive ratio):

(Rext)(330)ń(Rext)330)+Requivalent Requivalentń(Requivalent)330)+1ń4

Solve for Requivalent:

4(Requivalent)+Requivalent)330 3(Requivalent)+330 Requivalent+110

Substitute for Requivalent and solve for Rext:

330(Rext)+110(Rext))(330)(110) Rext+(330)(110)ń220 Rext+165

W

The IF is 10.7 MHz although any IF between 10 to 20 MHz could be used. The value of the coil is lowered from that used in the quadrature circuit because the unloaded Q must be maintained in a surface mount component. A standard value component having an unloaded Q = 100 at 10.7 MHz is 330 nH; therefore the capacitor is 669 pF. Standard values have been chosen for these components;

Rext+150

W

C+680 pF L+330 nH

330

V

CC

R

in

330

Computation of the loaded Q of this LCR network is

Q+RequivalentńX

L

where: XL = 2πfL and Requivalent is 103 Ω

Thus, Q+4.65

The total system loss is

20 log (103ń433)+–12.5 dB

Quadrature Detector

The quadrature detector is coupled to the IF with an internal 5.0 pF capacitor between Pins 12 and 13. For wideband data applications, the drive to the detector can be increased with an additional external capacitor between these pins; thus, the recovered signal level output is increased for a given bandwidth

The wideband performance of the detector is controlled by the loaded Q of the LC tank circuit. The following equation defines the components which set the detector circuit’s bandwidth:

Q+RTń

X

L

[1]

where RT is the equivalent shunt resistance across the LC Tank XL is the reactance of the quadrature inductor at the IF frequency (XL = 2πfL).

The inductor and capacitor are chosen to form a resonant LC tank with the PCB and parasitic device capacitance at the desired IF center frequency as predicted by

fc+[2p(LCp)

1ń2

]

[2]

–1

where L is the parallel tank inductor Cp is the equivalent parallel capacitance of the parallel resonant tank circuit.

The following is a design example for a wideband detector at 10.7 MHz and a loaded Q of 18. The loaded Q of the quadrature detector is chosen somewhat less than the Q of the IF bandpass. For an IF frequency of 10.7 MHz and an IF bandpass of 600 kHz, the IF bandpass Q is approximately

6.4.

Example:

Let the external Cext = 139 pF. (The minimum value here should be much greater than the internal device and PCB parasitic capacitance, Cint ≈ 3.0 pF). Thus, Cp = Cint + Cext = 142 pF.

Rewrite equation (2) and solve for L:

L = (0.159)2/(Cpfc2) L = 1.56 µH; Thus, a standard value is

choosen:

L = 1.56 µH (tunable shielded inductor)

The value of the total damping resistor to obtain the required loaded Q of 18 can be calculated by rearranging equation (1):

RT+

Q(2pfL)

RT+

18(2p)(10.7)(1.5)+1815

W

MOTOROLA ANALOG IC DEVICE DATA

17

MC13158

The internal resistance, Rint at the quadrature tank Pin 13 is approximately 13 kΩ and is considered in determining the external resistance, Rext which is calculated from

Rext

+((R

Rext+2110; Rext+2.2 k

)(

Rint))ń(Rint – R

T

Thus, choose the standard value:

W

It is important to set the DC level of the detector output at Pin 17 to center the peak to peak swing of the recovered signal. In the equivalent internal circuit shown in the Pin Function Description, the reference voltage at the positive terminal of the inverting op amp buffer amplifier is set at

1.0 VBE. The detector DC level, V17 is determined by the following equation:

V17+

[((

R15ń

)

R

)

1)ń(R15ń

17

Thus, for a 1:1 ratio of R15/R17, V17 = 2.0 VBE = 1.4 Vdc. Similarly for a 2:1, V17 = 1.5 VBE = 1.05 Vdc; and for 3:1, V

= 1.33 VBE = 0.93 Vdc.

17

Figure 19 shows the detector “S–Curves”, in which the resistor ratio is varied while maintaining a constant gain (R is held at 62 k). R15 is 62 k for a 1:1 ratio; while R15 = 120 k and 180 k to produce the 2:1 and 3:1 ratios. The IF signal into the detector is swept ± 500 kHz about the 10.7 MHz IF center frequency. The resulting curve show how the resistor ratio and the supply voltage effects the symmetry of the “S–curve” (Figure 21 Test Setup). For the 3:1 and 2:1 ratio, symmetry is maintained with VS from 2.0 to 5.0 Vdc; however, for the 1:1 ratio, symmetry is lost at 2.0 Vdc

)

T

)

R

]V

17

BE

17

.

Data Slicer Circuit

C20 at the input of the data slicer is chosen to maintain a time constant long enough to hold the charge on the capacitor for the longest strings of bits at the same polarity. For a data rate at 576 kHz a bit stream of 15 bits at the same polarity would equate to an apparent data rate of approximately 77 kbps or 38 kHz. The time constant would be approximately 26 µs. The following expression equates the time constant, t, to the external components:

t+2p(R18)(C20)

Solve for C20:

C20+tń2p

(R18)

where the effective resistance R18 is a complex function of the demodulator feedback resistance and the data slicer input circuit. In the data input network the back to back diodes form a charge and discharge path for the capacitor at Pin 20; however, the diodes create a non–linear response. This resistance is loaded by the ß, beta of the detector output transistor; beta =100 is a typical value (see Figure 21). Thus, the apparent value of the resistance at Pin 18 (DS IN1) is approximately equal to:

R18Y

R17ń

100

where R17 is 82 kΩ, the feedback resistor from Pin 17 to 15. Therefore, substituting for R18 and solving for C20:

C20+

15.9 (t)ńR17+

5.04 nF

Figure 19. Detector Output Voltage versus

Frequency Deviation

2.5

R15:R17 = 1:1 VS = 2.0 Vdc

17

2.0

1.5 R15:R17 = 2:1

VS = 2.0 to 5.0 Vdc

1.0

R15:R17 = 3:1 VS = 2.0 to 5.0 Vdc

0.5

DETECTOR OUTPUT VOLTAGE, V (Vdc)

0

– 600

– 400 – 200 0 200 400 600

FREQUENCY DEVIA TION (kHz)

R15:R17 = 1:1 VS = 3.5 to 5.0 Vdc

fc = 10.7 MHz R17 = 62 k Test Setup – Figure 20

Figure 20. Demodulator “S–Curve” Test Setup

Wavetek Signal

Generator

Model 134

Sweep Out

50

Ω

Output

EXT

MOD In

Signal Generator

Fluke 6082A

fc = 10.7 MHz

∆

f = ±500 kHz

RF Out

The closest standard value is 4.7 nF.

Figure 21. Data Slicer Equivalent Input Circuit

R

18

R

17/

β

C

20

V

CC

18

20

18

X Input

Oscilloscope

TEK 475

Y Input

DET

Out

Lim In

MC13158

MOTOROLA ANALOG IC DEVICE DATA

MC13158

SYSTEM PERFORMANCE DATA

RSSI

In Figure 22, the RSSI versus RF Input Level shows the linear response of RSSI over a 65 dB range but it has extended capability over 80 dB from – 80 dBm to +10 dBm. The RSSI is measured in the application circuit (Figure 12) in which a SAW filter is used before the mixer; thus, the overall sensitivity is compromised for the sake of selectivity. The curves are shown for three filters having different bandwidths:

1) LCR Filter with 2.3 MHz 3.0 dB BW (Circuit and

Component Placement is shown in Figure 12)

2) Series–Parallel Ceramic Filter with 650 kHz 3.0 dB BW (Murata Part # KMFC–545)

3) Ceramic Filter with 280 kHz 3.0 dB BW.

Figure 22. RSSI Output Voltage versus

Signal Input Level

3.0 VCC = 4.0 Vdc

2.7

fRF = 112 MHz fLO = 122.7 MHz

2.4 fIF = 10.7 MHz

2.1

See Figure 12 for LCR filter

1.8

Series–Parallel

1.5

Ceramic Filter

1.2

0.9

RSSI OUTPUT VOLTAGE (Vdc)

0.6

0.3

0

–90

LCR; Rext = 150

–80 –70 –60 –50 –40 –30 –20 20–10 100

SIGNAL INPUT LEVEL (dBm)

Ceramic Filter

Ω

Figure 23. RSSI Output Rise and Fall Times

versus RF Input Signal Level

35

µ

30

rf

25 20 15 10

5.0

RSSI RISE AND FALL TIMES, t & t ( s)

0

0 – 20 – 60 – 80

RF INPUT SIGNAL LEVEL (dBm)

–40

tr @ 22 k tf @ 22 k tr @ 47 k

tf @ 47 k tr @ 100 k tf @ 100 k

SINAD Performance

Figure 24 shows a test setup for a narrowband demodulator output response in which a C–message filter and an active de–emphasis filter is used following the demodulator. The input is matched using a 1:4 impedance transformer. The SINAD performance is shown in Figure 25 with no preamp and in Figure 26 with a preamp (Preamp – Figure 16). The 12 dB SINAD sensitivity is –101 dBm with no preamp and –113 dBm with the preamp.

Figure 24. Test Setup for Narrowband SINAD

HP8657B

fc = 112 MHz

f

= 1.0 kHz

mod

∆

f = ±125 kHz

HP8657B

fc = 122.7 MHz

PLO = –6.0 dBm

LO Output

Input

Match

LO

HP334

Distortion

Analyzer

N+D N

MC13158 IF 3.0 dB BW = 280 kHz

Detector Out

In

C–Message

Filter

Active

De–emphasis

RF

Voltmeter

MOTOROLA ANALOG IC DEVICE DATA

19

MC13158

versus Input Signal Level (without preamp)

10

0

–10 –20 –30 –40

S+N+D, N+D, N (dB)

–50 –60

–70

50

–120

–100 – 80 – 60 – 40 – 20 0

Mini–Circuits ZSFC–4

4 Way Zero Degree

Figure 25. S+N+D, N+D, N

S+N+D

VS = 3.0 Vdc

±

125 kHz

f

=

dev

f

= 1.0 kHz

mod

fRF = 112 MHz

N+D

N

RF INPUT SIGNAL (dBm)

IF 3.0 dB BW = 280 kHz

Figure 27. Input IP3, 1.0 dB Compression Pt. T est Setup

112 MHz

100 p

Combiner

0.8–10 p

Figure 26. S+N+D, N+D, N versus

Input Signal Level (with preamp)

10

0

–10 –20 –30 –40

S+N+D, N+D, N (dB)

–50 –60

100 p

–70

–120

–100 – 80 – 60 – 40 – 20 0

MIXER

G3 = 18 dB NF3 = 14 dB

S+N+D

VS = 3.0 Vdc f

=

dev

f

= 1.0 kHz

mod

fRF = 112 MHz

N +D

N

RF INPUT SIGNAL (dBm)

270

47

IF 3.0 dB BW = 280 kHz

±

125 kHz

FET Probe TEK P6201

To Spectrum Analyzer

50

112.1 MHz

–10 –20 –30 –40 –50

S+N+D, N+D, N (dB)

–60

–70 –80

Local Oscillator HP8657B

Figure 28. –1.0 dB Compression Pt. and Input

Third Order Intercept

1.0 dB Comp. Pt. = –39 dBm IP3 = –32 dBm

VS = 3.5 Vdc f

= 112 kHz

RF1

f

= 112.1 kHz

RF2

fLO = 122.7 MHz PLO = –6.0 dBm See Figure 27

–60

–50

RF INPUT SIGNAL LEVEL (dBm)

–40 –30 –20

fLO –

122.7 MHz @ –6.0 dBm

20

MOTOROLA ANALOG IC DEVICE DATA

MC13158

Figure 29. Circuit Side View

V

CC

3.8

″

MOTOROLA ANALOG IC DEVICE DATA

21

MC13158

Figure 30. Ground Side View

V

EE

10.7 P

CERAMIC

FILTER

10.7 P

CERAMIC

FILTER

10.7 S CERAMIC FILTER

SAW

FILTER

QUAD

COIL

XTAL

LO

V

CC

DS OFF

DS OPEN/ IN2

DS OUT

RSSI OUT

22

RF INPUT

MC13158

MOTOROLA ANALOG IC DEVICE DATA

L

24 17

25

MC13158

OUTLINE DIMENSIONS

FTB SUFFIX

PLASTIC PACKAGE

CASE 873–01

(Thin QFP)

16

L

–C–

SEATING PLANE

–A–

C

S S

–B–

B

V

M

B

0.20 (0.008) C A–B D

DETAIL A

32

9

81

0.05 (0.002) A–B

–D–

A

0.20 (0.008) A–B D

0.05 (0.002)

M

A–B

S S

C

S

0.20 (0.008) A–B D

M

E

H

G

S S

H

M

DETAIL C

DATUM

–H–

PLANE

0.01 (0.004)

0.20 (0.008) H A–B D

BASE METAL

B

–A–,–B–,–D–

DETAIL A

F

J

N

D

0.20 (0.008) A–B D

M

S S

C

P

SECTION B–B

VIEW ROTATED 90° CLOCKWISE

–H–

DATUM PLANE

K

X

DETAIL C

MOTOROLA ANALOG IC DEVICE DATA

U

NOTES:

T

R

Q

1. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982.

2. CONTROLLING DIMENSION: MILLIMETER.

3. DATUM PLANE –H– IS LOCATED AT BOTTOM OF LEAD AND IS COINCIDENT WITH THE LEAD WHERE THE LEAD EXITS THE PLASTIC BODY AT THE BOTTOM OF THE PARTING LINE.

4. DATUMS –A–, –B– AND –D– TO BE DETERMINED AT DATUM PLANE –H–.

5. DIMENSIONS S AND V TO BE DETERMINED AT SEATING PLANE –C–.

6. DIMENSIONS A AND B DO NOT INCLUDE MOLD PROTRUSION. ALLOWABLE PROTRUSION IS 0.25 (0.010) PER SIDE. DIMENSIONS A AND B DO INCLUDE MOLD MISMATCH AND ARE DETERMINED AT DATUM PLANE –H–.

7. DIMENSION D DOES NOT INCLUDE DAMBAR PROTRUSION. ALLOWABLE DAMBAR PROTRUSION SHALL BE 0.08 (0.003) TOTAL IN EXCESS OF THE D DIMENSION AT MAXIMUM MATERIAL CONDITION. DAMBAR CANNOT BE LOCATED ON THE LOWER RADIUS OR THE FOOT.

MILLIMETERS INCHES

MIN MINMAX MAX

DIM

A

B C D

E

F G H

J

K

L M N

P Q R

S

T U

V

X

7.10

6.95

7.10

6.95

1.60

1.40

0.373

0.273

1.50

1.30 –

0.273

0.80 BSC

0.20

–

0.197

0.119

0.57

0.33

5.6 REF 8

6

°

0.135

0.119

0.40 BSC

°

10

5

0.25

0.15

9.15

8.85

0.25

0.15 11

5

°

9.15

8.85

1.0 REF 0.039 REF

°

0.274

0.055

0.010

0.051

0.010

0.031 BSC

–

0.005

0.013

0.220 REF

6

°

0.005

0.016 BSC

°

5

0.006

0.348

0.006 5

°

0.348

0.280

0.063

0.015

0.059 –

0.008

0.022

8

°

0.005

°

10

0.010

0.360

0.010

11

°

0.360

23

MC13158

NOTES

Motorola reserves the right to make changes without further notice to any products herein. Motorola makes no warranty , representation or guarantee regarding the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation consequential or incidental damages. “T ypical” parameters which may be provided in Motorola data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including “Typicals” must be validated for each customer application by customer’s technical experts. Motorola does not convey any license under its patent rights nor the rights of others. Motorola products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the Motorola product could create a situation where personal injury or death may occur. Should Buyer purchase or use Motorola products for any such unintended or unauthorized application, Buyer shall indemnify and hold Motorola and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that Motorola was negligent regarding the design or manufacture of the part. Motorola and are registered trademarks of Motorola, Inc. Motorola, Inc. is an Equal Opportunity/Affirmative Action Employer.

How to reach us: USA/EUROPE/Locations Not Listed: Motorola Literature Distribution; JAPAN: Nippon Motorola Ltd.; T atsumi–SPD–JLDC, 6F Seibu–Butsuryu–Center,

P.O. Box 20912; Phoenix, Arizona 85036. 1–800–441–2447 or 602–303–5454 3–14–2 Tatsumi Koto–Ku, Tokyo 135, Japan. 03–81–3521–8315

MFAX: RMF AX0@email.sps.mot.com – TOUCHT ONE 602–244–6609 ASIA/PACIFIC: Motorola Semiconductors H.K. Ltd.; 8B Tai Ping Industrial Park, INTERNET: http://Design–NET.com 51 Ting Kok Road, Tai Po, N.T., Hong Kong. 852–26629298

24

◊

MOTOROLA ANALOG IC DEVICE DATA

MC13158/D

*MC13158/D*

Datasheet MC13158FTB Datasheet (Motorola)

Specifications and Main Features

Frequently Asked Questions

User Manual