Datasheet MAX17020 Datasheet (Maxim Integrated)

Page 1

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

________________________________________________________________

Maxim Integrated Products

Ordering Information

19-4118; Rev 2; 2/09

General Description

The MAX17020 is a dual Quick-PWM™ step-down power-supply (SMPS) controller with synchronous rectification, intended for main 5V/3.3V or I/O 1.5V/1.05V power generation in battery-powered systems. Lowside MOSFET sensing provides a simple low-cost, highly efficient current sense for valley current-limit protection. Combined with the output overvoltage and undervoltage protection features, this current limit ensures robust output supplies.

The 5V/3.3V or 1.5V/1.05V SMPS outputs can save power by operating in pulse-skipping mode or in ultrasonic mode to avoid audible noise. Ultrasonic mode forces the controller to maintain switching frequencies greater than 20kHz at light loads.

An internal 100mA linear regulator can be used to either generate the 5V bias needed for power-up or other lower power “always-on” suspend supplies. An independent bypass input allows automatic bypassing of the linear regulator when the SMPS is active.

This main controller also includes a secondary feedback input that triggers an ultrasonic pulse (DL1 turned on) if the SECFB voltage drops below its threshold voltage. This refreshes an external charge pump driven by DL1 without overcharging the output voltage.

The device includes independent shutdown controls to simplify power-up and power-down sequencing. To prevent current surges at startup, the internal voltage target is slowly ramped up from zero to the final target over a 1ms period. To prevent the output from ringing below ground in shutdown, the internal voltage target is ramped down from its previous value to zero over a 1ms period. Two independent power-good outputs simplify the interface with external controllers.

The MAX17020 is a pin-for-pin replacement of the MAX8778.

Features

Applications

Denotes a lead(Pb)-free/RoHS-compliant package.

EP = Exposed pad.

Dual Quick-PWM Internal 100mA 5V or Adjustable Linear Regulator Independent LDO Bypass Input Internal Boost Diodes Secondary Feedback Input Maintains Charge Pump

3.3V 5mA RTC Power (Always On) OUT1: 5V or 1.5V Fixed or 0.7V Adjustable

Feedback OUT2: 3.3V or 1.05V Fixed or Dynamic Adjustable Dynamic 0V to 2V REFIN2 Input on Second SMPS 2V ±1% 50µA Reference 6V to 24V Input Range (28V max) Ultrasonic Mode Independent SMPS and LDO Enable Controls Independent SMPS Power-Good Outputs Minimal Component Count

Notebook Computers

Main System Supply (5V and 3.3V Supplies)

I/O System Supply (1.5V and 1.05V Supplies)

Graphic Cards

DDR1, DDR2, DDR3 Power Supplies

Game Consoles

Low-Power I/O and Chipset Supplies

Two-to-Four Li+ Cell Battery-Powered Devices

PDAs and Mobile Communicators

Telecommunication

MAX17020

THIN QFN (T3255-4)

5mm x 5mm

TOP VIEW

A "+" SIGN FIRST-PIN INDICATOR DENOTES A LEAD-FREE PACKAGE.

TON

ONLDO

RTC

LDO

REF

DL2

AGND

SECFB

BST2

DL1

PGOOD2

4567

2324 22 20 19 18

SKIP

OUT2

ON1

PGOOD1

ILIM1

FB1

PGND

ILIM2

OUT1

REFIN2

BYP

ON2

DH1

DH2

LX1

LDOREFIN

BST1

LX2

Pin Configuration

Quick-PWM is a trademark of Maxim Integrated Products, Inc.

EVALUATION KIT

AVAILABLE

PART TEMP RANGE PIN-PACKAGE

M AX 17020E TJ+ -40°C to +85°C 32 TQFN

Page 2

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 3)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

IN, ONLDO to GND ................................................-0.3V to +28V

, VCCto GND .....................................................-0.3V to +6V

RTC, LDO to GND ....................................................-0.3V to +6V

OUT_ to GND ...........................................................-0.3V to +6V

ON1, ON2 to GND....................................................-0.3V to +6V

PGOOD_ to GND........................................-0.3V to (V

+ 0.3V)

REF, ILIM_, TON, SKIP to GND..................-0.3V to (V

+ 0.3V)

FB1, REFIN2, LDOREFIN to GND ............................-0.3V to +6V

SECFB to GND .........................................................-0.3V to +6V

BYP to GND..............................................-0.3V to (V

LDO

+ 0.3V)

GND to PGND .......................................................-0.3V to +0.3V

DL_ to PGND..............................................-0.3V to (V

+ 0.3V)

BST_ to GND ..........................................................-0.3V to +34V

BST_ to V

............................................................-0.3V to +28V

DH1 to LX1 ..............................................-0.3V to (V

BST1

+ 0.3V)

BST1 to LX1..............................................................-0.3V to +6V

DH2 to LX2 ..............................................-0.3V to (V

BST2

+ 0.3V)

BST2 to LX2..............................................................-0.3V to +6V

LDO, RTC, REF Short Circuit to GND.........................Momentary

RTC Current Continuous.....................................................+5mA

LDO Current (Internal Regulator)

Continuous..................................................................+100mA

LDO Current (Switched Over) Continuous .....................+200mA

Continuous Power Dissipation (T

= +70°C) 32-Pin 5mm x 5mm TQFN

(derate 34.5mW/°C above +70°C).................................2.76W

Operating Temperature Range ...........................-40°C to +85°C

Junction Temperature......................................................+150°C

Storage Temperature Range .............................-65°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

Dual Mode is a trademark of Maxim Integrated Products, Inc.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INPUT SUPPLIES

IN Standby Supply Current I

IN Shutdown Supply Current I

IN Supply Current I

VCC Supply C urrent I

PWM CONTROLLERS

OUT1 Output Voltage Accuracy (Note 1)

OUT1 Voltage Adjust Range 0.7 5.5 V

FB1 Dual-Mode™ Threshold Voltage Leve ls

FB1 Input B ia s Current I

IN(STBY)

IN(SHDN)

OUT1

FB1

VIN = 6V to 24V, ON1 = ON2 = GND, ONLDO = V

VIN = 4.5V to 24V, ON1 = ON2 = ONLDO = GND

ON1 = ON2 = REFIN2 = VCC, SKIP = FB1 = GND, V

OUT2

ON1 = ON2 = REFIN2 = VCC, SKIP = FB1 = GND, V

OUT2

5V preset output: FB1 = GND,

= 12V, SKIP = V

1.5V preset output: FB1 = VCC (5V),

= 12V, SKIP = V

Adju stable feedback output,

= 12V, SKIP = V

Low 0.04 0.110

High

FB1

= 3.5V, V

= 0.8V, TA = +25°C -0.2 +0.2 µA

OUT1

= 5.3V

85 175 µA

50 70 µA

0.1 0.2 mA

1.0 1.5 mA

4.95 5.00 5.05

1.485 1.50 1.515

0.693 0.700 0.707

1.6V

VCC -

0.7V

Page 3

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 3)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

3.3V preset output: REFIN2 = VCC (5V), V

= 12V, SKIP = V

OUT2 Output Voltage Accuracy (Note 1)

OUT2

1.05V preset output: REFIN2 = RTC (3.3V), V

= 12V, SKIP = V

Tracking output: V

= 12V, SKIP = V

REFIN2

= 1.0V,

OUT2 Voltage-Adjust Range 0 2 V

REFIN2 Voltage-Adjust Range 0 2 V

= 2.2V, TA = +25°C -0.1 +0.1

REFIN2 Input Bias Current I

REFIN2 Dual-Mode Threshold Voltage Leve ls

Load Regulation Error

REFIN2

= 0, TA = +25°C -0.5 +0.1

REFIN2

Low (REFIN2 = RTC) 2.2 3.0

High (REFIN2 = V

Either SMPS, SKIP = VCC, I Either SMPS, SKIP = REF, I Either SMPS, SKIP = GND, I

)

= 0 to 5A -0.1

LOAD

= 0 to 5A -1.7

LOAD

= 0 to 5A -1.5

LOAD

Line Regulation Error Either SMPS, VIN = 6V to 24V 0.005 %/V

TON = GND or REF (400kHz)

TON = V

(200kHz) 1895 2105 2315

TON = GND (500kH z) 475 555 635

TON = REF or V

(300kHz)

DH1 On-Time t

DH2 On-Time t

Minimum Off-Time t

OFF(MIN)

Soft-Start/Stop Slew Rate t

Dynamic REFIN2 Slew Rate t

Ultrasonic Operating Frequency f

SW(USONIC)

SECFB Threshold Voltage V

SECFB Input Bias Current I

VIN = 12V,

ON1

OUT1

(Note 2)

VIN = 12V, V

ON2

OUT2

(Note 2)

(Note 2) 250 400 ns

DYN

Risi ng/falling edge on ON1 or ON2 (preset)

Risi ng/falling edge on ON2 (REFIN2 AD J)

Risi ng edge on REFIN2 8 mV/µs SKIP = open (REF) 20 27 kHz

SECFB

SECFB VSECFB

1.94 2.0 2.06 V

= 5.0V

= 3.3V

= 2.2V, TA = +25°C -0.2 +0.2 µA

LINEAR REGULATOR (LDO)

VIN = 24V, LDOREFIN = BYP = GND,

< 100mA

LDO

< 100mA

LDO

= 0.5V,

LDO

= 2 x V

< 100mA

LDOR EFI N

0.3 2.0 V

= 0 or 2V, TA = +25°C -0.5 +0.5 µA

LDO Output-Voltage Accuracy V

LDOREFIN Input Range V

LDOREFIN Leakage Current I

LDOREFIN VLDO

LDOR EFI N VLDOREFIN

LDOREFIN Dual-Mode Threshold Voltage

LDO

0mA < I VIN = 24V, LDOREFIN = VCC, BYP = GND,

0mA < I VIN = 24V, BYP = GND, V 0mA < I

LDOREFIN low threshold 0.1 0.15 0.20

LDOREFIN high threshold

3.267 3.30 3.333

1.040 1.050 1.060

0.995 1.00 1.005

µA

1.0V

0.4V

895 1052 1209

833 925 1017

1 mV/µs

4.90 5.0 5.10

3.23 3.3 3.37

0.960 1.0 1.040

VCC -

1.5V

VCC -

0.9V

Page 4

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

4 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 3)

)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

LDO Short-Circuit Current I

LDO Regulation Reduction/ Bypass Switchover Threshold

LDO Bypass Switchover Threshold

LDO Bypass Switchover Startup Timeout

LDO Bypass Switch Resistance LDO to BYP, V

VCC Undervoltage-Lockout (UVLO) Threshold

Thermal-Shutdown Threshold T

3.3V ALWAYS-ON LINEAR REGULATOR (RTC)

RTC Output-Voltage Accuracy V

RTC Short-Circuit Current I

REFERENCE (REF)

Reference Voltage V

Reference Load-Regulation Error ΔV

REF Lockout Voltage V

OUT1 FAULT DETECTION

OUT1 Overvoltage Trip Threshold

OU T1 Over voltag e Faul tP r op ag ati on D el ay

OUT1 Undervoltage-Protection Trip Threshold

OUT1 Output-Undervoltage Fault-Propagation Delay

PGOOD1 Lower Trip Threshold

PGOOD1 Propagation Delay t

PGOOD1 Output Low Voltage

PGOOD1 Leakage Current I

ILIM(LDO

BYP

U V L O( V C C )

SHDN

RTC

ILIM(RTC

REF

R E F ( U V L O)

OV P ( OU T 1)

OVP

U V P ( OU T 1)

UVP

PGOOD1

LDO = GND 100 260 mA

With respect to the LDO voltage, falling edge of BYP

With respect to the LDO voltage, rising edge of BYP

Rising edge of BYP to bypass gate pulled low 500 μs

= 5V (Note 4) 1.2 4.5 Ω

BYP

Falling edge of VCC, PWM disabled below this threshold

Rising edge of V

Hysteresis = 10°C +160 °C

ON1 = ON2 = GND, VIN = 6V to 24V, 0 < I

ON1 = ON2 = ONLDO = GND, V

RTC = GND 5 30 mA

VCC = 4.5V to 5.5V, I

REFIREF

Rising edge, 350mV (typ) hysteresis 1.95 V

With respect to error-comparator threshold 13 16 19 %

FB1 forced 50mV above trip threshold 10 μs

With respect to error-comparator threshold 65 70 75 %

With respect to error-comparator threshold, falling edge, hysteresis = 1%

FB1 forced 50mV beyond PGOOD1 trip threshold, falling edge

V I

V PGOOD1 forced to 5.5V, T

< 5mA

RTC

= 6V to 24V, 0 < I

= -20μA to 50μA -10 +10 mV

= 0.56V (PGOOD1 low impedance),

FB1

= 4mA

SINK

= 0.70V (PGOOD1 high impedance),

FB1

< 5mA

RTC

= 0 1.980 2.00 2.020 V

REF

= +25°C

-11.0 -8.5 -6.0 %

-6.5 %

3.8 4.0 4.3

4.2

3.23 3.33 3.43

3.19 3.47

10 μs

-19 -16 -13 %

10 μs

0.3 V

1μA

Page 5

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

_______________________________________________________________________________________ 5

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 3)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

OUT2 FAULT DETECTION

OUT2 Overvoltage Tr ip Threshold

OUT2 Overvoltage Fault-Propagation Delay

OUT2 Undervoltage-Protection Trip Threshold

OUT2 Overvoltage Fault-Propagation Delay

OUT2 Output Undervoltage Fault-Propagation Delay

Dynamic REFIN2 Transition PGOOD Blanking Threshold

PGOOD2 Lower Trip Threshold

PGOOD2 Propagation Delay t

PGOOD2 Output-Low Voltage

PGOOD2 Leakage Current I

CURRENT LIMIT

ILIM_ Adjustment Range V

ILIM_ Current I

Valley Current-Limit Threshold (Adjustable)

Current-Limit Threshold (Negative)

Ultrasonic Current-Limit Threshold V

Current-Limit Threshold (Zero Crossing)

OVP(OUT2)

OVP

UVP(OUT2)

OVP

UVP

PGOOD2

ILIM

5 µA

ILIM

VALLEY VAGND

NEG

NEG(US)

Preset mode (REFIN2 = RTC or VCC): with respect to error-comparator threshold

Dynamic transition, SKIP = REF or V

Tracking mode: with respect to REFIN2 voltage 170 200 230 mV

Minimum overvoltage threshold 0.7 V

OUT2 forced 50mV above trip threshold 10 µs

Preset mode: with respect to error-comparator threshold

Tracking mode: with respect to REFIN2 voltage -250 -300 -350 mV

OUT2 forced 50mV above trip threshold 10 µs

OUT2 forced 50mV below trip threshold 10 µs

Blanking initiated; REFIN2 deviation from the internal target voltage (error-comparator threshold); hysteresis = 5mV

Preset mode: with respect to error-comparator threshold, falling edge, hysteresis = 1%

Tracking mode: with respect to REFIN2 voltage, falling edge, hysteresis = 12mV

OUT2 forced 50mV beyond PGOOD1 trip threshold, falling edge

= V

OUT2

impedance), I

OUT2 = REFIN2 (PGOOD2 high impedance), PGOOD2 forced to 5.5V, T

0.2 2.0 V

With respect to valley current-limit threshold, SKIP = V

OUT1

AGND

REFIN2

- VLX_

= V

OUT2

- VLX_, SKIP = GND or OPEN/REF 3 mV

and OUT2 > REFIN2

- 150mV (PGOOD2 low = 4mA

SINK

= +25°C

_ = 100k 44 50 56

ILIM

_ = 200k 90 100 110

ILIM

_ = 400k 180 200 220

ILIM

= V

= 0.77V, V

FB1

REFIN2

= 0.70V 25 mV

13 16 19 %

REF

0.20

65 70 75 %

±25 mV

-19 -16 -13 %

-175 -150 -125 mV

10 µs

0.3 V

1 µA

-120 %

Page 6

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

6 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)

(Note 3)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

GATE DRIVERS

DH_ Gate Driver On-Resistance R

DL_ Gate Driver On-Resistance R

DH_ Gate Driver Source/Sink Current

DL_ Gate Driver Source Current

DL_ Gate Driver Sink Current I

Internal BST_ Switch On-Resistance

BST_ Leakage Current I

INPUTS AND OUTPUTS

TON Input Logic Level s

SKIP Input Logic Level s

SKIP, TON Leakage Current I

ON_ Input Logic Levels 68mV hysteresis

ON_ Leakage Current I

ONLDO Input Logic Levels 68mV hysteres is

ONLDO Leakage Current I

(SOURCE)

DL (SINK)

BST IBST

BST

SKIP, ITON VSKIP

ON_

ONLDO VONLDO

BST1 - LX1 and BST2 - LX2 forced to 5V 1.5 3.5 

DL1, DL2; high state 2.2 4.5

DL1, DL2; low state 0.6 1.5

DH1, DH2 forced to 2.5V, BST1 - LX1 and BST2 - LX2 forced to 5V

DL1, DL2 forced to 2.5V 1.7 A

DL1, DL2 forced to 2.5V 3.3 A

_ = 10mA, VDD = 5V 5 

_ = 26V, TA = +25°C,

BST

OUT2 and FB1 above regulation threshold

High

REF or open 1.6 3.0

Low 0.4

High (forced-PWM)

Open (ultrasonic) 1.6 3.0

Low (skip) 0.4

ON1

= V

= 0 or 5V, TA = +25°C -2 +2 µA

TON

High (SMPS on) 2.4

Low (SMPS off) 0.8

= 0 or 5V, TA = +25°C -2 +2 µA

ON2

High (SMPS on) 2.4

Low (SMPS off) 0.8

= 0 or 24V, TA = +25°C -1 +1 µA

2 A

0.1 5 µA

0.4V



Page 7

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

_______________________________________________________________________________________ 7

ELECTRICAL CHARACTERISTICS

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= -40°C to +85°C, unless otherwise noted.) (Note 3)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

INPUT SUPPLIES

IN Standby Supply Current I

IN Shutdown Supply Current I

IN(STBY)

IN( SHDN) VIN

IN Supply Current I

VCC Supply Current I

VIN = 6V to 24V, ON1 = ON2 = GND, ONLDO = V

= 4.5V to 24V, ON1 = ON2 = ONLDO = GND 70 µA

ON1 = ON2 = REFIN2 = VCC,

SKIP = FB1 = GND, V

ON1 = ON2 = REFIN2 = VCC, SKIP = FB1 = GND, V

OUT2

= 3.5V, V

OUT1

= 5.3V

PWM CONTROLLERS

5V preset output: FB1 = GND,

= 12V, SKIP = V

OUT1 Output-Voltage Accuracy (Note 1)

OUT1

FB1

1.5V preset output: FB1 = VCC (5V), V

= 12V, SKIP = V

Adjustable feedback output, V

= 12V, SKIP = V

OUT1 Voltage-Adjust Range 0.7 5.5 V

FB1 Dual-Mode Threshold Voltage

Low 0.040 0.125

High

3.3V preset output: REFIN2 = VCC(5V), V

= 12V, SKIP = V

OUT2 Output-Voltage Accuracy (Note 1)

OUT2

1.05V preset output: REFIN2 = RTC (3.3V), V

= 1.2V, SKIP = V

Tracking output: V

= 12V, SKIP = V

REFIN2

= 1.0V,

OUT2 Voltage-Adjust Range 0 2 V

REFIN2 Voltage-Adjust Range 0 2 V

REFIN2 Dual-Mode Threshold Voltage

DH1 On-Time t

DH2 On-Time t

Minimum Off-Time t

Ultrasonic Operating Frequency f

OFF(MIN)

SW(USONIC)

SECFB Threshold Voltage V

ON1

ON2

SECFB

Low (REFIN2 = RTC) 2.2 3.0

High (REFIN2 = V

VIN = 12V,

= 5.0V (Note 2)

OUT1

VIN = 12V,

= 3.3V (Note 2)

OUT2

)

TON = GND or REF (400kHz)

TON = V

(200kHz) 1895 2315

TON = GND (500kHz) 475 635

TON = REF or V

(300kHz)

(Note 2) 425 ns SKIP = open (REF) 18 kHz

1.92 2.08 V

200 µA

0.2 mA

1.5 mA

4.90 5.10

1.47 1.53

0.685 0.715

1.6V

0.7V

3.234 3.366

1.029 1.071

0.985 1.015

1.2V

0.4V

895 1209

833 1017

Page 8

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

8 _______________________________________________________________________________________

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= -40°C to +85°C, unless otherwise noted.) (Note 3)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

LINEAR REGULATOR (LDO)

LDO Output-Voltage Accuracy V

LDO

LDOREFIN Input Range V

LDOREFIN Dual-Mode Threshold Voltage

LDO Short-Circuit Current I

LDO Regulation Reduction/ Bypass Switchover Threshold

VCC Undervoltage-Lockout Threshold

3.3V ALWAYS-ON LINEAR REGULATOR (RTC)

RTC Output-Voltage Accuracy V

RTC Short-Circuit Current I

REFERENCE (REF)

Reference Voltage V

Reference Load-Regulation Error V

OUT1 FAULT DETECTION

OUT1 Overvoltage Trip Threshold

OUT1 Undervoltage-Protection Trip Threshold

PGOOD1 Lower Trip Threshold

PGOOD1 Output-Low Voltage

OUT2 FAULT DETECTION

OUT2 Overvoltage Trip Threshold

LDOREFIN VLDO

ILIM(LDO)

UVLO(VCC)

RTC

ILIM(RTC)

REF

REF IREF

OVP(OUT1)

UVP(OUT1)

OVP(OUT2)

VIN = 24V, LDOREFIN = BYP = GND, 0mA < I

VIN = 24V, LDOREFIN = VCC, BYP = GND, 0mA < I

VIN = 24V, BYP = GND, V 0mA < I

LDOREFIN low threshold 0.10 0.25

LDOREFIN high threshold

LDO = GND 260 mA

Fal ling edge of BYP -12 -5 %

Fal ling edge of VCC, PWM disabled below this threshold

ON1 = ON2 = GND, VIN = 6V to 24V, 0 < I

ON1 = ON2 = ONLDO = GND,

= 6V to 24V, 0 < I

RTC = GND 5 30 mA

VCC = 4.5V to 5.5V, I

With respect to error-comparator thresho ld 12 20 %

With respect to error-comparator thresho ld 63 77 %

With respect to error-comparator threshold, falling edge, hysteresis = 1%

FB1

SINK

Preset mode (REFIN2 = RTC or VCC): with respect to error-comparator threshold

Tracking mode: with respect to REFIN2 voltage 160 240 mV

< 100mA

LDO

< 100mA

LDO

< 100mA

LDO

= 2x V

LDOR EFI N

< 5mA

RTC

= -20µA to 50µA -10 +10 mV

= 0.56V (PGOOD1 low impedance),

= 4mA

LDOR EFI N

0.3 2.0 V

< 5mA

RTC

= 0 1.975 2.025 V

REF

= 0.5V,

4.85 5.15

3.20 3.40

0.960 1.040

3.8 4.3 V

3.18 3.45

3.16 3.50

-20 -12 %

0.4 V

12 20 %

VCC -

0.9V

Page 9

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

_______________________________________________________________________________________ 9

ELECTRICAL CHARACTERISTICS (continued)

(Circuit of Figure 1, no load on LDO, RTC, OUT1, OUT2, and REF, VIN= 12V, VDD= VCC= V

SECFB

= 5V, V

REFIN2

= 1.0V, BYP =

LDOREFIN = GND, ONLDO = IN, ON1 = ON2 = V

, TA= -40°C to +85°C, unless otherwise noted.) (Note 3)

)

Note 1: DC output accuracy specifications refer to the threshold of the error comparator. When the inductor is in continuous conduc-

tion, the MAX17020 regulates the valley of the output ripple, so the actual DC output voltage is higher than the trip level by 50% of the output ripple voltage. In discontinuous conduction (I

OUT

< I

LOAD(SKIP)

), the output voltage has a DC regulation

level higher than the error-comparator threshold by approximately 1.5% due to slope compensation.

Note 2: On-time and off-time specifications are measured from 50% point to 50% point at the DH pin with LX = PGND, V

BST

= 5V, and a 500pF capacitor from DH to LX to simulate external MOSFET gate capacitance. Actual in-circuit times might be different due to MOSFET switching speeds.

Note 3: Limits are 100% production tested at T

= +25°C. Maximum and minimum limits over temperature are guaranteed by design

and characterization.

Note 4: Specifications increased by 1Ω to account for test measurement error.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

OUT2 Undervoltage-Protection Trip Threshold

PGOOD2 Lower Trip Threshold

PGOOD2 Output-Low Voltage

CURRENT LIMIT

ILIM_ Adjustment Range V

Valley Current-Limit Threshold (Adjustable)

GATE DRIVERS

DH_ Gate Driver On-Resistance R

DL_ Gate Driver On-Resistance R

INPUTS AND OUTPUTS

TON Input Logic Levels

SKIP Input Logic Levels

ON_ Input Logic Levels

ONLDO Input Logic Levels

UVP(OUT2

ILIM

VALLEYV AGN D

Preset mode: with respect to error-comparator threshold

Tr acki ng m od e: w i th r esp ect to RE FIN 2 vol tag e -230 -370 mV

Preset mode: with respect to error-comparator threshold, falling edge, hysteresis = 1%

Tracking mode: with respect to REFIN2 voltage, falling edge, hysteresis = 12mV

= V

OUT2

impedance), I

- V

BST1 - LX1 and BST2 - LX2 forced to 5V 3.5 Ω

DL1, DL2; high state 4.5

DL1, DL2; low state 1.5

High

REF or open 1.6 3.0

Low 0.4

High (forced-PWM)

Open (ultrasonic) 1.6 3.0

Low (skip) 0.4

High (SMPS on) 2.4

Low (SMPS off) 0.8

High (LDO on) 2.4

Low (LDO off) 0.8

- 150mV (PGOOD2 low

REFIN2

= 4mA

SINK

_ = 100kΩ 40 60

ILIM

_ = 200kΩ 85 115

ILIM

_ = 400kΩ 164 236

ILIM

63 77 %

-20 -12 %

-185 -115 mV

0.2 2.0 V

0.4V

0.4 V

Page 10

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

10 ______________________________________________________________________________________

Typical Operating Characteristics

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TON = REF, TA= +25°C, unless otherwise noted.)

5V OUTPUT EFFICIENCY

vs. LOAD CURRENT

100

EFFICIENCY (%)

0.01 10

LOAD CURRENT (A)

3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT

100

5V SMPS ENABLED

SKIP MODE

ULTRASONIC

EFFICIENCY (%)

MODE

0.01 10 LOAD CURRENT (A)

12V

20V

SKIP MODE PWM MODE

10.1

PWM MODE

10.1

12V

100

SKIP MODE

MAX17020 toc01

EFFICIENCY (%)

0.01 10

SMPS OUTPUT VOLTAGE DEVIATION

MAX17020 toc04

-1

SKIP MODE

OUTPUT VOLTAGE DEVIATION (%)

-2

-3

0.01 10

5V OUTPUT EFFICIENCY

vs. LOAD CURRENT

PWM MODE

ULTRASONIC MODE

10.1

LOAD CURRENT (A)

vs. LOAD CURRENT

LOW-NOISE ULTRASONIC

PWM MODE

10.1

LOAD CURRENT (A)

12V

100

5V SMPS ENABLED

MAX17020 toc02

EFFICIENCY (%)

0.01 10

1000

MAX17020 toc05

100

SWITCHING FREQUENCY (kHz)

0.01 10

3.3V OUTPUT EFFICIENCY vs. LOAD CURRENT

20V

12V

SKIP MODE PWM MODE

10.1

LOAD CURRENT (A)

SWITCHING FREQUENCY

vs. LOAD CURRENT

PWM MODE

LOW-NOISE ULTRASONIC MODE

SKIP MODE

10.1

LOAD CURRENT (A)

MAX17020 toc03

MAX17020 toc06

12V

5V LDO OUTPUT VOLTAGE

vs. LOAD CURRENT

5.2

5.1

5.0

4.9

OUTPUT VOLTAGE (V)

4.8

4.7 020 140

LOAD CURRENT (mA)

100 12040 60 80

MAX17020 toc07

OUTPUT VOLTAGE (V)

3.3V RTC OUTPUT VOLTAGE vs. LOAD CURRENT

3.5

3.4

3.3

3.2

3.1

3.0 02 12

LOAD CURRENT (mA)

10468

MAX17020 toc08

NO-LOAD INPUT SUPPLY CURRENT

vs. INPUT VOLTAGE

100

PWM MODE

SKIP MODE

SUPPLY CURRENT (mA)

0.1

0.01 025

INPUT VOLTAGE (V)

LOW-NOISE

ULTRASONIC

15 20510

MAX17020 toc09

Page 11

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 11

Typical Operating Characteristics (continued)

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TON = REF, TA= +25°C, unless otherwise noted.)

STANDBY AND SHUTDOWN INPUT

SUPPLY CURRENT vs. INPUT VOLTAGE

STANDBY (ONLDO = VIN)

0.1

SUPPLY CURRENT (mA)

SHUTDOWN

(ONLDO = ON1 = ON2 = GND)

0.01 025

INPUT VOLTAGE (V)

15 20510

MAX17020 toc10

SAMPLE PERCENTAGE (%)

-20 -4-12 4 12 20

100mV ILIM THRESHOLD VOLTAGE DISTRIBUTION

+85°C

+25°C

SAMPLE PERCENTAGE (%)

90 9894 102 106 110

ILIM THRESHOLD VOLTAGE (mV)

SAMPLE SIZE = 150

12V

MAX17020 toc13

REFERENCE OFFSET

VOLTAGE DISTRIBUTION

+85°C

+25°C

2V REF OFFSET VOLTAGE (mV)

LDO AND RTC POWER-UP

A. INPUT SUPPLY, 5V/div B. 5V LDO, 2V/div

200μs/div

SAMPLE SIZE = 150

MAX17020 toc14

C. 3.3V RTC, 2V/div D. 2.0V REF, 1V/div

MAX17020 toc11

SAMPLE PERCENTAGE (%)

12V

12V B

3.3V

2.0V

3.3V

REFIN2 OFFSET

VOLTAGE DISTRIBUTION

+85°C

+25°C

-5 -1-3 1 3 5

REFIN2 OFFSET VOLTAGE (mV)

LDO AND RTC POWER REMOVAL

A. INPUT SUPPLY, 5V/div B. 5V LDO, 2V/div

SAMPLE SIZE = 150

200μs/div

MAX17020 toc15

C. 3.3V RTC, 2V/div D. 2.0V REF, 1V/div

MAX17020 toc12

A 12V

B 5V

3.3V D

2.0V

Page 12

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

12 ______________________________________________________________________________________

Typical Operating Characteristics (continued)

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TON = REF, TA= +25°C, unless otherwise noted.)

5V LDO LOAD TRANSIENT

0.1A

A. LDO OUTPUT, 100mV/div

4μs/div

0V 5V

MAX17020 toc16

B. LOAD CURRENT, 100mA/div

SHUTDOWN WAVEFORMS

(SWITCHING REGULATORS)

5V SMPS STARTUP AND SHUTDOWN

A. 5V LDO OUTPUT, 0.2V/div B. 5V SMPS OUTPUT, 2V/div C. ON1, 5V/div

MAX17020 toc19

200μs/div

3.1A

MAX17020 toc17

A 5V

B 5V

5V SMPS LOAD TRANSIENT

(PWM MODE)

(SWITCHING REGULATORS)

A. ON1, 2V/div B. 5V SMPS OUTPUT, 2V/div

STARTUP WAVEFORMS

100μs/div

C. PGOOD1, 5V/div D. INDUCTOR CURRENT, 5A/div

MAX17020 toc20

MAX17020 toc18

B 5V

0V 5V

A. ON1, 5V/div B. 5V SMPS OUTPUT, 2V/div

200μs/div

C. PGOOD1, 2V/div D. INDUCTOR CURRENT, 5A/div

A. LOAD CURRENT, 2A/div B. 5V SMPS OUTPUT, 100mV/div C. INDUCTOR CURRENT, 2A/div

40μs/div

Page 13

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 13

Typical Operating Characteristics (continued)

(Circuit of Figure 1, VIN= 12V, VDD= VCC= 5V, TON = REF, TA= +25°C, unless otherwise noted.)

Pin Description

3.3V SMPS LOAD TRANSIENT

6.5A

0.5A

3.3V

A. LOAD CURRENT, 5A/div B. 3.3V SMPS OUTPUT, 100mV/div C. INDUCTOR CURRENT, 5A/div

40μs/div

MAX17020 toc21

A. INPUT VOLTAGE, 5V/div B. 5V LDO OUTPUT, 2V/div

PIN NAME FUNCTION

2V Reference-Voltage Output. Bypass REF to AGND with a 0.1µF or greater ceramic capacitor. The

1 REF

reference can source up to 50µA for external loads. Loading REF degrades output-voltage accuracy according to the REF load-regulation error. The reference shuts down when ON1, ON2, and ONLDO are all pulled low.

POWER REMOVAL

(SMPS UVLO RESPONSE)

10ms/div

MAX17020 toc22

C. 5V SMPS, 2V/div D. PGOOD1, 5V/div

Switching-Frequency Setting Input. Select the OUT1/OUT2 switching frequencies by connecting TON as follows for:

2 TON

High (V Open (REF) = 400kHz/300kHz GND = 400kH z/500kHz

3 V

4 ONLDO

Analog Supply Voltage Input. Connect VCC to the system supply voltage with a series 50 resistor, and bypass to analog ground using a 1µF or greater ceramic capacitor.

Enable Input for LDO. Drive ONLDO high to enable the linear regulator (LDO) output. Drive ONLDO low to shut down the linear regulator output.

3.3V Always-On Linear Regulator Output for RTC Power. Bypass RTC with a 1µF or greater ceramic

5 RTC

capacitor to analog ground. RTC can source at lea st 5mA for external load s upport. RTC power-up is required for controller operation.

Power-Input Supply. IN powers the linear regulators (RTC and LDO) and senses the input voltage for the

6 IN

Quick-PWM on-time one-shot timers. The high-side MOSFET’s on-time is inversely proportional to the input voltage. Bypass IN with a 0.1μF or greater ceramic capacitor to PGND close to the MAX17020.

Linear Regulator Output. Bypa ss LDO with a 4.7µF or greater ceramic capacitor. LDO can source at lea st

7 LDO

100mA for external load support. LDO is powered from IN and its regulation threshold is set by LDOREFIN. For preset 5V operation, connect LDOREFIN directl y to GND. For preset 3.3V operation, connect LDOREFIN directly to V

) = 200kHz/300kHz

. When LDO is used for 5V operation, LDO must supply VCC and VDD.

Page 14

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

14 ______________________________________________________________________________________

Pin Description (continued)

PIN NAME FUNCTION

External Reference Input for the Linear Regulator. LDOREFIN sets the LDO regulation voltage (V V

8 LDOREFIN

9 BYP

10 OUT1

11 FB1

12 ILIM1

13 PGOOD1

14 ON1 Enable Input for SMPS1. Drive ON1 high to enable SMPS1. Drive ON1 low to shut down SMPS1.

15 DH1 High-Side Gate-Driver Output for SMPS1. DH1 swings from LX1 to BST1.

16 LX1

17 BST1

18 DL1 Low-Side Gate-Driver Output for SMPS1. DL1 swings from PGND to V

19 V

20 SECFB

21 AGND Analog Ground. Connect the backside exposed pad to AGND.

22 PGND Power Ground

23 DL2 Low-Side Gate-Driver Output for SMPS2. DL2 swings from PGND to V

24 BST2

25 LX2

LDOR EFI N

output voltage, or connect LDOREFIN to V to 5V and is enabled, LDO must supply V

Linear Regulator Bypass Input. When BYP voltage exceeds 93.5% of the LDO voltage, the controller bypasses the LDO output to the BYP input. The bypass switch is disabled if the LDO voltage drops by 8.5% from its nominal regulation threshold. When not being used, connect BYP to GND.

Output Voltage-Sense Input for SMPS1. OUT1 is an input to the Quick-PWM on-time one-shot timer. OUT1 also serves as the feedback input for the preset 5V (FB1 = GND) and 1.5V (FB1 = V

Adjustable Feedback Voltage-Sense Connection for SMPS1. Connect FB1 to GND for fixed 5V operation. Connect FB1 to V to analog ground to ad just the output voltage between 0.7V and 5.5V.

Valley Current-Limit Adjustment for SMPS1. The GND - LX1 current-limit threshold is 1/10 the voltage present on ILIM1 over a 0.2V to 2V range. An internal 5µA current source allows this voltage to be set with a single resistor between ILIM1 and analog ground.

Open-Drain Power-Good Output for S MPS1. PGOOD1 is low when the output voltage is more than 16% ( typ) below the nominal regulation threshold, during soft-start, in shutdown, and after the fault latch has been tripped. After the soft-start circuit has terminated, PGOOD1 becomes high impedance if the output is in regulation.

Inductor Connection for SMPS1. Connect LX1 to the switched side of the inductor. LX1 is the lower supply rail for the DH1 high-side gate driver.

Boost Flying-Capacitor Connection for SMPS1. Connect to an external capacitor as shown in Figure 1. An optional resistor in series with BST1 allows the DH1 turn-on current to be adjusted.

Supply-Voltage Input for the DL_ Gate Drivers. Connect to a 5V supply. Also connect to the drain of the BST diode switch.

Secondary Feedback Input. The secondary feedback input forces the SMPS1 output into ultrasonic mode when the SECFB voltage drops below its 2V threshold voltage. This forces DL1 and DH1 to switch, allowing the system to refresh an external low-power charge pump being driven by DL1 (see Figure 1). Connect SECFB to V

Boost Flying-Capacitor Connection for SMPS2. Connect to an external capacitor as shown in Figure 1. An optional resistor in series with BST2 allows the DH2 turn-on current to be adjusted.

Inductor Connection for SMPS2. Connect LX2 to the switched side of the inductor. LX2 is the lower supply rail for the DH2 high-side gate driver.

) for a 0.3V to 2V LDOREFIN range. Connect LDOREFIN to GND for a fixed 5V linear-regulator

for a fixed 3.3V linear-regulator output voltage. When LDO is set

and VDD.

for fixed 1.5V operation. Connect FB1 to an external resistive voltage-divider from OUT1

DD.

to the 5V bias supply to disable secondary feedback.

DD.

= 2 x

LDO

) output voltage settings.

Page 15

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 15

Pin Description (continued)

PIN NAME FUNCTION

26 DH2 High-Side Gate-Driver Output for SMPS2. DH2 swings from LX2 to BST2.

27 ON2 Enable Input for SMPS2. Drive ON2 high to enable SMPS2. Drive ON2 low to shut down SMPS2.

Open-Drain Power-Good Output for SMPS2. PGOOD2 is low when the output voltage is more than 150mV (typ) below the REFIN2 voltage or more than 16% below the preset voltage, during soft-start, in shutdown,

28 PGOOD2

29 SKIP

30 OUT2

31 ILIM2

32 REFIN2

— EP Exposed Pad. Connect the backside exposed pad to AGND.

and when the fault latch has been tripped. After the soft-start circuit has terminated, PGOOD2 becomes high impedance if the output is in regulation. PGOOD2 is blanked—forced high-impedance state—when a dynamic REFIN transition is detected.

Pulse-skipping Control Input. This three-level input determines the operating mode for the switching regulators: High (VCC) = Forced-PWM operation Open/REF (2V) = Ultrasonic mode GND = Pulse-skipping mode

Output Voltage-Sense Input for SMPS2. OUT2 is an input to the Quick-PWM on-time one-shot timer. OUT2 also serves as the feedback input for the preset 3.3V (REFIN2 = V

Valley Current-Limit Adjustment for SMPS2. The GND - LX2 current-limit threshold is 1/10 the voltage present on ILIM2 over a 0.2V to 2V range. An internal 5µA current source allows this voltage to be set with a single resistor between ILIM2 and analog ground.

External Reference Input for SMPS2. REFIN2 sets the feedback-regulation voltage (V MAX17020 includes an internal window comparator to detect when the REFIN2 voltage changes, allowing the controller to blank PGOOD2 and the fault protection. Connect REFIN2 to RTC for fixed 1.05V operation. Connect REFIN2 to V

for fixed 3.3V operation.

) and 1.05V (REFIN2 = RTC).

= V

OUT2

REFIN2

). The

Page 16

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

16 ______________________________________________________________________________________

Figure 1. Standard Application Circuit—Main Supply

NOTE: PLACE C22 BETWEEN IN AND PGND AS CLOSE AS POSSIBLE TO THE MAX17020.

5V OUTPUT

OUT1

12V TO 15V

CHARGE

PUMP

0.1μF

500kΩ

100kΩ

BST1

0.1μF

10nF

C22

0.1μF

DH1

BST1

LX1

DL1

OUT1 BYP

MAX17020

SECFB

FB1

DH2

BST2

LX2

DL2

PGND AGND

OUT2

PGOOD1 PGOOD2

RTC

REF

SKIP

0.1μF

RGND

100kΩ

C3 1μF

0Ω

BST2

0.1μF

R7 100kΩ

4 x 10μF 25V

INPUT (V

7V TO 24V

3.3V OUTPUT

OUT2

5V SMPS OUTPUT (OUT1)

POWER-GOOD

}

RTC SUPPLY

LDOREFIN

5V LDO OUTPUT

POWER GROUND

ANALOG GROUND

*LOWER INPUT VOLTAGES REQUIRE ADDITIONAL INPUT CAPACITANCE. IF OPERATING NEAR DROPOUT, COMPONENT SELECTION MUST BE CAREFULLY DONE TO ENSURE PROPER OPERATION.

47Ω

1.0μF

4.7μF

LDO

REFIN2

ILIM1

PAD

ON1 ON2

ONLDO

TON

ILIM2

OFFON

OUT1/OUT2 SWITCHING FREQUENCY

OPEN (REF): 400kHz/300kHz

ILIM2

Page 17

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 17

Figure 2. Functional Diagram Overview

TON

SKIP

RTC

SECFB

ILIM1

OUT1

BST1

DH1

LX1

DL1

PGND

FB1

ON1

PGOOD1

ONLDO

5V LINEAR

REGULATOR

3.3V LINEAR REGULATOR

PWM1

CONTROLLER

(FIGURE 3)

FB SELECT

(PRESET vs. ADJ)

UVLO

POWER-GOOD

AND FAULT

PROTECTION

FAULT1

LDO BYPASS

CIRCUITRY

CONTROLLER

(FIGURE 3)

FAULT2

POWER-GOOD

AND FAULT

PROTECTION

PWM2

FB SELECT

(PRESET vs. ADJ)

UVLO

LDOREFIN

LDO

BYP

ILIM2

OUT2

BST2

DH2

LX2

DL2

REFIN2

ON2

PGOOD2

PAD

MAX17020

REF

GND

Page 18

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

18 ______________________________________________________________________________________

Table 1. Component Selection for Standard Applications

Table 2. Component Suppliers

400kHz/300kHz

COMPONENT

Input Voltage VIN = 7V to 24V VIN = 7V to 24V VIN = 7V to 24V

Input Capacitor

)

SMPS 1

Output Capacitor

)

OUT1

Inductor (L1)

High-Side MOSFET (N

)

Low-Side MOSFET (N

)

Current-Limit Resistor (R

)

ILIM1

SMPS 2

Output Capacitor

)

OUT2

Inductor (L2)

High-Side MOSFET

)

Low-Side MOSFET

)

Current-Limit Resistor

)

ILIM2

SMPS 1: 5V AT 5A

SMPS 2: 3.3V AT 8A

(4x) 10μF, 25V Taiyo Yuden TMK432BJ106KM

330μF, 6V, 18mΩ SANYO 6TPE330MIL

4.3μH, 11.4mΩ, 11A Sumida CEP125U

Fairchild Semiconductor FDS6612A 26mΩ/30mΩ, 30V

Fairchild Semiconductor FDS6670S 9mΩ/11.5mΩ, 30V

200kΩ 150kΩ 49.9kΩ

470μF, 4V, 15mΩ SANYO 4TPE470MFL

4.3μH, 11.4mΩ, 11A Sumida CEP125U

Fairchild Semiconductor FDS8690

8.6mΩ/11.4mΩ, 30V

Fairchild Semiconductor FDMS8660S

2.6mΩ/3.5mΩ, 30V

200kΩ 200kΩ 49.9kΩ

400kHz/500kHz

SMPS 1: 5V AT 3A

SMPS 2: 3.3V AT 5A

(2x) 10μF, 25V Taiyo Yuden TMK432BJ106KM

330μF, 6V, 18mΩ SANYO 6TPE330MIL

4.7μH, 9.8mΩ, 7A Sumida CDRH10D68

Vishay Siliconix Si4814DY Dual 30V MOSFET High side: 19mΩ/23mΩ Low side: 18mΩ/22mΩ

330μF, 6V, 18mΩ SANYO 6TPE330MIL

4.7μH, 9.8mΩ, 7A Sumida CDRH10D68

Vishay Siliconix Si4814DY Dual 30V MOSFET High side: 19mΩ/23mΩ Low side: 18mΩ/22mΩ

(4x) 10μF, 25V Taiyo Yuden TMK432BJ106KM

(2x) 330μF, 2V, 7mΩ SANYO 2TPF330M7

1.5μH, 12A, 7mΩ NEC/Tokin MPLC1040L1R5

Fairchild Semiconductor FDS8690

8.6mΩ/11.4mΩ, 30V

Fairchild Semiconductor FDMS8660S

2.6mΩ/3.5mΩ, 30V

330μF, 2V, 7mΩ SANYO 2TPF330M7

1.5μH, 12A, 7mΩ NEC/Tokin MPLC1040L1R5

Fairchild Semiconductor FDS8690

8.6mΩ/11.4mΩ, 30V

Fairchild Semiconductor FDMS8660S

2.6mΩ/3.5mΩ, 30V

400kHz/300kHz

SMPS 1: 1.5V AT 8A

SMPS 2: 1.05V AT 5A

SUPPLIER WEBSITE

AVX Corp. www.avxcorp.com

C entr al S emi cond uctor Cor p . www.centralsemi.com

Fairchild Semiconductor www.fairchildsemi.com

International Rectifier www.irf.com

KEMET Corp www.kemet.com

NEC/Tokin America, Inc. www.nec-tokinamerica.com

Panasonic Corp. www.panasonic.com

Philips/nxp Semiconductor www.semiconductors.philips.com

Pulse Engineering www.pulseeng.com

SUPPLIER WEBSITE

Renesas Technol og y C or p . www.renesas.com

SANYO Electric Co., Ltd. www.sanyodevice.com

Sumida Corp. www.sumida.com

Taiyo Yuden www.t-yuden.com

TDK Corp. www.component.tdk.com

TOKO America, Inc. www.tokoam.com

Vishay (Dale, Siliconix) www.vishay.com

Würth Elektronik GmbH & Co. KG

www.we-online.com

Page 19

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 19

Detailed Description

The MAX17020 step-down controller is ideal for highvoltage, low-power supplies for notebook computers. Maxim’s Quick-PWM pulse-width modulator in the MAX17020 is specifically designed for handling fast load steps while maintaining a relatively constant operating frequency and inductor operating point over a wide range of input voltages. The Quick-PWM architecture circumvents the poor load-transient timing problems of fixed-frequency current-mode PWMs, while also avoiding the problems caused by widely varying switching frequencies in conventional constant-on-time and constant-off-time PWM schemes. Figure 2 is a functional diagram overview. Figure 3 is the functional diagram—Quick-PWM core.

The MAX17020 includes several features for multipurpose notebook functionality, allowing this controller to be used two or three times in a single notebook—main, I/O chipset, and graphics. The MAX17020 includes a 100mA LDO that can be configured for preset 5V operation—ideal for initial power-up of the notebook and main supply—or can be adjusted for lower voltage operation—ideal for low-power I/O or graphics supply requirements. Additionally, the MAX17020 includes a

3.3V, 5mA RTC supply that remains always enabled, which can be used to power the RTC supply and system pullups when the notebook shuts down. The MAX17020 also includes an optional secondary feedback input that allows an unregulated charge pump or secondary winding to be included on a supply—ideal for generating the low-power 12V to 15V load switch supply. Finally, the MAX17020 includes a reference input on SMPS 2 that allows dynamic voltage transitions when driven by an adjustable resistive voltage-divider or DAC—ideal for the dynamic graphics core requirements.

3.3V RTC Power

The MAX17020 includes a low-current (5mA) linear regulator that remains active as long as the input supply (IN) exceeds 2V (typ). The main purpose of this “always-enabled” linear regulator is to power the realtime clock (RTC) when all other notebook regulators are disabled. RTC also serves as the main bias supply of the MAX17020 so it powers up before the LDO and switching regulators. The RTC regulator sources at least 5mA for external loads.

Adjustable 100mA Linear Regulator

The MAX17020 includes a high-current (100mA) linear regulator that can be configured for preset 5V or 3.3V operation or adjusted between 0.6V to 4V. When the MAX17020 is configured as a main supply, this LDO is required to generate the 5V bias supply necessary to power up the switching regulators. Once the switching regulators are enabled, the LDO can be bypassed using the dedicated BYP input. The adjustable linear regulator allows generation of the 3.3V suspend supply or buffered low-power chipset and GPU reference supplies. The MAX17020 LDO sources at least 100mA of supply current.

Bypass Switch

The MAX17020 includes an independent LDO bypass input that allows the LDO to be bypassed by either switching regulator output or from a different regulator all together. When the bypass voltage (BYP) exceeds 93.5% of the LDO output voltage for 500μs, the MAX17020 reduces the LDO regulation threshold and turns on an internal p-channel MOSFET to short BYP to LDO. Instead of disabling the LDO when the MAX17020 enables the bypass switch, the controller reduces the LDO regulation voltage, which effectively places the linear regulator in a standby state while switched over, yet allows a fast recovery if the bypass supply drops.

Connect BYP to GND when not used to avoid unintentional conduction through the body diode (BYP to LDO) of the p-channel MOSFET.

5V Bias Supply (VCC/VDD)

The MAX17020 requires an external 5V bias supply (VDDand VCC) in addition to the battery. Typically, this 5V bias supply is generated by either the internal 100mA LDO (when configured for a main supply) or from the notebook’s 95%-efficient 5V main supply (when configured for an I/O chipset, DDR, or graphics). Keeping these bias supply inputs independent improves the overall efficiency and allows the internal linear regulator to be used for other applications as well.

The VDDbias supply input powers the internal gate drivers and the VCCbias supply input powers the analog control blocks. The maximum current required is dominated by the switching losses of the drivers and can be estimated as follows:

BIAS(MAX)

= I

CC(MAX)

+ fSWQG≈ 30mA to 60mA (typ)

Page 20

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

20 ______________________________________________________________________________________

Free-Running Constant-On-Time PWM

Controller with Input Feed-Forward

The Quick-PWM control architecture is a pseudo-fixedfrequency, constant on-time, current-mode regulator with voltage feed-forward. This architecture relies on the output filter capacitor’s ESR to act as a currentsense resistor, so the feedback ripple voltage provides the PWM ramp signal. The control algorithm is simple: the high-side switch on-time is determined solely by a one-shot whose pulse width is inversely proportional to input voltage and directly proportional to output voltage. Another one-shot sets a minimum off-time (400ns typ). The on-time one-shot is triggered if the error comparator is low, the low-side switch current is below the valley current-limit threshold, and the minimum off-time one-shot has timed out.

On-Time One-Shot

The heart of the PWM core is the one-shot that sets the high-side switch on-time. This fast, low-jitter, adjustable one-shot includes circuitry that varies the on-time in response to battery and output voltage. The high-side switch on-time is inversely proportional to the battery voltage as sensed by the IN input, and proportional to the output voltage:

On-Time = K (V

OUT/VIN

)

where K (switching period) is set by the tri-level TON input (see the

Pin Description

section). High-frequency (400kHz/500kHz) operation optimizes the application for the smallest component size, trading off efficiency due to higher switching losses. This might be acceptable in ultra-portable devices where the load currents are lower and the controller is powered from a lower voltage supply. Low-frequency (200kHz/300kHz) operation offers the best overall efficiency at the expense of component size and board space.

For continuous conduction operation, the actual switching frequency can be estimated by:

where V

DROP1

is the sum of the parasitic voltage drops in the inductor discharge path, including synchronous rectifier, inductor, and PCB resistances; V

DROP2

is the sum of the voltage drops in the charging path, including the high-side switch, inductor, and PCB resistances; and tONis the on-time calculated by the MAX17020.

Table 3. Approximate K-Factor Errors

tVV V

ON IN DROP DROP

OUT DROP

+−

()

SWITCHING

REGULATOR

SMPS 1

SMPS 2

TON SETTING

(kHz)

200kHz

TON = V

TON = REF or GND

TON = REF or V

TON = GND

400kHz

300kHz

500kHz

TYPICAL K-FACTOR

(µs)

5.0 ±10 Use for absolute best efficiency.

2.5 ±12.5

3.3 ±10

2.0 ±12.5

K-FACTOR ERROR

(%)

COMMENTS

Useful in 3-cell systems for lighter loads than the CPU core or where size is key.

Considered mainstream by current standards.

Good operating point for compound buck designs or desktop circuits.

Page 21

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 21

Figure 3. Functional Diagram—Quick-PWM Core

INTEGRATOR

REF

AGND

ILIM

VCC

NEG CURRENT

LIMIT

VALLEY

CURRENT LIMIT

GND

AGND

SLOPE COMP

OFF(MIN)

TRIG

ONE-SHOT

* RESET DOMINATE

ONE-SHOT

ON-TIME

COMPUTE

INT PRESET OR EXT ADJ

ANALOG

SOFT-

START/STOP

Q DH DRIVER

TRIG

REFIN

TON

ZERO

CROSSING

GND

ULTRASONIC

THRESHOLD

GND

SKIP

REFIN

THREE-LEVEL

DECODE

ULTRASONIC

ONE-SHOT

TRIG

DL DRIVER

Page 22

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

22 ______________________________________________________________________________________

Modes of Operation

Forced-PWM Mode (

SSKKIIPP

= VCC)

The low-noise forced-PWM mode (SKIP = VCC) disables the zero-crossing comparator, which controls the low-side switch on-time. This forces the low-side gatedrive waveform to constantly be the complement of the high-side gate-drive waveform, so the inductor current reverses at light loads while DH maintains a duty factor of V

OUT/VIN

. The benefit of forced-PWM mode is to keep the switching frequency fairly constant. However, forced-PWM operation comes at a cost: the no-load 5V bias current remains between 20mA to 60mA depending on the switching frequency and MOSFET selection.

The MAX17020 automatically uses forced-PWM operation during all transitions—dynamic REFIN, startup, and shutdown—regardless of the SKIP configuration.

Automatic Pulse-Skipping Mode (

SSKKIIPP

= GND)

In skip mode (SKIP = GND), an inherent automatic switchover to PFM takes place at light loads. This switchover is affected by a comparator that truncates the low-side switch on-time at the inductor current’s zero crossing. The zero-crossing comparator threshold is set by the differential across LX and AGND.

DC output-accuracy specifications refer to the integrated threshold of the error comparator. When the inductor is in continuous conduction, the MAX17020 regulates the valley of the output ripple and the internal integrator removes the actual DC output-voltage error caused by the output-ripple voltage and internal slope compensation. In discontinuous conduction (SKIP = GND and I

OUT

< I

LOAD(SKIP)

), the integrator cannot correct for the low-frequency output ripple error, so the output voltage has a DC regulation level higher than the error comparator threshold by approximately 1.5% due to slope compensation and output ripple voltage.

Ultrasonic Mode (

SSKKIIPP

= Open or REF)

Leaving SKIP unconnected or connecting SKIP to REF (2V) activates a unique pulse-skipping mode with a guaranteed minimum switching frequency of 20kHz. This ultrasonic pulse-skipping mode eliminates audiofrequency modulation that would otherwise be present when a lightly loaded controller automatically skips pulses. In ultrasonic mode, the controller automatically transitions to fixed-frequency PWM operation when the load reaches the same critical conduction point (I

LOAD(SKIP)

) that occurs when normally pulse skipping.

An ultrasonic pulse occurs (Figure 4) when the controller detects that no switching has occurred within the last 37μs or when SECFB drops below its feedback threshold. Once triggered, the ultrasonic circuitry pulls

DL high, turning on the low-side MOSFET to induce a negative inductor current. After the inductor current reaches the negative ultrasonic current threshold, the controller turns off the low-side MOFET (DL pulled low) and triggers a constant on-time (DH driven high). When the on-time has expired, the controller reenables the low-side MOSFET until the inductor current drops below the zero-crossing threshold. Starting with a DL pulse greatly reduces the peak output voltage when compared to starting with a DH pulse.

The output voltage at the beginning of the ultrasonic pulse determines the negative ultrasonic current threshold, resulting in the following equation:

NEG(US)

= ILRCS= (V

NOM

- VFB) x 0.385V

where V

NOM

is the nominal feedback-regulation voltage, and VFBis the actual feedback voltage (VFB> V

NOM

), and RCSis the current-sense resistance seen

across LX to AGND.

Secondary Feedback: SECFB—OUT1 ONLY

When the controller skips pulses (SKIP = GND or REF), the long time between pulses (especially if the output is sinking current) allows the external charge-pump voltage or transformer secondary winding voltage to drop. When the SECFB voltage drops below its 2V feedback threshold, the MAX17020 issues an ultrasonic pulse (regardless of the ultrasonic one-shot state). This forces a switching cycle, allowing the external unregulated charge pump (or transformer secondary winding) to be refreshed. See the

Ultrasonic Mode (SKIP = Open or REF)

section for

switching cycle sequence/specifications.

Figure 4. Ultrasonic Waveforms

37μs (typ)

INDUCTOR CURRENT

ZERO-CROSSING DETECTION

SONIC

ON-TIME (tON)

Page 23

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 23

Dynamic Output Voltage—OUT2 Only

The MAX17020 regulates OUT2 to the voltage set at REFIN2, so the MAX17020 supports applications that require dynamic output-voltage changes between two set points by adjusting the REFIN2 voltage. For a stepvoltage change at REFIN2, the rate of change of the output voltage is limited either by the internal slew-rate circuit, by the REFIN2 slew rate, or by the component selection—inductor current ramp, the total output capacitance, the current limit, and the load during the transition—whichever is the slowest. The total output capacitance determines how much current is needed to change the output voltage, while the inductor limits the current ramp rate. Additional load current slows down the output voltage change during a positive REFIN2 voltage change, and speeds up the output voltage change during a negative REFIN2 voltage change. Figure 5 is the dynamic REFIN transition.

Automatic Fault Blanking

When the MAX17020 automatically detects that the internal target and REFIN2 are more than ±25mV (typ) apart, the controller automatically blanks PGOOD2, blanks the UVP protection, and sets the OVP threshold to REF + 200mV. The blanking remains until 1) the internal target and REFIN2 are within ±20mV of each other and 2) an edge is detected on the error amplifier signifying that the output is in regulation. This prevents the system or internal fault protection from shutting down the controller during transitions.

Valley Current-Limit Protection

The current-limit circuit employs a unique “valley” current-sensing algorithm that senses the inductor current through the low-side MOSFET—across LX to AGND. If the current through the low-side MOSFET exceeds the valley current-limit threshold, the PWM controller is not allowed to initiate a new cycle. The actual peak current is greater than the valley current-limit threshold by an amount equal to the inductor ripple current. Therefore, the exact current-limit characteristic and maximum load capability are a function of the inductor value and battery voltage. When combined with the undervoltage protection circuit, this current-limit method is effective in almost every circumstance.

In forced-PWM mode, the MAX17020 also implements a negative current limit to prevent excessive reverse inductor currents when V

OUT

is sinking current. The negative current-limit threshold is set to approximately 120% of the positive current limit.

POR, UVLO

When VCCrises above the power-on reset (POR) threshold, the MAX17020 clears the fault latches, forces the low-side MOSFET to turn on (DL high), and resets the soft-start circuit, preparing the controller for power-up. However, the V

undervoltage lockout (UVLO) circuitry

inhibits switching until V

reaches 4.2V (typ). When VCCrises above 4.2V and the controller has been enabled (ON_ pulled high), the controller activates the enabled PWM controllers and initializes soft-start.

Figure 5. Dynamic REFIN Transition

REFIN

OUTPUT

VOLTAGE

PGOOD

OVP

20mV

BLANK HIGH-Z

REF + 140mV

±20mV WINDOW BETWEEN

INTERNAL TARGET AND REFIN2

INTERNAL EA TARGET = ACTUAL V

EA TARGET + 140mV

DYNAMIC REFIN WINDOW

20mV

OUT

BLANK HIGH-Z

EA TARGET + 140mV

Page 24

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

24 ______________________________________________________________________________________

When VCCdrops below the UVLO threshold (falling edge), the controller stops switching, and DH and DL are pulled low and a 10Ω switch discharges the outputs. When the 2V POR falling-edge threshold is reached, the DL state no longer matters since there is not enough voltage to force the switching MOSFETs into a low on-resistance state, so the controller pulls DL high, allowing a soft discharge of the output capacitors (damped response). However, if the VCCrecovers before reaching the falling POR threshold, DL remains low until the error comparator has been properly powered up and triggers an on-time. Only one enable input needs to be toggled to clear the fault latches and activate both outputs.

Soft-Start and Soft-Shutdown

The MAX17020 includes voltage soft-start and softshutdown—slowly ramping up and down the target voltage. During startup, the slew-rate control softly slews the preset/fixed target voltage over a 1ms startup period or its tracking voltage (REFIN2 < 2V) with a 1mV/μs slew rate. This long startup period reduces the inrush current during startup.

When ON1 or ON2 is pulled low or the output undervoltage fault latch is set, the respective output automatically enters soft-shutdown—the regulator enters PWM mode and ramps down its preset/fixed output voltage over a 1ms period or its tracking voltage (REFIN2 < 2V) with a 1mV/μs slew rate. After the output voltage drops below

0.1V, the MAX17020 pulls DL high, clamping the output and LX switching node to ground, preventing leakage currents from pulling up the output and minimizing the negative output voltage undershoot during shutdown.

Output Voltage

DC output-accuracy specifications in the

Electrical

Characteristics

table refer to the error comparator’s threshold. When the inductor continuously conducts, the MAX17020 regulates the valley of the output ripple, so the actual DC output voltage is lower than the slope-compensated trip level by 50% of the output ripple voltage. For PWM operation (continuous conduction), the output voltage is accurately defined by the following equation:

where V

NOM

is the nominal feedback voltage, A

CCV

the integrator’s gain, and V

RIPPLE

is the output ripple

voltage (V

RIPPLE

= ESR x ΔI

INDUCTOR

, as described in

the

Output Capacitor Selection

section).

In discontinuous conduction (I

OUT

< I

LOAD(SKIP)

), the longer off-times allow the slope compensation to increase the threshold voltage by as much as 1%, so

the output voltage regulates slightly higher than it would in PWM operation.

Internal Integrator

The internal integrator improves the output accuracy by removing any output accuracy errors caused by the slope compensation, output ripple voltage, and erroramplifier offset. Therefore, the DC accuracy (in forcedPWM mode) depends on the integrator’s gain, the integrator’s offset, and the accuracy of the integrator’s reference input.

Adjustable/Fixed Output Voltages

Connect FB1 to GND for fixed 5V operation. Connect FB1 to VCCfor fixed 1.5V operation. Connect FB1 to an external resistive voltage-divider from OUT1 to analog ground to adjust the output voltage between 0.7V and

5.5V. During soft-shutdown, application circuits configured for adjustable feedback briefly switch modes when FB1 drops below the 110mV dual-mode threshold.

Choose R

FBL

(resistance from FB1 to AGND) to be

approximately 49.9kΩ and solve for R

FBH

(resistance

from OUT1 to FB1) using the following equation:

Connect REFIN2 to VCCfor fixed 3.3V operation. Connect REFIN2 to RTC (3.3V) for fixed 1.05V operation. Connect REFIN2 to an external resistive voltage-divider from REF to analog ground to adjust the output voltage between 0V and 2V.

Choose R

REFINL

(resistance from REFIN2 to GND) to

be approximately 49.9kΩ and solve for R

REFINH

(resis-

tance from REF to REFIN2) using the equation:

Power-Good Outputs (PGOOD)

and Fault Protection

PGOOD is the open-drain output that continuously monitors the output voltage for undervoltage and overvoltage conditions. PGOOD_ is actively held low in shutdown (ON_ = GND), during soft-start or soft-shutdown. Approximately 20μs (typ) after the soft-start terminates, PGOOD_ becomes high impedance as long as the feedback voltage exceeds 85% of the nominal fixed-regulation voltage or within 150mV of the REFIN2 input voltage. PGOOD_ goes low if the feedback voltage drops 16% below the fixed target voltage, or if the output voltage drops 150mV below the dynamic REFIN2 voltage, or if the SMPS controller is shut down. For a

⎝

⎠

⎞ ⎠

⎟

⎛

OUT PWM NOM

()

RIPPLE

⎜ ⎝

CCV

=× −

FBH FBL

REFINH

=×−

REFINL

⎛

OUT

⎜

071.

⎛ ⎜

⎝

REF

OUT

⎞ ⎟

⎠

Page 25

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 25

logic-level PGOOD_ output voltage, connect an external pullup resistor between PGOOD_ and VDD. A 100kΩ pullup resistor works well in most applications.

Overvoltage Protection (OVP)

When the output voltage rises 16% above the fixed-regulation voltage or has risen 200mV above the dynamic REFIN2 input voltage, the controller immediately pulls the respective PGOOD_ low, sets the overvoltage fault latch, and immediately pulls the respective DL_ high— clamping the output to GND. Toggle either ON1 or ON2 input, or cycle VCCpower below its POR threshold to clear the fault latch and restart the controller.

Undervoltage Protection (UVP)

When the output voltage drops 30% below the fixedregulation voltage or has dropped 300mV below the dynamic REFIN2 input voltage, the controller immediately pulls the respective PGOOD_ low, sets the undervoltage fault latch, and begins the shutdown sequence. After the output voltage drops below 0.1V, the synchronous rectifier turns on, clamping the output to GND. Toggle either ON1 or ON2 input, or cycle VCCpower below its POR threshold to clear the fault latch and restart the controller.

Thermal-Fault Protection (TSHDN)

The MAX17020 features a thermal-fault protection circuit. When the junction temperature rises above +160°C, a thermal sensor activates the fault latch, pulls PGOOD1 and PGOOD2 low, enables the 10Ω discharge circuit, and disables the controller—DH and DL are pulled low. Toggle ONLDO or cycle IN power to reactivate the controller after the junction temperature cools by 15°C.

Design Procedure

Firmly establish the input-voltage range and maximum load current before choosing a switching frequency and inductor operating point (ripple-current ratio). The primary

design trade-off lies in choosing a good switching frequency and inductor operating point, and the following four factors dictate the rest of the design:

• Input Voltage Range: The maximum value (V

IN(MAX)

) must accommodate the worst-case, high

AC-adapter voltage. The minimum value (V

IN(MIN)

) must account for the lowest battery voltage after drops due to connectors, fuses, and battery-selector switches. If there is a choice at all, lower input voltages result in better efficiency.

• Maximum Load Current: There are two values to consider. The peak load current (I

LOAD(MAX)

) determines the instantaneous component stresses and filtering requirements and thus drives output capacitor selection, inductor saturation rating, and the design of the current-limit circuit. The continuous load current (I

LOAD

) determines the thermal stresses and thus drives the selection of input capacitors, MOSFETs, and other critical heat-contributing components.

• Switching Frequency: This choice determines the basic trade-off between size and efficiency. The optimal frequency is largely a function of maximum input voltage due to MOSFET switching losses that are proportional to frequency and V

. The optimum frequency is also a moving target due to rapid improvements in MOSFET technology that are making higher frequencies more practical.

• Inductor Operating Point: This choice provides trade-offs between size vs. efficiency and transient response vs. output ripple. Low inductor values provide better transient response and smaller physical size, but also result in lower efficiency and higher output ripple due to increased ripple currents. The minimum practical inductor value is one that causes the circuit to operate at the edge of critical conduction (where the inductor current just touches zero with every cycle at maximum load). Inductor values

Table 4. Fault Protection and Shutdown Operation Table

MODE CONTROLLER STATE DRIVER STATE

Shutdown (ON_ = High to Low); Output UVP (Latched)

Output OVP (Latched)

VCC UVLO Fa lli ng-Edge Thermal Fault (Latched)

VCC UVLO Rising Edge SMPS controller enabled (assum ing ON_ pulled high). DL driven high, DH pulled low.

VCC POR SMPS inactive, 10 output discharge active. DL driven high, DH pulled low.

Voltage soft-shutdown in it iated. Internal error-amplifier target slowly ramped down to GND and output active ly discharged (automatica lly enter s forced-PWM mode).

Controller shuts down and EA target internal ly slewed down. Controller remains off until ON_ toggled or V

SMPS controller disabled (assuming ON_ pulled high), 10 output discharge active.

power cycled.

DL driven high and DH pulled low after soft-shutdown completed (output < 0.1V).

DL immediately driven high, DH pulled low.

DL and DH pulled low.

Page 26

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

26 ______________________________________________________________________________________

lower than this grant no further size-reduction benefit. The optimum operating point is usually found between 20% and 50% ripple current. When pulse skipping (SKIP low and light loads), the inductor value also determines the load-current value at which PFM/PWM switchover occurs.

Inductor Selection

The switching frequency and inductor operating point determine the inductor value as follows:

For example: I

LOAD(MAX)

= 4A, VIN= 12V, V

OUT2

2.5V, fSW= 355kHz, 30% ripple current or LIR = 0.3:

Find a low-loss inductor having the lowest possible DC resistance that fits in the allotted dimensions. Ferrite cores are often the best choice, although powdered iron is inexpensive and can work well at 200kHz. The core must be large enough not to saturate at the peak inductor current (I

PEAK

Most inductor manufacturers provide inductors in standard values, such as 1.0μH, 1.5μH, 2.2μH, 3.3μH, etc. Also look for nonstandard values, which can provide a better compromise in LIR across the input voltage range. If using a swinging inductor (where the no-load inductance decreases linearly with increasing current), evaluate the LIR with properly scaled inductance values.

Transient Response

The inductor ripple current also impacts transientresponse performance, especially at low VIN- V

OUT

differentials. Low inductor values allow the inductor current to slew faster, replenishing charge removed from the output filter capacitors by a sudden load step. The amount of output sag is also a function of the maximum duty factor, which can be calculated from the ontime and minimum off-time:

where t

OFF(MIN)

is the minimum off-time (see the

Electrical Characteristics

table) and K is from Table 3.

The amount of overshoot during a full-load to no-load transient due to stored inductor energy can be calculated as:

Setting the Current Limit

The minimum current-limit threshold must be great enough to support the maximum load current when the current limit is at the minimum tolerance value. The valley of the inductor current occurs at I

LOAD(MAX)

minus

half the ripple current; therefore:

where I

LIM(VAL)

equals the minimum valley current-limit threshold voltage divided by the current-sense resistance (R

SENSE

). When using a 100kΩ ILIM resistor, the

minimum valley current-limit threshold is 40mV.

Connect a resistor between ILIM_ and analog ground (AGND) to set the adjustable current-limit threshold. The valley current-limit threshold is approximately 1/10 the ILIM voltage formed by the external resistance and internal 5μA current source. The 40kΩ to 400kΩ adjustment range corresponds to a 20mV to 200mV valley currentlimit threshold. When adjusting the current limit, use 1% tolerance resistors to prevent significant inaccuracy in the valley current-limit tolerance.

Output Capacitor Selection

The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and load-transient requirements, yet have high enough ESR to satisfy stability requirements.

For processor core voltage converters and other applications where the output is subject to violent load transients, the output capacitor’s size depends on how much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag due to finite capacitance:

In applications without large and fast load transients, the output capacitor’s size often depends on how much ESR is needed to maintain an acceptable level of output voltage ripple. The output ripple voltage of a stepdown controller equals the total inductor ripple current

()

VVV

Vf I LIR

IN SW LOAD MAX

×−

RI PPLE IN OUT

()

×× ×

VVV

×−

25 12 25

.( .)

VkHzA

×××

12 355 4 0 3

465

μH=

PEAK LOA D MAX

LIR

=×+

()

⎛ ⎜

⎝

⎞

⎟ ⎠

⎡

SAG

()

××

OUT OUT

⎛

() (2IIN

LOAD MAX

⎢

⎜

⎝

⎣

⎡

⎛

VV K

−

()

IN OUT

⎢

⎜ ⎝

⎢

⎣

⎞

OUT

⎟

⎠

+Δ

⎞ ⎟

⎠

OFF M

−2

OFF MIN()

)

⎤ ⎥ ⎦

SOA R

()

≈

()

LOAD MAX

××

OUT OUT

LIM VAL LOAD MAX

>−

() ( )

⎛ ⎜

⎝

ILIR

LOAD MAX

()

≤

LOAD MAX

STEP

()

ESR

⎤ ⎥ ⎥

⎦

⎞ ⎟

⎠

Page 27

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 27

multiplied by the output capacitor’s ESR. Therefore, the maximum ESR required to meet ripple specifications is:

The actual capacitance value required relates to the physical size needed to achieve low ESR, as well as to the chemistry of the capacitor technology. Thus, the capacitor is usually selected by ESR and voltage rating rather than by capacitance value (this is true of tantalums, OS-CONs, polymers, and other electrolytics).

When using low-capacity filter capacitors, such as ceramic capacitors, size is usually determined by the capacity needed to prevent V

and V

OAR

from causing problems during load transients. Generally, once enough capacitance is added to meet the overshoot requirement, undershoot at the rising load edge is no longer a problem (see the V

and V

OAR

equa-

tions in the

Transient Response

section). However, lowcapacity filter capacitors typically have high ESR zeros that could affect the overall stability (see the

Output

Capacitor Stability Considerations

section).

Output Capacitor Stability Considerations

For Quick-PWM controllers, stability is determined by the value of the ESR zero relative to the switching frequency. The boundary of instability is given by the following equation:

where:

For a typical 300kHz application, the ESR zero frequency must be well below 95kHz, preferably below 50kHz. Tantalum and OS-CON capacitors in widespread use at the time of publication have typical ESR zero frequencies of 25kHz. In the design example used for inductor selection, the ESR needed to support 25mV

P-P

ripple is 25mV/1.2A = 20.8mΩ. One 220μF/4V SANYO polymer (TPE) capacitor provides 15mΩ (max) ESR. This results in a zero at 48kHz, well within the bounds of stability.

Do not put high-value ceramic capacitors directly across the feedback sense point without taking precautions to ensure stability. Large ceramic capacitors can have a high ESR zero frequency and cause erratic, unstable operation. However, it is easy to add enough series resistance by placing the capacitors a couple of

inches downstream from the feedback sense point, which should be as close as possible to the inductor.

Unstable operation manifests itself in two related, but distinctly different ways: double-pulsing and fast-feedback loop instability. Double-pulsing occurs due to noise on the output or because the ESR is so low that there is not enough voltage ramp in the output voltage signal. This “fools” the error comparator into triggering a new cycle immediately after the 400ns minimum offtime period has expired. Double-pulsing is more annoying than harmful, resulting in nothing worse than increased output ripple. However, it can indicate the possible presence of loop instability due to insufficient ESR. Loop instability results in oscillations at the output after line or load steps. Such perturbations are usually damped, but can cause the output voltage to rise above or fall below the tolerance limits.

The easiest method for checking stability is to apply a very fast zero-to-max load transient and carefully observe the output voltage ripple envelope for overshoot and ringing. It can help to simultaneously monitor the inductor current with an AC current probe. Do not allow more than one cycle of ringing after the initial step-response under/overshoot.

Input Capacitor Selection

The input capacitor must meet the ripple current requirement (I

RMS

) imposed by the switching currents:

For most applications, nontantalum chemistries (ceramic, aluminum, or OS-CON) are preferred due to their resistance to power-up surge currents typical of systems with a mechanical switch or connector in series with the input. If the MAX17020 is operated as the second stage of a two-stage power conversion system, tantalum input capacitors are acceptable. In either configuration, choose a capacitor that has less than 10°C temperature rise at the RMS input current for optimal reliability and lifetime.

Power-MOSFET Selection

Most of the following MOSFET guidelines focus on the challenge of obtaining high load-current capability when using high-voltage (> 20V) AC adapters. Low-current applications usually require less attention.

The high-side MOSFET (NH) must be able to dissipate the resistive losses plus the switching losses at both V

IN(MIN)

and V

IN(MAX)

. Ideally, the losses at V

IN(MIN)

should be roughly equal to the losses at V

IN(MAX)

, with

lower losses in between. If the losses at V

IN(MIN)

are

≤

ESR

RI PPLE

ILIR

LOAD MAX

()

≤

ESR

2π

××

ESR O UT

RMS LOAD

⎛

VVV

=×

OUT IN OUT

⎜ ⎝

−

()

⎞ ⎟

⎠

Page 28

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

28 ______________________________________________________________________________________

significantly higher, consider increasing the size of NH. Conversely, if the losses at V

IN(MAX)

are significantly higher, consider reducing the size of NH. If VINdoes not vary over a wide range, maximum efficiency is achieved by selecting a high-side MOSFET (NH) that has conduction losses equal to the switching losses.

Choose a low-side MOSFET (NL) that has the lowest possible on-resistance (R

DS(ON)

), comes in a moder-

ate-sized package (i.e., 8-pin SO, DPAK, or D

PAK), and is reasonably priced. Ensure that the MAX17020 DL_ gate driver can supply sufficient current to support the gate charge and the current injected into the parasitic drain-to-gate capacitor caused by the high-side MOSFET turning on; otherwise, cross-conduction problems might occur. Switching losses are not an issue for the low-side MOSFET since it is a zero-voltage switched device when used in the step-down topology.

Power-MOSFET Dissipation

Worst-case conduction losses occur at the duty factor extremes. For the high-side MOSFET (NH), the worstcase power dissipation due to resistance occurs at minimum input voltage:

Generally, use a small, high-side MOSFET to reduce switching losses at high input voltages. However, the R

DS(ON)

required to stay within package power-dissipation often limits how small the MOSFET can be. The optimum occurs when the switching losses equal the conduction (R

DS(ON)

) losses. High-side switching losses do not become an issue until the input is greater than approximately 15V.

Calculating the power dissipation in high-side MOSFETs (NH) due to switching losses is difficult, since it must allow for difficult-to-quantify factors that influence the turn-on and turn-off times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PCB layout characteristics. The following switching loss calculation provides only a very rough estimate and is no substitute for breadboard evaluation, preferably including verification using a thermocouple mounted on N

where C

OSS

is the high-side MOSFET’s output capaci-

tance, Q

G(SW)

is the charge needed to turn on the high-

side MOSFET, and I

GATE

is the peak gate-drive

source/sink current (1A typ).

Switching losses in the high-side MOSFET can become a heat problem when maximum AC adapter voltages are applied due to the squared term in the switchingloss equation provided above. If the high-side MOSFET chosen for adequate R

DS(ON)

at low battery voltages becomes extraordinarily hot when subjected to V

IN(MAX)

, consider choosing another MOSFET with

lower parasitic capacitance.

For the low-side MOSFET (NL), the worst-case power dissipation always occurs at maximum battery voltage:

The absolute worst case for MOSFET power dissipation occurs under heavy overload conditions that are greater than I

LOAD(MAX)

, but are not high enough to exceed the current limit and cause the fault latch to trip. To protect against this possibility, “overdesign” the circuit to tolerate:

where I

VALLEY(MAX)

is the maximum valley current allowed by the current-limit circuit, including threshold tolerance and sense-resistance variation. The MOSFETs must have a relatively large heatsink to handle the overload power dissipation.

Choose a Schottky diode (DL) with a forward voltage drop low enough to prevent the low-side MOSFET’s body diode from turning on during the dead time. As a general rule, select a diode with a DC current rating equal to 1/3 the load current. This diode is optional and can be removed if efficiency is not critical.

Applications Information

Step-Down Converter Dropout

Performance

The output-voltage adjustable range for continuousconduction operation is restricted by the nonadjustable minimum off-time one-shot. For best dropout performance, use the slower (200kHz) on-time setting. When working with low input voltages, the duty-factor limit must be calculated using worst-case values for on- and off-times. Manufacturing tolerances and internal propagation delays introduce an error to the TON K-factor. This error is greater at higher frequencies (Table 3).

PD N sistive

(Re )=

⎛ ⎜

⎝

VIfQ

⎛

PD N Switching

()

() ()

⎜ ⎝

⎛

⎜ ⎝

⎞

OUT

()

⎟

⎠

×××

MAX LOAD SW G SW

××

IIN OSS SW

LOAD D

TTE

SSON()

⎞ ⎟

⎠

⎞ ⎟

⎠

PD N sistiv e

(Re )

⎡

⎛

⎢

=−

⎜

⎢

⎝

⎣

IN MAX

⎤

⎞

OUT

⎥⎥

⎟

⎥

⎠

()

⎦

()

LOAD DS ON2()

×IR

≈+

LOAD VALLEY MAX

()

ILIR

⎛

LOAD MAX

⎜ ⎝

()

⎞ ⎟

⎠

Page 29

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 29

Also, keep in mind that transient response performance of buck regulators operated too close to dropout is poor, and bulk output capacitance must often be added (see the V

equation in the

Transient Response

section).

The absolute point of dropout is when the inductor current ramps down during the minimum off-time (ΔI

DOWN

) as much as it ramps up during the on-time (ΔIUP). The ratio h = ΔIUP/ΔI

DOWN

indicates the controller’s ability to slew the inductor current higher in response to increased load, and must always be greater than 1. As h approaches 1, the absolute minimum dropout point, the inductor current cannot increase as much during each switching cycle, and V

greatly increases

unless additional output capacitance is used.

A reasonable minimum value for h is 1.5, but adjusting this up or down allows trade-offs between V

, output capacitance, and minimum operating voltage. For a given value of h, the minimum operating voltage can be calculated as:

where V

CHG

is the parasitic voltage drop in the charge

path (see the

On-Time One-Shot

section), t

OFF(MIN)

from the

Electrical Characteristics

table, and K (1/fSW) is taken from Table 3. The absolute minimum input voltage is calculated with h = 1.

If the calculated V

IN(MIN)

is greater than the required minimum input voltage, operating frequency must be reduced or output capacitance added to obtain an acceptable V

. If operation near dropout is antici-

pated, calculate V

to be sure of adequate transient

response.

Dropout Design Example:

OUT2

= 2.5V

fSW= 355kHz

K = 3.0μs, worst-case K

MIN

= 3.3μs

OFF(MIN)

= 500ns

CHG

= 100mV

h = 1.5:

Calculating again with h = 1 and the typical K-factor value (K = 3.3μs) gives the absolute limit of dropout:

Therefore, VINmust be greater than 3.06V, even with very large output capacitance, and a practical input voltage with reasonable output capacitance would be 3.47V.

Dynamic Output Voltage Settings

(OUT2 Only)

The second output (OUT2) of the MAX17020 works with applications that require multiple dynamic output voltages, easily supporting two to four output voltages with external resistors selected by control FETs or REFIN2 can be driven by a DAC for tight voltage control.

Figure 6 shows an application circuit providing four voltage levels using discrete components. Switching resistors in and out of the resistor network changes the voltage at REFIN2. The reference input automatically detects large input voltage transitions and blanks the fault and PGOOD2 comparators, allowing the system to perform the transition without tripping the fault protection.

IN MIN

) =

OUT CHG

⎛

OFF MIN(()

−

⎜

⎝

⎞ ⎟

⎠

25 01

IN MIN(

)

⎛

1 5 500

−

⎜ ⎝

25 01

⎛

1 500

−

⎜ ⎝

μs

⎞ ⎟

⎠

⎞ ⎟

⎠

3..47V

3066V

Page 30

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

30 ______________________________________________________________________________________

Figure 6. Dynamic Output Application Circuit—Graphics Supply

NOTE: PLACE C22 BETWEEN IN AND PGND AS CLOSE AS POSSIBLE TO THE MAX17020.

1.2V OUTPUT

OUT1

BST1

0.1μF

C22

0.1μF

DH1

BST1

LX1

DL1

OUT1

DH2

BST2

LX2

DL2

PGND AGND

OUT2

0.1μF

RGND

0Ω

BST2

INPUT (V

5V TO 24V

OUT2

0.8V/1.2V GPU SUPPLY

5V SYSTEM SUPPLY

POWER GROUND

49.9kΩ

7.15kΩ

R2 10kΩ

1μF

4.7μF

1.0μF

150kΩ

R10

47Ω

1.0μF

ILIM1

MAX17020

FB1

BYP

RTC

LDO

LDOREFIN

SECFB

ILIM1

PAD

REF

REFIN2

SKIP

PGOOD1 PGOOD2

TON

ON1 ON2

ONLDO

ILIM2

100kΩ

ILIM2

0.1μF

80.6kΩ

R4 118kΩ

OUT1/OUT2 SWITCHING FREQUENCY OPEN (REF): 400kHz/300kHz

100kΩ

R7 100kΩ

OFFON

SLEEP

3.3 SYSTEM SUPPLY

POWER-GOOD

}

ANALOG GROUND

Page 31

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 31

PCB Layout Guidelines

Careful PCB layout is critical to achieving low switching losses and clean, stable operation. The switching power stage requires particular attention. If possible, mount all the power components on the top side of the board, with their ground terminals flush against one another. Follow these guidelines for good PCB layout:

• Keep the high-current paths short, especially at the

ground terminals. This practice is essential for stable, jitter-free operation.

• Keep the power traces and load connections short.

This practice is essential for high efficiency. Using thick copper PCBs (2oz vs. 1oz) can enhance fullload efficiency by 1% or more. Correctly routing PCB traces is a difficult task that must be approached in terms of fractions of centimeters, where a single milliohm of excess trace resistance causes a measurable efficiency penalty.

• Minimize current-sensing errors by connecting LX_

directly to the drain of the low-side MOSFET.

• When trade-offs in trace lengths must be made, it is

preferable to allow the inductor charging path to be made longer than the discharge path. For example, it is better to allow some extra distance between the input capacitors and the high-side MOSFET than to allow distance between the inductor and the lowside MOSFET or between the inductor and the output filter capacitor.

• Route high-speed switching nodes (BST_, LX_,

DH_, and DL_) away from sensitive analog areas (REF, FB_, and OUT_).

A sample layout is available in the MAX17020 evaluation kit data sheet.

Layout Procedure

1) Place the power components first, with ground terminals adjacent (NL_source, CIN, C

OUT_

, and D

anode). If possible, make all these connections on the top layer with wide, copper-filled areas.

2) Mount the controller IC adjacent to the low-side MOSFET, preferably on the back side opposite N

and NH_to keep LX_, GND, DH_, and the DL_ gatedrive lines short and wide. The DL_ and DH_ gate traces must be short and wide (50 mils to 100 mils wide if the MOSFET is 1in from the controller IC) to keep the driver impedance low and for proper adaptive dead-time sensing.

3) Group the gate-drive components (BST_ capacitor, V

bypass capacitor) together near the controller IC.

4) Make the DC-DC controller ground connections as shown in Figures 1 and 6. This diagram can be viewed as having two separate ground planes: power ground, where all the high-power components go; and an analog ground plane for sensitive analog components. The analog ground plane and power ground plane must meet only at a single point directly at the IC.

5) Connect the output power planes directly to the output filter capacitor positive and negative terminals with multiple vias. Place the entire DC-DC converter circuit as close to the load as is practical.

Table 5. MAX17020 vs. MAX8778 Design Differences

MAX17020 MAX8778

RTC power-up required for controller operation. LDO and switching regulators independent of RTC operation.

Page 32

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

32 ______________________________________________________________________________________

Figure 7. Standard Output Application Circuit—Chipset Supply

NOTE: PLACE C22 BETWEEN IN AND PGND AS CLOSE AS POSSIBLE TO THE MAX17020.

1.5V OUTPUT

OUT1

5V SYSTEM SUPPLY

3.3V SMPS SUPPLY

11.3kΩ

10kΩ

R10

47Ω

1.0μF

0.1μF

4.7μF

1μF

0.1μF

BST1

C22

DH1

BST1

LX1

DL1

OUT1

DH2

BST2

LX2

DL2

PGND

AGND

OUT2

MAX17020

FB1

SECFB

LDOREFIN

BYP

PGOOD1 PGOOD2

REFIN2

RTC

0.1μF

REF

SKIP

100kΩ

0.1μF

RGND

0Ω

BST2

R7 100kΩ

C3 1μF

2x 10μF 25V

INPUT (V

7V TO 24V

1.05V OUTPUT

OUT2

3.3 SMPS SUPPLY

POWER-GOOD

}

RTC SUPPLY

3.3V LDO OUTPUT

POWER GROUND

ANALOG GROUND

4.7μF

ILIM1

LDO

ILIM1

PAD

ON1 ON2

ONLDO

TON

ILIM2

OFFON

OUT1/OUT2 SWITCHING FREQUENCY

OPEN (REF): 400kHz/300kHz

ILIM2

Page 33

MAX17020

Dual Quick-PWM Step-Down Controller

with Low-Power LDO, RTC Regulator

______________________________________________________________________________________ 33

Package Information

For the latest package outline information, go to

www.maxim-ic.com/packages

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

32 TQFN T3255-3

21-0140

Chip Information

PROCESS: BiCMOS

Page 34

MAX17020

Dual Quick-PWM Step-Down Controller with Low-Power LDO, RTC Regulator

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

Revision History

REVISION

NUMBER

0 5/08 Initial release —

1 9/08 Added three new TOCs, various changes throughout 3, 8, 12–15, 24, 25, 26, 33

2 2/09 Minor edits to EC table and text additions 1, 5, 7, 14, 15, 23

REVISION

DATE

DESCRIPTION

PAGES

CHANGED

Datasheet MAX17020 Datasheet (Maxim Integrated)

Specifications and Main Features

Frequently Asked Questions

User Manual