Datasheet LTC3716 Datasheet (LINEAR TECHNOLOGY)

FEATURES

LTC3716

2-Phase, 5-Bit VID,

Current Mode, High Efficiency,

Synchronous Step-Down

Switching Regulator

U

DESCRIPTIO

■

Output Stages Operate Antiphase Reducing Input
and Output Capacitance Requirements and Power
Supply Induced Noise

■

Dual Input Supply Capability for Load Sharing

■

5-Bit Mobile VID Code: V

■

±1% Output Voltage Accuracy

■

True Remote Sensing Differential Amplifier

■

Power Good Output Voltage Monitor

■

Supports Active Voltage Positioning

■

Current Mode Control Ensures Current Sharing

■

OPTI-LOOPTM Compensation Minimizes C

■

Three Operational Modes: PWM, Burst and Cycle Skip

■

Programmable Fixed Frequency: 150kHz to 300kHz

■

Wide VIN Range: 4V to 36V Operation

■

Adjustable Soft-Start Current Ramping

■

Internal Current Foldback and Short-Circuit Shutdown

■

Overvoltage Soft Latch Eliminates Nuisance Trips

■

Available in 36-Lead Narrow SSOP Package

= 0.6V to 1.75V

OUT

OUT

U

APPLICATIO S

■

Mobile Computer CPU Supply

, LTC and LT are registered trademarks of Linear Technology Corporation.

OPTI-LOOP and Burst Mode are trademarks of Linear Technology Corporation.

The LTC®3716 is a 2-phase, VID programmable, synchro­nous step-down switching regulator controller that drives
two N-channel external power MOSFET stages in a fixed fre­quency architecture. The 2-phase controller drives its two
output stages out of phase at frequencies up to 300kHz to
minimize the RMS ripple currents in both input and output
capacitors. The 2-phase technique effectively multiplies the
fundamental frequency by two, improving transient re­sponse while operating each channel at an optimum fre­quency for efficiency. Thermal design is also simplified.

An operating mode select pin (FCB) can be used to regu­late a secondary winding or select among three modes
including Burst ModeTM operation for highest efficiency. An
internal differential amplifier provides true remote sensing
of the regulated supply’s positive and negative output ter­minals as required in high current applications.

The RUN/SS pin provides soft-start and optional timed,
short-circuit shutdown. Current foldback limits MOSFET
dissipation during short-circuit conditions when the
overcurrent latchoff is disabled. OPTI-LOOP compensation
allows the transient response to be optimized for a wide
range of output capacitors and ESR values.

TYPICAL APPLICATIO

0.1µF

3.3k

5 VID BITS

220pF

FCB
RUN/SS

I

TH

SGND
PGOOD

VID0–VID4

EAIN
ATTENOUT
ATTENIN
V

DIFFOUT

V

OS

V

OS

U

10µF

V

IN

TG1

SW1

BG1

PGND

TG2

SW2

BG2

S

0.47µF

S

+
–

0.47µF

CC

+
–

+

10µF

–
+

LTC3716

BOOST1

SENSE1
SENSE1

BOOST2

INTV
SENSE2
SENSE2

Figure 1. High Current Dual Phase Step-Down Converter

35V
×4

1µH

D1

0.002Ω

1µH

D2

0.002Ω

V

IN

5V TO 28V

21

43

V

+

OUT

0.6V TO 1.75V
40A

C

OUT

1000µF
4V
×2

3716 F01

21

43

1

LTC3716

WW

W

U

ABSOLUTE AXI U RATI GS

(Note 1)

Input Supply Voltage (VIN).........................36V to –0.3V

Topside Driver Voltages (BOOST1,2).........42V to –0.3V

Switch Voltage (SW1, 2) .............................36V to –5 V

SENSE1+, SENSE2+, SENSE1–,

SENSE2– Voltages ...................(1.1)INTVCC to –0.3V

EAIN, V

V

VID0–VID4, Voltages ...............................7V to –0.3V

Boosted Driver Voltage (BOOST-SW) ..........7V to –0.3V

PLLFLTR, PLLIN, V

FCB Voltages ................................... INTVCC to –0.3V

ITH Voltage................................................2.7V to –0.3V

Peak Output Current <1µs(TG1, 2, BG1, 2)................ 3A

INTVCC RMS Output Current................................ 50mA

Operating Ambient Temperature Range

(Note 2) .............................................. –40°C to 85°C

Junction Temperature (Note 3)............................. 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)..................300°C

+

–

, V

OS

, ATTENIN, ATTENOUT, PGOOD, AMPMD,

BIAS

, EXTVCC, INTVCC, RUN/SS,

OS

DIFFOUT

,

UUW

PACKAGE/ORDER I FOR ATIO

RUN/SS
SENSE1
SENSE1

EAIN

PLLFLTR

PLLIN

FCB

I

SGND

V

DIFFOUT

V

OS

V

OS

SENSE2
SENSE2

ATTENOUT

ATTENIN

VID0
VID1

TOP VIEW

1

+

2

–

3
4
5
6
7
8

TH

9

10

–

11

+

12

–

13

+

14
15
16
17
18

G PACKAGE

36-LEAD PLASTIC SSOP

T

= 125°C, θJA = 85°C/W

JMAX

36
35
34
33
32
31
30
29
28
27
26
25
24
23
22
21
20
19

PGOOD
TG1
SW1
BOOST1
V

IN

BG1
EXTV

CC

INTV

CC

PGND
BG2
BOOST2
SW2
TG2
AMPMD
V

BIAS

VID4
VID3
VID2

ORDER PART

NUMBER

LTC3716EG

Consult factory for parts specified with wider operating temperature ranges.

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, V

The ● denotes the specifications which apply over the full operating

BIAS

= 5V, V

= 5V unless otherwise noted.

RUN/SS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Main Control Loop

V

EAIN

V

SENSEMAX

I

INEAIN

V

LOADREG

V

REFLNREG

V

FCB

I

FCB

V

BINHIBIT

Regulated Feedback Voltage ITH Voltage = 1.2V; Measured at V

(Note 4) ● 0.594 0.600 0.606 V

EAIN

Maximum Current Sense Threshold ● 62 75 88 mV
Feedback Current (Note 4) –5 –50 nA
Output Voltage Load Regulation (Note 4)

Measured in Servo Loop, ∆I
Measured in Servo Loop, ∆I

Voltage: 1.2V to 0.7V ● 0.1 0.5 %

TH

Voltage: 1.2V to 2V ● –0.1 –0.5 %

TH

Reference Voltage Line Regulation VIN = 3.6V to 30V (Note 4) 0.002 0.02 %/V
Forced Continuous Threshold ● 0.57 0.6 0.63 V
Forced Continuous Current – 0.17 –1 µA
Burst Inhibit (Constant Frequency) Measured at FCB pin 4.3 4.8 V

Threshold

V

OVL

Output Overvoltage Threshold Measured at V

EAIN

● 0.64 0.66 0.68 V

UVLO Undervoltage Lockout VIN Ramping Down 3 3.33 4 V
g
g

m

mOL

Transconductance Amplifier g

m

ITH = 1.2V, Sink/Source 5µA (Note 4) 3 mmho

Transconductance Amplifier Gain ITH = 1.2V, (gm • ZL; No Ext Load) (Note 4) 1.5 V/mV

2

LTC3716

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, V

The ● denotes the specifications which apply over the full operating

BIAS

= 5V, V

= 5V unless otherwise noted.

RUN/SS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

I

Q

Input DC Supply Current (Note 5)

Normal Mode 1.2 mA

= 0V 20 40 µA

RUN/SS

= 1.9V –0.5 –1.2 µA

RUN/SS

Rising 1.0 1.5 1.9 V

RUN/SS

Rising from 3V 4.1 4.5 V

RUN/SS

= 0.5V, V

EAIN

= 0.5V 1.6 5 µA

EAIN

SENSE1–, 2

– = V

SENSE1+, 2

= 4.5V 0.5 2 4 µA

RUN/SS

+ = 0V –85 –60 µA

I

RUN/SS

V

RUN/SS

V

RUN/SSLO

I

SCL

I

SDLHO

I

SENSE

DF

MAX

Shutdown V
Soft-Start Charge Current V
RUN/SS Pin ON Threshold V
RUN/SS Pin Latchoff Arming V
RUN/SS Discharge Current Soft Short Condition V
Shutdown Latch Disable Current V
Total Sense Pins Source Current Each Channel: V
Maximum Duty Factor In Dropout 98 99.5 %
Top Gate Transition Time: (Note 6)

TG1, 2 t
TG1, 2 t

Rise Time C

r

Fall Time C

f

= 3300pF 30 90 ns

LOAD

= 3300pF 40 90 ns

LOAD

Bottom Gate Transition Time: (Note 6)

BG1, 2 t

Rise Time C

r

BG1, 2 tf Fall Time C
TG/BG t

Top Gate Off to Bottom Gate On Delay C

1D

= 3300pF 30 90 ns

LOAD

= 3300pF 20 90 ns

LOAD

= 3300pF Each Driver (Note 6) 90 ns

LOAD

Synchronous Switch-On Delay Time

BG/TG t

Bottom Gate Off to Top Gate On Delay C

2D

= 3300pF Each Driver (Note 6) 90 ns

LOAD

Top Switch-On Delay Time

t

ON(MIN)

Minimum On-Time Tested with a Square Wave (Note 7) 180 ns

Internal VCC Regulator

V

INTVCC

V

LDO

V

LDO

V

EXTVCC

V

LDOHYS

Internal VCC Voltage 6V < VIN < 30V, V

INT INTVCC Load Regulation ICC = 0 to 20mA, V
EXT EXTVCC Voltage Drop ICC = 20mA, V

EXTVCC

EXTVCC Switchover Voltage ICC = 20mA, EXTV
EXTVCC Switchover Hysteresis ICC = 20mA, EXTV

= 4V 4.8 5.0 5.2 V

EXTVCC

= 4V 0.2 1.0 %

EXTVCC

= 5V 80 160 mV
Ramping Positive ● 4.5 4.7 V

CC

Ramping Negative 0.2 V

CC

VID Parameters

V

BIAS

R

ATTEN

Operating Supply Voltage Range 2.7 5.5 V
Resistance Between ATTENIN 10 kΩ

and ATTENOUT Pins

ATTEN
R

PULLUP

VID

THLOW

VID

THHIGH

VID

LEAK

Resistive Divider Error ● – 0.25 0.25 %

ERR

VID0 to VID4 Pull-Up Resistance (Note 8) 40 kΩ
VID0 to VID4 Logic Threshold Low 0.4 V
VID0 to VID4 Logic Threshold High 1.6 V
VID0 to VID4 Leakage V

< VID0–VID4 < 7V ±1 µA

BIAS

Oscillator and Phase-Locked Loop

f

NOM

f

LOW

f

HIGH

R

PLLIN

I

PLLFLTR

R

RELPHS

Nominal Frequency V
Lowest Frequency V
Highest Frequency V

= 1.2V 190 220 250 kHz

PLLFLTR

= 0V 120 140 160 kHz

PLLFLTR

≥ 2.4V 280 310 360 kHz

PLLFLTR

PLLIN Input Resistance 50 kΩ
Phase Detector Output Current

Sinking Capability f

Sourcing Capability f

PLLIN
PLLIN

< f
> f

OSC
OSC

–15 µA

15 µA

Controller 2-Controller 1 Phase 180 Deg

3

LTC3716

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VIN = 15V, V

The ● denotes the specifications which apply over the full operating

BIAS

= 5V, V

= 5V unless otherwise noted.

RUN/SS

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
PGOOD Output

V

PGL

I

PGOOD

V

PG

PGOOD Voltage Low I
PGOOD Leakage Current V
PGOOD Trip Level, Either Controller V

= 2mA 0.1 0.3 V

PGOOD

= 5V ±1 µA

PGOOD

with Respect to Set Output Voltage

EAIN

Ramping Negative –8 – 10 – 12 %

V

EAIN

Ramping Positive 8 10 12 %

V

EAIN

Differential Amplifier/Op Amp Gain Block

A

DA

CMRR
R

IN

V

OS

I

B

A

OL

V

CM

CMRR
PSRR
I

CL

V

OMAX

GBW Gain-Bandwidth Product Op Amp Mode; I
SR Slew Rate Op Amp Mode; RL = 2k, V

Differential Amplifier Gain V
Common Mode Rejection Ratio 0V < VCM < 5V; V

DA

= 0V 0.995 1 1.005 V/V

AMPMD

Input Resistance Measured at VOS+ Input; V
Input Offset Voltage Op Amp Mode; VCM = 2.5V, V

= 5V; I

V

DIFFOUT

Input Bias Current Op Amp Mode; V
Open-Loop DC Gain Op Amp Mode; 0.7V ≤ V
Common Mode Input Voltage Range Op Amp Mode; V
Common Mode Rejection Ratio Op Amp Mode; 0V < VCM < 3V, V

OA

Power Supply Rejection Ratio Op Amp Mode; 6V < VIN < 30V, V

OA

Maximum Output Current Op Amp Mode; V
Maximum Output Voltage Op Amp Mode; I

= 0V 46 55 dB

AMPMD

= 0V 80 kΩ

AMPMD

= 5V 6 mV

AMPMD

= 1mA

DIFFOUT

= 5V 30 200 nA

AMPMD

< 10V, V

DIFFOUT

= 5V 0 3 V

AMPMD

AMPMD

AMPMD

= 0V, V

DIFFOUT

DIFFOUT

DIFFOUT

AMPMD

= 1mA, V
= 1mA, V

AMPMD

AMPMD

= 5V 5 V/µs

AMPMD

= 5V 5000 V/mV

AMPMD

= 5V 70 90 dB

= 5V 70 90 dB

= 5V 10 35 mA

= 5V 10 11 V
= 5V 2 MHz

Note 1: Absolute Maximum Ratings are those values beyond which the
life of a device may be impaired.

Note 2: The LTC3716 is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.

Note 3: T
dissipation P

is calculated from the ambient temperature TA and power

J

according to the following formula:

D

LTC3716EG: TJ = TA + (PD • 85°C/W)

Note 4: The LTC3716 is tested in a feedback loop that servos V
specified voltage and measures the resultant V

EAIN

.

ITH

to a

Note 5: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.

Note 6: Rise and fall times are measured using 10% and 90% levels. Delay
times are measured using 50% levels.

Note 7: The minimum on-time condition corresponds to the on inductor
peak-to-peak ripple current ≥40% I

(see Minimum On-Time

MAX

Considerations in the Applications Information section).
Note 8: Each built-in pull-up resistor attached to the VID inputs also has a

series diode to allow input voltages higher than the VIDV

supply without

CC

damage or clamping (see the Applications Information section).

4

UW

TEMPERATURE (

°C)

–50

INTV

CC

AND EXTV

CC

SWITCH VOLTAGE (V)

4.95

5.00

5.05

25 75

3716 G06

4.90

4.85

–25 0

50 100 125

4.80

4.70

4.75

INTVCC VOLTAGE

EXTVCC SWITCHOVER THRESHOLD

TYPICAL PERFOR A CE CHARACTERISTICS

LTC3716

Efficiency vs Load Current
(3 Operating Modes) (Figure 13)

100

Burst Mode

90

OPERATION

80
70
60
50
40

EFFICIENCY (%)

30
20
10

0

0.01

FORCED
CONTINUOUS
MODE

CONSTANT
FREQUENCY
(BURST DISABLE)

0.1

1

LOAD CURRENT (A)

V

= 5V

IN

V

= 1.6V

OUT

FREQ = 200kHz

3716 G01

10010

Efficiency vs Load Current
(Figure 13)

100

80

60

40

EFFICIENCY (%)

20

0

0.1

Supply Current vs Input Voltage
and Mode EXTVCC Voltage Drop

1000

800

600

ON

250

200

150

VIN = 5V
VIN = 8V
VIN = 12V
VIN = 20V

V

= 1.6V

OUT

V

= 0V

EXTVCC

FREQ = 200kHz

= 0V

V

FCB

1 10 100

LOAD CURRENT (A)

3716 G02

Efficiency vs Input Voltage
(Figure 13)

100

I

= 20A

OUT

= 1.6V

V

OUT

90

80

70

EFFICIENCY (%)

60

50

5

10

INPUT VOLTAGE (V)

INTVCC and EXTVCC Switch
Voltage vs Temperature

15

20

3716 G03

400

SUPPLY CURRENT (µA)

200

0

05

5.1

5.0

4.9

4.8

VOLTAGE (V)

4.7

CC

INTV

4.6

4.5

4.4
0

10

INPUT VOLTAGE (V)

I

= 1mA

LOAD

510

INPUT VOLTAGE (V)

SHUTDOWN

20

15

20 30 35

15 25

100

VOLTAGE DROP (mV)

CC

EXTV

50

0

10

30

35

3716 G04

25

0

CURRENT (mA)

30

40

20

50

3716 G05

Maximum Current Sense Threshold
vs Percent of Nominal Output
Voltage (Foldback)Internal 5V LDO Line Reg

80

70

60

50

(mV)

40

SENSE

V

30

20

10

0

0

PERCENT OF NOMINAL OUTPUT VOLTAGE (%)

25

50

75

100

3716 G09

3716 G07

Maximum Current Sense Threshold
vs Duty Factor

75

50

(mV)

SENSE

V

25

0

0

20 40 60 80

DUTY FACTOR (%)

100

3716 G08

5

LTC3716

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Maximum Current Sense Threshold

(mV)

SENSE

V

20

80

60

40

0

vs V

V

SENSE(CM)

0

(Soft-Start)

RUN/SS

= 1.6V

1234

V

(V)

RUN/SS

56

3716 G10

Load Regulation V

0.0

–0.1

(%)

OUT

–0.2

NORMALIZED V

–0.3

FCB = 0V

= 15V

V

IN

FIGURE 1

Maximum Current Sense Threshold
vs Sense Common Mode Voltage

80

76

72

(mV)

SENSE

68

V

64

(V)

ITH

V

60

2.5

2.0

1.5

1.0

0.5

1

0

COMMON MODE VOLTAGE (V)

vs V

ITH

RUN/SS

V

= 0.7V

OSENSE

3

2

(Soft-Start)

Current Sense Threshold
vs ITH Voltage

90
80
70
60
50
40

(mV)

30
20

SENSE

V

10

0
–10
–20

4

5

3716 G11

–30

0.5

0

1.5

2

1

V

(V)

ITH

2.5

3716 G12

SENSE Pins Total Source Current

100

50

(µA)

0

SENSE

I

–50

–0.4

5

0

10

LOAD CURRENT (A)

Maximum Current Sense
Threshold vs Temperature

80

78

76

(mV)

SENSE

74

V

72

70

–50 –25

25

0

TEMPERATURE (°C)

15

20

25

3716 G13

0

0

234

1

V

RUN/SS

(V)

56

3716 G14

RUN/SS Current vs Temperature

1.8

1.6

1.4

1.2

1.0

0.8

0.6

RUN/SS CURRENT (µA)

0.4

0.2

50

75

100

125

3716 G16

0

–50 –25

0 25 125

TEMPERATURE (°C)

75 10050

3716 G17

–100

0

24

V

COMMON MODE VOLTAGE (V)

SENSE

Soft-Start Up (Figure 13)

V

ITH

1V/DIV

V

OUT

1V/DIV

V

RUN/SS

2V/DIV

6

3716 G15

100ms/DIV 3716 G18

6

UW

TEMPERATURE (

°C)

–50

200

250

350

25 75

3716 G22

150

100

–25 0

50 100 125

50

0

300

FREQUENCY (kHz)

V

FREQSET

= 5V

V

FREQSET

= OPEN

V

FREQSET

= 0V

TYPICAL PERFOR A CE CHARACTERISTICS

Load Step (Figure 13)

Burst Mode Operation (Figure 13)

LTC3716

Constant Frequency Mode
(Figure 13)

V

50mV/DIV

I

4A/DIV

VIN = 15V, V

CC

CC

FCB = 0V

= 1.25V, SLEW RATE = 30A/µs

OUT

25µs/DIV

Current Sense Pin Input Current
vs Temperature

–12

V

= 1.6V

OUT

–11

–10

–9

–8

CURRENT SENSE INPUT CURRENT (µA)

–7

–50 –25

0

TEMPERATURE (°C)

50

25

VIN = 15V, V

V

OUT(AC)

I

L1

1A/DIV

I

L2

1A/DIV

FCB = INTV

Oscillator Frequency
vs Temperature

3716 G19

V

OUT(AC)

20mV/DIV

I

1A/DIV

I

1A/DIV

VIN = 15V, V

L1

L2

FCB = OPEN

= 1.6V, IL = 200mA

OUT

10µs/DIV

RMS

20mV/DIV

R9, R21 = 0.01Ω

3716 G25

EXTVCC Switch Resistance
vs Temperature

10

8

6

4

SWITCH RESISTANCE (Ω)

CC

2

EXTV

100

125

3716 G20

75

0

–50 –25

50

25

0

TEMPERATURE (°C)

100

125

3716 G21

75

= 1.6V, IL = 400mA

OUT

CC

2µs/DIV

RMS

R9, R21 = 0.01Ω

3716 G26

Undervoltage Lockout
vs Temperature

3.50

3.45

3.40

3.35

3.30

UNDERVOLTAGE LOCKOUT (V)

3.25

3.20
–50

–25 0

50 100 125

25 75

TEMPERATURE (°C)

3716 G23

V

Shutdown Latch

RUN/SS

Thresholds vs Temperature

4.5

4.0

3.5

3.0

2.5

2.0

1.5

1.0

0.5

SHUTDOWN LATCH THRESHOLDS (V)

0

–50 –25

LATCH ARMING

LATCHOFF

THRESHOLD

0 25 125

TEMPERATURE (°C)

75 10050

3716 G24

7

LTC3716

UUU

PI FU CTIO S

RUN/SS (Pin 1): Combination of Soft-Start, Run Control
Input and Short-Circuit Detection Timer. A capacitor to
ground at this pin sets the ramp time to full current output.
Forcing this pin below 0.8V causes the IC to shut down all
internal circuitry. All functions are disabled in shutdown.

SENSE1+, SENSE2+ (Pins 2,14): The (+) Input to Each
Differential Current Comparator. The ITH pin voltage and
built-in offsets between SENSE– and SENSE+ pins in
conjunction with R

SENSE1–, SENSE2– (Pins 3,13): The (–) Input to the
Differential Current Comparators.

set the current trip threshold.

SENSE

–

V

OS

fier. Internal precision resistors configure it as a differen­tial amplifier whose output is V

ATTENOUT (Pin 15): Voltage Feedback Signal Resistively
Divided According to the VID Programming Code.

ATTENIN (Pin 16): The Input to the VID Controlled Resis­tive Divider.

VID0–VID4 (Pins 17,18, 19, 20, 21): VID Control Logic
Input Pins.

V

BIAS

+

, V

(Pins 11, 12): Inputs to an Operational Ampli-

OS

DIFFOUT

(Pin 22): Supply Pin for the VID Control Circuit.

.

EAIN (Pin 4): Input to the error amplifier that compares the
feedback voltage to the internal 0.6V reference voltage.
This pin is normally connected to a resistive divider from
the output of the differential amplifier (DIFFOUT).

PLLFLTR (Pin 5): The phase-locked loop’s lowpass filter
is tied to this pin. Alternatively, this pin can be driven with
an AC or DC voltage source to vary the frequency of the
internal oscillator. Do not apply voltage to this pin prior to
application of VIN.

PLLIN (Pin 6): External Synchronization Input to Phase
Detector. This pin is internally terminated to SGND with
50kΩ. The phase-locked loop will force the rising top gate
signal of controller 1 to be synchronized with the rising
edge of the PLLIN signal.

FCB (Pin 7): Forced Continuous Control Input. This input
acts on both output stages and can be used to regulate a
secondary winding. Pulling this pin below 0.6V will force
continuous synchronous operation. Do not leave this pin
floating without a decoupling capacitor.

ITH (Pin 8): Error Amplifier Output and Switching Regula­tor Compensation Point. Both current comparator’s thresh­olds increase with this control voltage. The normal voltage
range of this pin is from 0V to 2.4V

SGND (Pin 9): Signal Ground. This pin is common to both
controllers. Route separately to the PGND pin.

V

DIFFOUT

pin provides true remote output voltage sensing. V
normally drives an external resistive divider that sets the
output voltage.

(Pin 10): Output of a Differential Amplifier. This

DIFFOUT

AMPMD (Pin 23): This Logic Input pin controls the con­nections of internal precision resistors that configure the
operational amplifier as a unity-gain differential amplifier.

TG2, TG1 (Pins 24, 35): High Current Gate Drives for Top
N-Channel MOSFETS. These are the outputs of floating
drivers with a voltage swing equal to INTVCC superim­posed on the switch node voltage SW.

SW2, SW1 (Pins 25, 34): Switch Node Connections to
Inductors. Voltage swing at these pins is from a Schottky
diode (external) voltage drop below ground to VIN.

BOOST2, BOOST1 (Pins 26, 33): Bootstrapped Supplies
to the Topside Floating Drivers. External capacitors are con­nected between the BOOST and SW pins, and Schottky
diodes are connected between the BOOST and INTVCC pins.

BG2, BG1 (Pins 27, 31): High Current Gate Drives for
Bottom N-Channel MOSFETS. Voltage swing at these pins
is from ground to INTVCC.

PGND (Pin 28): Driver Power Ground. Connect to sources
of bottom N-channel MOSFETS and the (–) terminals of CIN.

INTVCC (Pin 29): Output of the Internal 5V Linear Low
Dropout Regulator and the EXTVCC Switch. The driver and
control circuits are powered from this voltage source.
Decouple to power ground with a 1µF ceramic capacitor
placed directly adjacent to the IC and minimum of 4.7µF
additional tantalum or other low ESR capacitor.

EXTVCC (Pin 30): External Power Input to an Internal
Switch. This switch closes and supplies INTV
ing the internal low dropout regulator whenever EXTVCC is
higher than 4.7V. See EXTVCC Connection in the Applica­tions Information section. Do not exceed 7V on this pin
and ensure V

EXTVCC

≤ V

INTVCC

.

bypass-

CC,

8

UUU

PI FU CTIO S

LTC3716

VIN (Pin 32): Main Supply Pin. Should be closely de­coupled to the IC’s signal ground pin.

UU

W

FU CTIO AL DIAGRA

f

IN

AMPMD

R

C

–

V

OS

+

V

OS

DIFFOUT

LP

5V

+

LP

PLLIN

PLLFLTR

PGOOD

FCB

V

IN

EXTV

INTV

SGND

50k

0V POSITION

0.18µA

CC

CC

PHASE DET

OSCILLATOR

3V

0.60V

4.8V

4.5V

DUPLICATE FOR SECOND
CLK1
CLK2

TO SECOND
CHANNEL

–

0.66V

+

EAIN

–
+

0.54V

–

A1

+

–
+

+

FCB

–

V

REF

+
–

5V
LDO
REG

INTERNAL

SUPPLY

CONTROLLER CHANNEL

0.86V
5V

FB

SLOPE

COMP

1.2µA

6V

SRQ

0.55V

I

1

DROP

OUT
DET

Q

+
–

–
+

45k

PGOOD (Pin 36): Open-Drain Logic Output. PGOOD is
pulled to ground when the voltage on the EAIN pin is not
within ±10% of its set point.

B

+–

+–

SHDN

RST

5V

BOT FCB

TOP ON

FB

SHDN

START

–
+

45k

2.4V

OV

RUN

SOFT-

SWITCH

I

2

EA

LOGIC

–
+

+
–

V

FB

0.60V

0.66V

TOP

BOT

INTV

INTV

CC

30k

30k

CC

BOOST

TG

SW

BG

PGND

SENSE

SENSE

EAIN

I

TH

RUN/SS

INTV

+

–

V

CC

C

IN

D

B

C

B

L

C

C

R

C2

C

C

SS

R

D1

SENSE

+

C

IN

C

OUT

+

21

43

V

OUT

ATTENIN

ATTENOUT

10k

5-BIT VID DECODER

TYPICAL ALL

VID PINS

R1

VID1 VID2 VID3 VID4

VID0

40k

V

BIAS

3716 FBD

9

LTC3716

OPERATIO

U

(Refer to Functional Diagram)

Main Control Loop

The LTC3716 uses a constant frequency, current mode
step-down architecture with the two output stages oper­ating 180 degrees out of phase. During normal operation,
each top MOSFET is turned on when the clock for that
channel sets the RS latch, and turned off when the main
current comparator, I1, resets the RS latch. The peak
inductor current at which I1 resets the RS latch is con­trolled by the voltage on the I
error amplifier EA. The EAIN pin receives the voltage
feedback signal, which is compared to the internal refer­ence voltage by the EA. When the load current increases,
it causes a slight decrease in V
reference, which in turn causes the ITH voltage to increase
until the average inductor current matches the new load
current. After the top MOSFET has turned off, the bottom
MOSFET is turned on until either the inductor current
starts to reverse, as indicated by current comparator I2, or
the beginning of the next cycle.

The top MOSFET drivers are biased from floating boot­strap capacitor CB, which normally is recharged during
each off cycle through an external diode when the top
MOSFET turns off. As VIN decreases to a voltage close to
V

, the loop may enter dropout and attempt to turn on

OUT

the top MOSFET continuously. The dropout detector de­tects this and forces the top MOSFET off for about 500ns
every tenth cycle to allow CB to recharge.

The main control loop is shut down by pulling the RUN/
SS pin low. Releasing RUN/SS allows an internal 1.2µA
current source to charge soft-start capacitor CSS. When
CSS reaches 1.5V, the main control loop is enabled with
the ITH voltage clamped at approximately 30% of its
maximum value. As CSS continues to charge, the I
voltage is gradually released allowing normal, full-current
operation.

Low Current Operation

The FCB pin is a multifunction pin providing two func­tions: 1) to provide regulation for a secondary winding by
temporarily forcing continuous PWM operation on
both controllers; and 2) select between

pin, which is the output of

TH

relative to the 0.6V

EAIN

TH

two

modes of low

pin

current operation. When the FCB pin voltage is below

0.6V, the controller forces continuous PWM current
mode operation. In this mode, the top and bottom
MOSFETs are alternately turned on to maintain the output
voltage independent of direction of inductor current.
When the FCB pin is below V

0.6V, the controller enters Burst Mode operation. Burst
Mode operation sets a minimum output current level
before inhibiting the top switch and turns off the synchro­nous MOSFET(s) when the inductor current goes nega­tive. This combination of requirements will, at low currents,
force the ITH pin below a voltage threshold that will
temporarily inhibit turn-on of both output MOSFETs until
the output voltage drops. There is 60mV of hysteresis in
the burst comparator B tied to the ITH pin. This hysteresis
produces output signals to the MOSFETs that turn them
on for several cycles, followed by a variable “sleep”
interval depending upon the load current. The resultant
output voltage ripple is held to a very small value by
having the hysteretic comparator after the error amplifier
gain block.

Constant Frequency Operation

When the FCB pin is tied to INTVCC, Burst Mode operation
is disabled and a forced minimum peak output current
requirement is removed. This provides constant frequency,
discontinuous (preventing reverse inductor current) cur­rent operation over the widest possible output current
range. This constant frequency operation is not as efficient
as Burst Mode operation, but does provide a lower noise,
constant frequency operating mode down to approxi­mately 1% of designed maximum output current.

Continuous Current (PWM) Operation

Tying the FCB pin to ground will force continuous current
operation. This is the least efficient operating mode, but
may be desirable in certain applications. The output can
source or sink current in this mode. When sinking current
while in forced continuous operation, current will be
forced back into the main power supply potentially boost­ing the input supply to dangerous voltage levels—
BEWARE!

␣ –␣ 2V but greater than

INTVCC

10

OPERATIO

LTC3716

U

(Refer to Functional Diagram)

Frequency Synchronization

The phase-locked loop allows the internal oscillator to be
synchronized to an external source via the PLLIN pin. The
output of the phase detector at the PLLFLTR pin is also the
DC frequency control input of the oscillator that operates
over a 140kHz to 310kHz range corresponding to a DC
voltage input from 0V to 2.4V. When locked, the PLL aligns
the turn on of the top MOSFET to the rising edge of the
synchronizing signal. When PLLIN is left open, the PLLFLTR
pin goes low, forcing the oscillator to minimum frequency.

Input capacitance ESR requirements and efficiency losses
are substantially reduced because the peak current drawn
from the input capacitor is effectively divided by two and
power loss is proportional to the RMS current squared. A
two stage, single output voltage implementation can
reduce input path power loss by 75% and radically reduce
the required RMS current rating of the input capacitor(s).

INTVCC/EXTVCC Power

Power for the top and bottom MOSFET drivers and most
of the IC circuitry is derived from INTVCC. When the
EXTVCC pin is left open, an internal 5V low dropout
regulator supplies INTVCC power. If the EXTVCC pin is
taken above 4.8V, the 5V regulator is turned off and an
internal switch is turned on connecting EXTVCC to INTVCC.
This allows the INTVCC power to be derived from a high
efficiency external source such as the output of the regu­lator itself or a secondary winding, as described in the
Applications Information section. An external Schottky
diode can be used to minimize the voltage drop from
EXTVCC to INTV
the specified INTVCC current. Voltages up to 7V can be
applied to EXTVCC for additional gate drive capability.

Differential Amplifier

This amplifier provides true differential output voltage
sensing. Sensing both V
tion in high current applications and/or applications hav­ing electrical interconnection losses. The AMPMD pin
allows selection of internal precision feedback resistors
for high common mode rejection differencing applica­tions, or direct access to the actual amplifier inputs with­out these internal feedback resistors for other applications.

in applications requiring greater than

CC

OUT

+

and V

–

benefits regula-

OUT

The AMPMD pin is grounded to connect the internal pre­cision resistors in a unity-gain differencing application or
tied to the INTVCC pin to bypass the internal resistors and
make the amplifier inputs directly available. The amplifier
is a unity-gain stable, 2MHz gain-bandwidth, >120dB
open-loop gain design. The amplifier has an output slew
rate of 5V/µs and is capable of driving capacitive loads
with an output RMS current typically up to 25mA. The
amplifier is not capable of sinking current and therefore
must be resistively loaded to do so.

Output Overvoltage Protection

An overvoltage comparator, 0V, guards against transient
overshoots (>10%) as well as other more serious condi­tions that may overvoltage the output. In this case, the top
MOSFET is turned off and the bottom MOSFET is turned on
until the overvoltage condition is cleared.

Power Good (PGOOD)

The PGOOD pin is connected to the drain of an internal
MOSFET. The MOSFET turns on when the output voltage
is not within ±10% of its nominal output level as deter­mined by the feedback divider. When the output is within
±10% of its nominal value, the MOSFET is turned off within
10µs and the PGOOD pin should be pulled up by an
external resistor to a source of up to 7V.

Short-Circuit Detection

The RUN/SS capacitor is used initially to limit the inrush
current from the input power source. Once the control­lers have been given time, as determined by the capacitor
on the RUN/SS pin, to charge up the output capacitors
and provide full-load current, the RUN/SS capacitor is
then used as a short-circuit timeout circuit. If the output
voltage falls to less than 70% of its nominal output
voltage the RUN/SS capacitor begins discharging as­suming that the output is in a severe overcurrent and/or
short-circuit condition. If the condition lasts for a long
enough period as determined by the size of the RUN/SS
capacitor, the controller will be shut down until the
RUN/SS pin voltage is recycled. This built-in latchoff can
be overidden by providing a current >5µA at a compli-
ance of 5V to the RUN/SS pin. This current shortens the

11

LTC3716

OPERATING FREQUENCY (kHz)

120 170 220 270 320

PLLFLTR PIN VOLTAGE (V)

3716 F02

2.5

2.0

1.5

1.0

0.5

0

OPERATIO

U

(Refer to Functional Diagram)

soft-start period but also prevents net discharge of the
RUN/SS capacitor during a severe overcurrent and/or
short-circuit condition. Foldback current limiting is acti-

U

WUU

APPLICATIO S I FOR ATIO

The basic LTC3716 application circuit is shown in
Figure␣ 1 on the first page. External component selection
begins with the selection of the inductors based on ripple
current requirements and continues with the current
sensing resistors using the calculated peak inductor
current and/or maximum current limit. Next, the power
MOSFETs, D1 and D2 are selected. The operating fre­quency and the inductor are chosen based mainly on the
amount of ripple current. Finally, CIN is selected for its
ability to handle the input ripple current (that PolyPhase
operation minimizes) and C

is chosen with low enough

OUT

ESR to meet the output ripple voltage and load step
specifications (also minimized with PolyPhase). Current
mode architecture provides inherent current sharing be­tween output stages. The circuit shown in Figure␣ 1 can be
configured for operation up to an input voltage of 28V
(limited by the external MOSFETs). Current mode control
allows the ability to connect the two output stages to two

different

take some power from
selection of the R

input power supply rails. A heavy output load can

each

input supply according to the

resistors.

SENSE

TM

vated when the output voltage falls below 70% of its
nominal level whether or not the short-circuit latchoff
circuit is enabled.

Operating Frequency

The LTC3716 uses a constant frequency, phase-lockable
architecture with the frequency determined by an internal
capacitor. This capacitor is charged by a fixed current
plus an additional current which is proportional to the
voltage applied to the PLLFLTR pin. Refer to Phase­Locked Loop and Frequency Synchronization for addi­tional information.

A graph for the voltage applied to the PLLFLTR pin vs
frequency is given in Figure␣ 2. As the operating frequency
is increased the gate charge losses will be higher, reducing
efficiency (see Efficiency Considerations). The maximum
switching frequency is approximately 310kHz.

R

R

Selection For Output Current

SENSE

SENSE1,2

are chosen based on the required peak output
current. The LTC3716 current comparator has a maxi­mum threshold of 75mV/R
mode range of SGND to 1.1(INTVCC). The current com­parator threshold sets the peak inductor current, yielding
a maximum average output current I
value less half the peak-to-peak ripple current, ∆IL.

Assuming a common input power source for each output
stage and allowing a margin for variations in the
LTC3716 and external component values yields:

R

PolyPhase is a registered trademark of Linear Technology Corporation.

12

SENSE

= 2(50mV/I

MAX

SENSE

)

and an input common

equal to the peak

MAX

Figure 2. Operating Frequency vs V

PLLFLTR

Inductor Value Calculation and Output Ripple Current

The operating frequency and inductor selection are inter­related in that higher operating frequencies allow the use
of smaller inductor and capacitor values. So why would
anyone ever choose to operate at lower frequencies with
larger components? The answer is efficiency. A higher
frequency generally results in lower efficiency because
MOSFET gate charge and transition losses increase

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

directly with frequency. In addition to this basic tradeoff,
the effect of inductor value on ripple current and low
current operation must also be considered. The PolyPhase
approach reduces both input and output ripple currents
while optimizing individual output stages to run at a lower
fundamental frequency, enhancing efficiency.

The inductor value has a direct effect on ripple current.
The inductor ripple current ∆IL per individual section, N,
decreases with higher inductance or frequency and
increases with higher VIN or V



V

∆I

OUT OUT

=−

L

fL

1







V





V

IN

where f is the individual output stage operating frequency.
In a 2-phase converter, the net ripple current seen by the

output capacitor is much smaller than the individual
inductor ripple currents due to ripple cancellation. The
details on how to calculate the net output ripple current
can be found in Application Note 77.

Figure 3 shows the net ripple current seen by the output
capacitors for 1- and 2-phase configurations. The output
ripple current is plotted for a fixed output voltage as the
duty factor is varied between 10% and 90% on the x-axis.
The output ripple current is normalized against the induc­tor ripple current at zero duty factor. The graph can be
used in place of tedious calculations, simplifying the
design process.

1.0

0.9

0.8

0.7

0.6

/fL

0.5

O

O(P-P)

V

∆I

0.4

0.3

0.2

0.1
0

0.1 0.2 0.3 0.4

Figure 3. Normalized Output Ripple Current
vs Duty Factor [I

Kool Mµ is a registered trademark of Magnetics, Inc.

DUTY FACTOR (V

≈ 0.3 (∆I

RMS

:

OUT

1-PHASE
2-PHASE

0.5 0.6 0.7 0.8 0.9

OUT/VIN

)

O(P–P)

3716 F03

)]

Accepting larger values of ∆IL allows the use of low
inductances, but can result in higher output voltage ripple.
A reasonable starting point for setting ripple current is
∆IL = 0.4(I

)/2, where I

OUT

is the total load current.

OUT

Remember, the maximum ∆IL occurs at the maximum
input voltage. The individual inductor ripple currents are
determined by the inductor, input and output voltages.

Inductor Core Selection

Once the values for L1 and L2 are known, the type of
inductor must be selected. High efficiency converters
generally cannot afford the core loss found in low cost
powdered iron cores, forcing the use of more expensive
ferrite, molypermalloy, or Kool Mµ® cores. Actual core
loss is independent of core size for a fixed inductor value,
but it is very dependent on inductor type selected. As
inductance increases, core losses go down. Unfortu­nately, increased inductance requires more turns of wire
and therefore copper losses will increase.

Ferrite designs have very low core loss and are preferred
at high switching frequencies, so design goals can con­centrate on copper loss and preventing saturation. Ferrite
core material saturates “hard,” which means that induc­tance collapses abruptly when the peak design current is
exceeded. This results in an abrupt increase in inductor
ripple current and consequent output voltage ripple.

Do

not allow the core to saturate!

Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive
than ferrite. A reasonable compromise from the same
manufacturer is Kool Mµ. Toroids are very space effi­cient, especially when you can use several layers of wire.
Because they lack a bobbin, mounting is more difficult.
However, designs for surface mount are available which
do not increase the height significantly.

Power MOSFET, D1 and D2 Selection

Two external power MOSFETs must be selected for each
output stage with the LTC3716: one N-channel MOSFET
for the top (main) switch, and one N-channel MOSFET for
the bottom (synchronous) switch.

The peak-to-peak drive levels are set by the INTV

CC

voltage. This voltage is typically 5V during start-up

13

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

(see EXTVCC Pin Connection). Consequently, logic-level
threshold MOSFETs must be used in most applications.
The only exception is if low input voltage is expected
(VIN < 5V); then, sublogic-level threshold MOSFETs
(V
BV
logic-level MOSFETs are limited to 30V or less.

Selection criteria for the power MOSFETs include the “ON”
resistance R
input voltage and maximum output current. When the
LTC3716 is operating in continuous mode the duty factors
for the top and bottom MOSFETs of each output stage are
given by:

The MOSFET power dissipations at maximum output
current are given by:

< 1V) should be used. Pay close attention to the

GS(TH)

specification for the MOSFETs as well; most of the

DSS

, reverse transfer capacitance C

DS(ON)

V

Main SwitchDuty Cycle

Synchronous SwitchDuty Cycle

OUT

=

V

IN



VV

–

IN OUT

=





V

IN






RSS

,

short-circuit when the synchronous switch is on close to
100% of the period.

The term (1 + δ) is generally given for a MOSFET in the
form of a normalized R

vs temperature curve, but

DS(ON)

δ = 0.005/°C can be used as an approximation for low
voltage MOSFETs. C

is usually specified in the

RSS

MOSFET characteristics. The constant k = 1.7 can be
used to estimate the contributions of the two terms in the
main switch dissipation equation.

The Schottky diodes, D1 and D2 shown in Figure 1
conduct during the dead-time between the conduction of
the two large power MOSFETs. This helps prevent the
body diode of the bottom MOSFET from turning on,
storing charge during the dead-time, and requiring a
reverse recovery period which would reduce efficiency. A
1A to 3A Schottky (depending on output current) diode is
generally a good compromise for both regions of opera­tion due to the relatively small average current. Larger
diodes result in additional transition losses due to their
larger junction capacitance.

CIN and C

Selection

OUT

I

I

MAX

2



+

1

δ

()





2



Cf

()()



RSS



2













2

R

DS ON

+

12δ

()

+

()

R

DS ON

()

DS(ON)

and k

V



P

MAIN

P

SYNC

OUTINMAX

=

kV

=





V



2

()



IN



VVVI

–

IN OUTINMAX

where δ is the temperature dependency of R
is a constant inversely related to the gate drive current.

Both MOSFETs have I2R losses but the topside N-channel
equation includes an additional term for transition losses,
which peak at the highest input voltage. For V

< 20V the

IN

high current efficiency generally improves with larger
MOSFETs, while for V
increase to the point that the use of a higher R
with lower C

RSS

> 20V the transition losses rapidly

IN

device

DS(ON)

actual provides higher efficiency. The
synchronous MOSFET losses are greatest at high input
voltage when the top switch duty factor is low or during a

In continuous mode, the source current of each top
N-channel MOSFET is a square wave of duty cycle V

OUT

/
VIN. A low ESR input capacitor sized for the maximum
RMS current must be used. The details of a closed form
equation can be found in Application Note 77. Figure 4
shows the input capacitor ripple current for a 2-phase
configuration with the output voltage fixed and input
voltage varied. The input ripple current is normalized
against the DC output current. The graph can be used in
place of tedious calculations. The minimum input ripple
current can be achieved when the input voltage is twice the
output voltage.

In the graph of Figure 4, the 2-phase local maximum input
RMS capacitor currents are reached when:

V

OUT

V

IN

k

−21

=

4

where k = 1, 2
These worst-case conditions are commonly used for

design because even significant deviations do not offer

14

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may also be paralleled to
meet size or height requirements in the design. Always
consult the capacitor manufacturer if there is any
question.

0.6

0.5

0.4

0.3

0.2

DC LOAD CURRENT

RMS INPUT RIPPLE CURRNET

0.1

0

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8

Figure 4. Normalized RMS Input Ripple Current
vs Duty Factor for 1 and 2 Output Stages

DUTY FACTOR (V

It is important to note that the efficiency loss is propor­tional to the input RMS current
2-phase implementation results in 75% less power loss
when compared to a single phase design. Battery/input
protection fuse resistance (if used), PC board trace and
connector resistance losses are also reduced by the
reduction of the input ripple current in a 2-phase system.
The required amount of input capacitance is further
reduced by the factor, 2, due to the effective increase in
the frequency of the current pulses.

The selection of C

OUT

series resistance (ESR). Typically once the ESR require­ment has been met, the RMS current rating generally far
exceeds the I
output ripple (∆V

∆∆V I ESR

OUT RIPPLE

RIPPLE(P-P)

OUT

≈+

Where f = operating frequency of each stage, C
output capacitance and ∆I
ripple currents.

1-PHASE
2-PHASE

0.9

)

OUT/VIN

squared

3716 F04

and therefore a

is driven by the required effective

requirements. The steady state

) is determined by:






16

RIPPLE



1





fC

OUT

OUT

= combined inductor

=

The output ripple varies with input voltage since ∆IL is a
function of input voltage. The output ripple will be less than
50mV at max VIN with ∆IL = 0.4I

C
C

required ESR < 4(R

OUT

> 1/(16f)(R

OUT

SENSE

SENSE

)

OUT(MAX)

) and

/2 assuming:

The emergence of very low ESR capacitors in small,
surface mount packages makes very physically small
implementations possible. The ability to externally com­pensate the switching regulator loop using the I

TH

pin(OPTI-LOOP compensation) allows a much wider se­lection of output capacitor types. OPTI-LOOP compensa­tion effectively removes constraints on output capacitor
ESR. The impedance characteristics of each capacitor
type are significantly different than an ideal capacitor and
therefore require accurate modeling or bench evaluation
during design.

Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance
through-hole capacitors. The OS-CON semiconductor
dielectric capacitor available from Sanyo and the Panasonic
SP surface mount types have the lowest (ESR)(size)
product of any aluminum electrolytic at a somewhat
higher price. An additional ceramic capacitor in parallel
with OS-CON type capacitors is recommended to reduce
the inductance effects.

In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec­trolytic and dry tantalum capacitors are both available in
surface mount configurations. New special polymer sur­face mount capacitors offer very low ESR also but have
much lower capacitive density per unit volume. In the case
of tantalum, it is critical that the capacitors are surge tested
for use in switching power supplies. Several excellent
choices are the AVX TPS, AVX TPSV or the KEMET T510
series of surface mount tantalums, available in case heights
ranging from 2mm to 4mm. Other capacitor types include
Sanyo OS-CON, POSCAPs, Panasonic SP caps, Nichicon
PL series and Sprague 595D series. Consult the manufac­turer for other specific recommendations. A combination
of capacitors will often result in maximizing performance
and minimizing overall cost and size.

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INTVCC Regulator

An internal P-channel low dropout regulator produces 5V
at the INTVCC pin from the VIN supply pin. The INTV
regulator powers the drivers and internal circuitry of the
LTC3716. The INTVCC pin regulator can supply up to 50mA
peak and must be bypassed to power ground with a
minimum of 4.7µF tantalum or electrolytic capacitor. An
additional 1µF ceramic capacitor placed very close to the
IC is recommended due to the extremely high instanta­neous currents required by the MOSFET gate drivers.

High input voltage applications in which large MOSFETs
are being driven at high frequencies may cause the maxi­mum junction temperature rating for the LTC3716 to be
exceeded. The supply current is dominated by the gate
charge supply current, in addition to the current drawn
from the differential amplifier output. The gate charge is
dependent on operating frequency as discussed in the
Efficiency Considerations section. The supply current can
either be supplied by the internal 5V regulator or via the
EXTVCC pin. When the voltage applied to the EXTVCC pin
is less than 4.7V, all of the INTVCC load current is supplied
by the internal 5V linear regulator. Power dissipation for
the IC is higher in this case by (IIN)(VIN – INTVCC) and
efficiency is lowered. The junction temperature can be
estimated by using the equations given in Note 1 of the
Electrical Characteristics. For example, the LTC3716 V
current is limited to less than 24mA from a 24V supply:

TJ = 70°C + (24mA)(24V)(85°C/W) = 119°C

Use of the EXTVCC pin reduces the junction temperature␣ to:

TJ = 70°C + (24mA)(5V)(85°C/W) = 80.2°C

The input supply current should be measured while the
controller is operating in continuous mode at maximum
V

and the power dissipation calculated in order to

IN

prevent the maximum junction temperature from being
exceeded.

EXTVCC Connection

The LTC3716 contains an internal P-channel MOSFET
switch connected between the EXTVCC and INTVCC pins.
When the voltage applied to EXTV
internal regulator is turned off and an internal switch
closes, connecting the EXTV

rises above 4.7V, the

CC

pin to the INTV

CC

CC

CC

IN

pin

thereby supplying internal and MOSFET gate driving power
to the IC. The switch remains closed as long as the voltage
applied to EXTVCC remains above 4.5V. This allows the
MOSFET driver and control power to be derived from the
output during normal operation (4.7V < V
from the internal regulator when the output is out of
regulation (start-up, short-circuit). Do not apply greater
than 7V to the EXTVCC pin and ensure that EXTV

0.3V when using the application circuits shown. If an
external voltage source is applied to the EXTVCC pin when
the VIN supply is not present, a diode can be placed in
series with the LTC3716’s V
between the EXTV
from backfeeding VIN.

Significant efficiency gains can be realized by powering
INTVCC from the output, since the VIN current resulting
from the driver and control currents will be scaled by the
ratio: (Duty Factor)/(Efficiency). For 5V regulators this
means connecting the EXTVCC pin directly to V
ever, for 3.3V and other lower voltage regulators, addi­tional supply circuitry is required to derive INTVCC power
from the output.

The following list summarizes the four possible connec­tions for EXTV

1. EXTVCC left open (or grounded). This will cause INTV
to be powered from the internal 5V regulator resulting in
a significant efficiency penalty at high input voltages.

2. EXTVCC connected directly to V
connection for a 5V regulator and provides the highest
efficiency.

3. EXTVCC connected to an external supply. If an external
supply is available in the 5V to 7V range, it may be used to
power EXTVCC providing it is compatible with the MOSFET
gate drive requirements.

4. EXTVCC connected to an output-derived boost network.
For 3.3V and other low voltage regulators, efficiency gains
can still be realized by connecting EXTVCC to an output­derived voltage which has been boosted to greater than

4.7V but less than 7V. This can be done with either the
inductive boost winding as shown in Figure 5a or the
capacitive charge pump shown in Figure 5b. The charge
pump has the advantage of simple magnetics.

CC:

and the V

CC

pin and a Schottky diode

IN

pin, to prevent current

IN

EXTVCC

. This is the normal

OUT

< 7V) and

< V

CC

IN

. How-

OUT

+

CC

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V

IN

BAT85 0.22µF

VN2222LL

R

SENSE

L1

3

. Therefore, it must be

BIAS

to get a digital high input. The

V

TG1

BG1

+

C

IN

IN

N-CH

N-CH

R6

R5

OPTIONAL EXTVCC
CONNECTION

< 7V

5V < V

SEC

LTC3716

EXTV

CC

V

TG1

SW1

BG1FCB

PGND

+

C

IN

IN

N-CH

N-CH

V

IN

1N4148

T1

V

SEC

+

1µF

R

SENSE

21

V

OUT

3

4

+

C

OUT

3716 F05a

EXTV

LTC3716

CC

SW1

PGND

Figure 5a. Secondary Output Loop with EXTVCC Connection Figure 5b. Capacitive Charge Pump for EXTV

Topside MOSFET Driver Supply (CB,DB) (Refer to
Functional Diagram)

External bootstrap capacitors CB1 and CB2 connected to
the BOOST1 and BOOST2 pins supply the gate drive
voltages for the topside MOSFETs. Capacitor CB in the
Functional Diagram is charged though diode DB from
INTVCC when the SW pin is low. When the topside MOSFET
turns on, the driver places the CB voltage across the gate­source of the desired MOSFET. This enhances the MOSFET
and turns on the topside switch. The switch node voltage,
SW, rises to VIN and the BOOST pin rises to VIN + V

INTVCC

The value of the boost capacitor CB needs to be 30 to 100
times that of the total input capacitance of the topside
MOSFET(s). The reverse breakdown of DB must be greater
than V

IN(MAX).

The final arbiter when defining the best gate drive ampli­tude level will be the input supply current. If a change is
made that decreases input current, the efficiency has
improved. If the input current does not change then the
efficiency has not changed either.

Output Voltage

The LTC3716 has a true remote voltage sense capablity.
The sensing connections should be returned from the load
back to the differential amplifier’s inputs through a com­mon, tightly coupled pair of PC traces. The differential
amplifier corrects for DC drops in both the power and
ground paths. The differential amplifier output signal is

divided down and compared with the internal precision

0.6V voltage reference by the error amplifier.

Output Voltage Programming

The output voltage is digitally programmed as defined in
Table 1 using the VID0 to VID4 logic input pins. The VID
logic inputs program a precision, 0.25% internal feedback
resistive divider. The LTC3716 has an output voltage
range of 0.6V to 1.75V in 25mV and 50mV steps.

Between the ATTENOUT pin and ground is a variable

.

resistor, R1, whose value is controlled by the five VID input
pins (VID0 to VID4). Another resistor, R2, between the
ATTENIN and the ATTENOUT pins completes the resistive
divider. The output voltage is thus set by the ratio of
(R1␣ +␣ R2) to R1.

Each VID digital input is pulled up by a 40k resistor in
series with a diode from V
grounded to get a digital low input, and can be either
floated or connected to V

BIAS

series diode is used to prevent the digital inputs from
being damaged or clamped if they are driven higher than
V

. The digital inputs accept CMOS voltage levels.

BIAS

V

is the supply voltage for the VID section. It is

BIAS

normally connected to INTVCC but can be driven from
other sources. If it is driven from another source, that
source

must

be in the range of 2.7V to 5.5V and

alive prior to enabling the LTC3716.

+

BAT85

BAT85

21

4

+

3716 F05b

CC

V

C

OUT

must

OUT

be

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Table 1. VID Output Voltage Programming

VID4 VID3 VID2 VID1 VID0 LTC3716

0 0 0 0 0 1.750V
0 0 0 0 1 1.700V
0 0 0 1 0 1.650V
0 0 0 1 1 1.600V
0 0 1 0 0 1.550V
0 0 1 0 1 1.500V
0 0 1 1 0 1.450V
0 0 1 1 1 1.400V
0 1 0 0 0 1.350V
0 1 0 0 1 1.300V
0 1 0 1 0 1.250V
0 1 0 1 1 1.200V
0 1 1 0 0 1.150V
0 1 1 0 1 1.100V
0 1 1 1 0 1.050V
0 1 1 1 1 1.000V
1 0 0 0 0 0.975V
1 0 0 0 1 0.950V
1 0 0 1 0 0.925V
1 0 0 1 1 0.900V
1 0 1 0 0 0.875V
1 0 1 0 1 0.850V
1 0 1 1 0 0.825V
1 0 1 1 1 0.800V
1 1 0 0 0 0.775V
1 1 0 0 1 0.750V
1 1 0 1 0 0.725V
1 1 0 1 1 0.700V
1 1 1 0 0 0.675V
1 1 1 0 1 0.650V
1 1 1 1 0 0.625V
1 1 1 1 1 0.600V

Soft-Start/Run Function

The RUN/SS pin provides three functions: 1) Run/Shut­down, 2) soft-start and 3) a defeatable short-circuit latchoff
timer. Soft-start reduces the input power sources’ surge
currents by gradually increasing the controller’s current
limit I

TH(MAX)

. The latchoff timer prevents very short,

extreme load transients from tripping the overcurrent
latch. A small pull-up current (>5µA) supplied to the RUN/
SS pin will prevent the overcurrent latch from operating.
The following explanation describes how the functions
operate.

An internal 1.2µA current source charges up the soft-start
capacitor, CSS. When the voltage on RUN/SS reaches 1.5V,
the controller is permitted to start operating. As the voltage
on RUN/SS increases from 1.5V to 3.0V, the internal
current limit is increased from 25mV/R
R

. The output current limit ramps up slowly, taking

SENSE

SENSE

to 75mV/

an additional 1.4s/µF to reach full current. The output
current thus ramps up slowly, reducing the starting surge
current required from the input power supply. If RUN/SS
has been pulled all the way to ground there is a delay before
starting of approximately:

V

15

t

DELAY SS SS

=

.

12

.

CsFC

=µ

125

./

A

µ

()

The time for the output current to ramp up is then:

t

IRAMP SS SS

315

=

.

VV

−

12

.

CsFC

A

µ

125

./

=µ

()

By pulling the RUN/SS pin below 0.8V the LTC3716 is put
into low current shutdown (IQ < 40µA). The RUN/SS pins
can be driven directly from logic as shown in Figure 6.
Diode D1 in Figure 6 reduces the start delay but allows
CSS to ramp up slowly providing the soft-start function.
The RUN/SS pin has an internal 6V zener clamp (see
Functional Diagram).

D1*

INTV

CC

RSS*

RUN/SS

C

3716 F06

SS

V

3.3V OR 5V RUN/SS

*OPTIONAL TO DEFEAT OVERCURRENT LATCHOFF

IN

RSS*

D1

C

SS

Figure 6. RUN/SS Pin Interfacing

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Fault Conditions: Overcurrent Latchoff

The RUN/SS pin also provides the ability to latch off the
controllers when an overcurrent condition is detected. The
RUN/SS capacitor, CSS, is used initially to limit the inrush
current of both controllers. After the controllers have been
started and been given adequate time to charge up the
output capacitors and provide full load current, the RUN/
SS capacitor is used for a short-circuit timer. If the output
voltage falls to less than 70% of its nominal value after C
reaches 4.1V, CSS begins discharging on the assumption
that the output is in an overcurrent condition. If the
condition lasts for a long enough period as determined by
the size of the CSS, the controller will be shut down until the
RUN/SS pin voltage is recycled. If the overload occurs
during start-up, the time can be approximated by:

t

≈ (CSS • 0.6V)/(1.2µA) = 5 • 105 (CSS)

LO1

If the overload occurs after start-up, the voltage on CSS will
continue charging and will provide additional time before
latching off:

t

≈ (CSS • 3V)/(1.2µA) = 2.5 • 106 (CSS)

LO2

This built-in overcurrent latchoff can be overridden by
providing a pull-up resistor, RSS, to the RUN/SS pin as
shown in Figure 6. This resistance shortens the soft-start
period and prevents the discharge of the RUN/SS capaci­tor during a severe overcurrent and/or short-circuit con­dition. When deriving the 5µA current from VIN as in the
figure, current latchoff is always defeated. The diode
connecting this pull-up resistor to INTVCC, as in Figure␣ 6,
eliminates any extra supply current during shutdown
while eliminating the INTV

loading from preventing

CC

controller start-up.
Why should you defeat current latchoff? During the

prototyping stage of a design, there may be a problem with
noise pickup or poor layout causing the protection circuit
to latch off the controller. Defeating this feature allows
troubleshooting of the circuit and PC layout. The internal
short-circuit and foldback current limiting still remains
active, thereby protecting the power supply system from
failure. A decision can be made after the design is com­plete whether to rely solely on foldback current limiting or
to enable the latchoff feature by removing the pull-up
resistor.

SS

The value of the soft-start capacitor CSS may need to be
scaled with output voltage, output capacitance and load
current characteristics. The minimum soft-start capaci­tance is given by:

CSS > (C

OUT

)(V

)(10-4)(R

OUT

SENSE

)

The minimum recommended soft-start capacitor of CSS =

0.1µF will be sufficient for most applications.

Phase-Locked Loop and Frequency Synchronization

The LTC3716 has a phase-locked loop comprised of an
internal voltage controlled oscillator and phase detector.
This allows the top MOSFET turn-on to be locked to the
rising edge of an external source. The frequency range of
the voltage controlled oscillator is ±50% around the
center frequency fO. A voltage applied to the PLLFLTR pin
of 1.2V corresponds to a frequency of approximately
220kHz. The nominal operating frequency range of the
LTC3716 is 140kHz to 310kHz.

The phase detector used is an edge sensitive digital type
which provides zero degrees phase shift between the
external and internal oscillators. This type of phase detec­tor will not lock up on input frequencies close to the
harmonics of the VCO center frequency. The PLL hold-in
range, ∆fH, is equal to the capture range, ∆f

C:

∆fH = ∆fC = ±0.5 fO (150kHz-300kHz)

The output of the phase detector is a complementary pair
of current sources charging or discharging the external
filter network on the PLLFLTR pin. A simplified block
diagram is shown in Figure 7.

2.4V

PHASE

50k

DETECTOR

DIGITAL

PHASE/

FREQUENCY

DETECTOR

EXTERNAL

OSC

PLLIN

Figure 7. Phase-Locked Loop Block Diagram

R

LP

10k

PLLFLTR

OSC

3716 F07

C

LP

19

LTC3716

VV

R
R

SEC MIN()

.≈+











06 1

6

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If the external frequency (f
lator frequency f
pulling up the PLLFLTR pin. When the external frequency
is less than f
down the PLLFLTR pin. If the external and internal fre­quencies are the same but exhibit a phase difference, the
current sources turn on for an amount of time correspond­ing to the phase difference. Thus the voltage on the
PLLFLTR pin is adjusted until the phase and frequency of
the external and internal oscillators are identical. At this
stable operating point the phase comparator output is
open and the filter capacitor CLP holds the voltage. The
LTC3716 PLLIN pin must be driven from a low impedance
source such as a logic gate located close to the pin.

The loop filter components (CLP, RLP) smooth out the
current pulses from the phase detector and provide a
stable input to the voltage controlled oscillator. The filter
components CLP and RLP determine how fast the loop
acquires lock. Typically R

0.1µF.

Minimum On-Time Considerations

Minimum on-time, t
that the LTC3716 is capable of turning on the top MOSFET.
It is determined by internal timing delays and the gate
charge required to turn on the top MOSFET. Low duty cycle
applications may approach this minimum on-time limit
and care should be taken to ensure that:

t

ON MIN

()

If the duty cycle falls below what can be accommodated by
the minimum on-time, the LTC3716 will begin to skip
cycles resulting in variable frequency operation. The out­put voltage will continue to be regulated, but the ripple
current and ripple voltage will increase.

0SC

<

, current is sourced continuously,

0SC

, current is sunk continuously, pulling

ON(MIN)

V

OUT

Vf

()

IN

) is greater than the oscil-

PLLIN

=10k and CLP is 0.01µF to

LP

, is the smallest time duration

significant amount of cycle skipping can occur with corre­spondingly larger ripple current and voltage ripple.

If an application can operate close to the minimum
on-time limit, an inductor must be chosen that has a low
enough inductance to provide sufficient ripple amplitude
to meet the minimum on-time requirement.

As a general
rule, keep the inductor ripple current of each phase equal
to or greater than 15% of I

FCB Pin Operation

The FCB pin can be used to regulate a secondary winding
or as a logic level input. Continuous operation is forced
when the FCB pin drops below 0.6V. During continuous
mode, current flows continuously in the transformer pri­mary. The secondary winding(s) supply current only when
the bottom, synchronous switch is on. When primary load
currents are low and/or the VIN/V
synchronous switch may not be on for a sufficient amount
of time to transfer power from the output capacitor to the
secondary load. Forced continuous operation will support
secondary windings providing there is sufficient synchro­nous switch duty factor. Thus, the FCB input pin removes
the requirement that power must be drawn from the
inductor primary in order to extract power from the
auxiliary winding(s). With the loop in continuous mode,
the auxiliary output(s) may nominally be loaded without
regard to the primary output load.

The secondary output voltage V
shown in Figure 5a by the turns ratio N of the transformer:

V

≅ (N + 1) V

SEC

However, if the controller goes into Burst Mode operation
and halts switching due to a light primary load current,
then V
V

SEC

will droop. An external resistive divider from

SEC

to the FCB pin sets a minimum voltage V

OUT

OUT(MAX)

at V

IN(MAX)

ratio is low, the

OUT

is normally set as

SEC

.

SEC(MIN)

:

The minimum on-time for the LTC3716 is generally less
than 200ns. However, as the peak sense voltage de­creases, the minimum on-time gradually increases. This is
of particular concern in forced continuous applications
with low ripple current at light loads. If the duty cycle drops
below the minimum on-time limit in this situation, a

20

where R5 and R6 are shown in Figure 5a.
If V

temporary continuous switching operation until V
again above its minimum.

drops below this level, the FCB voltage forces

SEC

SEC

is

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APPLICATIO S I FOR ATIO

In order to prevent erratic operation if no external connec­tions are made to the FCB pin, the FCB pin has a 0.18µA
internal current source pulling the pin high. Include this
current when choosing resistor values R5 and R6.

The following table summarizes the possible states avail­able on the FCB pin:

Table 2

FCB Pin Condition

0V to 0.55V Forced Continuous (Current Reversal

0.65V < V

Feedback Resistors Regulating a Secondary Winding
>4.8V Burst Mode Operation Disabled

< 4.3V (typ) Minimum Peak Current Induces

FCB

Active Voltage Positioning

Active voltage positioning can be used to minimize peak­to-peak output voltage excursion under worst-case tran­sient loading conditions. The open-loop DC gain of the
control loop is reduced depending upon the maximum
load step specifications. Active voltage positioning can
easily be added to the LTC3716 by loading the ITH pin with
a resistive divider having a Thevenin equivalent voltage
source equal to the midpoint operating voltage of the error
amplifier, or 1.2V (see Figure 8).

INTV

CC

R

T2

R

T1

Figure 8. Active Voltage Positioning Applied to the LTC3716

The resistive load reduces the DC loop gain while main­taining the linear control range of the error amplifier. The
worst-case peak-to-peak output voltage deviation due to
transient loading can theoretically be reduced to half or

Allowed—Burst Inhibited)

Burst Mode Operation
No Current Reversal Allowed

Constant Frequency Mode Enabled
No Current Reversal Allowed

No Minimum Peak Current

I

TH

R

C

C

C

LTC3716

3716 F08

alternatively the amount of output capacitance can
be reduced for a particular application. A complete
explanation is included in Design Solutions 10 or the
LTC1736 data sheet. (See www.linear-tech.com)

Efficiency Considerations

The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can be
expressed as:

%Efficiency = 100% – (L1 + L2 + L3 + ...)

where L1, L2, etc. are the individual losses as a percentage
of input power.

Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC3716 circuits: 1) I2R losses, 2) Topside
MOSFET transition losses, 3) INTVCC regulator current
and 4) LTC3716 VIN current (including loading on the
differential amplifier output).

1) I2R losses are predicted from the DC resistances of the
fuse (if used), MOSFET, inductor, current sense resistor,
and input and output capacitor ESR. In continuous mode
the average output current flows through L and R

SENSE

,
but is “chopped” between the topside MOSFET and the
synchronous MOSFET. If the two MOSFETs have approxi­mately the same R

, then the resistance of one

DS(ON)

MOSFET can simply be summed with the resistances of L,
R
R

and ESR to obtain I2R losses. For example, if each

SENSE

= 10mΩ, RL = 10mΩ, and R

DS(ON)

= 5mΩ, then the

SENSE

total resistance is 25mΩ. This results in losses ranging
from 2% to 8% as the output current increases from 3A to
15A per output stage for a 5V output, or a 3% to 12% loss
per output stage for a 3.3V output. Efficiency varies as the
inverse square of V

for the same external components

OUT

and output power level. The combined effects of increas­ingly lower output voltages and higher currents required
by high performance digital systems is not doubling but
quadrupling the importance of loss terms in the switching
regulator system!

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2) Transition losses apply only to the topside MOSFET(s),
and are significant only when operating at high input
voltages (typically 12V or greater). Transition losses can
be estimated from:

I



OMAX

TransitionLoss V

3) INTVCC current is the sum of the MOSFET driver and
control currents. The MOSFET driver current results from
switching the gate capacitance of the power MOSFETs.
Each time a MOSFET gate is switched from low to high to
low again, a packet of charge dQ moves from INTVCC to
ground. The resulting dQ/dt is a current out of INTVCC that
is typically much larger than the control circuit current. In
continuous mode, I
are the gate charges of the topside and bottom side
MOSFETs.

Supplying INTV
from an output-derived source will scale the V
required for the driver and control circuits by the ratio
(Duty Factor)/(Efficiency). For example, in a 20V to 5V
application, 10mA of INTVCC current results in approxi­mately 3mA of VIN current. This reduces the mid-current
loss from 10% or more (if the driver was powered directly
from VIN) to only a few percent.

4) The VIN current has two components: the first is the
DC supply current given in the Electrical Characteristics
table, which excludes MOSFET driver and control cur­rents; the second is the current drawn from the differential
amplifier output. VIN current typically results in a small
(<0.1%) loss.

Other “hidden” losses such as copper trace and internal
battery resistances can account for an additional 5% to
10% efficiency degradation in portable systems. It is very
important to include these “system” level losses in the
design of a system. The internal battery and input fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and a very low ESR at
the switching frequency. A 50W supply will typically
require a minimum of 200µF to 300µF of output capaci-
tance having a maximum of 10mΩ to 20mΩ of ESR. The
LTC3716 2-phase architecture typically halves the input

=

17

(.)

GATECHG

power through the EXTVCC switch input

CC

2

IN





= (QT + QB), where QT and Q



()

2

Cf





RSS

current

IN

B

and output capacitance requirements over competing
solutions. Other losses including Schottky conduction
losses during dead-time and inductor core losses gener­ally account for less than 2% total additional loss.

Checking Transient Response

The regulator loop response can be checked by looking at
the load transient response. Switching regulators take
several cycles to respond to a step in DC (resistive) load
current. When a load step occurs, V
amount equal to ∆I
series resistance of C
discharge C
forces the regulator to adapt to the current change and
return V
time V
ringing, which would indicate a stability problem.

OUT

OUT

generating the feedback error signal that

OUT

to its steady-state value. During this recovery

can be monitored for excessive overshoot or

(ESR), where ESR is the effective

LOAD

OUT

(∆I

) also begins to charge or

LOAD

shifts by an

OUT

The
availability of the ITH pin not only allows optimization of
control loop behavior but also provides a DC coupled and
AC filtered closed loop response test point. The DC step,
rise time, and settling at this test point truly reflects the
closed loop response.

order system, phase margin and/or damping factor can be
estimated using the percentage of overshoot seen at this
pin. The bandwidth can also be estimated by examining
the rise time at the pin. The ITH external components
shown in the Figure 1 circuit will provide an adequate
starting point for most applications.

The I
loop compensation. The values can be modified slightly
(from 0.2 to 5 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be decided
upon first because the various types and values determine
the loop gain and phase. An output current pulse of 20%
to 80% of full-load current having a rise time of <2µs will
produce output voltage and ITH pin waveforms that will
give a sense of the overall loop stability without breaking
the feedback loop. The initial output voltage step resulting
from the step change in output current may not be within
the bandwidth of the feedback loop, so this signal cannot
be used to determine phase margin. This is why it is
better to look at the Ith pin signal which is in the feedback

series RC-CC filter sets the dominant pole-zero

TH

Assuming a predominantly second

22

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

loop and is the filtered and compensated control loop
response. The gain of the loop will be increased by
increasing RC and the bandwidth of the loop will be
increased by decreasing CC. If RC is increased by the
same factor that CC is decreased, the zero frequency will
be kept the same, thereby keeping the phase the same in
the most critical frequency range of the feedback loop.
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance.

Automotive Considerations: Plugging into the
Cigarette Lighter

As battery-powered devices go mobile, there is a natural
interest in plugging into the cigarette lighter in order to
conserve or even recharge battery packs during opera­tion. But before you connect, be advised: you are plugging
into the supply from hell. The main battery line in an
automobile is the source of a number of nasty potential
transients, including load-dump, reverse-battery and
double-battery.

Load-dump is the result of a loose battery cable. When the
cable breaks connection, the field collapse in the alternator
can cause a positive spike as high as 60V which takes
several hundred milliseconds to decay. Reverse-battery is
just what it says, while double-battery is a consequence of
tow truck operators finding that a 24V jump start cranks
cold engines faster than 12V.

The network shown in Figure 9 is the most straightfor­ward approach to protect a DC/DC converter from the
ravages of an automotive power line. The series diode
prevents current from flowing during reverse-battery,

50A I

RATING

PK

12V

TRANSIENT VOLTAGE

SUPPRESSOR

GENERAL INSTRUMENT

1.5KA24A

Figure 9. Automotive Application Protection

V

IN

LTC3716

3716 F09

while the transient suppressor clamps the input voltage
during load-dump. Note that the transient suppressor
should not conduct during double-battery operation, but
must still clamp the input voltage below breakdown of the
converter. Although the LT3716 has a maximum input
voltage of 36V, most applications will be limited to 30V by
the MOSFET BV

DSS

.

Design Example

As a design example, assume VIN = 5V (nominal), VIN␣ =␣ 5.5V
(max), V

OUT

= 1.2V, I

= 20A, TA = 70°C and f␣ =␣ 300kHz.

MAX

The inductance value is chosen first based on a 30% ripple
current assumption. The highest value of ripple current
occurs at the maximum input voltage. Tie the FREQSET pin
to the INTVCC pin for 300kHz operation. The minimum
inductance for 30% ripple current is:

≥

fL

∆

()

≥

300 30 10

()()()

≥µ

104

.

−

1





12

kHz A

H



V

OUT OUT

L



V





V

IN

V

.

%



1





−

12

.

55

.

V






V

A 1µH inductor will produce 31% ripple current. The peak
inductor current will be the maximum DC value plus one
half the ripple current, or 11.5A. The minimum on-time
occurs at maximum VIN:

t

ON MIN

The R

V

OUT

==

()

SENSE

Vf

IN

5 5 300

.

()( )

resistors value can be calculated by using the

V

12

.

V kHz

073

.

s

=µ

maximum current sense voltage specification with some
accomodation for tolerances:

mV

R

SENSE

50

=≈Ω

11 5

0 004

.

A

.

The power dissipation on the topside MOSFET can be
easily estimated. Using a Siliconix Si4420DY for example;
R

= 0.013Ω, C

DS(ON)

= 300pF. At maximum input

RSS

voltage with TJ (estimated) = 110°C at an elevated ambient
temperature:

23

LTC3716

SENSE

+

SENSE

–

TRACE TO INDUCTOR

TRACE TO OUTPUT CAP (+)

PADS OF SENSE RESISTOR

3716 F10

WUUU

APPLICATIO S I FOR ATIO

V

.

P

MAIN

The worst-case power disipated by the synchronous
MOSFET under normal operating conditions at elevated
ambient temperature and estimated 50°C junction tem­perature rise is:

P

SYNC

A short-circuit to ground will result in a folded back current
of about:

I

SC

The worst-case power disipated by the synchronous
MOSFET under short-circuit conditions at elevated ambi­ent temperature and estimated 50°C junction temperature
rise is:

P

SYNC

which is less than normal, full-load conditions. Inciden­tally, since the load no longer dissipates power in the
shorted condition, total system power dissipation is de­creased by over 99%.

The duty factor for this application is:

12

=

.

55

...

0 013 1 7 5 5 10 300

300 0 45

()

..

55 12

=

.

=

15

mV

25

=

Ω

0 004

.

..

55 12

=
=

696

2

10 1 0 005 110 25

()+()

V

Ω

+

kHz W

=

VV

−

V

.

55

W



1

+



2



VV

−

V

.

55
mW

.

[]

2

VApF

()()( )

.

2

A

2 10 1 48 0 013

()()

200 5 5

()

.

ns V

()

µ

1

H

2

A

...

68 148 0013

()

CC

°− °

()

..

()



=

68




()

Ω

.

A

Ω

An input capacitor(s) with a 5A
required.

The output capacitor ripple current is calculated by using
the inductor ripple already calculated for each inductor
and multiplying by the factor obtained from Figure␣ 3
along with the calculated duty factor. The output ripple in
con

tinuous mode will be highest at the maximum input
voltage since the duty factor is <50%. The maximum
output current ripple is:

V

∆∆I

COUT

I

COUTMAX

VmAmV

OUTRIPPLE P P P P

PC Board Layout Checklist

When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC3716. Check the following in your layout:

1) Are the signal and power grounds separate? The signal
ground traces should return to Pin 9 first. Connect Pin 9
to Pin 28 through a wide and straight trace. Then the signal
ground joins the power ground plane beside Pin 28. It is
recommended that the Pin 28 return to the (–) plates of
CIN.

2) Does the LTC3716 V
load? Does the LTC3716 V
return?

OUT

=

fL

=

()

=

2

=Ω

20 2 40

at DF

05 24

.%

()

V

12

.

kHz H

300 1 0
A

PP

()

-

()

+

OS

ripple current rating is

RMS

05

.

µ

.

=

--

pin connect to the point of

–

pin connect to the load

OS

Using Figure 4, the RMS ripple current will be:

24

V

12

DF

I

INRMS

O

== =

V

IN

= (20A)(0.25) = 5A

V

5

V

024..

RMS

Figure 10. Proper Current Sense Connections

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

3) Are the SENSE– and SENSE+ leads routed together with
minimum PC trace spacing? The filter capacitors between
SENSE+ and SENSE– pin pairs should be as close as
possible to the LTC3716. Ensure accurate current sensing
with Kelvin connections at the current sense resistor. See
Figure 10.

4) Does the (+) plate of CIN connect to the drains of the
topside MOSFETs as closely as possible? This capacitor
provides the AC current to the MOSFETs. Keep the input
current path formed by the input capacitor, top and bottom
MOSFETs, and the Schottky diode on the same side of the
PC board in a tight loop to minimize conducted and
radiated EMI.

5) Is the INTVCC 1µF ceramic decoupling capacitor con-
nected closely between INTVCC and the PGND pin? This
capacitor carries the MOSFET driver peak currents. A
small value is recommended to allow placement immedi­ately adjacent to the IC.

6) Keep the switching nodes, SW1 (SW2), away from
sensitive small-signal nodes. Ideally the switch nodes
should be placed at the furthest point from the
LTC3716.

7) Use a low impedance source such as a logic gate to drive
the PLLIN pin and keep the lead as short as possible.

The diagram in Figure 11 illustrates all branch currents in
a 2-phase switching regulator. It becomes very clear after
studying the current waveforms why it is critical to keep
the high-switching-current paths to a small physical size.
High electric and magnetic fields will radiate from these
“loops” just as radio stations transmit signals. The output
capacitor ground should return to the negative terminal of
the input capacitor and not share a common ground path
with any switched current paths. The left half of the circuit
gives rise to the “noise” generated by a switching regula­tor. The ground terminations of the sychronous MOSFETs
and Schottky diodes should return to the negative plate(s)

V

IN

R

IN

+

C

IN

BOLD LINES INDICATE
HIGH, SWITCHING
CURRENT LINES.
KEEP LINES TO A
MINIMUM LENGTH.

SW1

SW2

L1

R

SENSE1

D1

V

OUT

C

OUT

+

L2

R

SENSE2

D2

3716 F11

R

L

Figure 11. Instantaneous Current Path Flow in a Multiple Phase Switching Regulator

25

LTC3716

U

WUU

APPLICATIO S I FOR ATIO

of the input capacitor(s) with a short isolated PC trace
since very high switched currents are present. A separate
isolated path from the negative plate(s) of the input
capacitor(s) should be used to tie in the IC power ground
pin (PGND) and the signal ground pin (SGND). This
technique keeps inherent signals generated by high cur­rent pulses from taking alternate current paths that have
finite impedances during the total period of the switching
regulator. External OPTI-LOOP compensation allows over­compensation for PC layouts which are not optimized but
this is not the recommended design procedure.

Simplified Visual Explanation of How a 2-Phase
Controller Reduces Both Input and Output RMS Ripple
Current

A multiphase power supply significantly reduces the
amount of ripple current in both the input and output
capacitors. The RMS input ripple current is divided by, and
the effective ripple frequency is multiplied up by the
number of phases used (assuming that the input voltage
is greater than the number of phases used times the output
voltage). The output ripple amplitude is also reduced by,
and the effective ripple frequency is increased by the
number of phases used. Figure 12 graphically illustrates
the principle.

The worst-case RMS ripple current for a single stage
design peaks at an input voltage of twice the output

DUAL PHASESINGLE PHASE

SW V

I

CIN

I

COUT

Figure 12. Single and 2-Phase Current Waveforms

SW1 V

SW2 V

I

I

COUT

I

I

CIN

L1

L2

RIPPLE

3216 F12

voltage. The worst-case RMS ripple current for a two stage
design results in peak outputs of 1/4 and 3/4 of input
voltage. When the RMS current is calculated, higher
effective duty factor results and the peak current levels are
divided as long as the currents in each stage are balanced.
Refer to Application Note 19 for a detailed description of
how to calculate RMS current for the single stage switch­ing regulator. Figures 3 and 4 illustrate how the input and
output currents are reduced by using an additional phase.
The input current peaks drop in half and the frequency is
doubled for this 2-phase converter. The input capacity
requirement is thus reduced theoretically by a factor of
four! Ceramic input capacitors with their unbeatably low
ESR characteristics can be used.

Figure 4 illustrates the RMS input current drawn from the
input capacitance vs the duty cycle as determined by the
ratio of input and output voltage. The peak input RMS
current level of the single phase system is reduced by 50%
in a 2-phase solution due to the current splitting between
the two stages.

An interesting result of the 2-phase solution is that the V

IN

which produces worst-case ripple current for the input
capacitor, V

= VIN/2, in the single phase design pro-

OUT

duces zero input current ripple in the 2-phase design.
The output ripple current is reduced significantly when

compared to the single phase solution using the same
inductance value because the V

/L discharge current

OUT

term from the stage that has its bottom MOSFET on
subtracts current from the (VIN – V

)/L charging current

OUT

resulting from the stage which has its top MOSFET on. The
output ripple current is:

∆I

RIPPLE



DD

−−

OUT

fL

12 1






V

2

=

()

D

−+

12 1








where D is duty factor.
The input and output ripple frequency is increased by the

number of stages used, reducing the output capacity
requirements. When VIN is approximately equal to 2(V

OUT

)
as illustrated in Figures 3 and 4, very low input and output
ripple currents result.

26

TYPICAL APPLICATIO

LTC3716

U

Figure 13 shows a typical application using LTC3716 to
power the mobile CPU core. The input can vary from 7V to
24V, the output voltage can be programmed from 0.6V to

1.75V with a maximum current of 30A. This power supply
receives three input signals to generate different output
voltage offsets based on the operation conditions. With
the AMPMD pin of LTC3716 tied to INTVCC, the LTC3716
provides a regular operational amplifier to implement

R4

10k

10k

R15

R17
22k

R1

1N4148

0.47µF

845k

R

c

432k

Q13
2N7002

249k

R

R11

1%

b

OPTIONAL

R12
1M

R

d,

R13
10k

15.4k, 1%

C15

47pF

VID0

VID1

C18

1000pF

C1

1000pF

1

2

3

4

5

6

7

8

9

10

11

12

13

14

15

16

17

18

LTC3716

RUN/SS

SENSE1

SENSE1

EAIN

PLLFLTR

PLLIN

FCB

i

TH

SGND

V

DIFFOUT

–

V

OS

+

V

OS

SENSE2

SENSE2

ATTENOUT

ATTENIN

VID0

VID1

1µF

+

–

–

+

R23

10Ω

C1

PGOOD

TG1

SW1

BOOST1

BG1

EXTV

INTV

PGND

BG2

BOOST2

SW2

TG2

AMPMD

V

BIAS

VID4

VID3

VID2

36

35

34

33

32

V

IN

31

30

5V (0PT)

CC

29

CC

R24
10Ω

C11
1µF

28

27

26

25

24

23

22

21

VID4

20

VID3

19

VID2

3.3V

R3

R2

10k

PWRGOOD

10k

Q1

FMMT3906

Q2

FMMT3904

DPSLP#

CIN: FOUR CERAMIC CAPS (10µF/35V)
C
L1, L2: SUMIDA CEP125-1ROMC-H

R8

10k

V

C7

C9

INTV

C8

330pF

CC

R29
100k

RON

75k, 1%

D3
BAT54A

DPRSLPVR

C5

0.1µF

R10

3.9k

0.01µF

1000pF

C12

0.01µF

Q14
FMMT3904

R30
10k

2N7002

: FOUR PANASONIC SP CAPS EEFUEOD271R (270µF/2V)

OUT

these offsets. When GMUXSEL is low, the output voltage
is offset –1.2% from the VID command voltage. The offset
equals Ra/Rb. When DPSLP# is low, the output voltage is
decreased by approximately 4%. This offset can be in­creased by decreasing Rc and will be disabled when the
DPRSLPVR signal is high. The optional filtering circuit (Q1
and Q2) is used to mask the PWRGOOD during the VID
transitions.

R1

10Ω

R5

10Ω

R6

10Ω

INTV

C6

0.47µF

+

2N7002

2N7002

BAT54A

1

C10
10µF
6V

C16

0.1µF

Q12

3

D1

C13, 0.47µF

GMUXSEL

2

845k

CC

4

44

4

R

b

44

4.99k

R

a

5 6 7 8

Q3
IRF7811

1 2 3

5 6 7 8

Q4
IRF7811

1 2 3

5 6 7 8

Q9
IRF7811

1 2 3

5 6 7 8

Q10
IRF7811

1 2 3

C3
1µF

5 6 7 8

Q5
IRF7811

1 2 3

C14
1µF

5 6 7 8

Q11
IRF7811

1 2 3

C

IN

L1

1µH

D2
MBRS130LT3

L2

1µH

D4
MBRS130LT3

0Ω

R9

3

4

0.003

1

2

R21

0.003

2

1

3

+

4

1µF

V

5V TO 22V

C

OUT

R25
10Ω

10Ω

IN

V

GND

FB

FB

3716 F13

OUT

+

–

Figure 13. 5V to 20V Input, 0.6V to 1.75V/30A IMVPII Compatible Power Supply with Active Voltage Positioning

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

27

LTC3716

PACKAGE DESCRIPTIO

5.20 – 5.38**
(.205 – .212)

U

G Package

36-Lead Plastic SSOP (5.3mm)

(Reference LTC DWG # 05-08-1640)

12345678 9 10 11 12 14 15 16 17 1813

° – 8°

0

12.67 – 12.93*
(.499 – .509)

2526 22 21 20 19232427282930313233343536

7.65 – 7.90

(.301 – .311)

1.73 – 1.99

(.068 – .078)

.13 – .22

(.005 – .009)

NOTE:

1. CONTROLLING DIMENSION: MILLIMETERS

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE
*

DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED .152mm (.006") PER SIDE

**

DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE

.55 – .95

(.022 – .037)

MILLIMETERS

(INCHES)

.65

(.0256)

BSC

.25 – .38

(.010 – .015)

.05 – .21

(.002 – .008)

G36 SSOP 0501

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OUT

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are a trademarks of Linear Technology Corporation.

SENSE

≤ 3.5V, Current Mode Ensures

OUT

≤ 36V, 0.925V ≤ V

IN

IN

≤ 36V

OUT

IN

≤ 2V

≤ 36V

28

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone] ● www.linear-tech.com

3716f LT/TP 0601 2K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 2001