Datasheet LTC1778, LTC1778-1 Datasheet (LINEAR TECHNOLOGY)

FEATURES

No R

DESCRIPTIO

SENSE

LTC1778/LTC1778-1

Wide Operating Range,

TM

Step-Down Controller

U

■

No Sense Resistor Required

■

True Current Mode Control

■

Optimized for High Step-Down Ratios

■

t

ON(MIN)

■

Extremely Fast Transient Response

■

Stable with Ceramic C

■

Dual N-Channel MOSFET Synchronous Drive

■

Power Good Output Voltage Monitor (LTC1778)

■

Adjustable On-Time (LTC1778-1)

■

Wide VIN Range: 4V to 36V

■

±1% 0.8V Voltage Reference

■

Adjustable Current Limit

■

Adjustable Switching Frequency

■

Programmable Soft-Start

■

Output Overvoltage Protection

■

Optional Short-Circuit Shutdown Timer

■

Micropower Shutdown: IQ < 30µA

■

Available in a 16-Pin Narrow SSOP Package

≤ 100ns

OUT

U

APPLICATIO S

■

Notebook and Palmtop Computers

■

Distributed Power Systems

The LTC®1778 is a synchronous step-down switching regulator controller optimized for CPU power. The controller uses a valley current control architecture to deliver very low duty cycles with excellent transient response without requiring a sense resistor. Operating frequency is selected by an external resistor and is compensated for variations in VIN.

Discontinuous mode operation provides high efficiency operation at light loads. A forced continuous control pin reduces noise and RF interference, and can assist secondary winding regulation by disabling discontinuous operation when the main output is lightly loaded.

Fault protection is provided by internal foldback current limiting, an output overvoltage comparator and optional short-circuit shutdown timer. Soft-start capability for supply sequencing is accomplished using an external timing capacitor. The regulator current limit level is user programmable. Wide supply range allows operation from 4V to 36V at the input and from 0.8V up to (0.9)VIN at the output.

, LTC and LT are registered trademarks of Linear Technology Corporation. No R All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5481178, 6100678, 6580258, 5847554, 6304066

is a trademark of Linear Technology Corporation.

SENSE

TYPICAL APPLICATIO

R

ON

1.4MΩ

I

BOOST

LTC1778

INTV

V

SW

BG

PGND

V

ON

IN

TG

CB 0.22µF

D

B

CMDSH-3

CC

+

C

4.7µF

FB

C

500pF

C

SS

0.1µF

RUN/SS

I

TH

R

C

20k

SGND

PGOOD

Figure 1. High Efficiency Step-Down Converter

VCC

U

M1 Si4884

M2 Si4874

L1

1.8µH

D1 B340A

+

R2

30.1k

R1 14k

1778 F01a

C

IN

10µF 50V ×3

C

OUT

180µF 4V ×2

V

IN

5V TO 28V

V

OUT

2.5V 10A

Efficiency vs Load Current

100

V

OUT

90

80

EFFICIENCY (%)

70

60

0.01

= 2.5V

0.1 LOAD CURRENT (A)

VIN = 5V

VIN = 25V

1

10

1778 F01b

1778fb

1

LTC1778/LTC1778-1

TOP VIEW

GN PACKAGE

16-LEAD PLASTIC SSOP

1

2

3

4

5

6

7

8

16

15

14

13

12

11

10

9

RUN/SS

V

ON

V

RNG

FCB

I

TH

SGND

I

ON

V

FB

BOOST

TG

SW

PGND

BG

INTV

CC

V

IN

EXTV

CC

WWWU

ABSOLUTE AXI U RATI GS

(Note 1)

Input Supply Voltage (VIN, ION)................. 36V to –0.3V

Boosted Topside Driver Supply Voltage

(BOOST) ................................................... 42V to –0.3V

SW Voltage .................................................. 36V to –5V

EXTV

, (BOOST – SW), RUN/SS,

CC

PGOOD Voltages....................................... 7V to – 0.3V

FCB, V I

TH

, V

ON

Voltages .......... INTVCC + 0.3V to –0.3V

RNG

, VFB Voltages...................................... 2.7V to –0.3V

UU

W

PACKAGE/ORDER I FOR ATIO

RUN/SS

1

2

PGOOD

3

V

RNG

4

FCB

5

I

TH

6

SGND

7

I

ON

8

V

FB

16-LEAD PLASTIC SSOP

T

= 125°C, θJA = 130°C/ W

JMAX

TOP VIEW

GN PACKAGE

BOOST

16

15

TG

14

SW

13

PGND

12

BG

11

INTV

CC

10

V

IN

9

EXTV

CC

ORDER PART

NUMBER

LTC1778EGN LTC1778IGN

GN PART MARKING

1778 1778I

TG, BG, INTVCC, EXTVCC Peak Currents.................... 2A

TG, BG, INTVCC, EXTVCC RMS Currents .............. 50mA

Operating Ambient Temperature Range (Note 4)

LTC1778E ........................................... – 40°C to 85°C

LTC1778I.......................................... – 40°C to 125°C

Junction Temperature (Note 2)............................ 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

NUMBER

LTC1778EGN-1

GN PART MARKING

17781

T

= 125°C, θJA = 130°C/ W

JMAX

Consult LTC Marketing for parts specified with wider operating temperature ranges.

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

Main Control Loop

I

Q

V

FB

∆V

FB(LINEREG)

∆V

FB(LOADREG)

I

FB

g

m(EA)

V

FCB

I

FCB

t

ON

t

ON(MIN)

2

Input DC Supply Current Normal 900 2000 µA Shutdown Supply Current 15 30 µA

Feedback Reference Voltage ITH = 1.2V (Note 3) LTC1778E ● 0.792 0.800 0.808 V

= 1.2V (Note 3) LTC1778I ● 0.792 0.800 0.812 V

I

TH

Feedback Voltage Line Regulation VIN = 4V to 30V, ITH = 1.2V (Note 3) 0.002 %/V

Feedback Voltage Load Regulation ITH = 0.5V to 1.9V (Note 3) ● –0.05 –0.3 %

Feedback Input Current VFB = 0.8V –5 ±50 nA

Error Amplifier Transconductance ITH = 1.2V (Note 3) ● 1.4 1.7 2 mS

Forced Continuous Threshold ● 0.76 0.8 0.84 V

Forced Continuous Pin Current V

On-Time ION = 30µA, VON = 0V (LTC1778-1) 198 233 268 ns

Minimum On-Time ION = 180µA 50 100 ns

= 0.8V –1 –2 µA

FCB

I

= 15µA, VON = 0V (LTC1778-1) 396 466 536 ns

ON

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LTC1778/LTC1778-1

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are TA = 25°C. VIN = 15V unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

t

OFF(MIN)

V

SENSE(MAX)

V

SENSE(MIN)

∆V

FB(OV)

V

FB(UV)

V

RUN/SS(ON)

V

RUN/SS(LE)

V

RUN/SS(LT)

I

RUN/SS(C)

I

RUN/SS(D)

V

IN(UVLO)

V

IN(UVLOR)

TG R

UP

TG R

DOWN

BG R

UP

BG R

DOWN

TG t

r

TG t

f

BG t

r

BG t

f

Internal VCC Regulator

V

INTVCC

∆V

LDO(LOADREG)

V

EXTVCC

∆V

EXTVCC

∆V

EXTVCC(HYS)

PGOOD Output (LTC1778 Only)

∆V

FBH

∆V

FBL

∆V

FB(HYS)

V

PGL

Minimum Off-Time ION = 30µA 250 400 ns

Maximum Current Sense Threshold V

– V

V

PGND

SW

Minimum Current Sense Threshold V

– V

V

PGND

SW

= 1V, VFB = 0.76V ● 113 133 153 mV

RNG

V

= 0V, VFB = 0.76V ● 79 93 107 mV

RNG

V

= INTVCC, VFB = 0.76V ● 158 186 214 mV

RNG

= 1V, VFB = 0.84V – 67 mV

RNG

V

= 0V, VFB = 0.84V – 47 mV

RNG

= INTVCC, VFB = 0.84V – 93 mV

V

RNG

Output Overvoltage Fault Threshold 5.5 7.5 9.5 %

Output Undervoltage Fault Threshold 520 600 680 mV

RUN Pin Start Threshold ● 0.8 1.5 2 V

RUN Pin Latchoff Enable Threshold RUN/SS Pin Rising 4 4.5 V

RUN Pin Latchoff Threshold RUN/SS Pin Falling 3.5 4.2 V

Soft-Start Charge Current V

Soft-Start Discharge Current V

= 0V –0.5 –1.2 –3 µA

RUN/SS

= 4.5V, VFB = 0V 0.8 1.8 3 µA

RUN/SS

Undervoltage Lockout VIN Falling ● 3.4 3.9 V

Undervoltage Lockout Release VIN Rising ● 3.5 4 V

TG Driver Pull-Up On Resistance TG High 2 3 Ω

TG Driver Pull-Down On Resistance TG Low 2 3 Ω

BG Driver Pull-Up On Resistance BG High 3 4 Ω

BG Driver Pull-Down On Resistance BG Low 1 2 Ω

TG Rise Time C

TG Fall Time C

BG Rise Time C

BG Fall Time C

Internal VCC Voltage 6V < VIN < 30V, V

Internal VCC Load Regulation ICC = 0mA to 20mA, V

EXTVCC Switchover Voltage ICC = 20mA, V

EXTVCC Switch Drop Voltage ICC = 20mA, V

= 3300pF 20 ns

LOAD

= 3300pF 20 ns

LOAD

= 3300pF 20 ns

LOAD

= 3300pF 20 ns

LOAD

= 4V ● 4.7 5 5.3 V

EXTVCC

= 4V –0.1 ±2%

EXTVCC

Rising ● 4.5 4.7 V

EXTVCC

= 5V 150 300 mV

EXTVCC

EXTVCC Switchover Hysteresis 200 mV

PGOOD Upper Threshold VFB Rising 5.5 7.5 9.5 %

PGOOD Lower Threshold VFB Falling –5.5 –7.5 –9.5 %

PGOOD Hysteresis VFB Returning 1 2 %

PGOOD Low Voltage I

= 5mA 0.15 0.4 V

PGOOD

Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired.

Note 2: T dissipation P

is calculated from the ambient temperature TA and power

J

as follows:

D

LTC1778E: TJ = TA + (PD • 130°C/W)

Note 3: The LTC1778 is tested in a feedback loop that adjusts V a specified error amplifier output voltage (I

).

TH

to achieve

FB

Note 4: The LTC1778E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC1778I is guaranteed over the full – 40°C to 125°C operating temperature range.

1778fb

3

LTC1778/LTC1778-1

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Transient Response

(Discontinuous Mode)

Start-Up

V

OUT

50mV/DIV

I

L

5A/DIV

LOAD STEP 0A TO 10A

= 15V

V

IN

V

= 2.5V

OUT

FCB = 0V FIGURE 9 CIRCUIT

Efficiency vs Load Current

100

DISCONTINUOUS

90

80

70

EFFICIENCY (%)

60

50

0.001

MODE

0.01 LOAD CURRENT (A)

20µs/DIV

= 10V

V

IN

V

OUT

EXTV FIGURE 9 CIRCUIT

0.1

CONTINUOUS MODE

= 2.5V

= 5V

CC

1

1778 G03

1778 G01

10

V

OUT

50mV/DIV

I

L

5A/DIV

LOAD STEP 1A TO 10A

= 15V

V

IN

V

= 2.5V

OUT

FCB = INTV

CC

FIGURE 9 CIRCUIT

Efficiency vs Input Voltage

100

FCB = 5V FIGURE 9 CIRCUIT

95

I

LOAD

90

I

= 10A

LOAD

EFFICIENCY (%)

85

80

0

5101520

INPUT VOLTAGE (V)

20µs/DIV

= 1A

1778 G02

25 30

1778 G04

RUN/SS

2V/DIV

V

OUT

1V/DIV

I

L

5A/DIV

VIN = 15V

= 2.5V

V

OUT

R

= 0.25Ω

LOAD

Frequency vs Input Voltage

300

FCB = 0V FIGURE 9 CIRCUIT

280

260

240

FREQUENCY (kHz)

220

200

5

I

10

INPUT VOLTAGE (V)

50ms/DIV

= 10A

OUT

I

= 0A

OUT

15

1778 G19

20

25

1778 G05

Frequency vs Load Current

300

CONTINUOUS MODE

250

200

150

100

FREQUENCY (kHz)

50

0

DISCONTINUOUS MODE

2468

LOAD CURRENT (A)

4

1778 G26

Load Regulation

0

–0.1

(%)

–0.2

OUT

∆V

–0.3

10

–0.4

2

0

LOAD CURRENT (A)

4

FIGURE 9 CIRCUIT

6

8

1778 G06

10

ITH Voltage vs Load Current

2.5 FIGURE 9 CIRCUIT

2.0

1.5

CONTINUOUS

VOLTAGE (V)

1.0

TH

I

0.5

0

MODE

DISCONTINUOUS MODE

5

LOAD CURRENT (A)

10

15

1778 G07

1778fb

UW

TEMPERATURE (°C)

–50

0.78

FEEDBACK REFERENCE VOLTAGE (V)

0.79

0.80

0.81

0.82

–25 0 25 50

1778 G12

75 100 125

TYPICAL PERFOR A CE CHARACTERISTICS

Current Sense Threshold vs ITH Voltage

300

RNG

2V

=

V

On-Time vs ION Current

10k

V

VON

= 0V

LTC1778/LTC1778-1

On-Time vs VON Voltage

1000

I

ION

= 30µA

200

100

0

–100

CURRENT SENSE THRESHOLD (mV)

–200

0

1.0 1.5 2.0

0.5 ITH VOLTAGE (V)

On-Time vs Temperature

300

I

= 30µA

ION

= 0V

V

VON

250

200

150

ON-TIME (ns)

100

50

1.4V

1V

0.7V

0.5V

2.5 3.0

1778 G08

1k

ON-TIME (ns)

100

10

1

ION CURRENT (µA)

Current Limit Foldback

150

125

100

= 1V

V

RNG

75

50

25

10 100

1778 G20

800

600

400

ON-TIME (ns)

200

0

1

VON VOLTAGE (V)

Maximum Current Sense Threshold vs V

300

250

200

150

100

50

RNG

2

Voltage

3

1778 G21

0

–50

–25 0

Maximum Current Sense Threshold vs RUN/SS Voltage

150

125

100

MAXIMUM CURRENT SENSE THRESHOLD (mV)

= 1V

V

RNG

75

50

25

0

1.5

2 2.5 3 3.5

RUN/SS VOLTAGE (V)

50 100 125

25 75

TEMPERATURE (°C)

1778 G22

1778 G23

MAXIMUM CURRENT SENSE THRESHOLD (mV)

0

0.2 0.4 0.6 0.8 VFB (V)

Maximum Current Sense Threshold vs Temperature

150

V

= 1V

RNG

140

130

120

110

MAXIMUM CURRENT SENSE THRESHOLD (mV)

100

–50 –25

0

TEMPERATURE (°C)

50

25

75

1778 G09

100

1778 G11

125

MAXIMUM CURRENT SENSE THRESHOLD (mV)

0

0.5

0.75

1.0 1.25 1.5 V

VOLTAGE (V)

RNG

1.75 2.0

Feedback Reference Voltage vs Temperature

1778 G10

1778fb

5

LTC1778/LTC1778-1

TEMPERATURE (C)

–50

2.0

UNDERVOLTAGE LOCKOUT THRESHOLD (V)

2.5

3.0

3.5

4.0

–25 0 25 50

1778 G18

75 100 125

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Input and Shutdown Currents

Error Amplifier g

2.0

1.8

vs Temperature

m

vs Input Voltage

1200

1000

EXTVCC OPEN

INTV

Load Regulation

CC

60

50

SHUTDOWN CURRENT (µA)

0

–0.1

1.6

(mS)

m

g

1.4

1.2

1.0 –50 –25

25

0

TEMPERATURE (°C)

EXTVCC Switch Resistance vs Temperature

10

8

6

4

SWITCH RESISTANCE (Ω)

CC

2

EXTV

800

600

400

INPUT CURRENT (µA)

200

50

75

100

125

1778 G13

0

510

SHUTDOWN

EXTVCC = 5V

20 30 35

15 25

INPUT VOLTAGE (V)

1778 G24

40

30

20

10

0

–0.2

(%)

CC

–0.3

∆INTV

–0.4

–0.5

10

0

INTVCC LOAD CURRENT (mA)

30

40

20

50

1778 G25

RUN/SS Pin Current

FCB Pin Current vs Temperature

0

–0.25

–0.50

–0.75

–1.00

FCB PIN CURRENT (µA)

–1.25

vs Temperature

3

2

1

0

FCB PIN CURRENT (µA)

–1

PULL-DOWN CURRENT

PULL-UP CURRENT

0

–50 –25

6

25

0

TEMPERATURE (°C)

RUN/SS THRESHOLD (V)

50

75

100

125

1778 G14

RUN/SS Latchoff Thresholds vs Temperature

5.0

4.5

LATCHOFF ENABLE

4.0

3.5

3.0 –50

LATCHOFF THRESHOLD

–25 0 25 50

TEMPERATURE (°C)

–1.50

–50

–25 0

75 100 125

1778 G17

50 100 125

25 75

TEMPERATURE (°C)

1778 G15

–2

–50 –25

25

0

TEMPERATURE (°C)

Undervoltage Lockout Threshold vs Temperature

50

75

100

125

1778 G16

1778fb

LTC1778/LTC1778-1

U

UU

PI FU CTIO S

RUN/SS (Pin 1): Run Control and Soft-Start Input. A capacitor to ground at this pin sets the ramp time to full output current (approximately 3s/µF) and the time delay for overcurrent latchoff (see Applications Information). Forcing this pin below 0.8V shuts down the device.

PGOOD (Pin 2, LTC1778): Power Good Output. Open drain logic output that is pulled to ground when the output voltage is not within ±7.5% of the regulation point.

(Pin 2, LTC1778-1): On-Time Voltage Input. Voltage

V

ON

trip point for the on-time comparator. Tying this pin to the output voltage or an external resistive divider from the output makes the on-time proportional to V comparator input defaults to 0.7V when the pin is grounded or unavailable (LTC1778) and defaults to 2.4V when the pin is tied to INTVCC. Tie this pin to INTVCC in high V applications to use a lower RON value.

V

(Pin 3): Sense Voltage Range Input. The voltage at

RNG

this pin is ten times the nominal sense voltage at maximum output current and can be set from 0.5V to 2V by a resistive divider from INTVCC. The nominal sense voltage defaults to 70mV when this pin is tied to ground, 140mV when tied to INTVCC.

FCB (Pin 4): Forced Continuous Input. Tie this pin to ground to force continuous synchronous operation at low load, to INTVCC to enable discontinuous mode operation at low load or to a resistive divider from a secondary output when using a secondary winding.

I

(Pin 5): Current Control Threshold and Error Amplifier

TH

Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.8V corresponding to zero sense voltage (zero current).

SGND (Pin 6): Signal Ground. All small-signal components and compensation components should connect to this ground, which in turn connects to PGND at one point.

OUT

. The

OUT

ION (Pin 7): On-Time Current Input. Tie a resistor from V to this pin to set the one-shot timer current and thereby set the switching frequency.

VFB (Pin 8): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistive divider from V

EXTVCC (Pin 9): External VCC Input. When EXTVCC ex- ceeds 4.7V, an internal switch connects this pin to INTV and shuts down the internal regulator so that controller and gate drive power is drawn from EXTVCC. Do not exceed 7V at this pin and ensure that EXTVCC < VIN.

VIN (Pin 10): Main Input Supply. Decouple this pin to PGND with an RC filter (1Ω, 0.1µF).

INTVCC (Pin 11): Internal 5V Regulator Output. The driver and control circuits are powered from this voltage. Decouple this pin to power ground with a minimum of 4.7µF low ESR tantalum capacitor.

BG (Pin 12): Bottom Gate Drive. Drives the gate of the bottom N-channel MOSFET between ground and INTVCC.

PGND (Pin 13): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (–) terminal of C

SW (Pin 14): Switch Node. The (–) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below ground up to VIN.

TG (Pin 15): Top Gate Drive. Drives the top N-channel MOSFET with a voltage swing equal to INTVCC superimposed on the switch node voltage SW.

BOOST (Pin 16): Boosted Floating Driver Supply. The (+) terminal of the bootstrap capacitor CB connects here. This pin swings from a diode voltage drop below INTVCC up to V

+ INTVCC.

IN

.

OUT

and the (–) terminal of CIN.

VCC

IN

CC

1778fb

7

LTC1778/LTC1778-1

U

W

FU CTIO AL DIAGRA

R

ON

**

I

7

ON

R

SQ

20k

+

–

+

I

REV

–

×

0.7V22.4V

1

tON = (10pF)

1.4V

V

RNG

3

0.7V

V

I

CMP

ON

I

VON ION

1µA

V

IN

V

0.8V REF

5V

REG

10

BOOST

16

TG

15

SW

14

INTV

CC

11

BG

12

PGND

13

PGOOD*

2

IN

+

C

IN

C

B

M1

L1

D

B

V

OUT

+

C

VCC

M2

OUT

FCB

4

–

F

0.8V

+

4.7V

+

–

SHDN

FCNT

ON

OV

9

SWITCH

EXTV

LOGIC

CC

1

240k

I

+ –

×4

LTC1778

*

LTC1778-1

**

THB

0.8V

3.3µA

0.74V

Q2

Q4

Q6

Q3

Q1

EA

–

+

0.8V

1V

Q5

0.6V

RUN

SHDN

1

RUN/SS

SS

–

+

–

0.6V

C

I

5

C1

TH

R

C

UV

OV

1.2µA

6V

+

–

+

0.86V

–

C

SS

V

FB

8

SGND

6

1778 FD

R2

R1

8

1778fb

OPERATIO

LTC1778/LTC1778-1

U

Main Control Loop

The LTC1778 is a current mode controller for DC/DC step-down converters. In normal operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator I ating the next cycle. Inductor current is determined by sensing the voltage between the PGND and SW pins using the bottom MOSFET on-resistance . The voltage on the I pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this voltage by comparing the feedback signal VFB from the output voltage with an internal 0.8V reference. If the load current increases, it causes a drop in the feedback voltage relative to the reference. The ITH voltage then rises until the average inductor current again matches the load current.

At low load currents, the inductor current can drop to zero and become negative. This is detected by current reversal comparator I discontinuous operation. Both switches will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.8V) to initiate another cycle. Discontinuous mode operation is disabled by comparator F when the FCB pin is brought below 0.8V, forcing continuous synchronous operation.

The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an ontime that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor RON.

Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exits a ±7.5% window around the regulation point.

trips, restarting the one-shot timer and initi-

CMP

which then shuts off M2, resulting in

REV

TH

Furthermore, in an overvoltage condition, M1 is turned off and M2 is turned on and held on until the overvoltage condition clears.

Foldback current limiting is provided if the output is shorted to ground. As VFB drops, the buffered current threshold voltage I level set by Q4 and Q6. This reduces the inductor valley current level to one sixth of its maximum value as V approaches 0V.

Pulling the RUN/SS pin low forces the controller into its shutdown state, turning off both M1 and M2. Releasing the pin allows an internal 1.2µA current source to charge up an external soft-start capacitor CSS. When this voltage reaches 1.5V, the controller turns on and begins switching, but with the ITH voltage clamped at approximately

0.6V below the RUN/SS voltage. As CSS continues to charge, the soft-start current limit is removed.

INTVCC/EXTVCC Power

Power for the top and bottom MOSFET drivers and most of the internal controller circuitry is derived from the INTVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is recharged from INTVCC through an external Schottky diode DB when the top MOSFET is turned off. When the EXTV pin is grounded, an internal 5V low dropout regulator supplies the INTVCC power from VIN. If EXTVCC rises above 4.7V, the internal regulator is turned off, and an internal switch connects EXTVCC to INTVCC. This allows a high efficiency source connected to EXTVCC, such as an external 5V supply or a secondary output from the converter, to provide the INTVCC power. Voltages up to 7V can be applied to EXTVCC for additional gate drive. If the input voltage is low and INTVCC drops below 3.5V, undervoltage lockout circuitry prevents the power switches from turning on.

is pulled down by clamp Q3 to a 1V

THB

FB

CC

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The basic LTC1778 application circuit is shown in Figure 1. External component selection is primarily determined by the maximum load current and begins with the selection of the sense resistance and power MOSFET switches. The LTC1778 uses the on-resistance of the synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and C

is chosen with

OUT

low enough ESR to meet the output voltage ripple and transient specification.

Choosing the LTC1778 or LTC1778-1

The LTC1778 has an open-drain PGOOD output that indicates when the output voltage is within ±7.5

% of the regulation point. The LTC1778-1 trades the PGOOD pin for a VON pin that allows the on-time to be adjusted. Tying the VON pin high results in lower values for RON which is useful in high V

applications. The VON pin also provides a

OUT

means to adjust the on-time to maintain constant frequency operation in applications where V

changes and

OUT

to correct minor frequency shifts with changes in load current. Finally, the VON pin can be used to provide additional current limiting in positive-to-negative converters and as a control input to synchronize the switching frequency with a phase locked loop.

Maximum Sense Voltage and V

RNG

Inductor current is determined by measuring the voltage across a sense resistance that appears between the PGND and SW pins. The maximum sense voltage is set by the voltage applied to the V mately (0.133)V

. The current mode control loop will

RNG

pin and is equal to approxi-

RNG

not allow the inductor current valleys to exceed (0.133)V

RNG/RSENSE

. In practice, one should allow some margin for variations in the LTC1778 and external component values and a good guide for selecting the sense resistance is:

V

R

SENSE

=

10 •

RNG

I

OUT MAX

()

An external resistive divider from INTVCC can be used to set the voltage of the V

pin between 0.5V and 2V

RNG

resulting in nominal sense voltages of 50mV to 200mV. Additionally, the V

pin can be tied to SGND or INTV

RNG

CC

in which case the nominal sense voltage defaults to 70mV or 140mV, respectively. The maximum allowed sense voltage is about 1.33 times this nominal value.

Power MOSFET Selection

The LTC1778 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage V threshold voltage V transfer capacitance C

, on-resistance R

(GS)TH

and maximum current I

RSS

DS(ON)

(BR)DSS

, reverse

DS(MAX)

,

.

The gate drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC1778 applications. If the input voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered.

When the bottom MOSFET is used as the current sense element, particular attention must be paid to its onresistance. MOSFET on-resistance is typically specified with a maximum value R

DS(ON)(MAX)

at 25°C. In this case, additional margin is required to accommodate the rise in MOSFET on-resistance with temperature:

R

DS ON MAX

()( )

=

SENSE

ρ

T

The ρT term is a normalization factor (unity at 25°C) accounting for the significant variation in on-resistance

2.0

1.5

1.0

0.5

NORMALIZED ON-RESISTANCE

T

ρ

0

–50

Figure 2. R

0

JUNCTION TEMPERATURE (°C)

50

vs. Temperature

DS(ON)

100

150

1778 F02

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f

V

VR pF

H

OUT

VON ON

Z

=

()

[]

10

APPLICATIO S I FOR ATIO

LTC1778/LTC1778-1

with temperature, typically about 0.4%/°C as shown in Figure 2. For a maximum junction temperature of 100°C, using a value ρT = 1.3 is reasonable.

The power dissipated by the top and bottom MOSFETs strongly depends upon their respective duty cycles and the load current. When the LTC1778 is operating in continuous mode, the duty cycles for the MOSFETs are:

V

D

TOP

BOT

OUT

=

V

IN

–

VV

IN OUT

=

V

IN

The resulting power dissipation in the MOSFETs at maximum output current are:

P

TOP

BOT

= D

TOP IOUT(MAX)

+ k V

IN

= D

BOT IOUT(MAX)

2

I

2

ρ

T(TOP) RDS(ON)(MAX)

OUT(MAX) CRSS

2

ρ

T(BOT) RDS(ON)(MAX)

f

Both MOSFETs have I2R losses and the top MOSFET includes an additional term for transition losses, which are largest at high input voltages. The constant k = 1.7A–1 can be used to estimate the amount of transition loss. The bottom MOSFET losses are greatest when the bottom duty cycle is near 100%, during a short-circuit or at high input voltage.

Operating Frequency

The choice of operating frequency is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance in order to maintain low output ripple voltage.

The operating frequency of LTC1778 applications is determined implicitly by the one-shot timer that controls the on-time tON of the top MOSFET switch. The on-time is set by the current into the ION pin and the voltage at the V

ON

pin (LTC1778-1) according to:

Tying a resistor RON from VIN to the ION pin yields an ontime inversely proportional to VIN. For a step-down converter, this results in approximately constant frequency operation as the input supply varies:

To hold frequency constant during output voltage changes, tie the VON pin to V when V

> 2.4V. The VON pin has internal clamps that

OUT

or to a resistive divider from V

OUT

limit its input to the one-shot timer. If the pin is tied below

0.7V, the input to the one-shot is clamped at 0.7V. Similarly, if the pin is tied above 2.4V, the input is clamped at

2.4V. In high V

applications, tying VON to INTVCC so

OUT

that the comparator input is 2.4V results in a lower value for RON. Figures 3a and 3b show how RON relates to switching frequency for several common output voltages.

1000

V

= 3.3V

OUT

V

= 1.5V

OUT

SWITCHING FREQUENCY (kHz)

100

Figure 3a. Switching Frequency vs R for the LTC1778 and LTC1778-1 (VON = 0V)

1000

V

OUT

SWITCHING FREQUENCY (kHz)

RON (kΩ)

= 3.3V

V

= 2.5V

OUT

1000 10000

V

OUT

V

OUT

1778 F03a

ON

= 12V

= 5V

t

ON

VON defaults to 0.7V in the LTC1778.

V

VON

= ()10

I

ION

pF

100

Figure 3b. Switching Frequency vs R for the LTC1778-1 (VON = INTVCC)

1000 10000

RON (kΩ)

1778 F03b

ON

1778fb

11

LTC1778/LTC1778-1

∆ =

⎛

⎝

⎜

⎞

⎠

⎟

−

⎛

⎝

⎜

⎞

⎠

⎟

I

V

fL

V

L

OUT OUT

IN

1

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APPLICATIO S I FOR ATIO

Because the voltage at the ION pin is about 0.7V, the current into this pin is not exactly inversely proportional to VIN, especially in applications with lower input voltages. To correct for this error, an additional resistor R

ON2

connected from the ION pin to the 5V INTVCC supply will further stabilize the frequency.

5

07=.

V

R

V

R

ON ON2

Changes in the load current magnitude will also cause frequency shift. Parasitic resistance in the MOSFET switches and inductor reduce the effective voltage across the inductance, resulting in increased duty cycle as the load current increases. By lengthening the on-time slightly as current increases, constant frequency operation can be maintained. This is accomplished with a resistive divider from the I

pin to the VON pin and V

TH

. The values

OUT

required will depend on the parasitic resistances in the specific application. A good starting point is to feed about 25% of the voltage change at the ITH pin to the VON pin as shown in Figure 4a. Place capacitance on the VON pin to filter out the ITH variations at the switching frequency. The resistor load on ITH reduces the DC gain of the error amp and degrades load regulation, which can be avoided by using the PNP emitter follower of Figure 4b.

due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is:

tt

+

VV

IN MIN OUT

=

()

ON OFF MIN

t

ON

()

A plot of maximum duty cycle vs frequency is shown in Figure 5.

Inductor Selection

Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current:

Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage

2.0

1.5

1.0

DROPOUT

REGION

Minimum Off-time and Dropout Operation

The minimum off-time t

OFF(MIN)

is the smallest amount of time that the LTC1778 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the MOSFET back off. This time is generally about 250ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + t

12

OFF(MIN)

). If the maximum duty cycle is reached,

R

VON1

30k

V

OUT

R

VON2

100k

R

C

VON

0.01µF

V

ON

LTC1778

I

TH

(4a) (4b)

Figure 4. Correcting Frequency Shift with Load Current Changes

0.5

SWITCHING FREQUENCY (MHz)

0

0 0.25 0.50 0.75

DUTY CYCLE (V

OUT/VIN

)

1.0

1778 F05

Figure 5. Maximum Switching Frequency vs Duty Cycle

R

VON1

INTV

3k

V

OUT

CC

10k

2N5087

R

VON2

10k

Q1

C

VON

0.01µF

R

C

V

ON

LTC1778

I

TH

1778 F04

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LTC1778/LTC1778-1

ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency.

A reasonable starting point is to choose a ripple current that is about 40% of I

OUT(MAX)

. The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance should be chosen according to:

⎛

=

⎜

fI

⎝

V

∆

L

⎞

⎛

OUT

⎟ ⎠

() ()

LMAX

V

OUT

−

1

⎜

V

⎝

IN MAX

⎞ ⎟

⎠

Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko.

Schottky Diode D1 Selection

The Schottky diode D1 shown in Figure 1 conducts during the dead time between the conduction of the power MOSFET switches. It is intended to prevent the body diode of the bottom MOSFET from turning on and storing charge during the dead time, which can cause a modest (about 1%) efficiency loss. The diode can be rated for about one half to one fifth of the full load current since it is on for only a fraction of the duty cycle. In order for the diode to be effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these components be placed adjacently. The diode can be omitted if the efficiency loss is tolerable.

CIN and C

Selection

OUT

The input capacitance CIN is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current.

II

≅

RMS OUT MAX

()

V

OUT

V

IN

V

IN

OUT

–1

This formula has a maximum at VIN = 2V I

RMS

= I

OUT(MAX)

/ 2. This simple worst-case condition is

OUT

, where

commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor.

The selection of C

is primarily determined by the ESR

OUT

required to minimize voltage ripple and load step transients. The output ripple ∆V

is approximately

OUT

bounded by:

∆≤∆ +

⎛

V I ESR

OUT L

⎜ ⎝

8

fC

1

OUT

⎞ ⎟

⎠

Since ∆IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating.

Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5µF to 50µF aluminum electrolytic capacitor with an ESR in the range of 0.5Ω to 2Ω. High performance through-hole capacitors may also be used,

Kool Mµ is a registered trademark of Magnetics, Inc.

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but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance.

Top MOSFET Driver Supply (C

An external bootstrap capacitor C

, DB)

B

connected to the BOOST

B

pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode D

from INTV

B

CC

when the switch node is low. When the top MOSFET turns on, the switch node rises to V to approximately V

+ INTVCC. The boost capacitor needs

IN

and the BOOST pin rises

IN

to store about 100 times the gate charge required by the top MOSFET. In most applications 0.1µF to 0.47µF, X5R or X7R dielectric capacitor is adequate.

Discontinuous Mode Operation and FCB Pin

The FCB pin determines whether the bottom MOSFET remains on when current reverses in the inductor. Tying this pin above its 0.8V threshold enables discontinuous operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and will vary with changes in VIN. Tying the FCB pin below the

0.8V threshold forces continuous synchronous operation, allowing current to reverse at light loads and maintaining high frequency operation.

In addition to providing a logic input to force continuous operation, the FCB pin provides a means to maintain a flyback winding output when the primary is operating in discontinuous mode. The secondary output V

OUT2

is nor-

mally set as shown in Figure 6 by the turns ratio N of the

V

IN

+

C

V

IN

OPTIONAL

EXTV

CONNECTION

5V < V

CC

< 7V

OUT2

Figure 6. Secondary Output Loop and EXTVCC Connection

EXTV

R4

FCB

R3

SGND

LTC1778

CC

TG

SW

BG

PGND

IN

1N4148

V

•

T1

1:N

+

•

+

OUT2

C

OUT2

1µF

V

OUT1

C

OUT

1778 F06

transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary load current, then V divider from V V

OUT2(MIN)

until V

OUT2

VV

OUT MIN2

OUT2

below which continuous operation is forced

has risen above its minimum.

()

08 1

will droop. An external resistor

OUT2

to the FCB pin sets a minimum voltage

⎛

.=+

⎜ ⎝

⎞

R

4

⎟

R

3

⎠

Fault Conditions: Current Limit and Foldback

The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC1778, the maximum sense voltage is controlled by the voltage on the V

pin. With valley current control,

RNG

the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is:

V

I

LIMIT

SNS MAX

=+∆

R

DS ON T

()

1

I

ρ

L

2

The current limit value should be checked to ensure that I

LIMIT(MIN)

> I

OUT(MAX)

. The minimum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power loss in the converter. Note that it is important to check for self-consistency between the assumed MOSFET junction temperature and the resulting value of I

which heats

LIMIT

the MOSFET switches.

Caution should be used when setting the current limit based upon the R

of the MOSFETs. The maximum

DS(ON)

current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and maximum values for R reasonable assumption is that the minimum R

, but not a minimum. A

DS(ON)

lies the same amount below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines.

To further limit current in the event of a short circuit to ground, the LTC1778 includes foldback current limiting. If the output falls by more than 25%, then the maximum sense voltage is progressively lowered to about one sixth of its full value.

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LTC1778/LTC1778-1

INTVCC Regulator

An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC1778. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 4.7µF low ESR tantalum capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. Applications using large MOSFETs with a high input voltage and high frequency of operation may cause the LTC1778 to exceed its maximum junction temperature rating or RMS current rating. Most of the supply current drives the MOSFET gates unless an external EXTVCC source is used. In continuous mode operation, this current is I + Q

). The junction temperature can be estimated

g(BOT)

GATECHG

= f(Q

g(TOP)

from the equations given in Note 2 of the Electrical Characteristics. For example, the LTC1778CGN is limited to less than 14mA from a 30V supply:

TJ = 70°C + (14mA)(30V)(130°C/W) = 125°C

For larger currents, consider using an external supply with the EXTVCC pin.

will start-up using the internal linear regulator until the boosted output supply is available.

External Gate Drive Buffers

The LTC1778 drivers are adequate for driving up to about 30nC into MOSFET switches with RMS currents of 50mA. Applications with larger MOSFET switches or operating at frequencies requiring greater RMS currents will benefit from using external gate drive buffers such as the LTC1693. Alternately, the external buffer circuit shown in Figure 7 can be used. Note that the bipolar devices reduce the signal swing at the MOSFET gate, and benefit from an increased EXTVCC voltage of about 6V.

BOOST

Q1 FMMT619

10Ω 10Ω

TG

SW

Figure 7. Optional External Gate Driver

Q2 FMMT720

GATE OF M1

BG

INTV

PGND

CC

Q3 FMMT619

Q4 FMMT720

GATE OF M2

1778 F07

EXTVCC Connection

The EXTVCC pin can be used to provide MOSFET gate drive and control power from the output or another external source during normal operation. Whenever the EXTV

CC

pin is above 4.7V the internal 5V regulator is shut off and an internal 50mA P-channel switch connects the EXTV

CC

pin to INTVCC. INTVCC power is supplied from EXTVCC until this pin drops below 4.5V. Do not apply more than 7V to the EXTVCC pin and ensure that EXTVCC ≤ VIN. The following list summarizes the possible connections for EXTVCC:

1. EXTVCC grounded. INTV

is always powered from the

CC

internal 5V regulator.

2. EXTVCC connected to an external supply. A high efficiency supply compatible with the MOSFET gate drive requirements (typically 5V) can improve overall efficiency.

3. EXTVCC connected to an output derived boost network. The low voltage output can be boosted using a charge pump or flyback winding to greater than 4.7V. The system

Soft-Start and Latchoff with the RUN/SS Pin

The RUN/SS pin provides a means to shut down the LTC1778 as well as a timer for soft-start and overcurrent latchoff. Pulling the RUN/SS pin below 0.8V puts the LTC1778 into a low quiescent current shutdown (IQ < 30µA). Releasing the pin allows an internal 1.2µA current source to charge up the external timing capacitor CSS. If RUN/SS has been pulled all the way to ground, there is a delay before starting of about:

V

15

t

DELAY SS SS

=

.

12

.

CsFC

=µ

13

./

A

µ

()

When the voltage on RUN/SS reaches 1.5V, the LTC1778 begins operating with a clamp on ITH of approximately

0.9V. As the RUN/SS voltage rises to 3V, the clamp on I

TH

is raised until its full 2.4V range is available. This takes an additional 1.3s/µF, during which the load current is folded back until the output reaches 75% of its final value. The pin can be driven from logic as shown in Figure 7. Diode D1

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APPLICATIO S I FOR ATIO

reduces the start delay while allowing CSS to charge up slowly for the soft-start function.

After the controller has been started and given adequate time to charge up the output capacitor, CSS is used as a short-circuit timer. After the RUN/SS pin charges above 4V, if the output voltage falls below 75% of its regulated value, then a short-circuit fault is assumed. A 1.8µA cur- rent then begins discharging C

. If the fault condition

SS

persists until the RUN/SS pin drops to 3.5V, then the controller turns off both power MOSFETs, shutting down the converter permanently. The RUN/SS pin must be actively pulled down to ground in order to restart operation.

The overcurrent protection timer requires that the soft-start timing capacitor C

be made large enough to guarantee

SS

that the output is in regulation by the time CSS has reached the 4V threshold. In general, this will depend upon the size of the output capacitance, output voltage and load current characteristic. A minimum soft-start capacitor can be estimated from:

CSS > C

OUT VOUT RSENSE

(10–4 [F/V s])

Generally 0.1µF is more than sufficient.

Overcurrent latchoff operation is not always needed or desired. Load current is already limited during a shortcircuit by the current foldback circuitry and latchoff operation can prove annoying during troubleshooting. The feature can be overridden by adding a pull-up current greater than 5µA to the RUN/SS pin. The additional current prevents the discharge of CSS during a fault and also shortens the soft-start period. Using a resistor to V

IN

as shown in Figure 8a is simple, but slightly increases shutdown current. Connecting a resistor to INTVCC as

INTV

CC

V

3.3V OR 5V RUN/SS

Figure 8. RUN/SS Pin Interfacing with Latchoff Defeated

IN

RSS*

D1

C

SS

(8a) (8b)

RSS*

RUN/SS

D2*

2N7002

*OPTIONAL TO OVERRIDE OVERCURRENT LATCHOFF

C

SS

1778 F08

shown in Figure 8b eliminates the additional shutdown current, but requires a diode to isolate CSS . Any pull-up network must be able to pull RUN/SS above the 4.2V maximum threshold of the latchoff circuit and overcome the 4µA maximum discharge current.

Efficiency Considerations

The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC1778 circuits:

1. DC I

2

R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows through L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same R

DS(ON)

, then the resistance of one MOSFET can simply be summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if R

= 0.01Ω and RL = 0.005Ω, the

DS(ON)

loss will range from 15mW to 1.5W as the output current varies from 1A to 10A.

2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from:

Transition Loss ≅ (1.7A–1) V

IN

2

I

OUT CRSS

f

3. INTVCC current. This is the sum of the MOSFET driver and control currents. This loss can be reduced by supplying INTVCC current through the EXTVCC pin from a high efficiency source, such as an output derived boost network or alternate supply if available.

4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries.

1778fb

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P

VV

V

AW

BOT

=

()()

Ω

()

=

28 2 5

28

12 15 0010 197

2

–.

.. .

APPLICATIO S I FOR ATIO

LTC1778/LTC1778-1

Other losses, including C conduction loss during dead time and inductor core loss generally account for less than 2% additional loss.

When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency. If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency.

Checking Transient Response

The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, V equal to ∆I resistance of C discharge C the regulator to return V During this recovery time, V overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 9 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Application Note 76.

LOAD

OUT

(ESR), where ESR is the effective series

. ∆I

OUT

generating a feedback error signal used by

ESR loss, Schottky diode D1

OUT

immediately shifts by an amount

also begins to charge or

LOAD

to its steady-state value.

OUT

can be monitored for

OUT

Selecting a standard value of 1.8µH results in a maximum ripple current of:

.

µ

Ω

⎛

1

–

⎜ ⎝

= 1.5:

1

+

2

25

.

∆ =

L

250 1 8

()

Next, choose the synchronous MOSFET switch. Choosing a Si4874 (R

θ

= 40°C/W) yields a nominal sense voltage of:

JA

V

SNS(NOM)

Tying V for a nominal value of 110mV with current limit occurring at 146mV. To check if the current limit is acceptable, assume a junction temperature of about 80°C above a 70°C ambient with ρ

I

LIMIT

and double check the assumed TJ in the MOSFET:

TJ = 70°C + (1.97W)(40°C/W) = 149°C

to 1.1V will set the current sense voltage range

RNG

≥

15 0010

()

V

kHz H

()

= 0.0083Ω (NOM) 0.010Ω (MAX),

DS(ON)

= (10A)(1.3)(0.0083Ω) = 108mV

150°C

mV

146

..

()

⎞

25

.

V

51

.

=I

⎟

28

V

⎠

AA

51 12

.

()

A

=

Design Example

As a design example, take a supply with the following specifications: VIN = 7V to 28V (15V nominal), V ±5%, I timing resistor with VON = V

and choose the inductor for about 40% ripple current at the maximum VIN:

OUT(MAX)

R

=

ON

L

250 0 4 10

()()()

= 10A, f = 250kHz. First, calculate the

:

OUT

V

25

.

V kHz pF

0 7 250 10

.

()( )()

V

25

.

kHz A

.

⎛

25

−

1

⎜

28

⎝

M

= Ω

142

.

⎞

V

.

=µ

23

⎟

V

⎠

= 2.5V

OUT

H=

.

Because the top MOSFET is on for such a short time, an Si4884 R 40°C/W will be sufficient. Checking its power dissipation at current limit with ρ

P

TJ = 70°C + (0.7W)(40°C/W) = 98°C

The junction temperatures will be significantly less at nominal current, but this analysis shows that careful attention to heat sinking will be necessary in this circuit.

DS(ON)(MAX)

.

25

=

TOP

V

28

.

1 7 28 12 100 250

()( )( )( )( )

...

=+=

030 040 07

= 0.0165Ω, C

= 1.4:

100°C

V

2

..

A

12 1 4 0 0165

()()

2

VApFkHz

WWW

()

= 100pF, θJA =

RSS

Ω

+

1778fb

17

LTC1778/LTC1778-1

WUUU

APPLICATIO S I FOR ATIO

CIN is chosen for an RMS current rating of about 5A at 85°C. The output capacitors are chosen for a low ESR of

0.013Ω to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage will be only:

∆V

OUT(RIPPLE)

= ∆I

L(MAX)

(ESR)

= (5.1A) (0.013Ω) = 66mV

However, a 0A to 10A load step will cause an output change of up to:

∆V

OUT(STEP)

= ∆I

(ESR) = (10A) (0.013Ω) = 130mV

LOAD

An optional 22µF ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 9.

PC Board Layout Checklist

When laying out a PC board follow one of the two suggested approaches. The simple PC board layout requires a dedicated ground plane layer. Also, for higher currents, it is recommended to use a multilayer board to help with heat sinking power components.

• The ground plane layer should not have any traces and

it should be as close as possible to the layer with power MOSFETs.

• Place CIN, C

, MOSFETs, D1 and inductor all in one

OUT

compact area. It may help to have some components on the bottom side of the board.

• Place LTC1778 chip with pins 9 to 16 facing the power

components. Keep the components connected to pins 1 to 8 close to LTC1778 (noise sensitive components).

•

Use an immediate via to connect the components to ground plane including SGND and PGND of LTC1778. Use several bigger vias for power components.

• Use compact plane for switch node (SW) to improve

cooling of the MOSFETs and to keep EMI down.

• Use planes for VIN and V

to maintain good voltage

OUT

filtering and to keep power losses low.

• Flood all unused areas on all layers with copper. Flood-

ing with copper will reduce the temperature rise of power component. You can connect the copper areas to any DC net (VIN, V

, GND or to any other DC rail in

OUT

your system).

C

SS

0.1µF

R3

R4

11k

39k

C

C1

R

C

500pF

20k

C

C2

100pF

R1

14.0k

C2

R2

6.8nF

30.1k

CIN: UNITED CHEMICON THCR60EIHI06ZT

: CORNELL DUBILIER ESRE181E04B

C

OUT1-2

L1: SUMIDA CEP125-1R8MC-H

R

PG

100k

R

1.4MΩ

1

RUN/SS

2

PGOOD

3

V

RNG

4

FCB

5

I

TH

6

SGND

7

I

ON

8

V

FB

ON



M1 Si4884

M2 Si4874

LTC1778

BOOST

PGND

INTV

EXTV

SW

V

TG

BG

D

B

16

15

14

13

12

11

CC

10

IN

9

CC

CMDSH-3

C

B

0.22µF

C

VCC

+

4.7µF

R

1Ω

C

F

0.1µF

F

Figure 9. Design Example: 2.5V/10A at 250kHz

L1

1.8µH

D1 B340A

+

1778 F09

C

IN

10µF 35V ×3

C

OUT1-2

180µF 4V ×2

C

OUT3

22µF

6.3V X7R

V

IN

5V TO 28V

V

OUT

2.5V 10A

1778fb

18

WUUU

APPLICATIO S I FOR ATIO

LTC1778/LTC1778-1

When laying out a printed circuit board, without a ground plane, use the following checklist to ensure proper operation of the controller. These items are also illustrated in Figure 10.

• Segregate the signal and power grounds. All small signal components should return to the SGND pin at one point which is then tied to the PGND pin close to the source of M2.

• Place M2 as close to the controller as possible, keeping the PGND, BG and SW traces short.

C

SS

C

C1

R

C

C2

R1

1

2

3

4

5

6

7

8

LTC1778

RUN/SS

PGOOD

V

RNG

FCB

I

TH

SGND

I

ON

V

FB

BOOST

SW

PGND

INTV

EXTV

16

15

TG

14

13

12

BG

11

CC

10

V

IN

9

CC

C

B

D

B

C

VCC

+

•

Connect the input capacitor(s) CIN close to the power MOSFETs. This capacitor carries the MOSFET AC current.

• Keep the high dV/dT SW, BOOST and TG nodes away from sensitive small-signal nodes.

• Connect the INTV to the INTV

CC

• Connect the top driver boost capacitor C

decoupling capacitor C

CC

and PGND pins.

B

closely

VCC

closely to the

BOOST and SW pins.

• Connect the V

pin decoupling capacitor CF closely to

IN

the VIN and PGND pins.

L

M1

D1

M2

C

F

R

F

C

OUT

C

IN

+

V

IN

–

V

OUT

+

R

R2

BOLD LINES INDICATE HIGH CURRENT PATHS

ON

1778 F10

Figure 10. LTC1778 Layout Diagram

1778fb

19

LTC1778/LTC1778-1

TYPICAL APPLICATIO S

U

1.5V/10A at 300kHz from 3.3V Input

C

SS

0.1µF

R

R1

R2

11k

39k

C

C1

R

C

680pF

20k

C

C2

100pF

R1

10k

R2

8.87k

C

: MURATA GRM42-2X5R226K6.3

IN1-2

: CORNELL DUBILIER ESRE271M02B

C

OUT

R

PG

100k

R

ON

576k

1

2

3

4

5

6

7

8

LTC1778

RUN/SS

PGOOD

V

RNG

FCB

I

TH

SGND

I

ON

V

FB

BOOST

PGND

INTV

EXTV

SW

V

TG

BG

D

B

16

15

14

13

12

11

CC

10

IN

9

CC

CMDSH-3

C

B

0.22µF

C

VCC

4.7µF



5V

M1 IRF7811A

L1, 0.68µH

M2 IRF7811A

D1 B320B

V

IN

C

IN1-2

+

22µF

6.3V ×2

C

OUT

+

270µF 2V ×2

1778 TA01

C

IN3

330µF

6.3V

3.3V

V

OUT

1.5V 10A

1.2V/6A at 300kHz

C

SS

0.1µF

R

PG

100k

C

C1

R

470pF

C

20k

C

C2

100pF

R1

20k

R

ON

R2

10k

2200pF

CIN: TAIYO YUDEN TMK432BJ106MM

: CORNELL DUBILIER ESRD181M02B

C

OUT1

: TAIYO YUDEN JMK316BJ106ML

C

OUT2

L1: TOKO 919AS-1R8N

510k

C2

1

2

3

4

5

6

7

8

LTC1778

RUN/SS

PGOOD

V

RNG

FCB

I

TH

SGND

I

ON

V

FB

BOOST

PGND

INTV

EXTV

SW

V

BG

D

B

16

15

TG

14

13

12

11

CC

10

IN

9

CC

CMDSH-3

C

B

0.22µF

C

VCC

4.7µF

R 1Ω

C

F

0.1µF

F



M1 1/2 FDS6982S

M2 1/2 FDS6982S

L1

1.8µH

1778 TA02

V

IN

C

OUT2

10µF

6.3V

5V TO 25V

V

OUT

1.2V 6A

1778fb

C

IN

10µF 25V ×2

+

C

OUT1

180µF 2V

20

TYPICAL APPLICATIO S

C

SS

C

C1

100pF

10k 1%

1nF

C

R1

140k

1

1%

R2

0.1µF

R

C

47k

C

220pF

C2

1

2

3

4

5

6

7

8

R

1.5M

RUN/SS

V

FCB

I

TH

SGND

I

ON

V

ON2

1%

LTC1778-1

ON

RNG

FB

R

1.5M 1%

U

Single Inductor, Positive Output Buck/Boost

D

B

CMDSH-3

C

B

0.22µF

C

VCC

4.7µF

C

F

0.1µF

PGND

1Ω

M1 IRF7811A

L1 4.8µH

M2 IRF7811A

R

F

C

: MARCON THER70EIH226ZT

IN

: AVX TPSV107M020R0085

C

OUT

L1: SCHOTT 36835-1

ON1

BOOST

PGND

INTV

EXTV

TG

SW

BG

V

16

15

14

13

12

11

CC

10

IN

9

CC

D1

B340A

LTC1778/LTC1778-1

V

I

IN

OUT

18V

6A

12V

5A

6V

3.3A V

IN

C

OUT

100µF 20V ×6

6V TO 18V

V

OUT

12V

C

IN

22µF 50V

IR 12CWQ03FN

×2

M3 Si4888

D2

+

1778 TA04

C

SS

0.1µF

C

C1

2.2nF

R1

10k

R2

140k

LTC1778-1

1

RUN/SS

2

V

3

V

4

R

C

20k

C

C2

100pF

R

1.6M

C2

2200pF

:

C

UNITED CHEMICON THCR70E1H226ZT (847) 696-2000

IN

:

C

SANYO 16SV220M (619) 661-6835

OUT

L1:

SUMIDA CDRH127-100 (847) 956-0667

M1, M2:

FAIRCHILD FDS6680A (408) 822-2126

D1:

DIODES, INC. B340A (805) 446-4800

FCB

5

I

TH

6

SGND

7

I

ON

8

V

ON

RNG

FB

BOOST

PGND

INTV

EXTV

SW

V

TG

BG

CC

IN

CC

12V/5A at 300kHz

D

B

16

15

14

13

12

11

10

9

CMDSH-3

C

B

0.22µF

C

VCC

+

4.7µF

R

1Ω

C

F

0.1µF

F



M1

L1 10µH

M2

V

IN

14V TO 28V

C

IN

22µF 50V

V

OUT

12V

C

OUT

220µF 16V

5A

1778fb

+

D1

1778 TA05

21

LTC1778/LTC1778-1

TYPICAL APPLICATIO S

C

SS

C

C1

4700pF

R1

10k

R2

52.3k

0.1µF

R

10k

C

100pF

C2

R

ON

698k

1

2

3

4

5

6

7

8

LTC1778-1

RUN/SS

V

ON

V

RNG

FCB

I

TH

SGND

I

ON

V

FB

U

Positive-to-Negative Converter, –5V/5A at 300kHz

D

B

CMDSH-3

C

B

0.22µF

C

VCC

4.7µF

RF

1Ω

C

F

0.1µF

M1 IRF7811A

L1 2.7µH

M2 IRF7822

D1 B340A

BOOST

PGND

INTV

EXTV

TG

SW

BG

V

16

15

14

13

12

11

CC

10

IN

9

CC

V

I

IN

OUT

20V

8A

10V

6.7A

5V

5A

V

IN

C

IN1

10µF 25V ×2

+

C

OUT

100µF 6V ×3

C

IN2

10µF 35V

5V TO 20V

V

OUT

–5V

C

: TAIYO YUDEN TMK432BJ106MM

IN1

: SANYO 35CV10GX

C

IN2

: PANASONIC EEFUD0J101R

C

OUT

L1: PANASONIC ETQPAF2R7H

1778 TA06

22

1778fb

PACKAGE DESCRIPTIO

LTC1778/LTC1778-1

U

GN Package

16-Lead Plastic SSOP (Narrow 0.150)

(LTC DWG # 05-08-1641)

.045 ±.005

.254 MIN

RECOMMENDED SOLDER PAD LAYOUT

.007 – .0098

(0.178 – 0.249)

.016 – .050

NOTE:

1. CONTROLLING DIMENSION: INCHES

2. DIMENSIONS ARE IN

3. DRAWING NOT TO SCALE *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH

SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD

FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

(0.406 – 1.270)

INCHES

(MILLIMETERS)

.150 – .165

.0250 BSC.0165 ± .0015

.015 ± .004

(0.38 ± 0.10)

0° – 8° TYP

× 45°

.229 – .244

(5.817 – 6.198)

.0532 – .0688

(1.35 – 1.75)

.008 – .012

(0.203 – 0.305)

TYP

16

15

12

.189 – .196*

(4.801 – 4.978)

14

12 11 10

13

5

4

3

678

.0250

(0.635)

BSC

.009

(0.229)

9

(0.102 – 0.249)

REF

.150 – .157**

(3.810 – 3.988)

.004 – .0098

GN16 (SSOP) 0204

Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

1778fb

23

LTC1778/LTC1778-1

TYPICAL APPLICATIO

U

Typical Application 2.5V/3A at 1.4MHz

C

SS

0.1µF

C

C1

R

470pF

R1

11.5k

C

33k

C

C2

100pF

R

R2

24.9k

2200pF

CIN: TAIYO YUDEN TMK432BJ106MM

: CORNELL DUBILIER ESRD121M04B

C

OUT

L1: TOKO A921CY-1R0M

220k

C2

R

100k

ON

D

B

LTC1778

1

RUN/SS

PG

2

PGOOD

3

V

RNG

4

FCB

5

I

TH

6

SGND

7

I

ON

8

EXTV

V

FB

BOOST

SW

PGND

INTV

16

15

TG

14

13

12

BG

11

CC

10

V

IN

9

CC

CMDSH-3

C

B

0.22µF

C

VCC

4.7µF

RF

1Ω

C

F

0.1µF



M1 1/2 Si9802

M2 1/2 Si9802

L1, 1µH

V

IN

9V TO 18V

C

IN

10µF 25V

V

OUT

2.5V

C

OUT

120µF 4V

1778 TA03

3A

+

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3.5V ≤ V

0.8V ≤ V

≤ 36V

IN

≤ 2V

OUT

≤ 36V

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≤ 2V; 3.5V ≤ VIN ≤ 36V

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1778fb

Datasheet LTC1778, LTC1778-1 Datasheet (LINEAR TECHNOLOGY)

Specifications and Main Features

Frequently Asked Questions

User Manual