Datasheet LTC1772 Datasheet (Linear Technology)

FEATURES

Final Electrical Specifications

■

High Efficiency: Up to 94%

■

High Output Currents Easily Achieved

■

Wide VIN Range: 2.5V to 9.8V

■

Constant Frequency 550kHz Operation

■

Burst Mode

■

Low Dropout: 100% Duty Cycle

■

0.8V Reference Allows Low Output Voltages

■

Current Mode Operation for Excellent Line and Load

TM

Operation at Light Load

Transient Response

■

Low Quiescent Current: 270µA

■

Shutdown Mode Draws Only 8µA Supply Current

■

±2.5% Reference Accuracy

■

Tiny 6-Lead SOT-23 Package

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APPLICATIONS

■

One or Two Lithium-Ion-Powered Applications

■

Cellular Telephones

■

Wireless Modems

■

Portable Computers

■

Distributed 3.3V, 2.5V or 1.8V Power Systems

■

Scanners

LTC1772

Constant Frequency

Current Mode Step-Down

DC/DC Controller in SOT-23

September 1999

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DESCRIPTION

The LTC®1772 is a constant frequency current mode step­down DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltage lockout feature that shuts down the LTC1772
when the input voltage falls below 2.0V.

The LTC1772 boasts a ±2.5% output voltage accuracy and
consumes only 270µA of quiescent current. For applica-
tions where efficiency is a prime consideration, the LTC1772
is configured for Burst Mode operation, which enhances
efficiency at low output current.

To further maximize the life of a battery source, the
external P-channel MOSFET is turned on continuously in
dropout (100% duty cycle). In shutdown, the device draws
a mere 8µA. High constant operating frequency of 550kHz
allows the use of a small external inductor.

The LTC1772 is available in a small footprint 6-lead
SOT-23.

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

TYPICAL APPLICATION

1

ITH/RUN

10k

220pF

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: IR10BQ015

LTC1772

2

GND

3

V

FB

Figure 1. High Efficiency Step-Down Converter

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V

IN

169k

78.7k

2.5V
TO 9.8V

V

OUT

2.5V
2A

1772 F01a

R1

0.03Ω

6

PGATE

5

V

IN

4

–

SENSE

L1: MURATA LQN6C-4R7
M1: Si3443DV
R1: DALE 0.25W

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

M1

L1

4.7µH

D1

C1
10µF
16V

+

C2
47µF
6V

100

90

80

70

EFFICIENCY (%)

60

50

40

Efficiency vs Load Current

VIN = 3.3V

1 100 1000 10000

VIN = 4.2V

VIN = 9.8V

10

LOAD CURRENT (mA)

VIN = 8.4V

VIN = 6V

V

OUT

R

SENSE

= 2.5V

= 0.03Ω

1772 F01b

1

LTC1772

ITH/RUN 1

GND 2

V

FB

3

6 PGATE
5 V

IN

4 SENSE

–

TOP VIEW

S6 PACKAGE

6-LEAD PLASTIC SOT-23

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ABSOLUTE MAXIMUM RATINGS

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PACKAGE/ORDER INFORMATION

(Note 1)

Input Supply Voltage (VIN).........................– 0.3V to 10V

SENSE–, PGATE Voltages.............–0.3V to (VIN + 0.3V)

VFB, ITH/RUN Voltages ..............................–0.3V to 2.4V

PGATE Peak Output Current (<10µs) ....................... 1A

ORDER PART

NUMBER

LTC1772CS6

Storage Ambient Temperature Range ... – 65°C to 150°C

Operating Temperature Range (Note 2) ....... 0°C to 70°C

S6 PART MARKING

Junction Temperature (Note 3)............................. 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

Consult factory for Industrial and Military grade parts.

ELECTRICAL CHARACTERISTICS

range, otherwise specifications are at TA = 25°C. VIN = 4.2V unless otherwise specified. (Note 2)

PARAMETER CONDITIONS MIN TYP MAX UNITS

Input DC Supply Current Typicals at VIN = 4.2V (Note 4)
Normal Operation 2.4V ≤ V
Sleep Mode 2.4V ≤ V
Shutdown 2.4V ≤ V
UVLO V

Undervoltage Lockout Threshold VIN Falling ● 1.6 2.0 2.3 V

V

IN

Shutdown Threshold (at ITH/RUN) ● 0.2 0.35 0.5 V
Start-Up Current Source V
Regulated Feedback Voltage (Note 5) ● 0.780 0.800 0.820 V
Output Voltage Line Regulation 2.4V ≤ VIN ≤ 9.8V (Note 5) 0.05 mV/V
Output Voltage Load Regulation ITH/RUN Sinking 5µA (Note 5) 2.5 mV/µA

VFB Input Current (Note 5) 10 50 nA
Overvoltage Protect Threshold Measured at V
Overvoltage Protect Hysteresis 20 mV
Oscillator Frequency VFB = 0.8V 500 550 650 kHz

Gate Drive Rise Time C
Gate Drive Fall Time C
Maximum Current Sense Voltage 120 mV

I

TH

V

ITH

FB

LOAD

LOAD

The ● denotes specifications that apply over the full operating temperature

≤ 9.8V 270 420 µA

IN

≤ 9.8V 230 370 µA

IN

≤ 9.8V, V

IN

< UVLO Threshold 6 10 µA

IN

Rising 1.85 2.3 2.5 V

/RUN = 0V 0.25 0.5 0.85 µA

/RUN Sourcing 5µA (Note 5) 2.5 mV/µA

= 0V 120 kHz

= 3000pF 40 ns
= 3000pF 40 ns

ITH

FB

T

= 150°C, θJA = 230°C/W

JMAX

/RUN = 0V 8 22 µA

0.820 0.860 0.895 V

LTIL

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Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Note 2: The LTC1772 is guaranteed to meet specified performance over
the 0°C to 70°C operating temperature range.

Note 3: T
dissipation P

is calculated from the ambient temperature TA and power

J

according to the following formula:

D

TJ = TA + (PD • θJ°C/W)

2

Note 4: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.

Note 5: The LTC1772 is tested in a feedback loop that servos V
output of the error amplifier.

to the

FB

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TYPICAL PERFORMANCE CHARACTERISTICS

LTC1772

Reference Voltage
vs Temperature

825

VIN = 4.2V

820
815
810
805
800
795

VOLTAGE (mV)

FB

V

790
785
780
775

–35 5

–15

–55

85

45 125

25

TEMPERATURE (°C)

105

65

1772 G01

Maximum (VIN – SENSE–) Voltage
vs Duty Cycle

130

120

110

100

90

80

TRIP VOLTAGE (mV)

70

60

50

20 30

40 50

60 70

DUTY CYCLE (%)

Normalized Oscillator Frequency
vs Temperature

10

VIN = 4.2V

8
6
4
2

0
–2
–4
–6

NORMALIZED FREQUENCY (%)

–8

–10

–35 5

–55

VIN = 4.2V
T

= 25°C

A

80 90

–15

TEMPERATURE (°C)

100

1772 G04

45 125

65

25

85

105

1772 G02

Shutdown Threshold
vs Temperature

600

VIN = 4.2V

560
520
480
440
400
360

/RUN VOLTAGE (mV)

320

TH

I

280
240
200

–35 5

–15

–55

TEMPERATURE (

Undervoltage Lockout Trip
Voltage vs Temperature

2.24
VIN = 4.2V

2.20

FALLING

V

IN

2.16

2.12

2.08

2.04

2.00

TRIP VOLTAGE (V)

1.96

1.92

1.88

1.84

–35 5

–15

–55

45 125

65

25

°C)

25

TEMPERATURE (°C)

85

105

1772 G05

85

45 125

105

65

1772 G03

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PIN FUNCTIONS

ITH/RUN (Pin 1): This pin performs two functions. It
serves as the error amplifier compensation point as well as
the run control input. The current comparator threshold
increases with this control voltage. Nominal voltage range
for this pin is 0.7V to 1.9V. Forcing this pin below 0.35V
causes the device to be shut down. In shutdown all
functions are disabled and the PGATE pin is held high.

GND (Pin 2): Ground Pin.

VFB (Pin 3): Receives the feedback voltage from an exter-

nal resistive divider across the output.
SENSE– (Pin 4): The Negative Input to the Current Com-

parator.
VIN (Pin 5): Supply Pin. Must be closely decoupled to GND

Pin 2.
PGATE (Pin 6): Gate Drive for the External P-Channel

MOSFET. This pin swings from 0V to VIN.

3

LTC1772

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FUNCTIONAL DIAGRA

–

SENSE

V

IN

5

4

4

+

ICMP

–

OSC

FREQ

FOLDBACK

SHORT-CIRCUIT

DETECT

V

IN

+

0.3V

GND

2

–

SLOPE
COMP

–

+

0.5µA

V

IN

VOLTAGE

REFERENCE

UNDERVOLTAGE

LOCKOUT

0.3V

0.15V

V

REF

0.8V

0.35V

V

RS1
R

Q

S

+

–

I

/RUN

1

TH

+

–

BURST

CMP

SHDN

CMP

SWITCHING
LOGIC AND

BLANKING

CIRCUIT

SLEEP

SHDN

UV

OVP

EAMP

IN

PGATE

6

+

V

REF

+

–

60mV

V

REF

+

0.8V

–

1.2V

V

FB

3

V

IN

1772FD

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OPERATIO

Main Control Loop

The LTC1772 is a constant frequency current mode switch­ing regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (ICMP) resets the latch. The peak
inductor current at which ICMP resets the RS latch is
controlled by the voltage on the ITH/RUN pin, which is the
output of the error amplifier EAMP. An external resistive
divider connected between V
EAMP to receive an output feedback voltage VFB. When the

(Refer to Functional Diagram)

and ground allows the

OUT

4

load current increases, it causes a slight decrease in V

FB

relative to the 0.8V reference, which in turn causes the
ITH/RUN voltage to increase until the average inductor
current matches the new load current.

The main control loop is shut down by pulling the ITH/RUN
pin low. Releasing ITH/RUN allows an internal 0.5µA
current source to charge up the external compensation
network. When the ITH/RUN pin reaches 0.35V, the main
control loop is enabled with the ITH/RUN voltage then
pulled up to its zero current level of approximately 0.7V.
As the external compensation network continues to charge

OPERATIO

LTC1772

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(Refer to Functional Diagram)

up, the corre

sponding output current trip level follows,

allowing normal operation.
Comparator OVP guards against transient overshoots

>7.5% by turning off the external P-channel power
MOSFET and keeping it off until the fault is removed.

Burst Mode Operation

The LTC1772 enters Burst Mode operation at low load
currents. In this mode, the peak current of the inductor is
set as if V

/RUN = 1V (at low duty cycles) even though

ITH

the voltage at the ITH/RUN pin is at a lower value. If the
inductor’s average current is greater than the load require­ment, the voltage at the ITH/RUN pin will drop. When the
ITH/RUN voltage goes below 0.85V, the sleep signal goes
high, turning off the external MOSFET. The sleep signal
goes low when the ITH/RUN voltage goes above 0.925V
and the LTC1772 resumes normal operation. The next
oscillator cycle will turn the external MOSFET on and the
switching cycle repeats.

Dropout Operation

Short-Circuit Protection

When the output is shorted to ground, the frequency of the
oscillator will be reduced to about 120kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s fre­quency will gradually increase to its designed rate when
the feedback voltage again approaches 0.8V.

Overvoltage Protection

As a further protection, the overvoltage comparator in the
LTC1772 will turn the external MOSFET off when the
feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 20mV.

Slope Compensation and Inductor’s Peak Current

The inductor’s peak current is determined by:

V

I

PK

=

ITH

R

10

SENSE

()

When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the external P-channel MOSFET will remain on for more
than one oscillator cycle since the inductor current has not
ramped up to the threshold set by EAMP. Further reduc­tion in input supply voltage will eventually cause the
P-channel MOSFET to be turned on 100%, i.e., DC. The
output voltage will then be determined by the input voltage
minus the voltage drop across the MOSFET, the sense
resistor and the inductor.

Undervoltage Lockout

To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo­rated into the LTC1772. When the input supply voltage
drops below approximately 2.0V, the P-channel MOSFET
and all circuitry is turned off except the undervoltage
block, which draws only several microamperes.

when the LTC1772 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com­pensation begins and effectively reduces the peak induc­tor current. The amount of reduction is given by the curves
in Figure 2.

110
100

90
80

(%)

70
60

OUT(MAX)

/I

50

OUT

SF = I

Figure 2. Maximum Output Current vs Duty Cycle

I

= 0.4I

RIPPLE

AT 5% DUTY CYCLE

40

I

= 0.2I

30
20
10

RIPPLE

AT 5% DUTY CYCLE

VIN = 4.2V

0 70 80 90 1006010 20 30 40 50

DUTY CYCLE (%)

PK

PK

1772 F02

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LTC1772

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APPLICATIONS INFORMATION

The basic LTC1772 application circuit is shown in Figure

1. External component selection is driven by the load
requirement and begins with the selection of L1 and
R
diode D1 is selected followed by CIN (= C1)and C

R

R
With the current comparator monitoring the voltage devel­oped across R
determines the inductor’s peak current. The output cur­rent the LTC1772 can provide is given by:

where I
(see Inductor Value Calculation section).

A reasonable starting point for setting ripple current is
I

RIPPLE

becomes:

However, for operation that is above 40% duty cycle, slope
compensation effect has to be taken into consideration to
select the appropriate value to provide the required amount
of current. Using Figure 2, the value of R

Inductor Value Calculation

The operating frequency and inductor selection are inter­related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.

(= R1). Next, the power MOSFET and the output

SENSE

OUT

Selection for Output Current

SENSE

is chosen based on the required output current.

SENSE

, the threshold of the comparator

SENSE

I

OUT

012.

=−

R

SENSE

is the inductor peak-to-peak ripple current

RIPPLE

= (0.4)(I

R

R

SENSE

SENSE

=

()( )

=

()( )( )

I

RIPPLE

). Rearranging the above equation, it

OUT

1

for Duty Cycle < 40%

I

12

OUT

SENSE

SF

12 100

I

OUT

(= C2).

is:

The inductance value also has a direct effect on ripple
current. The ripple current, I

, decreases with higher

RIPPLE

inductance or frequency and increases with higher VIN or
V

. The inductor’s peak-to-peak ripple current is given

OUT

by:

I

RIPPLE

VVfLVV

−

IN OUT OUT D

=

()




VV



IN D



+



+



where f is the operating frequency. Accepting larger values
of I

allows the use of low inductances, but results in

RIPPLE

higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
I

RIPPLE

= 0.4(I

OUT(MAX)

). Remember, the maximum I

RIPPLE

occurs at the maximum input voltage.
In Burst Mode operation on the LTC1772, the ripple

current is normally set such that the inductor current is
continuous during the burst periods. Therefore, the peak­to-peak ripple current must not exceed:

I

RIPPLE

≤

0 0288.

R

SENSE

This implies a minimum inductance of:

VV

−

L

MIN

(Use V

IN OUT

=



f





IN(MAX)

0 0288.
R

SENSE






= VIN)

A smaller value than L



VV

OUT D



VV



IN D

could be used in the circuit;

MIN



+



+



however, the inductor current will not be continuous
during burst periods.

Inductor Core Selection

Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy

6

LTC1772

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APPLICATIONS INFORMATION

or Kool Mu® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase. Ferrite designs have very low core losses and are
preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core material saturates “hard,” which means that
inductance collapses abruptly when the peak design cur­rent is exceeded. This results in an abrupt increase in
inductor ripple current and consequent output voltage
ripple. Do not allow the core to saturate!

Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manu­facturer is Kool Mu. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new designs for surface mount that do
not increase the height significantly are available.

Power MOSFET Selection

An external P-channel power MOSFET must be selected
for use with the LTC1772. The main selection criteria for
the power MOSFET are the threshold voltage V
the “on” resistance RDS(ON), reverse transfer capaci­tance C

and total gate charge.

RSS

Since the LTC1772 is designed for operation down to low
input voltages, a sublogic level threshold MOSFET (R
guaranteed at VGS = 2.5V) is required for applications that
work close to this voltage. When these MOSFETs are used,
make sure that the input supply to the LTC1772 is less than
the absolute maximum VGS rating, typically 8V.

The required minimum R

of the MOSFET is gov-

DS(ON)

erned by its allowable power dissipation. For applications
that may operate the LTC1772 in dropout, i.e., 100% duty
cycle, at its worst case the required R

P

R

DS ON

()

DC

100

=

%=

Ip

()

OUT MAX

P

2

()

()

+

1 δ

DS(ON)

is given by:

GS(TH)

DS(ON)

and

where PP is the allowable power dissipation and δp is the
temperature dependency of R
given for a MOSFET in the form of a normalized R

. (1 + δp) is generally

DS(ON)

DS(ON)

vs

temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.

In applications where the maximum duty cycle is less than
100% and the LTC1772 is in continuous mode, the R

DS(ON)

is governed by:

P

R

DS ON

≅

()

DC I

()

P

2

+

1 δ

p

()

OUT

where DC is the maximum operating duty cycle of the
LTC1772.

Output Diode Selection

The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches V

OUT

the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition the diode must
safely handle I

at close to 100% duty cycle. Therefore,

PEAK

it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.

Under normal load conditions, the average current con­ducted by the diode is:

I

=

D



VV

−

IN OUT



VV



IN D



I



+

OUT



The allowable forward voltage drop in the diode is calcu­lated from the maximum short-circuit current as:

P

≈

I

D

SC MAX

()

V

F

where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.

Kool Mu is a registered trademark of Magnetics, Inc.

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LTC1772

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APPLICATIONS INFORMATION

A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.

CIN and C

In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (V
(VIN + VD). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:

CI

Required I

IN MAX

This formula has a maximum at VIN = 2V
= I

/2. This simple worst-case condition is commonly

OUT

used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1772, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.

The selection of C
series resistance (ESR). Typically, once the ESR require­ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆V

Selection

OUT

+ VD)/

OUT

VVV

[]

≈

RMS

is driven by the required effective

OUT

) is approximated by:

OUT

OUT IN OUT

−

()

V

IN

, where I

OUT

12/

RMS

Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through­hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for C

has been

OUT

met, the RMS current rating generally far exceeds the
I

RIPPLE(P-P)

requirement.

In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec­trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON,
Nichicon PL series and Panasonic SP.

Low Supply Operation

Although the LTC1772 can function down to approxi­mately 2.0V, the maximum allowable output current is
reduced when VIN decreases below 3V. Figure 3 shows the
amount of change as the supply is reduced down to 2V.
Also shown in Figure 3 is the effect of VIN on V

REF

as V

IN

goes below 2.3V.

105

V

100

95

90

REF

V

ITH

∆V I ESR

≈+

OUT RIPPLE





4

where f is the operating frequency, C



capacitance and I

is the ripple current in the induc-

RIPPLE

fC

1

OUT






is the output

OUT

tor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.

8

85

NORMALIZED VOLTAGE (%)

80

75

2.0

2.2 2.4 2.6 2.8
INPUT VOLTAGE (V)

Figure 3. Line Regulation of V

REF

3.0

1772 F03

and V

ITH

LTC1772

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APPLICATIONS INFORMATION

Setting Output Voltage

The LTC1772 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the overall output voltage. The
regulated output voltage is determined by:

R

2

V

OUT



=+

081

.





For most applications, an 80k resistor is suggested for R1.
To prevent stray pickup, a 100pF capacitor is suggested
across R1 located close to LTC1772.

LTC1772






R

1

V

OUT

100pF

R2

R1

1772 F04

3

V

FB

1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.

2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge dQ moves from VIN to ground.
The resulting dQ/dt is a current out of VIN which is
typically much larger than the DC supply current. In
continuous mode, I

GATECHG

= f(Qp).

3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but
is “chopped” between the P-channel MOSFET in series
with R
plus R
with the resistances of L and R

and the output diode. The MOSFET R

SENSE

multiplied by duty cycle can be summed

SENSE

to obtain I2R

SENSE

DS(ON)

losses.

Figure 4. Setting Output Voltage

Efficiency Considerations

The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percent­age of input power.

Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1772 circuits: 1) LTC1772 DC bias current,

2) MOSFET gate charge current, 3) I2R losses and 4)
voltage drop of the output diode.

4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage times the diode duty cycle multiplied by the
load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of

0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.

5. Transition losses apply to the external MOSFET and
increase at higher operating frequencies and input
voltages. Transition losses can be estimated from:

Transition Loss = 2(VIN)2I

O(MAX)CRSS

Other losses including CIN and C

(f)

ESR dissipative

OUT

losses, and inductor core losses, generally account for
less than 2% total additional loss.

9

LTC1772

U

WUU

APPLICATIONS INFORMATION

Foldback Current Limiting

As described in the Output Diode Selection, the worst-case
dissipation occurs with a short-circuited output when the
diode conducts the current limit value almost continu­ously. To prevent excessive heating in the diode, foldback
current limiting can be added to reduce the current in
proportion to the severity of the fault.

Foldback current limiting is implemented by adding diodes
D

and D

FB1

shown in Figure 5. In a hard short (V
will be reduced to approximately 50% of the maximum
output current.

between the output and the ITH/RUN pin as

FB2

= 0V), the current

OUT

V

D

FB1

D

FB2

1772 F05

OUT

LTC1772

V

ITH/RUN

Figure 5. Foldback Current Limiting

R2

FB

+

R1

In the application, a 0.03Ω resistor is used. For the
inductor, the required value is:

L

MIN

42 25

..

kHz

−

0 0288

.






003

.






=

550

+

25 03

..






+

42 03

..



=

200

. µ





H

In the application, a 5.6µH inductor is used to reduce ripple
current.

For the selection of the external MOSFET, the R

DS(ON)

must be guaranteed at 2.5V since the LTC1772 has to work
down to 2.7V. Let’s assume that the MOSFET dissipation
is to be limited to PP = 250mW and its thermal resistance
is 50°C/W. Hence the junction temperature at TA = 25°C
will be 37.5°C and δp = 0.005 (37.5 – 25) = 0.0625. The
required R

R

DS ON

()

is then given by:

DS(ON)

P

P

DC I p

2

()

OUT

011δΩ

.≅

=

+

1

()

The P-channel MOSFET requirement can be met by an
Si6433DQ.

Design Example

Assume the LTC1772 is used in a single Lithium-Ion
battery-powered cellular phone application. The VIN will be
operating from a maximum of 4.2V down to a minimum of

2.7V. Load current requirement is a maximum of 1.5A but
most of the time it will be on standby mode, requiring only
2mA. Efficiency at both low and high load current is
important. Output voltage is 2.5V.

VV

+

Maximum

Duty Cycle =

OUT D

VV

IN MIN D

+

()

%93

=

From Figure 2, SF = 57%.

R

SENSE

=

SF

12 100

I

()( )( )=()( )

OUT

057

.

12 1 5

.

=

0 0317

. Ω

The requirement for the Schottky diode is the most strin­gent when V
R

resistor, the short-circuit current through the

SENSE

= 0V, i.e., short circuit. With a 0.03Ω

OUT

Schottky is 0.1/0.03 = 3.3A. An MBRS340T3 Schottky
diode is chosen. With 3.3A flowing through, the diode is
rated with a forward voltage of 0.4V. Therefore, the worst­case power dissipated by the diode is 1.32W. The addition
of D

FB1

and D

(Figure 5) will reduce the diode dissipa-

FB2

tion to approximately 0.66W.
The input capacitor requires an RMS current rating of at

least 0.75A at temperature, and C

will require an ESR

OUT

of 0.1Ω for optimum efficiency.

10

LTC1772

U

WUU

APPLICATIONS INFORMATION

PC Board Layout Checklist

When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1772. These items are illustrated graphically in the
layout diagram in Figure 6. Check the following in your
layout:

1. Is the Schottky diode closely connected between ground
(Pin 2) and drain of the external MOSFET?

2. Does the (+) plate of CIN connect to the sense resistor
as closely as possible? This capacitor provides AC
current to the MOSFET.

3. Is the input decoupling capacitor (0.1µF) connected
closely between VIN (Pin 5) and ground (Pin 2)?

4. Connect the end of R

as close to VIN (Pin 5) as

SENSE

possible. The VIN pin is the SENSE+ of the current
comparator.

5. Is the trace from SENSE– (Pin 4) to the Sense resistor
kept short? Does the trace connect close to R

SENSE

?

6. Keep the switching node PGATE away from sensitive
small signal nodes.

7. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of C

and signal

OUT

ground. The 100pF capacitor should be as close as
possible to the LTC1772.

V

IN

V

OUT

R2

1

ITH/RUN

LTC1772

R

ITH

C

ITH

2

GND

3

V

FB

C1

BOLD LINES INDICATE HIGH CURRENT PATHS

PGATE

SENSE

6

R

5

V

IN

4

–

S

0.1µF

M1

+

C

IN

L1

5W

+

D1

R1

1772 F06

C

OUT

Figure 6. LTC1772 Layout Diagram (See PC Board Layout Checklist)

11

LTC1772

TYPICAL APPLICATIONS

LTC1772 High Efficiency, High Output Current 2.5V/2A Regulator

R4
10k

C3
220pF

U

1

ITH/RUN

2

GND

3

V

LTC1772

FB

PGATE

SENSE

V

IN

R2
169k

2.5V
TO 9.8V

V

OUT

2.5V
2A

R1

0.03Ω

6

5

V

IN

4

–

M1

D1

L1

4.7µH

C1
10µF
10V

+

C2
47µF
6V

C1: TAIYO YUDEN CERAMIC

EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: IR10BQ015

L1: MURATA LQN6C-4R7
M1: Si3443DV
R1: DALE 0.25W

C4
100pF

LTC1772 High Efficiency, Small Footprint 1.8V/0.5A Regulator

C1
10µF
10V

+

C2
47µF
6V

R4

10k

C3
220pF

1

ITH/RUN

LTC1772

2

GND

3

V

FB

C1: TAIYO YUDEN CERAMIC

EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: IR10BQ015

PGATE

SENSE

6

5

V

IN

4

–

R1

0.03Ω

L1

M1

4.7µH

D1

L1: MURATA LQN6C-4R7
M1: Si3443DV
R1: DALE 0.25W

R3

78.7k

R2
100k

R3
80k

1772 TA01

V

IN

2.5V
TO 9.8V

V

OUT

1.8V

0.5A

1772 TA02

12

U

TYPICAL APPLICATIONS

V

IN

3.3V

1

R4
10k

C3
220pF

2
3

LTC1772 3.3V to 5V/1A Boost Regulator

R1

0.033Ω

ITH/RUN

LTC1772
GND
V

FB

C1
47µF
16V
×2

PGATE

SENSE

U1

6

5

V

IN

4

–

5

42

3

L1

4.7µH

M1

LTC1772

D1

C2

+

100µF
10V
×2

R2
420k

V

OUT

5V
1A

C1: AVXTPSE476M016R0047
C2: AVXTPSE107M010R0100
D1: MOTOROLA M516

L1: MURATA LQN6C-4R7
M1: Si9804DY
R1: DALE 0.25W

U1: FAIRCHILD NC7SZ04

R3
80k

1772 TA03

13

LTC1772

TYPICAL APPLICATIONS

R4
10k

C3
220pF

U

LTC1772 5V/500mA Flyback Regulator

R1

0.033Ω

1

2
3

ITH/RUN

LTC1772
GND
V

FB

PGATE

SENSE

R5

22Ω

V

IN

–

6

5
4

C4
100pF
CERAMIC

M1

T1

•

100Ω

10µH10µH

•

R6

150pF

CERAMIC

D1

V

IN

2.5V

C1

TO 9.8V

47µF
16V
×2

C5

V

OUT

C2

+

100µF
10V
×2

5V
500mA

R2

52.3k

C1: AVXTPSE476M016R0047
C2: AVXTPSE107M010R0100
D1: MOTOROLA M516

M1: Si9803
R1: DALE 0.25W
T1: COILTRONICS CTX10-4

C6
100pF

R3
10k

1772 TA04

14

PACKAGE DESCRIPTION

LTC1772

U

Dimensions in inches (millimeters) unless otherwise noted.

S6 Package

6-Lead Plastic SOT-23

(LTC DWG # 05-08-16XX)

2.80 – 3.00

(0.110 – 0.118)

(NOTE 3)

1.90

2.6 – 3.0

(0.110 – 0.118)

1.50 – 1.75

(0.059 – 0.069)

0.10 – 0.60

(0.004 – 0.024)

REF

NOTE:

1. DIMENSIONS ARE IN MILLIMETERS

2. DIMENSIONS ARE INCLUSIVE OF PLATING

3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR

4. MOLD FLASH SHALL NOT EXCEED 0.254mm

5. PACKAGE EIAJ REFERENCE IS SC-74 (EIAJ)

0.09 – 0.20

(0.004 – 0.008)

(NOTE 2)

(0.074)

REF

0.00 – 0.15

(0.00 – 0.006)

0.35 – 0.50

(0.014 – 0.020)

SIX PLACES (NOTE 2)

0.95

(0.037)

REF

0.90 – 1.45

(0.035 – 0.057)

0.90 – 1.30

(0.035 – 0.051)

S6 SOT-23 0898

15

LTC1772

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2.5V to 6V

Up to 4.5A,

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16

Linear T echnolog y Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com

1772is, sn1772 LT/TP 0999 4K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1999