Page 1
Final Electrical Specifications
FEATURES
■
Tiny 5-Lead SOT-23 Package
■
Uses Tiny Capacitors and Inductor
■
High Frequency Operation: 1MHz
■
High Output Current: 500mA
■
Low R
■
High Efficiency: Up to 94%
■
Current Mode Operation for Excellent Line
Internal Switch: 0.28Ω
DS(ON)
and Load Transient Response
■
Short-Circuit Protected
■
Low Quiescent Current: 135µA
■
Low Dropout Operation: 100% Duty Cycle
■
Ultralow Shutdown Current: IQ < 1µA
■
Peak Inductor Current Independent of Inductor Value
■
Output Voltages from 5V Down to 1.25V
U
APPLICATIO S
LTC1701
1MHz Step-Down
DC/DC Converter in SOT-23
December 1999
U
DESCRIPTIO
The LTC®1701 is the industry’s first 5-lead SOT-23 step
down, current mode, DC/DC converter. Intended for small
to medium power applications, it operates from 2.5V to
5.5V input voltage range and switches at 1MHz, allowing
the use of tiny, low cost capacitors and inductors 2mm or
less in height. The output voltage is adjustable from 1.25V
to 5V. A built-in 0.28Ω switch allows up to 0.5A of output
current at high efficiency. OPTI-LOOPTM compensation
allows the transient response to be optimized over a wide
range of loads and output capacitors.
The LTC1701 incorporates automatic power saving Burst
ModeTM operation to reduce gate charge losses when the
load current drops below the level required for continuous
operation. With no load, the converter draws only 135µA.
In shutdown, it draws less than 1µA, making it ideal for
current sensitive applications.
■
PDAs/Palmtop PCs
■
Digital Cameras
■
Cellular Phones
■
Portable Media Players
■
PC Cards
■
Handheld Equipment
TYPICAL APPLICATIO
V
IN
2.5V TO
5.5V
+
C1
10µF
C1: TAIYO YUDEN JMK316BJ106ML
C2: SANYO POSCAP 6TPA47M
D1: MBRM120L
L1: SUMIDA CD43-4R7
Figure 1. Step-Down 2.5V/500mA Regulator
R4
1M
R3
5.1k
C3
330pF
V
IN
LTC1701
ITH/RUN
GND
SW
V
FB
U
4.7µH
D1
In dropout, the internal P-channel MOSFET switch is
turned on continuously, thereby maximizing battery life.
Its small size and switching frequency enables the complete DC/DC converter function to consume less than 0.3
square inches of PC board area.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode and OPTI-LOOP are trademarks of Linear Technology Corporation.
L1
R2
121k
R1
121k
V
OUT
(2.5V/
500mA)
+
C2
47µF
1701 F01
100
95
90
85
EFFICIENCY (%)
80
75
70
1 10 100 1000
Efficiency Curve
VIN = 3.3V
LOAD CURRENT (mA)
1701 F01a
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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LTC1701
WW
W
ABSOLUTE AXI U RATI GS
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PACKAGE/ORDER I FOR A TIO
(Note 1)
(Voltages Referred to GND Pin)
VIN Voltage (Pin 5).......................................–0.3V to 6V
ITH/RUN Voltage (Pin 4) ..............................–0.3V to 3V
VFB Voltage (Pin 3) ......................................–0.3V to 3V
Peak Switch Current (Pin 1) ................................... 1.3A
VIN – SW (Max Switch Voltage)................8.5V to –0.3V
Operating Temperature Range (Note 2).. –40°C to 85°C
Junction Temperature (Note 5).............................125°C
TOP VIEW
SW 1
GND 2
3
V
FB
S5 PACKAGE
5-LEAD PLASTIC SOT-23
T
= 125°C, θJA = 110°C/W
JMAX
5 V
IN
4 ITH/RUN
ORDER PART
NUMBER
LTC1701ES5
S5 PART
MARKING
LTKG
Storage Temperature Range................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec)..................300°C
Consult factory for Industrial and Military grade parts.
ELECTRICAL CHARACTERISTICS
temperature range, otherwise specifications are at TA = 25°C. V
specified. (Note 2)
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
V
IN
I
FB
V
FB
∆V
LINE REG
∆V
LOAD REG
V
ITH/RUN
I
ITH/RUN
I
SW(PEAK)
R
DS(ON)
I
SW(LKG)
t
OFF
Operating Voltage Range 2.5 5.5 V
Feedback Pin Input Current (Note 3) ±0.1 µA
Feedback Voltage (Note 3) ● 1.22 1.25 1.28 V
Reference Voltage Line Regulation VIN = 2.5V to 5V (Note 3) 0.04 0.1 %/V
Output Voltage Load Regulation Measured in Servo Loop, V
Input DC Supply Current (Note 4)
Active Mode V
Sleep Mode V
Shutdown V
Run Threshold High I
Run Threshold Low I
Run Pullup Current V
Peak Switch Current Threshold VFB = 0V 0.9 1.1 A
Switch ON Resistance VIN = 5V, VFB = 0V 0.28 Ω
Switch Leakage Current VIN = 5V, V
Switch Off-Time 400 500 600 ns
The ● denotes the specifications which apply over the full operating
= 3.3V, R
IN
Measured in Servo Loop, V
= 0V 185 300 µA
FB
= 1.4V 135 200 µA
FB
= 0V 0.25 1 µA
ITH/RUN
Ramping Down 1.4 1.6 V
TH/RUN
Ramping Up 0.3 0.6 V
TH/RUN
= 1V 50 100 300 µA
ITH/RUN
= 3.3V, VFB = 0V 0.30 Ω
V
IN
V
= 2.5V, VFB = 0V 0.35 Ω
IN
= 0V, VFB = 0V 0.01 1 µA
ITH/RUN
= 1Meg (from VIN to ITH/RUN) unless otherwise
ITH/RUN
= 1.5V, (Note 3) 0.01 0.70 %
ITH
= 1.9V, (Note 3) –0.80 –1.50 %
ITH
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: The LTC1701E is guaranteed to meet performance specifications
from 0°C to 70°C. Specifications over the –40°C to 85°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls.
2
Note 3: The LTC1701 is tested in a feedback loop which servos VFB to the
midpoint for the error amplifier (V
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
Note 5: T
according to the following formula:
is calculated from the ambient TA and power dissipation P
J
LTC1701ES5: T
= TA + (PD•110°C/W)
J
= 1.7V unless otherwise specified).
ITH
D
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PI FU CTIO S
LTC1701
SW (Pin 1): The Switch Node Connection to the Inductor.
This pin swings from VIN to a Schottky diode (external)
voltage drop below ground. The cathode of the Schottky
diode must be closely connected to this pin.
GND (Pin 2): Ground Pin. Connect to the (–) terminal of
C
, the Schottky diode and (–) terminal of CIN.
OUT
VFB (Pin 3): Receives the feedback voltage from the
external resistive divider across the output. Nominal volt-
ITH/RUN (Pin 4): Combination of Error Amplifier Compensation Point and Run Control Input. The current comparator threshold increases with this control voltage. Nominal
voltage range for this pin is 1.25V to 2.25V. Forcing this
pin below 0.8V causes the device to be shut down. In
shutdown all functions are disabled.
VIN (Pin 5): Main Supply Pin and the (+) Input to the
Current Comparator. Must be closely decoupled to ground.
age for this pin is 1.25V.
Pin Limit Table
NOMINAL (V) ABSOLUTE MAX (V)
PIN NAME DESCRIPTION MIN TYP MAX MIN MAX
1 SW Switch Node –0.3 V
2 GND Ground Pin 0
3VFBOutput Feedback Pin 0 1.25 1.35 –0.3 3
4I
/RUN Error Amplifier Compensation and RUN Pin 0 2.25 –0.3 3
TH
5VINMain Power Supply 2.5 5.5 –0.3 6
–0.3 VIN + 0.3
IN
BLOCK DIAGRA
REF
REF
+
I
TH
CLAMP
–
+
ERROR
AMP
–
1.4V
/REF
+
OVER
VOLTAGE
COMP
–
ITH/RUN
V
V
V
FB
W
V
IN
50µA
SHDN
(1.25V TO 2.25V)
PULSE
STRETCHER
<0.6V
V
FB
1.5V
V
IN
1.25V
BANDGAP
REFERENCE
+
–
I
TH
COMP
V
REF
(1.25V)
CURRENT
COMP
+
–
CONTROL LOGIC
V
REF
OFF-TIMER
AND GATE
CURRENT
SENSE
AMP
GATE
DRIVER
V
IN
+
–
SW
GND
1701 BD
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LTC1701
fO=
−
+
VV
VV T
IN OUT
IN D OFF
1
OPERATIO
U
The LTC1701 uses a contant off-time, current mode
architecture. The operating frequency is then determined
by the off-time and the difference between VIN and V
To optimize efficiency, the LTC1701 automatically switches
between continuous and Burst Mode operation.
The output voltage is set by an external divider returned to
the VFB pin. An error amplfier compares the divided output
voltage with a reference voltage of 1.25V and adjusts the
peak inductor current accordingly.
Main Control Loop
During normal operation, the internal PMOS switch is
turned on when the VFB voltage is below the reference
voltage. The current into the inductor and the load increases until the current limit is reached. The switch turns
off and energy stored in the inductor flows through the
external Schottky diode into the load. After the constant
off-time interval, the switch turns on and the cycle repeats.
The peak inductor current is controlled by the voltage on
the ITH/RUN pin, which is the output of the error
amplifier.This amplifier compares the VFB pin to the 1.25V
reference. When the load current increases, the FB voltage
decreases slightly below the reference. This decrease
causes the error amplifier to increase the ITH/RUN voltage
until the average inductor current matches the new load
current.
OUT
.
The main control loop is shut down by pulling the ITH/RUN
pin to ground. When the pin is released an external resistor
is used to charge the compensation capacitor. When the
voltage at the ITH/RUN pin reaches 0.8V, the main control
loop is enabled and the error amplifier drives the ITH/RUN
pin. Soft-start can be implemented by ramping the voltage
on the ITH/RUN pin (see Applications Information section).
Low Current Operation
When the load is relatively light, the LTC1701 automatically switches to Burst Mode operation in which the
internal PMOS switch operates intermittently based on
load demand. The main control loop is interrupted when
the output voltage reaches the desired regulated value.
The hysteretic voltage comparator trips when ITH/RUN is
below 1.5V, shutting off the switch and reducing the
power consumed. The output capacitor and the inductor
supply the power to the load until the output voltage drops
slightly and the ITH/RUN pin exceeds 1.5V, turning on the
switch and the main control loop which starts another
cycle.
Dropout Operation
In dropout, the internal PMOS switch is turned on continuously (100% duty cycle) providing low dropout operation
with V
an under voltage lockout, care should be taken to shut
down the LTC1701 for VIN < 2.5V.
at VIN. Since the LTC1701 does not incorporate
OUT
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APPLICATIO S I FOR ATIO
The basic LTC1701 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement and begins with the selection of L1. Once
L1 is chosen, the Schottky diode D1 can be selected
followed by CIN and C
L Selection and Operating Frequency
The operating frequency is fixed by VIN, V
constant off-time of about 500ns. The complete expression for operating frequency is given by:
4
OUT
.
OUT
and the
Although the inductor does not influence the operating
frequency, the inductor value has a direct effect on ripple
current. The inductor ripple current ∆IL decreases with
higher inductance and increases with higher VIN or V
∆=
I
L
−
VVfLVV
IN OUT OUT D
VV
IN D
+
+
OUT
:
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LTC1701
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APPLICATIO S I FOR ATIO
where VD is the output Schottky diode forward drop.
Accepting larger values of ∆IL allows the use of low
inductances, but results in higher output voltage ripple
and greater core losses. A reasonable starting point for
setting ripple current is ∆IL = 0.4A.
The inductor value also has an effect on low current
operation. Lower inductor values (higher ∆IL) will cause
Burst Mode operation to begin at higher load currents,
which can cause a dip in efficiency in the upper range of
low current operation. In Burst Mode operation, lower
inductance values will cause the burst frequency to decrease.
Inductor Core Selection
Once the value for L is selected, the type of inductor must
be chosen. Basically, there are two kinds of losses in an
inductor —core and copper losses.
Core losses are dependent on the peak-to-peak ripple
current and core material. However, it is independent of
the physical size of the core. By increasing inductance, the
peak-to-peak inductor ripple current will decrease, therefore reducing core loss. Unfortunately, increased inductance requires more turns of wire and, therefore, copper
losses will increase. When space is not a premium, larger
wire can be used to reduce the wire resistance. This also
prevents excessive heat dissipation in the inductor.
High efficiency converters generally cannot afford the core
loss found in low cost powdered iron cores, forcing the
use of more expensive ferrite, molypermalloy or Kool Mµ
cores. These low core loss materials allow the user to
concentrate on reducing copper loss and preventing saturation.
Ferrite designs have very low core loss and are preferred
at high switching frequencies. Ferrite core material saturates “hard,” which means that inductance collapses
abruptly when the peak design current is exceeded. This
results in an abrupt increase in inductor ripple current and
consequent output voltage ripple. Do not allow the core to
saturate!
Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manu-
®
facturer is Kool Mµ core material. Toroids are very space
efficient, expecially when you can use several layers of
wire. Because they generally lack a bobbin, mounting is
more difficult. However, surface mount designs that do
not increase the height significantly are available
Catch Diode Selection
The diode D1 shown in Figure 1 conducts during the offtime. It is important to adequately specify the diode peak
current and average power dissipation so as not to exceed
the diode ratings.
Losses in the catch diode depend on forward drop and
switching times. Therefore, Schottky diodes are a good
choice for low drop and fast switching times.
Since the catch diode carries the load current during the
off-time, the average diode current is dependent on the
switch duty cycle. At high input voltages, the diode conducts most of the time. As VIN approaches V
conducts only a small fraction of the time. The most
stressful condition for the diode is when the regulator
output is shorted to ground.
Under short-circuit conditions (V
must safely handle I
Under normal load conditions, the average current conducted by the diode is simply:
II
DIODE avg LOAD avg
Remember to keep lead lengths short and observe proper
grounding (see Board Layout Considerations) to avoid
ringing and increased dissipation.
The forward voltage drop allowed in the diode is calculated
from the maximum short-circuit current as:
V
≈
D
where PD is the allowable diode power dissipation and will
be determined by efficiency and/or thermal requirements
(see Efficiency Considerations).
Kool Mµ is a registered trademark of Magnetics, Inc.
=
() ()
I
SC avg
P
D
()
at close to 100% duty cycle.
SC(PK)
VV
−
IN OUT
VV
IN D
VV
IN D
+
V
IN
= 0V), the diode
OUT
+
, the diode
OUT
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APPLICATIO S I FOR ATIO
Most LTC1701 circuits will be well served by either an
MBR0520L or an MBRM120L. An MBR0520L is a good
choice for I
doesn’t need to sustain a continuous short.
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is
a square wave with a duty cycle of approximately V
VIN. To prevent large voltage transients, a low equivalent
series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS
capacitor current is given by:
II
RMS MAX
where the maximum average output current I
the peak current (1 Amp) minus half the peak-to-peak
ripple current, I
This formula has a maximum at VIN = 2V
= I
OUT
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature
than required. Several capacitors may also be paralleled to
meet the size or height requirements of the design. An
additional 0.1µF to 1µF ceramic capacitor is also recom-
mended on VIN for high frequency decoupling.
Output Capacitor (C
The selection of C
Typically, once the ESR requirement is satisfied, the
capacitance is adequate for filtering. The output ripple
(∆V
OUT
∆≈∆ +
V I ESR
where f = operating frequency, C
and ∆IL = ripple current in the inductor. With ∆IL = 0.4
I
OUT(MAX)
OUT(MAX)
≈
/2. This simple worst-case is commonly used to
) is determined by:
OUT L
the output ripple will be less than 100mV with:
≤ 500mA, as long as the output
VVV
OUT IN OUT
= 1 – ∆IL/2.
MAX
OUT
OUT
−
()
V
IN
MAX
, where I
OUT
) Selection
is driven by the required ESR.
1
fC
OUT
= output capacitance
OUT
8
OUT
equals
RMS
/
ESR
Once the ESR requirements for C
RMS current rating generally far exceeds the I
requirement.
When the capacitance of C
output ripple at low frequencies will be large enough to trip
the ITH comparator. This causes Burst Mode operation to
be activated when the LTC1701 would normally be in
continuous mode operation. The effect can be improved at
higher frequencies with lower inductor values.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Aluminum electrolyte and dry tantulum capacitors are both
available in surface mount configurations. The OS-CON
semiconductor dielectric capacitor available from Sanyo
has the lowest ESR(size) product of any aluminum electrolytic at a somewhat higher price. In the case of tantalum, it is critical that the capacitors are surge tested for use
in switching power supplies. An excellent choice is the
AVX TPS, AVX TPSV and KEMET T510 series of surface
mount tantalums, avalable in case heights ranging from
2mm to 4mm. Other capacitor types include Nichicon PL
series, Sanyo POSCAP and Panasonic SP.
Ceramic Capacitors
Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempting for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumental in giving acceptable loop phase margin. Ceramic capacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
Also, ceramic caps are prone to temperature effects which
requires the designer to check loop stability over the
operating temperature range.
For these reasons, most of the input and output capacitance should be composed of tantalum capacitors for
stability combined with about 0.1µF to 1µF of ceramic
capacitors for high frequency decoupling.
COUT
< 100mΩ
have been met, the
OUT
RIPPLE(P-P)
is made too small, the
OUT
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Setting the Output Voltage
The LTC1701 develops a 1.25V reference voltage between
the feedback pin, VFB, and the signal ground as shown in
Figure 2. The output voltage is set by a resistive divider
according to the following formula:
VV
=+
125 1
OUT
.
To prevent stray pickup, a capacitor of about 5pF can be
added across R1, located close to the LTC1701. Unfortunately, the load step response is degraded by this capacitor. Using a good printed circuit board layout eliminates
the need for this capacitor. Great care should be taken to
route the VFB line away from noise sources, such as the
inductor or the SW line.
LTC1701
SGND
Figure 2. Setting the Output Voltage
Transient Response
The OPTI-LOOP compensation allows the transient response to be optimized for a wide range of loads and
output capacitors. The availability of the ITH pin not only
allows optimization of the control loop behavior but also
provides a DC coupled and AC filtered closed-loop response test point. The DC step, rise time and settling at this
test point truly reflects the closed-loop response. Assuming a predominately second order system, phase margin
and/or damping factor can be estimated using the percentage of overshoot seen at this pin. The bandwidth can also
be estimated by examining the rise time at the pin.
The ITH external components shown in the Figure 1 circuit
will provide an adequate starting point for most applications. The series R3-C3 filter sets the dominant pole-zero
loop compensation. The values can be modified slightly
R
2
R
1
V
OUT
R2
C
F
1%
V
FB
5pF
R1
100k
1%
1701 F02
(from 0.5 to 2 times their suggested values) to optimize
transient response once the final PC layout is done and the
particular output capacitor type and value have been
determined. The output capacitors need to be selected
because the various types and values determine the loop
feedback factor gain and phrase. An output current pulse
of 20% to 100% of full-load current having a rise time of
1µs to 10µs will produce output voltage and ITH pin
waveforms that will give a sense of the overall loop
stability without breaking the feedback loop.
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard secondorder overshoot/DC ratio cannot be used to determine
phase margin. The gain of the loop increases with R3 and
the bandwidth of the loop increases with decreasing C3. If
R3 is increased by the same factor that C3 is decreased,
the zero frequency will be kept the same, thereby keeping
the phase the same in the most critical frequency range of
the feedback loop. In addition, a feed-forward capacitor,
CF, can be added to improve the high frequency response,
as shown in Figure 2. Capacitor CF provides phase lead by
creating a high frequency zero with R2 which improves the
phase margin.
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to Application Note 76.
RUN Function
The ITH/RUN pin is a dual purpose pin that provides the
loop compensation and a means to shut down the LTC1701.
Soft-start can also be implemented with this pin. Soft-start
reduces surge currents from VIN by gradually increasing
the internal peak inductor current. Power supply sequencing can also be accomplished using this pin.
An external pull-up is required to charge the external
capacitor C3 in Figure 1. Typically, a 1M resistor between
VIN and ITH/RUN is used. When the voltage on ITH/RUN
reaches about 0.8V the LTC1701 begins operating. At this
point the error amplifier pulls up the ITH/RUN pin to the
normal operating range of 1.25V to 2.25V.
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APPLICATIO S I FOR ATIO
Soft-start can be implemented by ramping the voltage on
ITH/RUN during start-up as shown in Figure 3(c). As the
voltage on ITH/RUN ramps through its operating range the
internal peak current limit is also ramped at a proportional
linear rate.
During normal operation the voltage on the ITH/RUN pin
will vary from 1.25V to 2.25V depending on the load
current. Pulling the ITH/RUN pin below 0.8V puts the
LTC1701 into a low quiescent current shutdown mode
(IQ < 1µA). This pin can be driven directly from logic as
shown in Figures 3(a) and 3(b).
3.3V OR 5V ITH/RUN
D1
(a) (b)
C
C
R
C
ITH/RUN
R1
D1
C
C1
C
R
C
ITH/RUN
C
C
R
C
1) The VIN current is the DC supply current given in the
electrical characteristics which excludes MOSFET driver
and control currents. VIN current results in a small (<0.1%)
loss that increases with VIN, even at no load.
2) The switching current is the sum of the internal MOSFET
driver and control currents. The MOSFET driver current
results from switching the gate capacitance of the power
MOSFET. Each time a MOSFET gate is switched from low
to high to low again, a packet of charge dQ moves from V
IN
to ground. The resulting dQ/dt is a current out of VIN that
is typically much larger than the control circuit current. In
continuous mode, I
GATECHG
= f • QP, where QP is the gate
charge of the internal MOSFET switch.
3) I2R Losses are predicted from the DC resistances of the
MOSFET and inductor. In continuous mode the average
output current flows through L, but is “chopped” between
the topside internal MOSFET and the Schottky diode. At
low supply voltages where the switch on-resistance is
higher and the switch is on for longer periods due to the
higher duty cycle, the switch losses will dominate. Using
a larger inductance helps minimize these switch losses. At
high supply voltages, these losses are proportional to the
load. I2R losses cause the efficiency to drop at high output
currents.
(c)
Figure 3. ITH/RUN Pin Interfacing
1701 F03
Efficiency Considerations
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and what change would
produce the most improvement. Percent efficiency can be
expressed as:
%Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc. are the individual losses as a percentage
of input power.
Although all dissipative elements in the circuit produce
losses, 4 main sources usually account for most of the
losses in LTC1701 circuits: 1) LTC1701 VIN current,
2) switching losses, 3) I2R losses, 4) Schottky diode
losses.
4) The Schottky diode is a major source of power loss at
high currents and gets worse at low output voltages. The
diode loss is calculated by multiplying the forward voltage
drop times the diode duty cycle multiplied by the load
current.
Other “hidden” losses such as copper trace and internal
battery resistances can account for additional efficiency
degradations in portable systems. It is very important to
include these “system” level losses in the design of a
system. The internal battery and fuse resistance losses
can be minimized by making sure that CIN has adequate
charge storage and very low ESR at the switching frequency. Other losses including Schottky conduction losses
during dead-time and inductor core losses generally account for less than 2% total additional loss.
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THERMAL CONSIDERATIONS
The power handling capability of the device at high ambient temperatures will be limited by the maximum rated
junction temperature (125°C). It is important to give
careful consideration to all sources of thermal resistance
from junction to ambient. Additional heat sources mounted
nearby must also be considered.
For surface mount devices, heat sinking is accomplished
by using the heat spreading capabilities of the PC board
and its copper traces. Copper board stiffeners and plated
through-holes can also be used to spread the heat generated by power devices.
The following table lists thermal resistance for several
different board sizes and copper areas. All measurements
were taken in still air on 3/32" FR-4 board with one ounce
copper.
Table 1. Measured Thermal Resistance
COPPER AREA THERMAL RESISTANCE
TOPSIDE* BACKSIDE BOARD AREA θ
2500mm
2
2
2
2
2
2500mm22500mm
1000mm22500mm
225mm22500mm
100mm22500mm
2
50mm
*Device is mounted on topside.
2500mm
2500mm
2500mm
2500mm
2500mm
2
2
2
2
2
Calculating Junction Temperature
JA
125°C/W
125°C/W
130°C/W
135°C/W
150°C/W
The junction temperature is given by:
TJ = T
RISE
+ T
AMBIENT
As an example, consider the case when the LTC1701 is in
dropout at an input voltage of 3.3V with a load current of
0.5A. The ON resistance of the P-channel switch is approximately 0.30Ω. Therefore, power dissipated by the
part is:
PD = I2 • R
DS(ON)
= 75mW
The SOT package junction-to-ambient thermal resistance,
θJA, will be in the range of 125°C/W to 150°C/W. There-
fore, the junction temperature of the regulator operating in
a 25°C ambient temperature is approximately:
TJ = 0.075 • 150 + 25 = 36°C
Remembering that the above junction temperature is
obtained from a R
junction temperature based on a higher R
at 25°C, we might recalculate the
DS(ON)
DS(ON)
since it
increases with temperature. However, we can safely assume that the actual junction temperature will not exceed
the absolute maximum junction temperature of 125°C.
Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1701. These items are also illustrated graphically in
the layout diagram of Figure 4. Check the following in your
layout:
In a majority of applications, the LTC1701 does not
dissipate much heat due to its high efficiency. However, in
applications where the switching regulator is running at
high duty cycles or the part is in dropout with the switch
turned on continuously (DC), some thermal analysis is
required. The goal of the thermal analysis is to determine
whether the power dissipated by the regulator exceeds the
maximum junction temperature. The temperature rise is
given by:
T
= PD • θ
RISE
where PD is the power dissipated by the regulator and θ
JA
JA
is the thermal resistance from the junction of the die to the
ambient temperature.
1. Does the capacitor CIN connect to the power VIN (Pin 5)
and GND (Pin 2) as close as possible? This capacitor
provides the AC current to the internal P-channel MOSFET
and its driver.
2. Is the Schottky diode closely connected between the
ground (Pin 2) and switch output (Pin 1)?
3. Are the C
, L1 and D1 closely connected? The
OUT
Schottky anode should connect directly to the input capacitor ground.
4. The resistor divider, R1 and R2, must be connected
between the (+) plate of C
and a ground line terminated
OUT
near GND (Pin 2). The feedback signal FB should be routed
away from noisy components and traces, such as the SW
line (Pin 1).
9
Page 10
LTC1701
SW
V
FB
V
IN
ITH/RUN
GND
LTC1701
D1
L1
R2
R1
C
OUT
1701 F04
V
OUT
V
IN
1
2
3
5
4
C
IN
C
C
R
C
R
S
BOLD LINES INDICATE HIGH CURRENT PATHS
++
U
WUU
APPLICATIO S I FOR ATIO
5. Keep sensitive components away from the SW pin. The
input capacitor CIN, the compensation capacitor CC and all
the resistors R1, R2, RC and RS should be routed away
from the SW trace and the components L1 and D1.
U
TYPICAL APPLICATIO S
3- to 4-Cell NiCd/NiMH to 2.5V Converter
V
2.7V TO 5.5V
IN
C1
15µF
R4
+
1M
C4
0.1µF
R3
5.1k
V
ITH/RUN
C3
330pF
IN
LTC1701
GND
Figure 4. LTC1701 Layout Diagram (See Board Layout Checklist)
L1
SW
V
4.7µH
D1
R2
121k
FB
R1
121k
+
V
OUT
(2.5V/0.5A)
C2
15µF
10
C1, C2: AVX TAJA156M010R
C4: TAIYO YUDEN EMK107BJ104MA
D1: MBR0520
L1: MURATA LQH3C4R7M24 OR TAIYO YUDEN LEMC3225B4R7M
2mm Nominal Height 1.5V Converter
V
IN
15µF
+
C1
2.5V TO 5.5V
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
C4: TAIYO YUDEN LMK212BJ105MG
C5: TAIYO YUDEN JMK212BJ475MG
D1: MBRM120L
L1: MURATA LQH3C4R7M24
C4
1µF
1M
5.1k
R4
R3
V
IN
ITH/RUN
C3
330pF
LTC1701
GND
SW
V
1701 TA01
L1
4.7µH
D1
22µF
+
C2
R2
R1
100k
20k
FB
V
OUT
(1.5V/0.5A)
C5
4.7µF
1701 TA02
Page 11
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TYPICAL APPLICATIO S
V
IN
5V
+
C1
15µF
C1: AVX TAJA156M010R
C2: AVX TAJA226M006R
C4: TAIYO YUDEN LMK212BJ105MG
C5: TAIYO YUDEN JMK212BJ475MG
D1: MBRM120L
L1: MURATA LQH3C4R7M24
C4
1µF
OFF
5V to 3.3V Converter with Push-Button On/Off
V
IN
ON
R5
5.1M
1M
R4
R3
5.1k
LTC1701
ITH/RUNSWV
GND
C3
330pF
FB
D1
R1
20.5k
L1
4.7µH
R2
34k
22µF
LTC1701
V
OUT
(3.3V/0.5A)
+
C2
C5
4.7µF
1701 TA03
PACKAGE DESCRIPTIO
0.35 – 0.55
(0.014 – 0.022)
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. DIMENSIONS ARE INCLUSIVE OF PLATING
3. DIMENSIONS ARE EXCLUSIVE OF MOLD FLASH AND METAL BURR
4. MOLD FLASH SHALL NOT EXCEED 0.254mm
5. PACKAGE EIAJ REFERENCE IS SC-74A (EIAJ)
U
Dimensions in inches (millimeters) unless otherwise noted.
S5 Package
5-Lead Plastic SOT-23
(LTC DWG # 05-08-1633)
2.80 – 3.00
(0.110 – 0.118)
(NOTE 3)
0.95
(0.037)
REF
0.90 – 1.45
(0.035 – 0.057)
0.90 – 1.30
(0.035 – 0.051)
S5 SOT-23 0599
2.60 – 3.00
(0.102 – 0.118)
1.50 – 1.75
(0.059 – 0.069)
(0.00 – 0.006)
0.09 – 0.20
(0.004 – 0.008)
(NOTE 2)
1.90
(0.074)
REF
0.00 – 0.15
0.35 – 0.50
(0.014 – 0.020)
FIVE PLACES (NOTE 2)
11
Page 12
LTC1701
TYPICAL APPLICATIO
U
Single Cell Li-Ion to 3.3V Buck Boost Converter
V
2.5V TO 4.2V
C1: AVX TPSB336K006R0600
C2: TAJA156M010R
C6: TAIYO YUDEN JMK212BJ475MG
D1: MBR0520L
L1, L2: COILTRONICS CTX5-1
IN
1µF
CERAMIC
R4
1M
+
C4
C1
33µF
R3
5.1k
V
IN
LTC1701
ITH/RUNSWV
GND
C3
330pF
C6
4.7µF
+
L2 D1
FB
R1
20.5k
L1
4.7µH
R2
34k
V
OUT
(3.3V)
+
C2
V
15µF
1701 TA04
INIOUT(MAX)
2.5V 200mA
3.0V 225mA
3.5V 250mA
4.0V 280mA
4.2V 290mA
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
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LTC1174-5 I
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LT1375/LT1376 1.5A, 500kHz Step-Down Switching Regulator High Frequency, Small Inductor, High Efficiency, SO-8
LTC1435/LTC1435A High Efficiency, Low Noise, Synchronous Step-Down Converter 3.5V ≤ VIN ≤ 36V, 16-Pin Narrow SO and SSOP
LTC1474/LTC1475 Low Quiescent Current High Efficiency Step-Down Converter 10µA IQ, 8-Pin MSOP and SO Packages
LTC1622 Low Input Voltage Current Mode Step-Down DC/DC Controller High Frequency, High Efficiency, 8-Pin MSOP
LTC1627 Monolithic Synchronous Step-Down Switching Regulator SO-8, 2.65V ≤ VIN ≤ 10V, I
LTC1707 Monolithic Synchronous Step-Down Switching Regulator SO-8, 2.95V ≤ VIN ≤ 10V, V
LTC1772 Low Input Voltage Current Mode Step-Down DC/DC Controller 550kHz, 6-Pin SOT-23, I
Linear Technology Corporation
12
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear-tech.com
Up to 300mA, SO-8
OUT
Up to 500mA
OUT
Output
REF
Up to 5A, 2.2V < VIN < 10V
OUT
1701i LT/TP 1299 4K • PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1999
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