Datasheet LTC1622 Datasheet (Linear Technology)

FEATURES

LTC1622

Low Input Voltage

Current Mode Step-Down

DC/DC Controller

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DESCRIPTIO

■

High Efficiency

■

Constant Frequency 550kHz Operation

■

VIN Range: 2V to 10V

■

Multiampere Output Currents

■

OPTI-LOOPTM Compensation Minimizes C

■

Selectable, Burst Mode Operation

■

Low Dropout Operation: 100% Duty Cycle

■

Synchronizable up to 750kHz

■

Current Mode Operation for Excellent Line and Load

OUT

Transient Response

■

Low Quiescent Current: 350µA

■

Shutdown Mode Draws Only 15µA Supply Current

■

±1.9% Reference Accuracy

■

Available in 8-Lead MSOP

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APPLICATIO S

■

1- or 2-Cell Li-Ion Powered Applications

■

Cellular Telephones

■

Wireless Modems

■

Portable Computers

■

Distributed 3.3V, 2.5V or 1.8V Power Systems

■

Scanners

■

Battery-Powered Equipment

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode and OPTI-LOOP are a trademarks of Linear Technology Corporation.

The LTC®1622 is a constant frequency current mode step­down DC/DC controller providing excellent AC and DC load
and line regulation. The device incorporates an accurate
undervoltage feature that shuts the LTC1622 down when
the input voltage falls below 2V.

The LTC1622 boasts a ±1.9% output voltage accuracy and
consumes only 350µA of quiescent current. For applica-
tions where efficiency is a prime consideration and the
load current varies from light to heavy, the LTC1622 can
be configured for Burst ModeTM operation. Burst Mode
operation enhances low current efficiency and extends
battery run time. Burst Mode operation is inhibited during
synchronization or when the SYNC/MODE pin is pulled low
to reduce noise and possible RF interference.

High constant operating frequency of 550kHz allows the
use of a small inductor. The device can also be synchro­nized up to 750kHz for special applications. The high
frequency operation and the available 8-lead MSOP pack­age create a high performance solution in an extremely
small amount of PCB area.

To further maximize the life of the battery source, the
P-channel MOSFET is turned on continuously in dropout
(100% duty cycle). In shutdown, the device draws a mere
15µA.

TYPICAL APPLICATIO

V

IN

2.5V TO 8.5V

8

V

IN

2

I

4

TH

LTC1622

SYNC/MODE

RUN/SS

V

3

R1
10k

C3
220pF

470pF

C1: TAIYO YUDEN CERAMIC EMK325BJ106MNT
C2: SANYO POSCAP 6TPA47M
D1: INTERNATIONAL RECTIFIER IR10BQ015

SENSE

FB

PDRV

GND

1

–

7

5

6

Figure 1. High Efficiency Step-Down Converter

R2

0.03Ω

Si3443DV

L1: MURATA LQN6C-4R7
R2: DALE WSL-1206 0-03Ω

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4.7µH

D1
IR10BQ015

Efficiency vs Load Current with
Burst Mode Operation Enabled

100

C1
10µF
10V

L1

R3
159k

R4
75k

V

OUT

2.5V

1.5A

+

C2
47µF
6V

1622 F01a

90

80

70

EFFICIENCY (%)

60

50

40

1 100 1000 5000

VIN = 3.3V

10

VIN = 4.2V

VIN = 8.4V

V

OUT

R

SENSE

LOAD CURRENT (mA)

VIN = 6V

= 2.5V

= 0.03Ω

1622 F01b

1

LTC1622

WW

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ABSOLUTE MAXIMUM RATINGS

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(Note 1)

Input Supply Voltage (VIN).........................–0.3V to 10V

RUN/SS Voltage .......................................–0.3V to 2.4V

SYNC/MODE Voltage ................................. –0.3V to V

SENSE– Voltage .......................................... 2.4V to V

IN
IN

PDRV Peak Output Current (<10µs) ......................... 1A

Storage Ambient Temperature Range ... –65°C to 150°C

U

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PACKAGE/ORDER INFORMATION

ORDER PART

NUMBER

LTC1622CMS8

MS8 PART MARKING

LTDB

–

SENSE

1
2

I

TH

3

V

FB

4

RUN/SS

MS8 PACKAGE

8-LEAD PLASTIC MSOP

T

= 125°C, θJA = 250°C/W

JMAX

TOP VIEW

8

V

IN

7

PDRV

6

GND

5

SYNC/MODE

Operating Temperature Range

Commercial ............................................ 0°C to 70°C

Industrial ........................................... –45°C to 85°C

Junction Temperature (Note 2)............................. 125°C

Lead Temperature (Soldering, 10 sec).................. 300°C

ORDER PART

SENSE

I

V

RUN/SS

TOP VIEW

–

1

2

TH

3

FB

4

S8 PACKAGE

8-LEAD PLASTIC SO

T

= 125°C, θJA = 150°C/ W

JMAX

V

8

IN

PDRV

7

GND

6

SYNC/MODE

5

NUMBER

LTC1622CS8
LTC1622IS8

S8 PART MARKING

1622
1622I

Consult factory for Military grade parts.

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

I

VFB

V

FB

V

OVL

∆V

OSENSE

V

LOADREG

I

S

V

RUN/SS

I

RUN/SS

f

OSC

V

SYNC/MODE

V

UVLO

Feedback Current (Note 3) VFB = 0.8V 10 70 nA
Regulated Feedback Voltage (Note 3) Commercial Grade ● 0.785 0.8 0.815 V

(Note 3) Industrial Grade
Output Overvoltage Lockout Referenced to Nominal V
Reference Voltage Line Regulation VIN = 4.2V to 8.5V (Note 3) 0.04 0.08 %/V
Output Voltage Load Regulation Measured in Servo Loop; V

Measured in Servo Loop; V
Input DC Supply Current (Note 4)

Burst Mode Inhibited V
Sleep Mode V
Shutdown V
Shutdown V

RUN/SS Threshold Commercial Grade ● 0.4 0.7 0.9 V

Soft-Start Current Source V
Oscillator Frequency VFB = 0.8V 475 550 625 kHz

SYNC/MODE Threshold V
Undervoltage Lockout VIN Ramping Down ● 1.55 1.92 2.3 V

= 2.3V 450 µA

IN

= 0V, V

ITH
RUN/SS
RUN/SS

Industrial Grade

RUN/SS

V

FB

SYNC/MODE

Ramping Up 1.97 2.36 V

V

IN

SYNC/MODE

= 0V 15 30 µA
= 0V, VIN = V

= 0V 1 2.5 5 µA

= 0V 75 110 140 kHz

Ramping Down 1 1.5 V

OUT

= 0.2V to 0.625V 0.3 0.5 %

ITH

= 0.9V to 0.625V – 0.3 –0.5 %

ITH

= 2.4V 350 400 µA

– 0.1V 4 10 µA

UVLO

● 0.780 0.8 0.820 V

4 7.5 10.5 %

● 0.3 0.7 1.0 V

2

LTC1622

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating

temperature range, otherwise specifications are at TA = 25°C. VIN = 4.2V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

PDRV t

r

PDRV t

f

∆V

SENSE(MAX)

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Note 2: T
dissipation P

Gate Drive Rise Time C
Gate Drive Fall Time C

= 3000pF 80 140 ns

LOAD

= 3000pF 100 140 ns

LOAD

Maximum Current Sense Voltage ● 80 110 140 mV

Note 3: The LTC1622 is tested in a feedback loop that servos V

= 0.8V).

ITH

is calculated from the ambient temperature TA and power

J

according to the following formula:

D

feedback point for the error amplifier (V
Note 4: Dynamic supply current is higher due to the gate charge being

delivered at the switching frequency.

to the

FB

LTC1622CS8; TJ = TA + (PD • 150°C/W),
LTC1622CMS8; TJ = TA + (PD • 250°C/W)

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TYPICAL PERFORMANCE CHARACTERISTICS

Shutdown Current
vs Supply Voltage

45

40

35

30

25

20

15

SHUTDOWN CURRENT (µA)

10

5

0

23

45

67

SUPPLY VOLTAGE (V)

89

1622 G01

RUN/SS Current vs Supply Voltage

3.50

3.00

2.50

2.00

SOFT-START CURRENT (µA)

1.50

10

1.00
3579

2

46 10

SUPPLY VOLTAGE (V)

8

1622 G02

Maximum Current Sense Voltage
vs Duty Cycle

110

VIN = 4.2V

100

90

80

70

60

TRIP VOLTAGE (mV)

50

40

30

20 30

40 50

DUTY CYCLE (%)

UNSYNC

60 70

80 90

100

1622 G03

Normalized Oscillator Frequency
vs Temperature

10.0
VIN = 4.2V

7.5

5.0

2.5

0

–2.5

–5.0

NORMALIZED FREQUENCY (%)

–7.5

–10.0

–55 –35

–15 5

25 45 65

TEMPERATURE (°C)

85 105

1622 G04

125

Reference Voltage
vs Temperature

0.810
VIN = 4.2V

0.805

0.800

0.795

0.790

0.785

REFERENCE VOLTAGE (V)

0.780

0.775

–55 –35

–15 5

TEMPERATURE (°C)

25 45 65

85 105

1622 G05

125

Undervoltage Lockout Voltage
vs Temperature

2.10

2.05

2.00

1.95

1.90

1.85

1.80

UNDERVOLTAGE LOCKOUT VOLTAGE (V)

1.75
–55 –35

–15 5

25 45 65

TEMPERATURE (°C)

85 105

125

1622 G06

3

LTC1622

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TYPICAL PERFORMANCE CHARACTERISTICS

Efficiency vs Load Current for
Figure 1 with Burst Mode
Operation Defeated

100

90

80

VIN = 4.2V

VIN = 3.3V

Load Step Transient Response
Burst Enabled

Load Step Transient Response
Burst Inhibited

70

EFFICIENCY (%)

60

50

40

1

VIN = 6V

VIN = 8.4V

V

OUT

R

SENSE

10 100

LOAD CURRENT (mA)

= 2.5V

= 0.03Ω

1000

1622 G07

100mV/DIV

I

= 50mA TO 1.2A

LOAD

VIN = 4.2V

UUU

PIN FUNCTIONS

SENSE– (Pin 1): The Negative Input to the Current Com­parator.

ITH (Pin 2): Error Amplifier Compensation Point. The
current comparator threshold increases with this control
voltage. Nominal voltage range for this pin is 0V to 1.2V.

VFB (Pin 3): Receives the feedback voltage from an exter­nal resistive divider across the output capacitor.

100mV/DIV

I

= 50mA TO 1.2A

LOAD

1622 G08

VIN = 4.2V

1622 G09

SYNC/MODE (Pin 5): This pin performs three functions.
Greater than 2V on this pin allows Burst Mode operation
at low load currents, while grounding or applying a clock
signal on this pin defeats Burst Mode operation. An
external clock between 625kHz and 750kHz applied to this
pin forces the LTC1622 to operate at the external clock
frequency.

Do not attempt to synchronize below 625kHz

.

Pin 5 has an internal 1µA pull-up current source.

RUN/SS (Pin 4): Combination of Soft-Start and Run
Control Inputs. A capacitor to ground at this pin sets the
ramp time to full output current. The time is approximately

0.45s/µF. Forcing this pin below 0.4V causes all circuitry
to be shut down.

4

GND (Pin 6): Ground Pin.
PDRV (PIN 7): Gate Drive for the External P-Channel

MOSFET. This pin swings from 0V to VIN.
VIN (Pin 8): Main Supply Pin. Must be closely decoupled

to ground Pin 6.

LTC1622

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FUNCTIONAL DIAGRA

V

SYNC/
MODE

V

FB

V

REF

0.8V

IN

1µA

5

0.3V

3

REFERENCE

TRIP = 1.97V

SHUTDOWN

–
+

V

IN

0.8V

UVLO

BURST DEFEAT

X

6

GND

Y

OSC

FREQ

SHIFT

V

IN

RUN/SS

4

V

REF

Y = “0” ONLY WHEN X IS A CONSTANT “1”
OTHERWISE Y = “1”

SLOPE
COMP

–

0.8V

V

2.5µA

+ 60mV

REF

+

+

EA

–

g

= 0.5m

m

RUN/

SOFT-START

+

OV

–

Ω

2

I

TH

0.12V

EN

–
+

BURST

S

RQ

R

S1

V

CC

SLEEP

SWITCHING

BLANKING

LOGIC

AND

CIRCUIT

0.36V

SENSE

1622 BD

–

1

+

ICOMP

V

8

IN

–

V

IN

PDRV

7

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OPERATIO

Main Control Loop

The LTC1622 is a constant frequency current mode switch­ing regulator. During normal operation, the external
P-channel power MOSFET is turned on each cycle when
the oscillator sets the RS latch (RS1) and turned off when
the current comparator (I
inductor current at which I
controlled by the voltage on the ITH pin, which is the output
of the error amplifier EA. An external resistive divider
connected between V
an output feedback voltage VFB. When the load current
increases, it causes a slight decrease in VFB relative to the

0.8V reference, which in turn causes the ITH voltage to
increase until the average inductor current matches the
new load current.

The main control loop is shut down by pulling the RUN/SS
pin low. Releasing RUN/SS allows an internal 2.5µA

(Refer to Functional Diagram)

) resets the latch. The peak

COMP

resets the RS latch is

COMP

and ground allows EA to receive

OUT

current source to charge up the soft-start capacitor CSS.
When CSS reaches 0.7V, the main control loop is enabled
with the ITH voltage clamped at approximately 5% of its
maximum value. As CSS continues to charge, ITH is gradu­ally released allowing normal operation to resume.

Comparator OV guards against transient overshoots
>7.5% by turning off the P-channel power MOSFET and
keeping it off until the fault is removed.

Burst Mode Operation

The LTC1622 can be enabled to go into Burst Mode
operation at low load currents simply by leaving the SYNC/
MODE pin open or connecting it to a voltage of at least 2V.
In this mode, the peak current of the inductor is set as if
V

= 0.36V (at low duty cycles) even though the voltage

ITH

at the ITH pin is at lower value. If the inductor’s average
current is greater than the load requirement, the voltage at

5

LTC1622

OPERATIO

U

(Refer to Functional Diagram)

the ITH pin will drop. When the ITH voltage goes below

0.12V, the sleep signal goes high, turning off the external
MOSFET. The sleep signal goes low when the ITH voltage
rises above 0.22V and the LTC1622 resumes normal
operation. The next oscillator cycle will turn the external
MOSFET on and the switching cycle repeats.

Frequency Synchronization

The LTC1622 can be externally driven by a TTL/CMOS
compatible clock signal up to 750kHz.

Do not

synchronize
the LTC1622 below its maximum default operating fre­quency of 625kHz as this may cause abnormal operation
and an undesired frequency spectrum. The LTC1622 is
synchronized to the rising edge of the clock. The external
clock pulse width must be at least 100ns and not more
than the period minus 200ns.

Synchronization is inhibited when the feedback voltage is
below 0.3V. This is to prevent inductor current buildup
under short-circuit conditions. Burst Mode operation is
deactivated when the LTC1622 is externally driven by a
clock.

Dropout Operation

Short-Circuit Protection

When the output is shorted to ground, the frequency of the
oscillator will be reduced to about 110kHz. This lower
frequency allows the inductor current to safely discharge,
thereby preventing current runaway. The oscillator’s fre­quency will gradually increase to its nominal value when
the feedback voltage increases above 0.65V. Note that
synchronization is inhibited until the feedback voltage
goes above 0.3V.

Overvoltage Protection

As a further protection, the overvoltage comparator in the
LTC1622 will turn the external MOSFET off when the
feedback voltage has risen 7.5% above the reference
voltage of 0.8V. This comparator has a typical hysteresis
of 35mV.

Slope Compensation and Peak Inductor Current

The inductor’s peak current is determined by:

V

I

PK

=

ITH

R

10

SENSE

()

When the input supply voltage decreases towards the
output voltage, the rate of change of inductor current
during the ON cycle decreases. This reduction means that
the P-channel MOSFET will remain on for more than one
oscillator cycle since the inductor current has not ramped
up to the threshold set by EA. Further reduction in input
supply voltage will eventually cause the P-channel MOSFET
to be turned on 100%, i.e., DC. The output voltage will then
be determined by the input voltage minus the voltage drop
across the MOSFET, the sense resistor and the inductor.

Undervoltage Lockout

To prevent operation of the P-channel MOSFET below safe
input voltage levels, an undervoltage lockout is incorpo­rated into the LTC1622. When the input supply voltage
drops below 2V, the P-channel MOSFET and all circuitry is
turned off except the undervoltage block, which draws
only several microamperes.

when the LTC1622 is operating below 40% duty cycle.
However, once the duty cycle exceeds 40%, slope com­pensation begins and effectively reduces the peak induc­tor current. The amount of reduction is given by the curves
in Figure 2.

110
100

90
80

(%)

70
60

OUT(MAX)

/I

OUT

SF = I

Figure 2. Maximum Output Current vs Duty Cycle

I

= 0.4I

50
40
30
20
10

RIPPLE

AT 5% DUTY CYCLE
I

= 0.2I

RIPPLE

AT 5% DUTY CYCLE

VIN = 4.2V
UNSYNC

0 70 80 90 1006010 20 30 40 50

DUTY CYCLE (%)

PK

PK

1622 F02

6

LTC1622

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APPLICATIONS INFORMATION

The basic LTC1622 application circuit is shown in Figure

1. External component selection is driven by the load
requirement and begins with the selection of L and R
Next, the Power MOSFET and the output diode D1 are
selected followed by CIN and C

R

R

Selection for Output Current

SENSE

is chosen based on the required output current.

SENSE

OUT

.

With the current comparator monitoring the voltage devel­oped across R

, the threshold of the comparator

SENSE

determines the inductor’s peak current. The output cur­rent the LTC1622 can provide is given by:

I

OUT

where I

0082.

=−

R

SENSE

is the inductor peak-to-peak ripple current

RIPPLE

I

RIPPLE

(see Inductor Value Calculation section).
A reasonable starting point for setting ripple current is

I

RIPPLE

= (0.4)(I

). Rearranging the above equation, it

OUT

becomes:

SENSE

.

V

. The inductor’s peak-to-peak ripple current is given

OUT

by:

I

RIPPLE

VVfLVV

−

IN OUT OUT D

=

()




VV



IN D



+



+



where f is the operating frequency. Accepting larger values
of I

allows the use of low inductances, but results in

RIPPLE

higher output voltage ripple and greater core losses. A
reasonable starting point for setting ripple current is
I

RIPPLE

= 0.4(I

OUT(MAX)

). Remember, the maximum I

RIPPLE

occurs at the maximum input voltage.
With Burst Mode operation selected on the LTC1622, the

ripple current is normally set such that the inductor
current is continuous during the burst periods. Therefore,
the peak-to-peak ripple current should not exceed:

I

RIPPLE

≤

0 036.

R

SENSE

This implies a minimum inductance of:

R

SENSE

=

1

for Duty Cycle < 40%

I

15

OUT

()( )

However, for operation that is above 40% duty cycle, slope
compensation has to be taken into consideration to select
the appropriate value to provide the required amount of
current. Using Figure 2, the value of R

R

SENSE

=

SF

15

I

OUT

100

()( )( )

SENSE

is:

Inductor Value Calculation

The operating frequency and inductor selection are inter­related in that higher operating frequencies permit the use
of a smaller inductor for the same amount of inductor
ripple current. However, this is at the expense of efficiency
due to an increase in MOSFET gate charge losses.

The inductance value also has a direct effect on ripple
current. The ripple current, I

, decreases with higher

RIPPLE

inductance or frequency and increases with higher VIN or

VV

−

L

MIN

(Use V

IN OUT

=



f



R



IN(MAX)

0 036.

SENSE

= VIN)






A smaller value than L



VV

OUT D



VV



IN D

could be used in the circuit;

MIN



+



+



however, the inductor current will not be continuous
during burst periods.

Inductor Core Selection

Once the value for L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mu® cores. Actual core loss is independent of core
size for a fixed inductor value, but it is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, increased inductance requires
more turns of wire and therefore copper losses will
increase. Ferrite designs have very low core losses and are

Kool Mu is a registered trademark of Magnetics, Inc.

7

LTC1622

I

VV

VV

I

D

IN OUT

IN D

OUT

=

−
+











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APPLICATIONS INFORMATION

preferred at high switching frequencies, so design goals
can concentrate on copper loss and preventing saturation.
Ferrite core materials saturate “hard,” which means that
the inductance collapses abruptly when the peak design
current is exceeded. This results in an abrupt increase in
inductor ripple current and consequently, output voltage
ripple. Do not allow the core to saturate!

Molypermalloy (from Magnetics, Inc.) is a very good, low
loss core material for toroids, but it is more expensive than
ferrite. A reasonable compromise from the same manu­facturer is Kool Mu. Toroids are very space efficient,
especially when you can use several layers of wire.
Because they generally lack a bobbin, mounting is more
difficult. However, new surface mountable designs that do
not increase the height significantly are available.

Power MOSFET Selection

An external P-channel power MOSFET must be selected
for use with the LTC1622. The main selection criteria for
the power MOSFET are the threshold voltage V
the “on” resistance R
C

and total gate charge.

RSS

Since the LTC1622 is designed for operation down to low
input voltages, a sublogic level threshold MOSFET (R
guaranteed at VGS = 2.5V) is required for applications that
work close to this voltage. When these MOSFETs are used,
make sure that the input supply to the LTC1622 is less than
the absolute maximum MOSFET VGS rating, typically 8V.
The gate drive voltage levels are from ground to VIN.

The required minimum R
erned by its allowable power dissipation. For applications
that may operate the LTC1622 in dropout, i.e., 100% duty
cycle, at its worst case the required R

R

DS ON

()

DC

where PP is the allowable power dissipation and δp is the
temperature dependency of R
given for a MOSFET in the form of a normalized R
temperature curve, but δp = 0.005/°C can be used as an
approximation for low voltage MOSFETs.

100

=

%=

,reverse transfer capacitance

DS(ON)

of the MOSFET is gov-

DS(ON)

is given by:

DS(ON)

P

P

2

+

Ip

()

OUT MAX

()

1 δ

()

. (1 + δp) is generally

DS(ON)

GS(TH)

DS(ON)

DS(ON)

and

vs

In applications where the maximum duty cycle is less than
100% and the LTC1622 is in continuous mode, the R
is governed by:

P

R

DS ON

where DC is the maximum operating duty cycle of the
LTC1622.

When the LTC1622 is operating in continuous mode, the
MOSFET power dissipation is:

P

MOSFET

where K is a constant inversely related to gate drive
current. Because of the high switching frequency, the
second term relating to switching loss is important not to
overlook. The constant K = 3 can be used to estimate the
contributions of the two terms in the MOSFET dissipation
equation.

Output Diode Selection

The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches V
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short circuited. Under this condition the diode must
safely handle I
it is important to adequately specify the diode peak current
and average power dissipation so as not to exceed the
diode ratings.

Under normal load conditions, the average current con­ducted by the diode is:

≅

()

=

+

DC I

()

VV

VV

KV I C f

()( )( )()

PEAK

P

2

+

1 δ

p

()

OUT

+

OUT D

+

IN D

2

IN OUT RSS

at close to 100% duty cycle. Therefore,

2

IpR

OUT DS ON

()

+

1 δ

()

DS(ON)

()

OUT

8

LTC1622

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APPLICATIONS INFORMATION

The allowable forward voltage drop in the diode is calcu­lated from the maximum short-circuit current as:

P

V

≈

F

where PD is the allowable power dissipation and will be
determined by efficiency and/or thermal requirements.

A fast switching diode must also be used to optimize
efficiency. Schottky diodes are a good choice for low
forward drop and fast switching times. Remember to keep
lead length short and observe proper grounding (see
Board Layout Checklist) to avoid ringing and increased
dissipation.

CIN and C

In continuous mode, the source current of the P-channel
MOSFET is a square wave of duty cycle (V
(VIN + VD). To prevent large voltage transients, a low ESR
input capacitor sized for the maximum RMS current must
be used. The maximum RMS capacitor current is given by:

CI

IN MAX

This formula has a maximum at VIN = 2V
= I

OUT

used for design because even significant deviations do not
offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Several capacitors may be paralleled to meet the
size or height requirements in the design. Due to the high
operating frequency of the LTC1622, ceramic capacitors
can also be used for CIN. Always consult the manufacturer
if there is any question.

The selection of C
series resistance (ESR). Typically, once the ESR require­ment is satisfied, the capacitance is adequate for filtering.
The output ripple (∆V

D

I

SC MAX

()

Selection

OUT

+ VD)/

OUT

12/

VVV

OUT IN OUT

Required I

/2. This simple worst-case condition is commonly

≈

RMS

is driven by the required effective

OUT

OUT

[]

) is approximated by:

−

()

V

IN

, where I

OUT

RMS

∆V I ESR

where f is the operating frequency, C
capacitance and I
tor. The output ripple is highest at maximum input voltage
since ∆IL increases with input voltage.

The choice of using a smaller output capacitance in­creases the output ripple voltage due to the frequency
dependent term, but can be compensated for by using
capacitors of very low ESR to maintain low ripple voltage.
The ITH pin OPTI-LOOP compensation components can be
optimized to provide stable, high performance transient
response regardless of the output capacitors selected.

Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance through­hole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest ESR (size)
product of any aluminum electrolytic at a somewhat
higher price. Once the ESR requirement for C
met, the RMS current rating generally far exceeds the
I

RIPPLE(P-P)

In surface mount applications, multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec­trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS, AVX TPSV and KEMET T510 series of surface mount
tantalum, available in case heights ranging from 2mm to
4mm. Other capacitor types include Sanyo OS-CON, Sanyo
POSCAP, Nichicon PL series and the Panasonic SP series.

Low Supply Operation

Although the LTC1622 can function down to 2V, the
maximum allowable output current is reduced when V
decreases below 3V. Figure 3 shows the amount of change
as the supply is reduced down to 2V. Also shown in
Figure 3 is the effect of VIN on V
Remember the maximum voltage on the ITH pin defines

≈+

OUT RIPPLE

requirement.






RIPPLE

8

is the ripple current in the induc-



1



fC



OUT

is the output

OUT

has been

OUT

as VIN goes below 2.3V.

REF

IN

9

LTC1622

U

WUU

APPLICATIONS INFORMATION

101

V

100

99

98

97

NORMALIZED VOLTAGE (%)

96

95

2.0

Figure 3. Line Regulation of V

the maximum current sense voltage that sets the maxi­mum output current.

Setting Output Voltage

The LTC1622 develops a 0.8V reference voltage between
the feedback (Pin 3) terminal and ground (see Figure 4). By
selecting resistor R1, a constant current is caused to flow
through R1 and R2 to set the output voltage. The regulated
output voltage is determined by:

REF

V

ITH

2.2 2.4 2.6 2.8
INPUT VOLTAGE (V)

REF

3.0

1622 F03

and V

ITH

is limiting the efficiency and which change would produce
the most improvement. Efficiency can be expressed as:

Efficiency = 100% – (η1 + η2 + η3 + ...)

where η1, η2, etc. are the individual losses as a percent­age of input power.

Although all dissipative elements in the circuit produce
losses, four main sources usually account for most of the
losses in LTC1622 circuits: 1) LTC1622 DC bias current,

2) MOSFET gate charge current, 3) I2R losses, 4) voltage
drop of the output diode and 5) transition losses.

1. The VIN current is the DC supply current, given in the
electrical characteristics, that excludes MOSFET driver
and control currents. VIN current results in a small loss
which increases with VIN.

2. MOSFET gate charge current results from switching
the gate capacitance of the power MOSFET. Each time
a MOSFET gate is switched from low to high to low
again, a packet of charge dQ moves from VIN to ground.
The resulting dQ/dt is a current out of VIN which is
typically much larger than the DC supply current. In
continuous mode, I

GATECHG

= f(Qp).

V

=+

08 1

OUT



.







R

2



R

1



For most applications, a 30k resistor is suggested for R1.
To prevent stray pickup, an optional 100pF capacitor is
suggested across R1 located close to LTC1622.

V

OUT

100pF

R2

R1

1622 F04

LTC1622

Figure 4. Setting Output Voltage

3

V

FB

Efficiency Considerations

The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what

3. I2R losses are predicted from the DC resistances of the
MOSFET, inductor and current shunt. In continuous
mode the average output current flows through L but
is “chopped” between the P-channel MOSFET in series
with R
plus R

and the output diode. The MOSFET R

SENSE

multiplied by duty cycle can be summed

SENSE

DS(ON)

with the resistance of the inductor to obtain I2R losses.

4. The output diode is a major source of power loss at
high currents and gets worse at high input voltages.
The diode loss is calculated by multiplying the forward
voltage drop times the diode duty cycle multiplied by
the load current. For example, assuming a duty cycle of
50% with a Schottky diode forward voltage drop of

0.4V, the loss increases from 0.5% to 8% as the load
current increases from 0.5A to 2A.

5. Transition losses apply to the external MOSFET and
increase with higher operating frequencies and input
voltages. Transition losses can be estimated from:

10

LTC1622

U

WUU

APPLICATIONS INFORMATION

Transition Loss = 3(VIN)2I

O(MAX)CRSS

Other losses including CIN and C
losses, and inductor core losses, generally account for
less than 2% total additional loss.

Run/Soft-Start Function

The RUN/SS pin is a dual purpose pin that provides the
soft-start function and a means to shut down the LTC1622.
Soft-start reduces input surge current from VIN by gradu­ally increasing the internal current limit. Power supply
sequencing can also be accomplished using this pin.

An internal 2.5µA current source charges up an external
capacitor CSS. When the voltage on the RUN/SS reaches

0.7V the LTC1622 begins operating. As the voltage on
RUN/SS continues to ramp from 0.7V to 1.8V, the internal
current limit is also ramped at a proportional linear rate.
The current limit begins near 0A (at V
ends at 0.1/R

SENSE

(V

≥ 1.8V). The output current

RUN/SS

thus ramps up slowly, reducing the starting surge current
required from the input power supply. If the RUN/SS has
been pulled all the way to ground, there will be a delay
before the current limit starts increasing and is given by:

(f)

ESR dissipative

OUT

= 0.7V) and

RUN/SS

V

D

FB

1622 F05

OUT

LTC1622

V

I

TH

Figure 5. Foldback Current Limiting

R2

FB

+

R1

Design Example

Assume the LTC1622 is used in a single lithium-ion
battery-powered cellular phone application. The VIN will be
operating from a maximum of 4.2V down to a minimum of

2.7V. Load current requirement is a maximum of 1.5A but

most of the time it will be on standby mode, requiring only
2mA. Efficiency at both low and high load current is
important. Output voltage is 2.5V.

In the above application, Burst Mode operation is enabled
by connecting Pin 5 to VIN.

VV

+

Maximum

Duty Cycle =

OUT D

VV

IN MIN D

+

()

%93

=

t

= 2.8 • 105 • CSS in seconds

DELAY

Pulling the RUN/SS pin below 0.4V puts the LTC1622 into
a low quiescent current shutdown (IQ < 15µA).

Foldback Current Limiting

As described in the Output Diode Selection, the worst­case dissipation occurs with a short-circuited output
when the diode conducts the current limit value almost
continuously. To prevent excessive heating in the diode,
foldback current limiting can be added to reduce the
current in proportion to the severity of the fault.

Foldback current limiting is implemented by adding diode
DFB (1N4148 or equivalent) between the output and the I
pin as shown in Figure 5. In a hard short (V

= 0V), the

OUT

TH

current will be reduced to approximately 50% of the
maximum output current.

From Figure 2, SF = 57%.
Use the curve of Figure 2 since the operating frequency is

the free running frequency of the LTC1622.

R

SENSE

=

SF

15 100

I

()( )( )=()( )

OUT

057

.

15 1 5

.

A

=

0 0253

. Ω

In the application, a 0.025Ω resistor is used. For the
inductor, the required value is:

L

MIN

=

42 25

kHz

550

..

−


0 036

.



0 025

.





25 03

..





42 03

..









+
+





=

133

.µ

H

In the application, a 3.9µH inductor is used to reduce
inductor ripple current and thus, output voltage ripple.

For the selection of the external MOSFET, the R

DS(ON)

must be guaranteed at 2.5V since the LTC1622 has to work

11

LTC1622

U

WUU

APPLICATIONS INFORMATION

down to 2.7V. Let’s assume that the MOSFET dissipation
is to be limited to PP = 250mW and its thermal resistance
is 50°C/W. Hence the junction temperature at TA = 25°C
will be 37.5°C and δp = 0.005 (37.5 – 25) = 0.0625. The
required R

R

DS ON

()

The P-channel MOSFET requirement can be met by an
Si6433DQ.

The requirement for the Schottky diode is the most strin­gent when V
R

resistor, the short-circuit current through the

SENSE

Schottky is 0.1/0.025 = 4A. An MBRS340T3 Schottky
diode is chosen. With 4A flowing through, the diode is
rated with a forward voltage of 0.4V. Therefore, the worst­case power dissipated by the diode is 1.6W. The addition
of DFB (Figure 5) will reduce the diode dissipation to
approximately 0.8W.

The input capacitor requires an RMS current rating of at
least 0.75A at temperature, and C
of 0.1Ω for optimum efficiency.

PC Board Layout Checklist

When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1622. These items are illustrated graphically in the

is then given by:

DS(ON)

P

P

DC I p

OUT

2

+

()

OUT

1

()

= 0V, i.e., short circuit. With a 0.025Ω

.≅

=

011δΩ

will require an ESR

OUT

layout diagram in Figure 6. Check the following in your
layout:

1. Is the Schottky diode closely connected between ground
at (–) lead of CIN and drain of the external MOSFET?

2. Does the (+) plate of CIN connect to the sense resistor
as closely as possible? This capacitor provides AC
current to the MOSFET.

3. Is the input decoupling capacitor (0.1µF) connected
closely between VIN (Pin 8) and ground (Pin 6)?

4. Connect the end of R

as close to VIN (Pin 8) as

SENSE

possible. The VIN pin is the SENSE+ of the current
comparator.

5. Is the trace from the SENSE– (Pin 1) to the Sense
resistor kept short? Does the trace connect close to
R

SENSE

?

6. Keep the switching node, SW, away from sensitive
small signal nodes.

7. Does the VFB pin connect directly to the feedback
resistors? The resistive divider R1 and R2 must be
connected between the (+) plate of C

and signal

OUT

ground. Optional capacitor C1 should be located as
close as possible to the LTC1622.

R1 and R2 should be located as close as possible to the
LTC1622. R2 should connect to the output as close to
the load as practicable.

12

+

R

SENSE

1

–

SENSE

2

I

TH

LTC1622

3

V

4

RUN/
SS

C

SS

FB

QUIET SGND

R

ITH

C

ITH

R1

BOLD LINES INDICATE HIGH CURRENT PATHS

C1

R2

Figure 6. LTC1622 Layout Diagram (See PC Board Layout Checklist)

V

PDRV

GND

SYNC/

MODE

8

IN

7

6

5

0.1µF

M1

SW

V

IN

C

IN

L1

+

1622 F06

V

OUT

C

OUT

U

TYPICAL APPLICATIONS

LTC1622 1.8V/1.5A Regulator with Burst Mode Operation Disabled

LTC1622

V

IN

2.5V TO

8.5V

R1
10K

C3
220pF

C4

560pF

C1: AVX TPSD476M016R0150
C2: AVX TPSD227M006R0100
L1: MURATA LQN6C3R3

R1
10k

C1

1

2

3

4

SENSE

I

TH

V

FB

RUN/
SS

–

LTC1622

47µF

8

V

IN

7

PDRV

6

GND

5

SYNC/

MODE

R2: DALE WSL-1206 0.025Ω
U1: INTERNATIONAL RECTIFIER

16V

+

R2

0.025Ω

1

2

3

4

FETKY

U1

TM

IRF7422D2

8

7

6

5

LTC1622 2.5V/2A Regulator with Burst Mode Operation Enabled

+

C1
47µF
16V
× 2

C2

+

150µF
6V
× 2

C3
220pF

1

2

3

4

C4
560pF

SENSE

I

TH

V

FB

RUN/
SS

–

LTC1622

V

PDRV

GND

SYNC/

MODE

8

IN

7

D1

6

5

R2

0.02Ω

M1

L1

4.7µH

3.3µH

R3
158k

R4
75k

L1

R3

93.1k

R4
75k

V

IN

3.3V TO

8.5V

V

OUT

2.5V
2A

+

C2
220µF
6V

1622 TA01

V

OUT

1.8V

1.5A

C1: AVX TPSD476M016R0150
C2: SANYO POSCAP 6TPA47M
D1: MOTOROLA MBR320T3

L1: COILCRAFT D03316-472
M1: SILICONIX Si3443DV
R2: DALE WSL-2010 0.02Ω

FETKY

is a trademark of International Rectifier Corporation.

1622 TA02

13

LTC1622

TYPICAL APPLICATIONS

LTC1622 2.5V/3A Regulator with External Frequency Synchronization

1

2

R1
10k

C3
220pF

3

4

C4
560pF

U

SENSE

I

TH

V

FB

RUN/
SS

–

LTC1622

V

PDRV

GND

SYNC/
MODE

V

IN

R3
158k

R4
75k

3.3V TO

8.5V

V

OUT

2.5V
3A

8

IN

7

6

5

650kHz

1.5V

P-P

R2

0.01Ω

M1

D1

L1

4.7µH

47µF
16V
× 2

C2

+

100µF

6.3V
× 2

C1

+

C1: AVX TPSD476M016R0150
C2: AVX TPSD107M010R0065
D1: MOTOROLA MBR320T3

Zeta Converter with Foldback Current Limit

D2
1N4818

R1
47k

C3
470pF

0.1µF

C1: AVX TPSD476M016R0150
C2: AVX TPSD107M010R0080
D1: MOTOROLA MBRS320T3
L1A, L1B: BH ELECTRONICS BH511-1012
R2: DALE WSL-1206 0.04Ω

1

–

SENSE

2

I

TH

LTC1622

3

V

FB

4

RUN/
SS

C4

V

PDRV

GND

SYNC/
MODE

IN

L1B

8

7

6

5

L1: COILCRAFT D03316-472
M1: SILICONIX Si3443DV
R2: DALE WSL-2512 0.01Ω

R2

0.04Ω

Si3441DV

47µF

L1A

6.2µH

3
TOP VIEW

4

16V

2

L1A

1

•

1622 TA03

V

IN

R3
232k

R4
75k

2.5V TO

8.5V

V

OUT

3.3V

1622 TA04

C1

+

47µF
16V
× 2

L1B

6.2µH

+

D1

+

C2
100µF
10V

VINI

OUT(MAX)

(V) (A)

2.5 0.45

3.3 0.70

5.0 0.95

6.0 1.00

8.4 1.05

14

PACKAGE DESCRIPTION

U

Dimensions in inches (millimeters) unless otherwise noted.

MS8 Package

8-Lead Plastic MSOP

(LTC DWG # 05-08-1660)

0.118 ± 0.004*
(3.00 ± 0.102)

8

7

6

5

LTC1622

0.193 ± 0.006
(4.90 ± 0.15)

12

0.040

± 0.006

SEATING

PLANE

(1.02 ± 0.15)

0.012
(0.30)

0.0256

REF

(0.65)

BSC

0.007
(0.18)

0.021

± 0.006

(0.53 ± 0.015)

* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,

PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

° – 6° TYP

0

0.118 ± 0.004**

4

3

0.034 ± 0.004
(0.86 ± 0.102)

(3.00 ± 0.102)

0.006 ± 0.004

(0.15 ± 0.102)

MSOP (MS8) 1098

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*
(4.801 – 5.004)

7

8

5

6

0.228 – 0.244

(5.791 – 6.197)

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

× 45°

0°– 8° TYP

0.016 – 0.050

(0.406 – 1.270)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.150 – 0.157**
(3.810 – 3.988)

1

3

2

4

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

BSC

SO8 1298

15

LTC1622

LOAD CURRENT (mA)

1

EFFICIENCY (%)

100

90

80

70

60

50

10 100 1000

1622 TA05b

V

OUT

= 3.3V

R

SENSE

= 0.025Ω

VIN = 3.5V

VIN = 4.2V

VIN = 6V

TYPICAL APPLICATION

U

Small Footprint 3.3V/1A Regulator

1

–

SENSE

2

I

TH

3

V

R1
10k

C3
220pF

C1: MURATA CERAMIC GRM235Y5V106Z
C2: SANYO POSCAP 6TPA47M
D1: MOTOROLA MBRS120LT3

C3
470pF

R1
33k

C5

150pF

4

C4
560pF

RUN/
SS

1

2

3

4

C4

0.1µF

FB

SENSE

I

TH

V

RUN/
SS

LTC1622

LTC1622

FB

–

V

PDRV

GND

SYNC/
MODE

PDRV

SYNC/
MODE

GND

R2

8

IN

7

6

5

D1

L1: COILCRAFT D01608C-222
M1: SILICONIX Si3443DY
R2: DALE WSL-2010 0.025Ω

0.025Ω

M1

L1

2.2µH

47µF

C2

6V

Boost Converter 3.3V/2.5A

8

V

IN

7

6

5

+

Si6801DQ

C1
100µF
10V

+

+

C6

0.1µF

M1

C1
10µF
16V
CERAMIC

R3
232k

R4
75k

R2

0.015Ω

L1

4.6µH

D1

V

IN

3.3V TO

8.5V

V

OUT

3.3V
1A

1622 TA05

R3
105k

R4
20k

Efficiency vs Load Current

Efficiency vs Load Current With LTC1622

Configured as Boost Converter

100

V

= 5V

V

IN

3.3V

V

OUT

5V

2.5A

C2

+

220µF
10V
×2

OUT

R

SENSE

90

80

70

EFFICIENCY (%)

60

= 0.015Ω

VIN = 3.3V

C1, C2: SANYO POSCAP TPB SERIES
D1: MOTOROLA MBRD835L
L1: SUMIDA CEP123-4R6

M1: SILICONIX Si3442DV
R2: DALE WS-L2512 0.015Ω

1622 TA06a

50

0.001

0.01 0.1 1
LOAD CURRENT (mA)

1622 TA06b

RELATED PARTS

PART NUMBER DESCRIPTION COMMENTS

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LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 8-Pin N-Channel Drive, 3.5V ≤ VIN ≤ 36V
LTC1626 Low Voltage, High Efficiency Step-Down DC/DC Converter Monolithic, Constant Off-Time, 2.5V ≤ VIN ≤ 6V
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LTC1628 Dual High Efficiency 2-Phase Step-Down Controller Antiphase Drive, 3.5V ≤ VIN ≤ 36V, Protection
LTC1772 SOT-23 Current Mode Step-Down Controller 6-Lead SOT-23, 2.5V ≤ VIN ≤ 9.8V, 550kHz
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Linear T echnology Corporation

16

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com

= 10µA

OUT

1622f LT/TP 0100 4K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1998

IN