Datasheet LTC1474 Datasheet (LINEAR TECHNOLOGY)

LOAD CURRENT (mA)

EFFICIENCY (%)

100

90

80

70

60

50

0.03 3 300

1474/75 TA01

0.3 30

VIN = 5V

VIN = 10V

VIN = 15V

L = 100µH

V

OUT

= 3.3V

R

SENSE

= 0Ω

FEATURES

■

High Efficiency: Over 92% Possible

■

Very Low Standby Current: 10µA Typ

■

Available in Space Saving 8-Lead MSOP Package

■

Internal 1.4Ω Power Switch (VIN = 10V)

■

Wide VIN Range: 3V to 18V Operation

■

Very Low Dropout Operation: 100% Duty Cycle

■

Low-Battery Detector Functional During Shutdown

■

Programmable Current Limit with Optional
Current Sense Resistor (10mA to 400mA Typ)

■

Short-Circuit Protection

■

Few External Components Required

■

Active Low Micropower Shutdown: IQ = 6µA Typ

■

Pushbutton On/Off (LTC1475 Only)

■

3.3V, 5V and Adjustable Output Versions

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APPLICATIONS

■

Cellular Telephones and Wireless Modems

■

4mA to 20mA Current Loop Step-Down Converter

■

Portable Instruments

■

Battery-Operated Digital Devices

■

Battery Chargers

■

Inverting Converters

■

Intrinsic Safety Applications

LTC1474/LTC1475

Low Quiescent Current

High Efficiency Step-Down

Converters

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DESCRIPTION

The LTC®1474/LTC1475 series are high efficiency step­down converters with internal P-channel MOSFET power
switches that draw only 10µA typical DC supply current at
no load while maintaining output voltage. The LTC1474
uses logic-controlled shutdown while the LTC1475 fea­tures pushbutton on/off.

The low supply current coupled with Burst ModeTM opera­tion enables the LTC1474/LTC1475 to maintain high effi­ciency over a wide range of loads. These features, along
with their capability of 100% duty cycle for low dropout
and wide input supply range, make the LTC1474/LTC1475
ideal for moderate current (up to 300mA) battery-powered
applications.

The peak switch current is user-programmable with an
optional sense resistor (defaults to 325mA minimum if not
used) providing a simple means for optimizing the design
for lower current applications. The peak current control
also provides short-circuit protection and excellent start­up behavior. A low-battery detector that remains functional
in shutdown is provided .

The LTC1474/LTC1475 series availability in 8-lead MSOP
and SO packages and need for few additional components
provide for a minimum area solution.

, LTC and LT are registered trademarks of Linear Technology Corporation.

Burst Mode is a trademark of Linear Technology Corporation.

TYPICAL APPLICATION

V

IN

4V TO 18V

+

10µF
25V

LOW BATTERY IN

RUN SHDN

L1 = SUMIDA CDRH74-101

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LOW BATTERY OUT

0.1µF

6

SENSE

3

LBI

100k

8

RUN

Figure 1. High Efficiency Step-Down Converter

7

V

IN

LTC1474-3.3

GND

4

V

LBO

SW

1

FB

2

5

L1
100µH

D1
MBR0530

+

V

OUT

3.3V AT 250mA

100µF

6.3V

1474/75 F01

LTC1474 Efficiency

1

LTC1474/LTC1475

1
2
3
4

V

OUT/VFB

LBO

LBI

GND

8
7
6
5

RUN
V

IN

SENSE
SW

TOP VIEW

MS8 PACKAGE

8-LEAD PLASTIC MSOP

WW

W

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ABSOLUTE MAXIMUM RATINGS

Input Supply Voltage (VIN).........................–0.3V to 20V

Switch Current (SW, SENSE).............................. 750mA

Switch Voltage (SW) ............. (VIN – 20V) to (VIN + 0.3V)

VFB (Adjustable Versions) ..........................–0.3V to 12V

V

(Fixed Versions)................................ –0.3V to 20V

OUT

LBI, LBO ....................................................–0.3V to 20V

RUN, SENSE ..................................–0.3V to (VIN + 0.3V)

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PACKAGE/ORDER INFORMATION

TOP VIEW

8

V

1

OUT/VFB

LBO

2

LBI/OFF

3

GND

4

MS8 PACKAGE

8-LEAD PLASTIC MSOP

T

= 125°C, θJA = 150°C/W T

JMAX

= 125°C, θJA = 150°C/W

JMAX

7
6
5

ON
V

IN

SENSE
SW

Operating Ambient Temperature Range

Commercial ............................................ 0°C to 70°C

Industrial ............................................ –40°C to 85°C

Junction Temperature (Note 1)............................ 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec).................. 300°C

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TOP VIEW

V

1

OUT/VFB

LBO

2

3

LBI

4

GND

S8 PACKAGE

8-LEAD PLASTIC SO

= 125°C, θJA = 110°C/W T

JMAX

8

7

6

5

RUN

V

IN

SENSE

SW

V

OUT/VFB

LBI/OFF

LBO

GND

TOP VIEW

1

2

3

4

S8 PACKAGE

8-LEAD PLASTIC SO

= 125°C, θJA = 110°C/WT

JMAX

8

7

6

5

ON

V

IN

SENSE

SW

ORDER PART NUMBER ORDER PART NUMBER ORDER PART NUMBERORDER PART NUMBER

LTC1474CMS8
LTC1474CMS8-3.3
LTC1474CMS8-5

MS8 PART MARKING

LTBW
LTCR
LTCS

Consult factory for Military grade parts.

LTC1475CMS8
LTC1475CMS8-3.3
LTC1475CMS8-5

MS8 PART MARKING S8 PART MARKING

LTBK
LTCP
LTCQ

LTC1474CS8
LTC1474IS8
LTC1474CS8-3.3
LTC1474CS8-5
LTC1474IS8-3.3
LTC1474IS8-5

1474
1474I
147433
14745
474I33
1474I5

LTC1475CS8
LTC1475IS8
LTC1475CS8-3.3
LTC1475CS8-5

S8 PART MARKING

1475
1475I
147533
14755

2

LTC1474/LTC1475

ELECTRICAL CHARACTERISTICS

TA = 25°C, VIN = 10V, V

= open, R

RUN

= 0, unless otherwise noted.

SENSE

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

V

FB

Feedback Voltage I

= 50mA ● 1.205 1.230 1.255 V

LOAD

LTC1474/LTC1475

V

OUT

I

FB

Regulated Output Voltage I

LTC1474-3.3/LTC1475-3.3
LTC1474-5/LTC1475-5

LOAD

= 50mA

● 3.234 3.300 3.366 V

● 4.900 5.000 5.100 V

Feedback Current ● 030 nA

LTC1474/LTC1475 Only

I

SUPPLY

∆V

OUT

I

Q

R

ON

I

PEAK

V

SENSE

V

HYST

t

OFF

V

LBI, TRIP

V

RUN

V

LBI, OFF

I

LBO, SINK

I

RUN, SOURCE

I

SW, LEAK

I

LBI, LEAK

I

LBO, LEAK

The ● denotes specifications which apply over the full operating
temperature range.

Note 1: T
dissipation P

LTC1474CS8/LTC1475CS8: TJ = TA + (PD • 110°C/W)
LTC1474CMS8/LTC1475CMS8: T

No Load Supply Current (Note 3) I
Output Voltage Line Regulation VIN = 7V to 12V, I
Output Voltage Load Regulation I
Output Ripple I

= 0 (Figure 1 Circuit) 10 µA

LOAD

= 50mA 5 20 mV

LOAD

= 0mA to 50mA 2 15 mV

LOAD

= 10mA 50 mV

LOAD

P-P

Input DC Supply Current (Note 2) (Exclusive of Driver Gate Charge Current)

Active Mode (Switch On) V
Sleep Mode (Note 3) V
Shutdown V

= 3V to 18V 100 175 µA

IN

= 3V to 18V 9 15 µA

IN

= 3V to 18V, V

IN

= 0V 6 12 µA

RUN

Switch Resistance ISW = 100mA 1.4 1.6 Ω
Current Comp Max Current Trip Threshold R

= 0Ω 325 400 mA

SENSE

= 1.1Ω 70 76 85 mA

R

SENSE

Current Comp Sense Voltage Trip Threshold ● 90 100 110 mV
Voltage Comparator Hysteresis 5mV
Switch Off-Time V

at Regulated Value 3.5 4.75 6.0 µs

OUT

V

= 0V 65 µs

OUT

Low Battery Comparator Threshold ● 1.16 1.23 1.27 V
Run/ON Pin Threshold 0.4 0.7 1.0 V
OFF Pin Threshold (LTC1475 Only) 0.4 0.7 1.0 V
Sink Current into Pin 2 V
Source Current from Pin 8 V
Switch Leakage Current VIN = 18V, VSW = 0V, V
Leakage Current into Pin 3 V
Leakage Current into Pin 2 V

= 0V, V

LBI

= 0V 0.4 0.8 1.2 µA

RUN

= 18V, VIN = 18V 0 0.1 µA

LBI

= 2V, V

LBI

= 0.4V 0.45 0.70 mA

LBO

= 0V 0.015 1 µA

RUN

= 5V 0 0.5 µA

LBO

Note 2: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency. See Applications Information.

is calculated from the ambient temperature TA and power

J

according to the following formulas:

D

Note 3: No load supply current consists of sleep mode DC current (9µA
typical) plus a small switching component (about 1µA for Figure 1 circuit)
necessary to overcome Schottky diode and feedback resistor leakage.

= TA + (PD • 150°C/W)

J

3

LTC1474/LTC1475

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TYPICAL PERFORMANCE CHARACTERISTICS

Efficiency vs Input Voltage Line Regulation

100

95

90

85

EFFICIENCY (%)

80

75

70

0

I

LOAD

I

LOAD

4

8

INPUT VOLTAGE (V)

= 25mA

Current Trip Threshold vs
Temperature

500

VIN = 10V

400

300

200

100

CURRENT TRIP THRESHOLD (mA)

FIGURE 1 CIRCUIT
L: CDRH73-101

I

= 200mA

LOAD

= 1mA

12

1474/75 G01

R

= 0Ω

SENSE

R

= 1.1Ω

SENSE

16

40

FIGURE 1 CIRCUIT
I

LOAD

30

20

(mV)

10

OUT

∆V

0

–10

–20

0

Switch Resistance vs
Input Voltage

5

4

3

(Ω)

DS(ON)

2

R

1

= 100mA

4

8

INPUT VOLTAGE (V)

R

SENSE

T = 70°C

T = 25°C

R

12

= 0Ω

SENSE

= 0.33Ω

16

1474/75 G02

Load Regulation

40

FIGURE 1 CIRCUIT

30

20

10

(mV)

OUT

0

∆V

–10

–20

–30

0

50

150 200 300

100

LOAD CURRENT (mA)

Supply Current in Shutdown

10.0

7.5

5.0

SUPPLY CURRENT (µA)

2.5

VIN = 15V

VIN = 10V

VIN = 5V

250

1474/75 G03

0

0

20

TEMPERATURE (°C)

Switch Leakage Current vs
Temperature

1.0
VIN = 18V

0.8

0.6

0.4

LEAKAGE CURRENT (µA)

0.2

0

20

0

40 60 80 100

TEMPERATURE (°C)

40

60

80

1474/75 G04

0

0

5

10

INPUT VOLTAGE (V)

15

20

1474/75 G05

VIN DC Supply Current

120

ACTIVE MODE

SLEEP MODE

4

8

INPUT VOLTAGE (V)

16 20

12

1474/75 G08

1474/75 G07

100

80

60

40

SUPPLY CURRENT (µA)

20

0

0

0

0

5

10

INPUT VOLTAGE (V)

Off-Time vs Output Voltage

80

60

40

OFF-TIME (µs)

20

0

0

% OF REGULATED OUTPUT VOLTAGE (%)

40

20

15

VIN = 10V

60 80

20

1474/75 G06

100

1474/75 G09

4

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PIN FUNCTIONS

LTC1474/LTC1475

V

OUT/VFB

(Pin 1): Feedback of Output Voltage. In the fixed

versions, an internal resistive divider divides the output
voltage down for comparison to the internal 1.23V refer­ence. In the adjustable versions, this divider must be
implemented externally.

LBO (Pin 2): Open Drain Output of the Low Battery
Comparator. This pin will sink current when Pin 3 is below

1.23V.
LBI/OFF (Pin 3): Input to Low Battery Comparator. This

input is compared to the internal 1.23V reference. For the
LTC1475, a momentary ground on this pin puts regulator
in shutdown mode.

GND (Pin 4): Ground Pin.

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FUNCTIONAL DIAGRA

LBI/OFF

100mV

–

+

–

V

+

1.23V

1.23V

REFERENCE

LTC1474: RUN

LTC1475: ON

LBO

1µA

×

8

1-SHOT

TRIGGER OUT

STRETCH

2

CONNECTION NOT PRESENT IN LTC1474 SERIES

×

CONNECTION PRESENT IN LTC1474 SERIES ONLY

WAKEUP

LB

4.75µs

+

–

3

LTC1474: LBI
LTC1475: LBI/OFF

READY

ON

ON

C

SW (Pin 5): Drain of Internal PMOS Power Switch. Cath­ode of Schottky diode must be closely connected to this
pin.

SENSE (Pin 6): Current Sense Input for Monitoring Switch
Current and Source of Internal PMOS Power Switch.
Maximum switch current is programmed with a resistor
between SENSE and VIN pins.

V

(Pin 7): Main Supply Pin.

IN

RUN/ON (Pin 8): On LTC1474, voltage level on this pin

controls shutdown/run mode (ground = shutdown, open/
high = run). On LTC1475, a momentary ground on this pin
puts regulator in run mode. A 100k series resistor must be
used between Pin 8 and the switch or control voltage.

V

IN

7

R

SENSE

(OPTIONAL)

V

CC

6

1M

20×

3M

SENSE

SW

5

V

OUT/VFB

1

OUTPUT DIVIDER IS
IMPLEMENTED EXTERNALLY IN
ADJUSTABLE VERSIONS

5Ω

1×

(5V VERSION)

1.68M

(3.3V VERSION)

GND

4

V

IN

+

V

OUT

+

1474/75 FD

5

LTC1474/LTC1475

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OPERATIO

(Refer to Functional Diagram)

The LTC1474/LTC1475 are step-down converters with
internal power switches that use Burst Mode operation to
keep the output capacitor charged to the proper output
voltage while minimizing the quiescent current. Burst
Mode operation functions by using short “burst” cycles to
ramp the inductor current through the internal power
switch and external Schottky diode, followed by a sleep
cycle where the power switch is off and the load current is
supplied by the output capacitor. During sleep mode, the
LTC1474/LTC1475 draw only 9µA typical supply current.
At light loads, the burst cycles are a small percentage of
the total cycle time; thus the average supply current is very
low, greatly enhancing efficiency.

Burst Mode Operation

At the beginning of the burst cycle, the switch is turned on
and the inductor current ramps up. At this time, the internal
current comparator is also turned on to monitor the switch
current by measuring the voltage across the internal and
optional external current sense resistors. When this volt­age reaches 100mV, the current comparator trips and
pulses the 1-shot timer to start a 4.75µs off-time during
which the switch is turned off and the inductor current
ramps down. At the end of the off-time, if the output
voltage is less than the voltage comparator threshold, the
switch is turned back on and another cycle commences. To
minimize supply current, the current comparator is turned
on only during the switch-on period when it is needed to
monitor switch current. Likewise, the 1-shot timer will only
be on during the 4.75µs off-time period.

The average inductor current during a burst cycle will
normally be greater than the load current, and thus the
output voltage will slowly increase until the internal volt­age comparator trips. At this time, the LTC1474/LTC1475
go into sleep mode, during which the power switch is off
and only the minimum required circuitry is left on: the
voltage comparator, reference and low battery compara­tor. During sleep mode, with the output capacitor supply­ing the load current, the VFB voltage will slowly decrease
until it reaches the lower threshold of the voltage com­parator (about 5mV below the upper threshold). The
voltage comparator then trips again, signaling the LTC1474/
LTC1475 to turn on the circuitry necessary to begin a new
burst cycle.

Peak Inductor Current Programming

The current comparator provides a means for program­ming the maximum inductor/switch current for each switch
cycle. The 1X sense MOSFET, a portion of the main power
MOSFET, is used to divert a sample (about 5%) of the
switch current through the internal 5Ω sense resistor. The
current comparator monitors the voltage drop across the
series combination of the internal and external sense
resistors and trips when the voltage exceeds 100mV. If the
external sense resistor is not used (Pins 6 and 7 shorted),
the current threshold defaults to the 400mA maximum due
to the internal sense resistor.

Off-Time

The off-time duration is 4.75µs when the feedback voltage
is close to the reference; however, as the feedback voltage
drops, the off-time lengthens and reaches a maximum
value of about 65µs when this voltage is zero. This ensures
that the inductor current has enough time to decay when
the reverse voltage across the inductor is low such as
during short circuit.

Shutdown Mode

Both LTC1474 and LTC1475 provide a shutdown mode
that turns off the power switch and all circuitry except for
the low battery comparator and 1.23V reference, further
reducing DC supply current to 6µA typical. The LTC1474’s
run/shutdown mode is controlled by a voltage level at the
RUN pin (ground = shutdown, open/high = run). The
LTC1475’s run/shutdown mode, on the other hand, is
controlled by an internal S/R flip-flop to provide pushbutton
on/off control. The flip-flop is set (run mode) by a momen­tary ground at the ON pin and reset (shutdown mode) by
a momentary ground at the LBI/OFF pin.

Low Battery Comparator

The low battery comparator compares the voltage on the
LBI pin to the internal reference and has an open drain
N-channel MOSFET at its output. If LBI is above the
reference, the output FET is off and the LBO output is high
impedance. If LBI is below the reference, the output FET is
on and sinks current. The comparator is still active in
shutdown.

6

LTC1474/LTC1475

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APPLICATIONS INFORMATION

The basic LTC1474/LTC1475 application circuit is shown
in Figure 1, a high efficiency step-down converter. External
component selection is driven by the load requirement and
begins with the selection of R
known, L can be chosen. Finally D1, CIN and C
selected.

R

The current sense resistor (R

SENSE

Selection

SENSE

program the maximum inductor/switch current to opti­mize the inductor size for the maximum load. The LTC1474/
LTC1475 current comparator has a maximum threshold of
100mV/(R
current I

MAX

+ 0.25). The maximum average output

SENSE

is equal to this peak value less half the peak-

to-peak ripple current ∆IL.
Allowing a margin for variations in the LTC1474/LTC1475

and external components, the required R
calculated from Figure 2 and the following equation:

R

for 10mA < I

For I

= (0.067/I

SENSE

< 200mA.

MAX

5

4

FOR LOWEST NOISE

3

(Ω)

SENSE

2

R

1

0

0

above 200mA, R

MAX

) – 0.25 (1)

MAX

FOR BEST EFFICIENCY

100 150 200

50

MAXIMUM OUTPUT CURRENT (mA)

Figure 2. R

SENSE

is set to zero by shorting

SENSE

Pins 6 and 7 to provide the maximum peak current of
400mA (limited by the fixed internal sense resistor). This
400mA default peak current can be used for lower I
desired to eliminate the need for the sense resistor and
associated decoupling capacitor. However, for optimal
performance, the peak inductor current should be set to no
more than what is needed to meet the load current require-

. Once R

SENSE

SENSE

OUT

) allows the user to

can be

SENSE

250 300

1474/75 F02

Selection

are

MAX

is

if

ments. Lower peak currents have the advantage of lower
output ripple (∆V

OUT

= I

• ESR), lower noise, and less

PEAK

stress on alkaline batteries and other circuit components.
Also, lower peak currents allow the use of inductors with
smaller physical size.

Peak currents as low as 10mA can be programmed with
the appropriate sense resistor. Increasing R
10Ω, however, gives no further reduction of I

For R

values less than 1Ω, it is recommended that

SENSE

SENSE

PEAK

above

.

the user parallel standard resistors (available in values ≥
1Ω) instead of using a special low valued shunt resistor.
Although a single resisor could be used with the desired
value, these low valued shunt resistor types are much
more expensive and are currently not available in case
sizes smaller than 1206. Three or four 0603 size standard
resistors require about the same area as one 1206 size
current shunt resistor at a fraction of the cost.

At higher supply voltages and lower inductances, the peak
currents may be slightly higher due to current comparator
overshoot and can be predicted from the second term in
the following equation:

−

7





VV

−

()



IN OUT



(2)

L

I

=

PEAK

Note that R

25 10

.

01

.

025

.

R

+

SENSE

only sets the maximum inductor current

SENSE

()

+

peak. At lower dI/dt (lower input voltages and higher
inductances), the observed peak current at loads less than
I

may be less than this calculated peak value due to the

MAX

voltage comparator tripping before the current ramps up
high enough to trip the current comparator. This effect
improves efficiency at lower loads by keeping the I2R
losses down (see Efficiency Considerations section).

Inductor Value Selection

Once R

SENSE

and I

are known, the inductor value can

PEAK

be determined. The minimum inductance recommended
as a function of I



075.

MIN



≥



L



where t

= 4.75µs.

OFF

and I

PEAK

VVt

+

()

OUT D OFF

II

−

PEAK MAX

can be calculated from:

MAX







(3)

7

LTC1474/LTC1475

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APPLICATIONS INFORMATION

If the L
be used. Although the above equation provides the mini­mum, better performance (efficiency, line/load regulation,
noise) is usually gained with higher values. At higher
inductances, peak current and frequency decrease (im­proving efficiency) and inductor ripple current decreases
(improving noise and line/load regulation). For a given
inductor type, however, as inductance is increased, DC
resistance (DCR) increases, increasing copper losses,
and current rating decreases, both effects placing an
upper limit on the inductance. The recommended range of
inductances for small surface mount inductors as a func­tion of peak current is shown in Figure 3. The values in this
range are a good compromise between the trade-offs
discussed above. If space is not a premium, inductors with
larger cores can be used, which extends the recom­mended range of Figure 3 to larger values.

calculated is not practical, a larger I

MIN

1000

500

PEAK

should

section, increased inductance requires more turns of wire
and therefore copper losses will increase.

Ferrite and Kool Mµ designs have very low core loss and
are preferred at high switching frequencies, so design
goals can concentrate on copper loss and preventing
saturation. Ferrite core material saturates “hard,” which
means that inductance collapses abruptly when the peak
design current is exceeded. This results in an abrupt
increase in inductor current above I
increase in voltage ripple.

rate!

Coiltronics, Coilcraft, Dale and Sumida make high

Do not allow the core to satu-

and consequent

PEAK

performance inductors in small surface mount packages
with low loss ferrite and Kool Mµ cores and work well in
LTC1474/LTC1475 regulators.

Catch Diode Selection

The catch diode carries load current during the off-time.
The average diode current is therefore dependent on the
P-channel switch duty cycle. At high input voltages the
diode conducts most of the time. As VIN approaches V

OUT

the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short-circuited. Under this condition, the diode must
safely handle I

at close to 100% duty cycle.

PEAK

INDUCTOR VALUE (µH)

100

50

10

PEAK INDUCTOR CURRENT (mA)

Figure 3. Recommended Inductor Values

100 1000

1474/75 F03

Inductor Core Selection

Once the value of L is known, the type of inductor must be
selected. High efficiency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mµ® cores. Actual core loss is independent of core
size for a fixed inductor value, but is very dependent on
inductance selected. As inductance increases, core losses
go down. Unfortunately, as discussed in the previous

To maximize both low and high current efficiency, a fast
switching diode with low forward drop and low reverse
leakage should be used. Low reverse leakage current is
critical to maximize low current efficiency since the leak­age can potentially approach the magnitude of the LTC1474/
LTC1475 supply current. Low forward drop is critical for
high current efficiency since loss is proportional to for­ward drop. These are conflicting parameters (see Table 1),
but a good compromise is the MBR0530 0.5A Schottky
diode specified in the application circuits.

Table 1. Effect of Catch Diode on Performance

FORWARD NO LOAD

DIODE (D1) LEAKAGE DROP SUPPLY CURRENT EFFICIENCY*

BAS85 200nA 0.6V 9.7µA 77.9%
MBR0530 1µA 0.4V 10µA 83.3%
MBRS130 20µA 0.3V 16µA 84.6%
*Figure 1 circuit with VIN = 15V, I

Kool Mµ is a registered trademark of Magnetics, Inc.

OUT

= 0.1A

8

LTC1474/LTC1475

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APPLICATIONS INFORMATION

CIN and C

At higher load currents, when the inductor current is
continuous, the source current of the P-channel MOSFET
is a square wave of duty cycle V
voltage transients, a low ESR input capacitor sized for the
maximum RMS current must be used. The maximum
capacitor current is given by:

C

Required I =

IN

This formula has a maximum at VIN = 2V
I

= I

RMS

monly used for design because even significant deviations
do not offer much relief. Note that capacitor manufacturer’s
ripple current ratings are often based on 2000 hours of life.
This makes it advisable to further derate the capacitor, or
to choose a capacitor rated at a higher temperature than
required. Do not underspecify this component. An addi­tional 0.1µF ceramic capacitor is also required on VIN for
high frequency decoupling.

The selection of C
series resistance (ESR) to meet the output voltage ripple
and line regulation requirements. The output voltage ripple
during a burst cycle is dominated by the output capacitor
ESR and can be estimated from the following relation:

25mV < ∆V

where ∆IL ≤ I
voltage comparator hysteresis. Line regulation can also
vary with C
voltage range and high peak currents.

ESR is a direct function of the volume of the capacitor.
Manufacturers such as Nichicon, AVX and Sprague should
be considered for high performance capacitors. The
OS-CON semiconductor dielectric capacitor available from
SANYO has the lowest ESR for its size at a somewhat
higher price. Typically, once the ESR requirement is satis­fied, the capacitance is adequate for filtering. For lower
current applications with peak currents less than 50mA,
10µF ceramic capacitors provide adequate filtering and
are a good choice due to their small size and almost

Selection

OUT

OUT/VIN

VVV

I

RMS

/2. This simple worst-case condition is com-

OUT

is driven by the required effective

OUT

OUT, RIPPLE

and the lower limit of 25mV is due to the

PEAK

ESR in applications with a large input

OUT

OUT IN OUT

MAX

[]

= ∆IL • ESR

. To prevent large

12/

−

()

V

IN

, where

OUT

negligible ESR. AVX and Marcon are good sources for
these capacitors.

In surface mount applications multiple capacitors may
have to be paralleled to meet the ESR or RMS current
handling requirements of the application. Aluminum elec­trolytic and dry tantalum capacitors are both available in
surface mount configurations. In the case of tantalum, it is
critical that the capacitors are surge tested for use in
switching power supplies. An excellent choice is the AVX
TPS series of surface mount tantalums, available in case
heights ranging from 2mm to 4mm. Other capacitor types
include SANYO OS-CON, Nichicon PL series and Sprague
595D series. Consult the manufacturer for other specific
recommendations.

To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
worst-case ripple current in the output capacitor is given
by:

I

= I

RMS

Once the ESR requirement for C
RMS current rating generally far exceeds the I
requirement.

Efficiency Considerations

The efficiency of a switching regulator is equal to the
output power divided by the input power times 100%. It is
often useful to analyze individual losses to determine what
is limiting efficiency and which change would produce the
most improvement. Efficiency can be expressed as:

Efficiency = 100% – (L1 + L2 + L3 + ...)

where L1, L2, etc. are the individual losses as a percentage
of input power.

Although all dissipative elements in the circuit produce
losses, three main sources usually account for most of the
losses in LTC1474/LTC1475 circuits: VIN current, I2R
losses and catch diode losses.

1. The VIN current is due to two components: the DC bias
current and the internal P-channel switch gate charge
current. The DC bias current is 9µA at no load and
increases proportionally with load up to a constant
100µA during continuous mode. This bias current is so

PEAK

/2

has been met, the

OUT

RIPPLE(P-P)

9

LTC1474/LTC1475

V

R
R

TRIP

=+











123 1

4
3

.

U

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APPLICATIONS INFORMATION

small that this loss is negligible at loads above a
milliamp but at no load accounts for nearly all of the
loss. The second component, the gate charge current,
results from switching the gate capacitance of the
internal P-channel switch. Each time the gate is switched
from high to low to high again, a packet of charge dQ
moves from VIN to ground. The resulting dQ/dt is the
current out of VIN which is typically much larger than the
DC bias current. In continuous mode, I

GATECHG

where QP is the gate charge of the internal switch. Both
the DC bias and gate charge losses are proportional to
VIN and thus their effects will be more pronounced at
higher supply voltages.

2. I2R losses are predicted from the internal switch,
inductor and current sense resistor. At low supply
voltages where the switch on-resistance is higher and
the switch is on for longer periods due to higher duty
cycle, the switch losses will dominate. Keeping the peak
currents low with the appropriate R

SENSE

larger inductance helps minimize these switch losses.
At higher supply voltages, these losses are proportional
to load and result in the flat efficiency curves seen in
Figure 1.

3. The catch diode loss is due to the VDID loss as the diode
conducts current during the off-time and is more pro­nounced at high supply voltage where the on-time is
short. This loss is proportional to the forward drop.
However, as discussed in the Catch Diode section,
diodes with lower forward drops often have higher
leakage current, so although efficiency is improved, the
no load supply current will increase.

Adjustable Applications

For adjustable versions, the output voltage is programmed
with an external divider from V

to VFB (Pin 1) as shown

OUT

in Figure 4. The regulated voltage is determined by:

V

OUT



=1.23 1+





R2

R1






= fQ

and with

(4)

P

To minimize no-load supply current, resistor values in the
megohm range should be used. The increase in supply
current due to the feedback resistors can be calculated
from:

VIN






RRVV

∆=

I

V

OUT OUT

12










+



IN






A 10pF feedforward capacitor across R2 is necessary due
to the high impedances to prevent stray pickup and
improve stability.

V

OUT

R2

LTC1474
LTC1475

GND

4

Figure 4. LTC1474/LTC1475 Adjustable Configuration

1

V

FB

10pF

R1

1474/75 F04

Low Battery Comparator

The LTC1474/LTC1475 have an on-chip low battery com­parator that can be used to sense a low battery condition
when implemented as shown in Figure 5. The resistive
divider R3/R4 sets the comparator trip point as follows:

The divided down voltage at the LBI pin is compared to the
internal 1.23V reference. When V

< 1.23V, the LBO

LBI

output sinks current. The low battery comparator is active
all the time, even during shutdown mode.

V

IN

R4

LBI

R3

LTC1474/LTC1475

–

+

1.23V
REFERENCE

LBO

1474/75 F05

10

Figure 5. Low Battery Comparator

LTC1474/LTC1475

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APPLICATIONS INFORMATION

LTC1475 Pushbutton On/Off and
Microprocessor Interface

The LTC1475 provides pushbutton control of power on/off
for use with handheld products. A momentary ground on
the ON pin sets an internal S/R latch to run mode while a
momentary ground on the LBI/OFF pin resets the latch to
shutdown mode. See Figure 6 for a comparsion of on/off
operation of the LTC1474 and LTC1475 and Figure 7 for a
comparison of the circuit implementations.

In the LTC1475, the LBI/OFF pin has a dual function as
both the shutdown control pin and the low battery com­parator input. Since the “OFF” pushbutton is normally
open, it does not affect the normal operation of the low
battery comparator. In the unpressed state, the LBI/OFF
input is the divided down input voltage from the resistive
divider to the internal low battery comparator and will
normally be above or just below the trip threshold of

1.23V. When shutdown mode is desired, the LBI/OFF pin

is pulled below the 0.7V threshold to invoke shutdown.

the depressed switch state is detected by the microcon­troller through its input. The microcontroller then pulls the
LBI/OFF pin low with the connection to one of its ouputs.
With the LBI/OFF pin low, the LTC1475 powers down
turning the microcontroller off. Note that since the I/O pins
of most microcontrollers have a reversed bias diode
between input and supply, a blocking diode with less than
1µA leakage is necessary to prevent the powered down
microcontroller from pulling down on the ON pin.

Figure 19 in the Typical Applications section shows how to
use the low battery comparator to provide a low battery
lockout on the “ON” switch. The LBO output disconnects
the pushbutton from the ON pin when the comparator has
tripped, preventing the LTC1475 from attempting to start
up again until VIN is increased.

100k

V

ON

IN

LTC1475

LBI/OFF

RUN

100k

RUN

LTC1474

ON

LTC1474

LTC1475

RUN

MODE

ON

LBI/OFF

MODE

Figure 6. Comparison of LTC1474 and LTC1475
Run/Shutdown Operation

RUN SHUTDOWN RUN

ON OVERRIDES LBI/OFF

WHILE ON IS LOW

RUN

SHUTDOWN

RUN

1474/75 F06

The ON pin has precedence over the LBI/OFF pin. As seen
in Figure 6, if both pins are grounded simultaneously, run
mode wins.

Figure 18 in the Typical Applications section shows an
example for the use of the LTC1475 to control on/off of a
microcontroller with a single pushbutton. With both the
microcontroller and LTC1475 off, depressing the
pushbutton grounds the LTC1475 ON pin and starts up the
LTC1475 regulator which then powers up the microcon­troller. When the pushbutton is depressed a second time,

OFF

1474/75 F07

Figure 7. Simplified Implementation of
LTC1474 and LTC1475 On/Off

Absolute Maximum Ratings and Latchup Prevention

The absolute maximum ratings specify that SW (Pin 5) can
never exceed VIN (Pin 7) by more than 0.3V. Normally this
situation should never occur. It could, however, if the
output is held up while the supply is pulled down. A
condition where this could potentially occur is when a
battery is supplying power to an LTC1474 or LTC1475
regulator and also to one or more loads in parallel with the
the regulator’s VIN. If the battery is disconnected while the
LTC1474 or LTC1475 regulator is supplying a light load
and one of the parallel circuits is a heavy load, the input
capacitor of the LTC1474 or LTC1475 regulator could be
pulled down faster than the output capacitor, causing the
absolute maximum ratings to be exceeded. The result is
often a latchup which can be destructive if V

is reapplied.

IN

Battery disconnect is possible as a result of mechanical
stress, bad battery contacts or use of a lithium-ion battery

11

LTC1474/LTC1475

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APPLICATIONS INFORMATION

with a built-in internal disconnect. The user needs to
assess his/her application to determine whether this situ­ation could occur. If so, additional protection is necessary.

Prevention against latchup can be accomplished by sim­ply connecting a Schottky diode across the SW and V
pins as shown in Figure 8. The diode will normally be
reverse biased unless VIN is pulled below V
time the diode will clamp the (V

– VIN) potential to less

OUT

at which

OUT

than the 0.6V required for latchup. Note that a low leakage
Schottky should be used to minimize the effect on no-load
supply current. Schottky diodes such as MBR0530, BAS85
and BAT84 work well. Another more serious effect of the
protection diode leakage is that at no load with nothing to
provide a sink for this leakage current, the output voltage
can potentially float above the maximum allowable toler­ance. To prevent this from occuring, a resistor must be
connected between V

and ground with a value low

OUT

enough to sink the maximum possible leakage current.

LATCHUP
PROTECTION
SCHOTTKY

IN

where P is the power dissipated by the regulator and θ

JA

is the thermal resistance from the junction of the die to the
ambient temperature.

The junction temperature is given by:

TJ = TA + T

R

As an example consider the LTC1474/LTC1475 in dropout
at an input voltage of 3.5V, a load current of 300mA, and
an ambient temperature of 70°C. From the typical perfor­mance graph of switch resistance, the on-resistance of the
P-channel switch at 70°C is 3.5Ω. Therefore, power dissi­pated by the part is:

P = I2 • R

For the MSOP package, the θ

DS(ON)

= 0.315W

is 150°C/W. Thus the

JA

junction temperature of the regulator is:

TJ = 70°C + (0.315)(150) = 117°C

which is near the maximum junction temperature of 125oC.
Note that at higher supply voltages, the junction tempera­ture is lower due to reduced switch resistance.

V

SW

IN

LTC1474
LTC1475

Figure 8. Preventing Absolute Maximum
Ratings from Being Exceeded

V

OUT

+

1474/75 F08

Thermal Considerations

In the majority of the applications, the LTC1474/LTC1475
do not dissipate much heat due to their high efficiency.
However, in applications where the switching regulator is
running at high ambient temperature with low supply
voltage and high duty cycles, such as dropout with the
switch on continuously, the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated by the regulator
exceeds the maximum junction temperature of the part.
The temperature rise is given by:

TR = P • θ

JA

PC Board Layout Checklist

When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC1474/LTC1475. These items are also illustrated
graphically in the layout diagram of Figure 9. Check the
following in your layout:

1. Is the Schottky diode cathode

closely

connected to SW

(Pin 5)?

2. Is the 0.1µF input decoupling capacitor

closely

con­nected between VIN (Pin 7) and ground (Pin 4)? This
capacitor carries the high frequency peak currents.

3. When using adjustable version, is the resistive divider
closely connected to the (+) and (–) plates of C

OUT

with

a 10pF capacitor connected across R2?

4. Is the 1000pF decoupling capacitor for the current
sense resistor connected as close as possible to Pins 6
and 7? If no current sense resistor is used, Pins 6 and
7 should be shorted.

12

LTC1474/LTC1475

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APPLICATIONS INFORMATION

OUTPUT DIVIDER REQUIRED WITH
ADJUSTABLE VERSION ONLY

10pF

V

OUT

+

Figure 9. LTC1474/LTC1475 Layout Diagram (See Board Layout Checklist)

R2

R1

C

OUT

0.1µF

BOLD LINES INDICATE HIGH PATH CURRENTS

5. Are the signal and power grounds segregated? The
signal ground consists of the (–) plate of C

, Pin 4 of

OUT

the LTC1474/LTC1475 and the resistive divider. The
power ground consists of the Schottky diode anode,
the (–) plate of CIN and the 0.1µF decoupling capacitor.

6. Is a 100k resistor connected in series between RUN
(Pin 8) and the RUN control voltage? The resistor
should be as close as possible to Pin 8.

Design Example (Refer to R

and Inductor

SENSE

Selection)

As a design example, assume VIN = 10V, V
a maximum average output current I

MAX

= 3V, and

OUT

= 100mA. With
this information, we can easily calculate all the important
components:

From the equation (1),

R

= (0.067/0.1) – 0.25 = 0.42Ω

SENSE

Using the standard resistors (1Ω, 1Ω and 2Ω) in parallel
provides 0.4Ω without having to use a more expensive
low value current shunt type resistor (see R

SENSE

Selec-

tion section).
With R

= 0.4Ω, the peak inductor current I

SENSE

PEAK

calculated from (2), neglecting the second term, to be

LTC1474

1

V

FB

2

LBO

3

LBI

4

GND

C

IN

+

is

RUN

SENSE

V

IN

7

1000pF

6

5

SW

D1

V

IN

R

SENSE

1474/75 F09

L

100k

8

150mA. The minimum inductance is, therefore, from the
equation (3) and assuming VD = 0.4V,

s

µ

264

H

µ

=

L

07533 04 475

....

()()

015 01

MIN

=

+

−

..

From Figure 3, an inductance of 270µH is chosen from the
recommended region. The CDRH73-271 or CD54-271 is a
good choice for space limited applications.

For the feedback resistors, choose R1 = 1M to minimize
supply current. R2 can then be calculated from the equa­tion (4) to be:



V

R

OUT

2



123





RM

11143=−

•=..





For the catch diode, the MBR0530 will work well in this
application.

For the input and output capacitors, AVX 4.7µF and 100µF,
respectively, low ESR TPS series work well and meet the
RMS current requirement of 100mA/2 = 50mA. They are
available in small “C” case sizes with 0.15Ω ESR. The

0.15Ω output capacitor ESR will result in 25mV of output
voltage ripple.

Figure 10 shows the complete circuit for this example.

13

LTC1474/LTC1475

U

TYPICAL APPLICATIONS

V

IN

3.5V TO 18V

* SUMIDA CDRH73-271

** 3 PARALLEL STANDARD RESISTORS

PROVIDE LEAST EXPENSIVE SOLUTION
(SEE R

†

AVX TPSC475M035

††

AVX TPSC107M006

4mA TO 20mA

4mA TO 20mA

+

4.7µF
35V

SELECTION SECTION)

SENSE

+

IN

–

IN

†

††

D2
12V

†

MBR0530

* COILCRAFT DO1608-334

** MARCON THCS50E1E106Z,

AVX 1206ZG106Z

†

OPTIONAL RESISTOR FOR SENSING LOOP CURRENT BY A/D CONVERTER

††

MOTOROLA MMBZ5242BL

10pF

0.1µF

1

2

5

1.43M
1%

1M
1%

RUN

1000pF

100k

6

3

8

SENSE

LBI

RUN

2Ω**1Ω** 1Ω**

7

V

IN

LTC1474

GND

4

V

LBO

SW

FB

Figure 10. High Efficiency 3V/100mA Regulator (Design Example)

TO A/D

1µF
× 3

7.5M

1M

2Ω

RUN

100k

1000pF

6

SENSE

3

LBI

8

RUN

7

V

IN

LTC1474-3.3

GND

4

V

OUT

LBO

SW

1

2

5

L*
330µH

D1
MBR0530

L*
270µH

D1
MBR0530

+

10µF**

100µF

6.3V

V

3.3V
10mA

1474/75 F11

††

1474/75 F10

OUT

V

OUT

3V
100mA

14

Figure 11. High Efficiency 3.3V/10mA Output from 4mA to 20mA Loop

U

TYPICAL APPLICATIONS

LTC1474/LTC1475

V

IN

3.5V TO 6V

* COILTRONICS CTX200-4

** AVX TPSC226M016

†

AVX TPSC106M025

††

AVX TPSD226M025

V

3.5V TO 12V

IN

* COILTRONICS CTX100-4

** AVX TPSC106MO25

†

AVX TPSC336M010

MBR0530

10pF

+

22µF**
16V

RUN

100k

6

3

8

SENSE

LBI

RUN

V

LTC1474

GND

0.1µF

7

V

LBO

SW

1

FB

2

5

IN

4

L*
200µH

536k
1%

4.7M
1%

10µF

25V

+

†

L*
200µH

D1
MBR0530

+

+

††

22µF
25V

††

22µF
25V

V

IN

3.5 30mA

V

OUT

–12V
70mA

V

OUT

12V
70mA

(V) I

LOAD(MAX)

4 50mA
5 70mA
6 90mA

1474/75 F12

Figure 12. 5V to ±12V Regulator

+

10µF**
25V

RUN

100k

6

3

8

SENSE

LBI

RUN

7

V

IN

LTC1474-5

GND

4

V

LBO

OUT

SW

0.1µF

V

1

2

5

10µF**

L*
100µH

25V

+

L*
100µH

D1
MBR0530

+

OUT

5V
200mA AT V

†

33µF
10V

V

(V) I

IN

3.5 70mA
4 95mA
5 125mA
8 180mA

10 200mA
12 225mA

IN

LOAD(MAX)

1474/75 F13

= 10V

Figure 13. 5V Buck-Boost Converter

15

LTC1474/LTC1475

U

TYPICAL APPLICATIONS

V

ON/OFF

IN

+

10µF**

††

TP0610

10M

* SUMIDA CDRH74-101

** AVX TPSC106M025

†

AVX TPSC336M010

††

RUN: ON/OFF = 0, SHUTDOWN: 0N/OFF = V

25V

3.5V TO 12V

6

3

8

SENSE

LBI

RUN

7

V

IN

LTC1474-5

GND

4

IN

V

OUT

LBO

SW

0.1µF

1

2

5

L*
100µH

D1
MBR0530

+

33µF
10V

†

VIN (V) I

V

OUT

–5V
140mA AT V

3.5 100mA

12 240mA

LOAD(MAX)

5 140mA
8 190mA

1474/75 F14

= 5V

IN

V

IN

8V TO 18V

* SUMIDA CDRH73-101

** AVX TPSC475M035

†

AVX TPSD476M016

+

4.7µF**
35V

CHARGER

ON/OFF

Figure 14. Positive-to-Negative (–5V) Converter

10pF

4.69M

1M

100k

6

3

8

SENSE

LBI

RUN

7

V

IN

LTC1474

GND

4

V

LBO

SW

0.1µF

1

FB

2

5

Figure 15. 4-NiCd Battery Charger

L*
100µH

D1
MBR0530

MBR0530

+

47µF
16V

†

V

OUT

4-NiCd
200mA

1474/75 F15

16

U

TYPICAL APPLICATIONS

V

IN

4V TO 18V

5.7V TO 18V

+

††

†

4.7µF
35V

* SUMIDA CDRH73-101

†

AVX TPSC475M035
AVX TPSC107M006

Figure 16. High Efficiency 3.3V Regulator with Low Battery Lockout

V

IN

+

4.7µF**
35V

OFF

Figure 17. Pushbutton On/Off 5V/250mA Regulator

2.2M

1M

3.65M

1M

RUN

ON

100k

100k

6

3

8

6

3

8

SENSE

LBI/OFF

ON

SENSE

LTC1474-3.3

LBI

RUN

7

V

IN

LTC1475-5

GND

4

V

GND

LTC1474/LTC1475

7

IN

V

OUT

LBO

SW

4

1

V

OUT

2

LBO

5

SW

0.1µF

V

1

+

2

5

0.1µF

L*
100µH

D1
MBR0530

L*
100µH

D1
MBR0530

100µF

6.3V

+

33µF
10V

* SUMIDA CDRH73-101

** AVX TPSC475M035

†

AVX TPSC336M010

OUT

3.3V
250mA

††

1474/75 F16

V

OUT

5V
250mA

†

1474/75 F17

V

V

CC

µC

IN

4V TO 18V

MMBD914LT1

* SUMIDA CDRH73-101

** AVX TPSC475M035

†

AVX TPSC107M006

0.1µF

+

ON/OFF

4.7µF**
35V

100k

2.2M

1M

8

2

3

SENSE

ON

LBO

LBI/OFF

6

V

LTC1475-3.3

GND

4

0.1µF

7

1

IN

V

OUT

SW

5

+

L*
100µH

D1
MBR0530

100µF

6.3V

V

3.3V
250mA

†

1474/75 F18

OUT

Figure 18. LTC1475 Regulator with 1-Button Toggle On/Off

17

LTC1474/LTC1475

(

PACKAGE DESCRIPTION

U

Dimensions in inches (millimeters) unless otherwise noted.

MS8 Package

8-Lead Plastic MSOP

(LTC DWG # 05-08-1660)

0.118 ± 0.004*
(3.00 ± 0.102)

8

7

6

5

0.192 ± 0.004

(4.88 ± 0.10)

12

0.040

± 0.006

SEATING

PLANE

(1.02 ± 0.15)

0.012

(0.30)

0.0256

REF

(0.65)

0.152mm) PER SIDE

TYP

0.007
(0.18)

0.021

± 0.006

(0.53 ± 0.015)

* DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH,

PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.

INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006"

° – 6° TYP

0

0.118 ± 0.004**

4

3

0.034 ± 0.004
(0.86 ± 0.102)

(3.00 ± 0.102)

0.006 ± 0.004
(0.15 ± 0.102)

MSOP (MS8) 1197

18

PACKAGE DESCRIPTION

LTC1474/LTC1475

U

Dimensions in inches (millimeters) unless otherwise noted.

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*
(4.801 – 5.004)

7

8

5

6

0.228 – 0.244

(5.791 – 6.197)

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

× 45°

0°– 8° TYP

0.016 – 0.050

0.406 – 1.270

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

0.150 – 0.157**
(3.810 – 3.988)

1

3

2

4

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

TYP

SO8 0996

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

19

LTC1474/LTC1475

N

TYPICAL APPLICATIO

U

V

IN

3.5V to 18V

1M

MMBT2N2222LT1

RELATED PARTS

10pF

+

4.7µF**
35V

ON

1.8M

1M

100k

OFF

6

8

3

SENSE

ON

LBI/OFF

7

V

IN

LTC1475

GND

4

V

LBO

SW

FB

0.1µF

1

2

5

1.02M
1%

1M
1%

MBR0530

V

OUT

2.5V

100µF

6.3V

†

250mA

1474/75 F19

+

L*
100µH

D1

* SUMIDA CDRH73-101

** AVX TPSC475M035

†

AVX TPSC107M006

Figure 19. Pushbutton On/Off with Low Battery Lockout

PART NUMBER DESCRIPTION COMMENTS

LTC1096/LTC1098 Micropower Sampling 8-Bit Serial I/O A/D Converter IQ = 80µA Max
LT1121/LT1121-3.3/LT1121-5 150mA Low Dropout Regulator Linear Regulator, IQ = 30µA
LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Selectable I

= 300mA or 600mA

PEAK

LTC1265 1.2A High Efficiency Step-Down DC/DC Converter Burst Mode Operation, Internal MOSFET
LT1375/LT1376 1.5A 500kHz Step-Down Switching Regulators 500kHz, Small Inductor, High

Efficiency Switchers, 1.5A Switch

LTC1440/LTC1441/LTC1442 Ultralow Power Comparator with Reference IQ = 2.8µA Max
LT1495/LT1496 1.5µA Precision Rail-to-Rail Op Amps IQ = 1.5µA Max
LT1521/LT1521-3/LT1521-3.3/ 300mA Low Dropout Regulator Linear Regulator, IQ = 12µA

LT1521-5
LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode
LT1634-1.25 Micropower Precision Shunt Reference I

sn14745 14745fas LT/TP 0398 4K REV A • PRINTED IN USA

20

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417 ● [phone]
FAX: [phone]

●

TELEX: 499-3977 ● www.linear-tech.com

= 10µA

Q(MIN)

 LINEAR TECHNOLOGY CORPORATION 1997