Page 1
LT3972
prop
33V, 3.5A, 2.4MHz
Step-Down Switching Regulator
with 75µA Quiescent Current
FEATURES
■
Wide Input Range:
Operation from 3.6V to 33V
Over-Voltage Lockout Protects Circuits
Through 62V Transients
■
3.5A Maximum Output Current
■
Low Ripple (<15mV
I
■
■
■
■
■
■
■
■
■
■
= 75μA at 12VIN to 3.3V
Q
Adjustable Switching Frequency: 200kHz to 2.4MHz
Low Shutdown Current: IQ < 1μA
Integrated Boost Diode
Synchronizable Between 250kHz to 2MHz
Power Good Flag
Saturating Switch Design: 95mΩ On-Resistance
Output Voltage: 0.79V to 30V
Thermal Protection
Soft-Start Capability
Small 10-Pin Thermally Enhanced MSOP and
) Burst Mode® Operation:
P-P
OUT
(3mm × 3mm) DFN Packages
APPLICATIONS
■
Automotive Battery Regulation
■
Distributed Supply Regulation
■
Industrial Supplies
■
Wall Transformer Regulation
DESCRIPTION
The LT®3972 is an adjustable frequency (200kHz to 2.4MHz)
monolithic buck switching regulator that accepts input
voltages up to 33V (62V Maximum). A high effi ciency
95mΩ switch is included on the die along with a boost
Schottky diode and the necessary oscillator, control, and
logic circuitry. Current mode topology is used for fast
transient response and good loop stability. Low ripple
Burst Mode operation maintains high effi ciency at low
output currents while keeping output ripple below 15mV
in a typical application. In addition, the LT3972 can further enhance low output current effi ciency by drawing
bias current from the output when V
Shutdown reduces input supply current to less than 1μA
while a resistor and capacitor on the RUN/SS pin provide a
controlled output voltage ramp (soft-start). A power good
fl ag signals when V
reaches 91% of the programmed
OUT
output voltage. The LT3972 is available in 10-Pin MSOP
and 3mm × 3mm DFN packages with exposed pads for
low thermal resistance.
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Burst Mode is a registered trademark of Linear Technology Corporation.
All other trademarks are the
erty of their respective owners.
is above 3V.
OUT
TYPICAL APPLICATION
5V Step-Down Converter
V
IN
6.3V TO 33V
TRANSIENT
TO 62V
10μF
OFF ON
15k
680pF
63.4k
V
RUN/SS BOOST
V
C
RT
PG
SYNC
IN
LT3972
GND
BD
SW
Effi ciency
V
OUT
5V
3.5A
0.47μF
4.7μH
100k
536k
47μF
3680 TA01
FB
100
90
80
70
EFFICIENCY (%)
60
V
OUT
L = 4.7μH
f = 600kHz
50
00.5
VIN = 24V
= 5V
1.5
1
OUTPUT CURRENT (A)
VIN = 12V
VIN = 30V
2
3.5
2.5
3
3680 TA01a
3972f
1
Page 2
LT3972
C
ABSOLUTE MAXIMUM RATINGS
(Note 1)
VIN, RUN/SS Voltage (Note 5) ...................................62V
BOOST Pin Voltage ...................................................56V
BOOST Pin Above SW Pin .........................................30V
FB, RT, V
Voltage .......................................................5V
C
PG, BD, SYNC Voltage ..............................................30V
PIN CONFIGURATION
TOP VIEW
10
BD
1
BOOST
2
11
3
SW
4
V
IN
5
RUN/SS
10-LEAD (3mm s 3mm) PLASTIC DFN
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
DD PACKAGE
θJA = 45°C/W, θJC = 10°C/W
RT
9
V
C
FB
8
7
PG
6
SYNC
Operating Junction Temperature Range (Note 2)
LT3972E .............................................–40°C to 125°C
LT3972I ..............................................– 40°C to 125°C
Storage Temperature Range ...................–65°C to 150°C
Lead Temperature (Soldering, 10 sec)
(MSE Only) ....................................................... 300°C
TOP VIEW
10
1
BD
2
BOOST
SW
V
IN
RUN/SS
10-LEAD PLASTIC MSOP
EXPOSED PAD (PIN 11) IS GND, MUST BE SOLDERED TO PCB
θJA = 45°C/W, θJC = 10°C/W
3
4
5
MSE PACKAGE
11
RT
9
V
C
FB
8
PG
7
SYN
6
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT3972EDD#PBF LT3972EDD#TRPBF LDXR
LT3972IDD#PBF LT3972IDD#TRPBF LDXR
10-Lead (3mm × 3mm) Plastic DFN
10-Lead (3mm × 3mm) Plastic DFN
LT3972EMSE#PBF LT3972EMSE#TRPBF LTDXS 10-Lead Plastic MSOP –40°C to 125°C
LT3972IMSE#PBF LT3972IMSE#TRPBF LTDXS 10-Lead Plastic MSOP – 40°C to 125°C
Consult LTC Marketing for parts specifi ed with wider operating temperature ranges. *The temperature grade is identifi ed by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based fi nish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifi cations, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T
= 25°C. VIN = 10V, V
A
RUN/SS
= 10V, V
= 15V, VBD = 3.3V unless otherwise
BOOST
noted. (Note 2)
PARAMETER CONDITIONS MIN TYP MAX UNITS
Minimum Input Voltage
Overvoltage Lockout
V
IN
Quiescent Current from V
IN
Quiescent Current from BD V
V
= 0.2V 0.01 0.5 μA
RUN/SS
= 3V, Not Switching
V
BD
= 0, Not Switching 120 160 μA
V
BD
= 0.2V 0.01 0.5 μA
RUN/SS
●
●
33 35 37 V
●
–40°C to 125°C
–40°C to 125°C
3 3.6 V
30 65 μA
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2
Page 3
LT3972
ELECTRICAL CHARACTERISTICS
The ● denotes the specifi cations which apply over the full operating
temperature range, otherwise specifi cations are at T
noted. (Note 2)
PARAMETER CONDITIONS MIN TYP MAX UNITS
Minimum Bias Voltage (BD Pin) 2.7 3 V
Feedback Voltage
FB Pin Bias Current (Note 3) V
FB Voltage Line Regulation 4V < V
Error Amp g
Error Amp Gain 2000
Source Current 60 μA
V
C
Sink Current 60 μA
V
C
Pin to Switch Current Gain 5.3 A/V
V
C
Clamp Voltage 2V
V
C
Switching Frequency R
Minimum Switch Off-Time
Switch Current Limit Duty Cycle = 5% 4.6 5.4 6.2 A
Switch V
Boost Schottky Reverse Leakage V
Minimum Boost Voltage (Note 4)
BOOST Pin Current I
RUN/SS Pin Current V
RUN/SS Input Voltage High 2.5 V
RUN/SS Input Voltage Low 0.2 V
PG Threshold Offset from Feedback Voltage VFB Rising 65 mV
PG Hysteresis 10 mV
PG Leakage V
PG Sink Current V
SYNC Low Threshold 0.5 V
SYNC High Threshold 0.7 V
SYNC Pin Bias Current V
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LT3972E is guaranteed to meet performance specifi cations
from 0°C to 125°C. Specifi cations over the –40°C to 125°C operating
temperature range are assured by design, characterization and correlation
with statistical process controls. The LT3972I specifi cations are
guaranteed over the –40°C to 125°C temperature range.
m
CESAT
= 25°C. VIN = 10V, V
A
= 3V, Not Switching
V
BD
= 0, Not Switching 15 μA
V
BD
= 0.8V, VC = 1.2V
FB
< 33V 0.002 0.01 %/V
IN
RUN/SS
= 10V, V
= 15V, VBD = 3.3V unless otherwise
BOOST
●
780
775
●
●
90 130 μA
790
790
10 40 nA
800
805
mV
mV
500 μMho
= 8.66k
T
R
= 29.4k
T
R
= 187k
T
2.2
1.0
200
●
2.45
1.1
230
2.7
1.25
260
MHz
MHz
60 150 nS
kHz
ISW = 3.5A 335 mV
= 0V 0.02 2 μA
BD
●
= 1A 35 50 mA
SW
= 2.5V 5 8 μA
RUN/SS
= 5V 0.1 1 μA
PG
= 0.4V
PG
= 0V 0.1 μA
SYNC
●
200 800 μA
1.5 2 V
Note 3: Bias current fl ows out of the FB pin.
Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
Note 5: Absolute Maximum at V
and RUN/SS Pins is 62V for non-
IN
repetitive 1 minute transients, and 40V for continuous operation.
3972f
3
Page 4
LT3972
TYPICAL PERFORMANCE CHARACTERISTICS
= 25°C unless otherwise noted.
T
A
Effi ciency
100
90
80
70
EFFICIENCY (%)
60
V
OUT
L = 4.7μH
f = 600kHz
50
0 0.5
VIN = 24V
= 5V
1.5
1
OUTPUT CURRENT (A)
No Load Supply Current
130
110
90
70
50
SUPPLY CURRENT (μA)
30
10
0
510 20
INPUT VOLTAGE (V)
VIN = 12V
15
VIN = 30V
2
2.5
25 30 35
3
3680 G01
3.5
Effi ciency
100
90
80
70
EFFICIENCY (%)
60
V
= 3.3V
OUT
L = 3.3μH
f = 600kHz
50
0 0.5
VIN = 12V
VIN = 30V
VIN = 24V
2
1.5
1
OUTPUT CURRENT (A)
2.5
3
3.5
3680 G02
Effi ciency
100
90
80
70
EFFICIENCY (%)
VIN = 12V
60
V
= 5V
OUT
L = 4.7μH
f = 600kHz
50
0 0.5
2
1.5
1
OUTPUT CURRENT (A)
2.5
3
3680 G03
3.0
2.5
TOTAL POWER LOSS (W)
2.0
1.5
1.0
0.5
3.5
No Load Supply Current Maximum Load Current
V
= 3.3V
OUT
3680 G04
400
CATCH DIODE: DIODES, INC. PDS360
350
VIN = 12V
= 3.3V
V
OUT
300
250
200
150
SUPPLY CURRENT (μA)
100
50
0
–50
INCREASED SUPPLY
CURRENT DUE TO CATCH
DIODE LEAKAGE AT
HIGH TEMPERATURE
–25 0 50
25
TEMPERATURE (°C)
75 100 150125
3680 G05
5.5
5.0
4.5
4.0
3.5
LOAD CURRENT (A)
V
= 3.3V
OUT
= 25°C
T
A
3.0
L = 4.7μH
f = 600kHz
2.5
10 20
5
TYPICAL
MINIMUM
15
INPUT VOLTAGE (V)
25 30
3680 G06
Maximum Load Current
5.5
5.0
4.5
4.0
LOAD CURRENT (A)
V
= 5V
OUT
3.5
= 25°C
T
A
L = 4.7μH
f = 600kHz
3.0
10 20
5
4
TYPICAL
MINIMUM
15
INPUT VOLTAGE (V)
25 30
3680 G07
Switch Current Limit
6.0
5.5
5.0
4.5
4.0
SWITCH CURRENT LIMIT(A)
3.5
3.0
20 60
0
40
DUTY CYCLE (%)
80 100
3680 G08
Switch Current Limit
6.5
6.0
5.5
5.0
4.5
4.0
3.5
3.0
SWITCH CURRENT LIMIT (A)
2.5
2.0
–50 25–25 0 50 75 100 150125
DUTY CYCLE = 10 %
DUTY CYCLE = 90 %
TEMPERATURE (°C)
3680 G09
3972f
Page 5
TYPICAL PERFORMANCE CHARACTERISTICS
= 25°C unless otherwise noted.
T
A
LT3972
Switch Voltage Drop
700
600
500
400
300
VOLTAGE DROP (mV)
200
100
0
12 45
0
SWITCH CURRENT (A)
3
Switching Frequency
1.20
1.15
1.10
1.05
1.00
0.95
FREQUENCY (MHz)
0.90
0.85
0.80
–50 25–25 0 50 75 100 150125
TEMPERATURE (°C)
3680 G10
3680 G13
Boost Pin Current Feedback Voltage
120
105
90
75
60
45
30
BOOST PIN CURRENT (mA)
15
0
0312 45
SWITCH CURRENT (A)
Frequency Foldback
1200
1000
800
600
400
SWITCHING FREQUENCY (kHz)
200
0
0
200 400
100 300
FB PIN VOLTAGE (mV)
500
700 900
600
3680 G11
800
3680 G14
840
820
800
780
FEEDBACK VOLTAGE (mV)
760
–50 25–25 0 50 75 100 150125
Minimum Switch On-Time
140
120
100
80
60
40
MINIMUM SWITCH ON TIME (ns)
20
0
–50 25–25 0 50 75 100 150125
TEMPERATURE (°C)
3680 G12
TEMPERATURE (°C)
3680 G15
Soft-Start
7
6
5
4
3
2
SWITCH CURRENT LIMIT (A)
1
0
0.5 1 2
0
1.5
RUN/SS PIN VOLTAGE (V)
2.5 3 3.5
3680 G16
RUN/SS Pin Current Boost Diode
12
10
8
6
4
RUN/SS PIN CURRENT (μA)
2
0
0
510
15 25
RUN/SS PIN VOLTAGE (V)
20 30 35
3680 G17
1.4
1.2
1.0
(V)
F
0.8
0.6
BOOST DIODE V
0.4
0.2
0
0
0.5 1.0 1.5
BOOST DIODE CURRENT (A)
2.0
3680 G18
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5
Page 6
LT3972
TYPICAL PERFORMANCE CHARACTERISTICS
Error Amp Output Current
50
40
30
20
10
0
–10
PIN CURRENT (μA)
C
–20
V
–30
–40
–50
–200
–100 100
FB PIN ERROR VOLTAGE (mV)
0 200
3680 G19
VC Voltages Power Good Threshold
2.50
2.00
1.50
VOLTAGE (V)
1.00
C
V
0.50
CURRENT LIMIT CLAMP
SWITCHING THRESHOLD
Minimum Input Voltage Minimum Input Voltage
5.0
4.5
4.0
3.5
3.0
INPUT VOLTAGE (V)
V
= 3.3V
OUT
2.5
= 25°C
T
A
L = 4.7μH
f = 800kHz
2.0
1
10 100 1000
LOAD CURRENT (mA)
95
90
85
80
THRESHOLD VOLTAGE (%)
= 25°C unless otherwise noted.
T
A
6.5
6.0
5.5
5.0
INPUT VOLTAGE (V)
V
= 5V
OUT
4.5
= 25 °C
T
A
L = 4.7μH
f = 800kHz
10000
3680 G20
4.0
1 1000010 100 1000
LOAD CURRENT (mA)
Switching Waveforms; Burst Mode
V
SW
5V/DIV
I
L
0.2A/DIV
V
OUT
10mV/DIV
3680 G21
0
–50 25–25 0 50 75 100 150125
TEMPERATURE (°C)
Switching Waveforms; Transition
from Burst Mode to Full Frequency
V
SW
5V/DIV
I
L
0.2A/DIV
V
OUT
10mV/DIV
VIN = 12V
V
= 3.3V
OUT
I
= 110mA
LOAD
3680 G22
1μs/DIV
75
–50 25–25 0 50 75 100 150125
TEMPERATURE (°C)
V
5V/DIV
0.5A/DIV
V
10mV/DIV
3680 G25
VIN = 12V
= 3.3V
V
OUT
= 10mA
I
3680 G23
LOAD
Switching Waveforms; Full
Frequency Continuous Operation
SW
I
L
OUT
VIN = 12V
V
= 3.3V
OUT
I
= 1A
LOAD
1μs/DIV
5μs/DIV
3680 G26
3680 G24
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6
Page 7
PIN FUNCTIONS
LT3972
BD (Pin 1): This pin connects to the anode of the boost
Schottky diode. BD also supplies current to the internal
regulator.
BOOST (Pin 2): This pin is used to provide a drive
voltage, higher than the input voltage, to the internal bipolar
NPN power switch.
SW (Pin 3): The SW pin is the output of the internal power
switch. Connect this pin to the inductor, catch diode and
boost capacitor.
(Pin 4): The VIN pin supplies current to the LT3972’s
V
IN
internal regulator and to the internal power switch. This
pin must be locally bypassed.
RUN/SS (Pin 5): The RUN/SS pin is used to put the
LT3972 in shutdown mode. Tie to ground to shut down
the LT3972. Tie to 2.5V or more for normal operation. If
the shutdown feature is not used, tie this pin to the V
IN
pin. RUN/SS also provides a soft-start function; see the
Applications Information section.
SYNC (Pin 6): This is the external clock synchronization
input. Ground this pin for low ripple Burst Mode operation at
low output loads. Tie to a clock source for synchronization.
Clock edges should have rise and fall times faster than 1μs.
Tie Pin to GND if not used. See Synchronization section
in Applications Information.
PG (Pin 7): The PG pin is the open collector output of an
internal comparator. PG remains low until the FB pin is
within 9% of the fi nal regulation voltage. PG output is valid
when V
is above 3.6V and RUN/SS is high.
IN
FB (Pin 8): The LT3972 regulates the FB pin to 0.790V.
Connect the feedback resistor divider tap to this pin.
(Pin 9): The VC pin is the output of the internal error
V
C
amplifi er. The voltage on this pin controls the peak switch
current. Tie an RC network from this pin to ground to
compensate the control loop.
RT (Pin 10): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.
Exposed Pad (Pin 11): Ground. The Exposed Pad must
be soldered to PCB.
BLOCK DIAGRAM
V
V
IN
IN
4
C1
INTERNAL 0.79V REF
RUN/SS
5
RT
10
R
T
SYNC
6
SOFT-START
PG
7
GND
11 8
–
BOOST
BD
1
2
C3
SW
3
V
C
9
L1
D1
C
C
C
F
R
C
C2
3680 BD
V
OUT
3972f
+
SLOPE COMP
3
OSCILLATOR
200kHzTO2.4MHz
ERROR AMP
0.73V
+
–
FB
R1
+
–
R2
SWITCH
LATCH
R
S
DISABLE
BurstMode
DETECT
CLAMP
V
C
Q
7
Page 8
LT3972
OPERATION
The LT3972 is a constant frequency, current mode stepdown regulator. An oscillator, with frequency set by RT,
enables an RS fl ip-fl op, turning on the internal power
switch. An amplifi er and comparator monitor the current
fl owing between the V
off when this current reaches a level determined by the
voltage at V
voltage through an external resistor divider tied to the FB
pin and servos the V
increases, more current is delivered to the output; if it
decreases, less current is delivered. An active clamp on the
pin provides current limit. The VC pin is also clamped to
V
C
the voltage on the RUN/SS pin; soft-start is implemented
by generating a voltage ramp at the RUN/SS pin using an
external resistor and capacitor.
An internal regulator provides power to the control circuitry.
The bias regulator normally draws power from the V
but if the BD pin is connected to an external voltage higher
than 3V bias power will be drawn from the external source
(typically the regulated output voltage). This improves
effi ciency. The RUN/SS pin is used to place the LT3972
in shutdown, disconnecting the output and reducing the
input current to less than 0.5μA.
The switch driver operates from either the input or from
the BOOST pin. An external capacitor and diode are used
. An error amplifi er measures the output
C
and SW pins, turning the switch
IN
pin. If the error amplifi er’s output
C
pin,
IN
to generate a voltage at the BOOST pin that is higher than
the input supply. This allows the driver to fully saturate
the internal bipolar NPN power switch for effi cient operation.
To further optimize effi ciency, the LT3972 automatically
switches to Burst Mode operation in light load situations.
Between bursts, all circuitry associated with controlling
the output switch is shut down, reducing the input supply
current to 75μA in a typical application.
The oscillator reduces the LT3972’s operating frequency
when the voltage at the FB pin is low. This frequency
foldback helps to control the output current during startup
and overload.
The LT3972 contains a power good comparator which trips
when the FB pin is at 91% of its regulated value. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the LT3972 is
enabled and V
The LT3972 has an overvoltage protection feature which
disables switching action when the V
typical (33V minimum). When switching is disabled, the
LT3972 can safely sustain input voltages up to 62V.
is above 3.6V.
IN
goes above 35V
IN
8
3972f
Page 9
APPLICATIONS INFORMATION
LT3972
FB Resistor Network
The output voltage is programmed with a resistor divider
between the output and the FB pin. Choose the 1% resistors according to:
RR
12
⎛
⎜
⎝
079
OUT
.
⎞
1=
–
⎟
⎠
V
V
Reference designators refer to the Block Diagram.
Setting the Switching Frequency
The LT3972 uses a constant frequency PWM architecture
that can be programmed to switch from 200kHz to 2.4MHz
by using a resistor tied from the RT pin to ground. A table
showing the necessary RT value for a desired switching
frequency is in Figure 1.
SWITCHING FREQUENCY (MHz) RT VALUE (kΩ)
0.2
0.3
0.4
0.5
0.6
0.7
0.8
0.9
1.0
1.2
1.4
1.6
1.8
2.0
2.2
2.4
Figure 1. Switching Frequency vs. RT Value
215
140
100
78.7
63.4
53.6
45.3
39.2
34
26.7
22.1
18.2
15
12.7
10.7
9.09
Operating Frequency Tradeoffs
Selection of the operating frequency is a tradeoff between
effi ciency, component size, minimum dropout voltage, and
maximum input voltage. The advantage of high frequency
operation is that smaller inductor and capacitor values may
be used. The disadvantages are lower effi ciency, lower
maximum input voltage, and higher dropout voltage. The
highest acceptable switching frequency (f
SW(MAX)
) for a
given application can be calculated as follows:
VV
+
D OUT
+
()
()
DINSW
–
f
SW MAX
=
()
tVVV
ON MIN
where VIN is the typical input voltage, V
voltage, V
is the catch diode drop (~0.5V) and VSW is the
D
is the output
OUT
internal switch drop (~0.5V at max load). This equation
shows that slower switching frequency is necessary to
safely accommodate high V
IN/VOUT
ratio. Also, as shown
in the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
switching frequency is because the LT3972 switch has fi nite
minimum on and off times. The switch can turn on for a
minimum of ~150ns and turn off for a minimum of ~150ns.
Typical minimum on time at 25°C is 80ns. This means
that the minimum and maximum duty cycles are:
DC f t
DC f t
where fSW is the switching frequency, the t
minimum switch on time (~150ns), and the t
=
MIN SW
MAX SW
ON MIN
=
1–
()
OFF MIN
()
ON(MIN)
OFF(MIN)
is the
is
the minimum switch off time (~150ns). These equations
show that duty cycle range increases when switching
frequency is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
Input Voltage Range
The maximum input voltage for LT3972 applications depends on switching frequency , Absolute Maximum Ratings
of the V
and BOOST pins, and the operating mode.
IN
The LT3972 can operate from input voltages of up to 33V,
and withstand voltages up to 62V. Note that while V
IN
is
above 35V typical (33V minimum and 37V maximum)
the part will keep the switch off and the output will not
be in regulation.
To safely allow inputs of up to 62V, be sure to choose a
Schottky diode, inductor size, and switching frequency
to allow safe operation at 37V according to the following
discussions.
While the output is in start-up, short-circuit, or other
overload conditions, the switching frequency should be
chosen according to the following equation:
3972f
9
Page 10
LT3972
APPLICATIONS INFORMATION
VV
+
V
IN MAX
()
where V
V
OUT
IN(MAX)
is the output voltage, VD is the catch diode drop
(~0.5V), V
load), f
t
ON(MIN)
SW
is the minimum switch on time (~100ns). Note that
=
ft
SW
OUT D
ON MIN
()
VV
+–
DSW
is the maximum operating input voltage,
is the internal switch drop (~0.5V at max
SW
is the switching frequency (set by RT), and
a higher switching frequency will depress the maximum
operating input voltage. Conversely, a lower switching
frequency will be necessary to achieve safe operation at
high input voltages.
If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage
transients of up to 33V are acceptable regardless of the
switching frequency. In this mode, the LT3972 may enter
pulse skipping operation where some switching pulses
are skipped to maintain output regulation. In this mode
the output voltage ripple and inductor current ripple will
be higher than in normal operation.
The minimum input voltage is determined by either the
LT3972’s minimum operating voltage of ~3.6V or by its
maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
VV
+
V
IN MIN
()
where V
IN(MIN)
=
1–
OUT D
ft
SW
OFF MIN
()
VV
+
–
DSW
is the minimum input voltage, and t
OFF(MIN)
is the minimum switch off time (150ns). Note that higher
switching frequency will increase the minimum input
voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
Inductor Selection
For a given input and output voltage, the inductor value
and switching frequency will determine the ripple current.
The ripple current ΔI
increases with higher VIN or V
L
OUT
and decreases with higher inductance and faster switching frequency. A reasonable starting point for selecting
the ripple current is:
= 0.4(I
ΔI
L
where I
OUT(MAX)
OUT(MAX)
)
is the maximum output load current. To
guarantee suffi cient output current, peak inductor current
must be lower than the LT3972’s switch current limit (I
LIM
).
The peak inductor current is:
I
L(PEAK)
where I
= I
OUT(MAX)
is the peak inductor current, I
L(PEAK)
the maximum output load current, and ΔI
ripple current. The LT3972’s switch current limit (I
+ ΔIL/2
OUT(MAX)
is the inductor
L
) is
LIM
is
5.5A at low duty cycles and decreases linearly to 4.5A at
DC = 0.8. The maximum output current is a function of
the inductor ripple current:
= I
I
OUT(MAX)
LIM
– ΔIL/2
Be sure to pick an inductor ripple current that provides
suffi cient maximum output current (I
OUT(MAX)
).
The largest inductor ripple current occurs at the highest
. To guarantee that the ripple current stays below the
V
IN
specifi ed maximum, the inductor value should be chosen
according to the following equation:
⎛
VV
OUT D
L
=
⎜
fI
Δ
⎝
SW L
⎛
⎞
+
VV
⎜
1–
⎟
⎜
⎠
⎝
OUT D
V
IN MAX
⎞
+
⎟
⎟
()
⎠⎠
where VD is the voltage drop of the catch diode (~0.4V),
V
IN(MAX)
voltage, f
is the maximum input voltage, V
is the switching frequency (set by RT), and
SW
is the output
OUT
L is in the inductor value.
The inductor’s RMS current rating must be greater than the
maximum load current and its saturation current should be
about 30% higher. For robust operation in fault conditions
(start-up or short circuit) and high input voltage (>30V),
the saturation current should be above 5A. To keep the
effi ciency high, the series resistance (DCR) should be less
than 0.1Ω, and the core material should be intended for
high frequency applications. Table 1 lists several vendors
and suitable types.
10
3972f
Page 11
APPLICATIONS INFORMATION
LT3972
Table 1. Inductor Vendors
VENDOR URL PART SERIES TYPE
Murata www.murata.com LQH55D Open
TDK www.componenttdk.com SLF10145 Shielded
Toko www.toko.com D75C
D75F
Sumida www.sumida.com CDRH74
CR75
CDRH8D43
NEC www.nec.com MPLC073
MPBI0755
Shielded
Open
Shielded
Open
Shielded
Shielded
Shielded
Of course, such a simple design guide will not always result in the optimum inductor for your application. A larger
value inductor provides a slightly higher maximum load
current and will reduce the output voltage ripple. If your
load is lower than 3.5A, then you can decrease the value
of the inductor and operate with higher ripple current. This
allows you to use a physically smaller inductor, or one
with a lower DCR resulting in higher effi ciency. There are
several graphs in the Typical Performance Characteristics
section of this data sheet that show the maximum load
current as a function of input voltage and inductor value
for several popular output voltages. Low inductance may
result in discontinuous mode operation, which is okay
but further reduces maximum load current. For details of
maximum output current and discontinuous mode operation, see Linear Technology Application Note 44. Finally,
for duty cycles greater than 50% (V
OUT/VIN
> 0.5), there
is a minimum inductance required to avoid subharmonic
oscillations. See AN19.
Input Capacitor
Bypass the input of the LT3972 circuit with a ceramic
capacitor of X7R or X5R type. Y5V types have poor
performance over temperature and applied voltage, and
should not be used. A 10μF to 22μF ceramic capacitor is
adequate to bypass the LT3972 and will easily handle the
ripple current. Note that larger input capacitance is required
when a lower switching frequency is used. If the input
power source has high impedance, or there is signifi cant
inductance due to long wires or cables, additional bulk
capacitance may be necessary. This can be provided with
a lower performance electrolytic capacitor.
Step-down regulators draw current from the input supply in pulses with very fast rise and fall times. The input
capacitor is required to reduce the resulting voltage
ripple at the LT3972 and to force this very high frequency
switching current into a tight local loop, minimizing EMI.
A 10μF capacitor is capable of this task, but only if it is
placed close to the LT3972 and the catch diode (see the
PCB Layout section). A second precaution regarding the
ceramic input capacitor concerns the maximum input
voltage rating of the LT3972. A ceramic input capacitor
combined with trace or cable inductance forms a high
quality (under damped) tank circuit. If the LT3972 circuit
is plugged into a live supply, the input voltage can ring to
twice its nominal value, possibly exceeding the LT3972’s
voltage rating. This situation is easily avoided (see the Hot
Plugging Safety section).
For space sensitive applications, a 4.7μF ceramic capacitor can be used for local bypassing of the LT3972 input.
However, the lower input capacitance will result in increased input current ripple and input voltage ripple, and
may couple noise into other circuitry. Also, the increased
voltage ripple will raise the minimum operating voltage
of the LT3972 to ~3.7V.
Output Capacitor and Output Ripple
The output capacitor has two essential functions. Along
with the inductor, it fi lters the square wave generated by the
LT3972 to produce the DC output. In this role it determines
the output ripple, and low impedance at the switching
frequency is important. The second function is to store
energy in order to satisfy transient loads and stabilize the
LT3972’s control loop. Ceramic capacitors have very low
equivalent series resistance (ESR) and provide the best
ripple performance. A good starting value is:
C
OUT
where fSW is in MHz, and C
100
=
Vf
OUT SW
is the recommended
OUT
output capacitance in μF. Use X5R or X7R types. This
choice will provide low output ripple and good transient
response. Transient performance can be improved with
a higher value capacitor if the compensation network is
also adjusted to maintain the loop bandwidth. A lower
3972f
11
Page 12
LT3972
APPLICATIONS INFORMATION
Table 2. Capacitor Vendors
VENDOR PHONE URL PART SERIES COMMANDS
Panasonic (714) 373-7366 www.panasonic.com Ceramic,
Kemet (864) 963-6300 www.kemet.com Ceramic,
Sanyo (408) 749-9714 www.sanyovideo.com Ceramic,
Murata (408) 436-1300 www.murata.com Ceramic
AVX www.avxcorp.com Ceramic,
Taiyo Yuden (864) 963-6300 www.taiyo-yuden.com Ceramic
Polymer,
Tantalum
Tantalum T494, T495
Polymer,
Tantalum
Tantalum TPS Series
EEF Series
POSCAP
value of output capacitor can be used to save space and
cost but transient performance will suffer. See the Frequency Compensation section to choose an appropriate
compensation network.
When choosing a capacitor, look carefully through the
data sheet to fi nd out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specifi ed
by the supplier, and should be 0.05Ω or less. Such a
capacitor will be larger than a ceramic capacitor and will
have a larger capacitance, because the capacitor must be
large to achieve low ESR. Table 2 lists several capacitor
vendors.
Catch Diode
The catch diode conducts current only during switch off
time. Average forward current in normal operation can
be calculated from:
I
D(AVG)
where I
= I
OUT
OUT
(VIN – V
OUT
)/V
IN
is the output load current. The only reason to
consider a diode with a larger current rating than necessary
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a Schottky diode with a
reverse voltage rating greater than the input voltage. The
overvoltage protection feature in the LT3972 will keep the
switch off when V
rated Schottky even when V
> 35V which allows the use of 40V
IN
ranges up to 62V. Table 3
IN
lists several Schottky diodes and their manufacturers.
Table 3. Diode Vendors
V
PART NUMBER
On Semiconductor
MBRA340 40 3 500
Diodes Inc.
PDS340
B340A
B340LA
R
(V)
40
40
40
I
(A)
AVE
3
3
3
V
AT 3 A
F
(mV)
500
500
450
Ceramic Capacitors
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the LT3972 due to their piezoelectric
nature. When in Burst Mode operation, the LT3972’s
switching frequency depends on the load current, and at
very light loads the LT3972 can excite the ceramic capacitor at audio frequencies, generating audible noise. Since
the LT3972 operates at a lower current limit during Burst
Mode operation, the noise is nearly silent to a casual ear.
If this is unacceptable, use a high performance tantalum
or electrolytic capacitor at the output.
12
3972f
Page 13
APPLICATIONS INFORMATION
LT3972
Frequency Compensation
The LT3972 uses current mode control to regulate the
output. This simplifi es loop compensation. In particular,
the LT3972 does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
pin, as shown in Figure 2. Generally a capacitor (CC)
V
C
and a resistor (R
) in series to ground are used. In addi-
C
tion, there may be lower value capacitor in parallel. This
capacitor (C
) is not part of the loop compensation but
F
is used to fi lter noise at the switching frequency, and is
required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with one of the circuits in this data
sheet that is similar to your application and tune the compensation network to optimize the performance. Stability
should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LT1375 data sheet contains a more thorough discussion of
loop compensation and describes how to test the stability using a transient load. Figure 2 shows an equivalent
circuit for the LT3972 control loop. The error amplifi er is a
transconductance amplifi er with fi nite output impedance.
The power section, consisting of the modulator, power
switch and inductor, is modeled as a transconductance
amplifi er generating an output current proportional to
the voltage at the V
integrates this current, and that the capacitor on the V
) integrates the error amplifi er output current, resulting
(C
C
pin. Note that the output capacitor
C
pin
C
in two poles in the loop. In most cases a zero is required
and comes from either the output capacitor ESR or from
a resistor R
in series with CC. This simple model works
C
well as long as the value of the inductor is not too high
and the loop crossover frequency is much lower than the
switching frequency. A phase lead capacitor (C
) across
PL
the feedback divider may improve the transient response.
Figure 3 shows the transient response when the load current is stepped from 1A to 3A and back to 1A.
LT3972
CURRENT MODE
POWER STAGE
= 5.3mho
g
m
3M
V
C
R
C
C
F
C
C
ERROR
AMPLIFIER
gm =
500μmho
Figure 2. Model for Loop Response
V
OUT
100mV/DIV
I
L
1A/DIV
V
= 12V
IN
= 3.3V
V
OUT
Figure 3. Transient Load Response of the LT3972 Front Page
Application as the Load Current is Stepped from 1A to 3A.
V
= 5V
OUT
–
+
GND
SW
FB
0.8V
10μs/DIV
R1
R2
C
PL
ESR
C1
POLYMER
OR
TANTALUM
3680 F03
OUTPUT
+
C1
CERAMIC
3680 F02
3972f
13
Page 14
LT3972
APPLICATIONS INFORMATION
Low-Ripple Burst Mode and Pulse-Skip Mode
The LT3972 is capable of operating in either Low-Ripple
Burst Mode or Pulse-Skip Mode which are selected using the SYNC pin. See the Synchronization section for
details.
To enhance effi ciency at light loads, the LT3972 can be
operated in Low-Ripple Burst Mode operation which keeps
the output capacitor charged to the proper voltage while
minimizing the input quiescent current. During Burst Mode
operation, the LT3972 delivers single cycle bursts of current
to the output capacitor followed by sleep periods where
the output power is delivered to the load by the output
capacitor. Because the LT3972 delivers power to the output
with single, low current pulses, the output ripple is kept
below 15mV for a typical application. In addition, V
IN
and
BD quiescent currents are reduced to typically 30μA and
80μA respectively during the sleep time. As the load current decreases towards a no load condition, the percentage
of time that the LT3972 operates in sleep mode increases
and the average input current is greatly reduced resulting
in high effi ciency even at very low loads. See Figure 4.
At higher output loads (above 140mA for the front page
application) the LT3972 will be running at the frequency
programmed by the R
resistor, and will be operating in
T
standard PWM mode. The transition between PWM and
Low-Ripple Burst Mode is seamless, and will not disturb
the output voltage.
If low quiescent current is not required the LT3972 can
operate in Pulse-Skip mode. The benefi t of this mode is
V
SW
5V/DIV
I
L
0.2A/DIV
V
OUT
10mV/DIV
VIN = 12V
= 3.3V
V
OUT
= 10mA
I
LOAD
Figure 4. Burst Mode Operation
5μs/DIV
3680 F04
that the LT3972 will enter full frequency standard PWM
operation at a lower output load current than when in Burst
Mode. The front page application circuit will switch at full
frequency at output loads higher than about 60mA.
BOOST and BIAS Pin Considerations
Capacitor C3 and the internal boost Schottky diode (see
the Block Diagram) are used to generate a boost voltage that is higher than the input voltage. In most cases
a 0.22μF capacitor will work well. Figure 2 shows three
ways to arrange the boost circuit. The BOOST pin must be
more than 2.3V above the SW pin for best effi ciency. For
outputs of 3V and above, the standard circuit (Figure 5a)
is best. For outputs between 2.8V and 3V, use a 1μF boost
capacitor. A 2.5V output presents a special case because it
is marginally adequate to support the boosted drive stage
while using the internal boost diode. For reliable BOOST pin
operation with 2.5V outputs use a good external Schottky
diode (such as the ON Semi MBR0540), and a 1μF boost
capacitor (see Figure 5b). For lower output voltages the
boost diode can be tied to the input (Figure 5c), or to
another supply greater than 2.8V. Tying BD to V
reduces
IN
the maximum input voltage to 28V. The circuit in Figure 5a
is more effi cient because the BOOST pin current and BD
pin quiescent current comes from a lower voltage source.
You must also be sure that the maximum voltage ratings
of the BOOST and BD pins are not exceeded.
The minimum operating voltage of an LT3972 application
is limited by the minimum input voltage (3.6V) and by the
maximum duty cycle as outlined in a previous section. For
proper startup, the minimum input voltage is also limited
by the boost circuit. If the input voltage is ramped slowly,
or the LT3972 is turned on with its RUN/SS pin when the
output is already in regulation, then the boost capacitor
may not be fully charged. Because the boost capacitor is
charged with the energy stored in the inductor, the circuit
will rely on some minimum load current to get the boost
circuit running properly. This minimum load will depend
on input and output voltages, and on the arrangement of
the boost circuit. The minimum load generally goes to
zero once the circuit has started. Figure 6 shows a plot
of minimum load to start and to run as a function of input
voltage. In many cases the discharged output capacitor
will present a load to the switcher, which will allow it to
3972f
14
Page 15
APPLICATIONS INFORMATION
LT3972
V
OUT
BD
BOOST
V
4.7μF
V
4.7μF
V
4.7μF
IN
IN
IN
V
LT3972
IN
GND
(5a) For V
BD
V
LT3972
IN
GND
(5b) For 2.5V < V
BD
V
LT3972
IN
GND
(5c) For V
OUT
SW
OUT
BOOST
SW
BOOST
SW
< 2.5V; V
C3
> 2.8V
C3
< 2.8V
OUT
IN(MAX)
C3
D2
3680 FO5
= 30V
V
OUT
V
OUT
Figure 5. Three Circuits For Generating The Boost Voltage
start. The plots show the worst-case situation where V
IN
is ramping very slowly. For lower start-up voltage, the
boost diode can be tied to V
; however, this restricts the
IN
input range to one-half of the absolute maximum rating
of the BOOST pin.
At light loads, the inductor current becomes discontinuous and the effective duty cycle can be very high. This
reduces the minimum input voltage to approximately
300mV above V
. At higher load currents, the inductor
OUT
current is continuous and the duty cycle is limited by the
maximum duty cycle of the LT3972, requiring a higher
input voltage to maintain regulation.
6.0
5.5
TO START
(WORST CASE)
5.0
4.5
4.0
TO RUN
3.5
INPUT VOLTAGE (V)
3.0
V
= 3.3V
OUT
= 25°C
T
A
2.5
L = 8.2μH
f = 700kHz
2.0
1
8.0
7.0
6.0
5.0
4.0
INPUT VOLTAGE (V)
3.0
2.0
1 1000010 100 1000
10 100 1000
LOAD CURRENT (mA)
TO START
(WORST CASE)
TO RUN
V
= 5V
OUT
= 25°C
T
A
L = 8.2μH
f = 700kHz
LOAD CURRENT (mA)
10000
3680 F06
Figure 6. The Minimum Input Voltage Depends on
Output Voltage, Load Current and Boost Circuit
Soft-Start
The RUN/SS pin can be used to soft-start the LT3972,
reducing the maximum input current during start-up.
The RUN/SS pin is driven through an external RC fi lter to
create a voltage ramp at this pin. Figure 7 shows the startup and shut-down waveforms with the soft-start circuit.
By choosing a large RC time constant, the peak start-up
current can be reduced to the current that is required to
regulate the output, with no overshoot. Choose the value
of the resistor so that it can supply 20μA when the RUN/SS
pin reaches 2.5V.
Synchronization
To select Low-Ripple Burst Mode operation, tie the SYNC
pin below 0.5V (this can be ground or a logic output).
3972f
15
Page 16
LT3972
APPLICATIONS INFORMATION
I
L
RUN
15k
RUN/SS
0.22μF
Figure 7. To Soft-Start the LT3972, Add a Resisitor
and Capacitor to the RUN/SS Pin
GND
2ms/DIV
Synchronizing the LT3972 oscillator to an external frequency can be done by connecting a square wave (with
20% to 80% duty cycle) to the SYNC pin. The square
wave amplitude should have valleys that are below 0.3V
and peaks that are above 0.8V (up to 6V).
The LT3972 will not enter Burst Mode at low output loads
while synchronized to an external clock, but instead will
skip pulses to maintain regulation.
The LT3972 may be synchronized over a 250kHz to 2MHz
range. The R
resistor should be chosen to set the LT3972
T
switching frequency 20% below the lowest synchronization
input. For example, if the synchronization signal will be
250kHz and higher, the R
should be chosen for 200kHz.
T
To assure reliable and safe operation the LT3972 will only
synchronize when the output voltage is near regulation
as indicated by the PG fl ag. It is therefore necessary to
choose a large enough inductor value to supply the required
output current at the frequency set by the R
Inductor Selection section. It is also important to note that
slope compensation is set by the R
value: When the sync
T
frequency is much higher than the one set by R
compensation will be signifi cantly reduced which may
require a larger inductor value to prevent subharmonic
oscillation.
Shorted and Reversed Input Protection
If the inductor is chosen so that it won’t saturate excessively, an LT3972 buck regulator will tolerate a shorted
output. There is another situation to consider in systems
where the output will be held high when the input to the
LT3972 is absent. This may occur in battery charging ap-
1A/DIV
V
RUN/SS
2V/DIV
V
OUT
2V/DIV
3680 F07
resistor. See
T
, the slope
T
plications or in battery backup systems where a battery
or some other supply is diode OR-ed with the LT3972’s
output. If the V
pin is allowed to fl oat and the RUN/SS
IN
pin is held high (either by a logic signal or because it is
tied to V
), then the LT3972’s internal circuitry will pull
IN
its quiescent current through its SW pin. This is fi ne if
your system can tolerate a few mA in this state. If you
ground the RUN/SS pin, the SW pin current will drop to
essentially zero. However, if the V
pin is grounded while
IN
the output is held high, then parasitic diodes inside the
LT3972 can pull large currents from the output through
the SW pin and the V
pin. Figure 8 shows a circuit that
IN
will run only when the input voltage is present and that
protects against a shorted or reversed input.
D4
MBRS340
V
IN
Figure 8. Diode D4 Prevents a Shorted Input from
Discharging a Backup Battery Tied to the Output. It Also
Protects the Circuit from a Reversed Input. The LT3972
Runs Only When the Input is Present
V
IN
RUN/SS
V
C
LT3972
GND FB
BOOST
SW
3680 F08
V
OUT
BACKUP
PCB Layout
For proper operation and minimum EMI, care must be
taken during printed circuit board layout. Figure 9 shows
the recommended component placement with trace,
ground plane and via locations. Note that large, switched
currents fl ow in the LT3972’s V
and SW pins, the catch
IN
diode (D1) and the input capacitor (C1). The loop formed
by these components should be as small as possible. These
components, along with the inductor and output capacitor,
should be placed on the same side of the circuit board,
and their connections should be made on that layer. Place
a local, unbroken ground plane below these components.
The SW and BOOST nodes should be as small as possible.
Finally, keep the FB and V
nodes small so that the ground
C
16
3972f
Page 17
APPLICATIONS INFORMATION
L1
V
OUT
OUT
C1
VIAS TO SYNC
D1
VIAS TO LOCAL GROUND PLANE
VIAS TO V
Figure 9. A Good PCB Layout Ensures Proper, Low EMI Operation
C2
GND
VIAS TO RUN/SS
VIAS TO PG
C
R
RT
R
PG
C
R
C
R2
R1
VIAS TO V
IN
OUTLINE OF LOCAL
GROUND PLANE
3680 F09
LT3972
traces will shield them from the SW and BOOST nodes.
The Exposed Pad on the bottom of the package must be
soldered to ground so that the pad acts as a heat sink. To
keep thermal resistance low, extend the ground plane as
much as possible, and add thermal vias under and near
the LT3972 to additional ground planes within the circuit
board and on the bottom side.
Hot Plugging Safely
The small size, robustness and low impedance of ceramic
capacitors make them an attractive option for the input
bypass capacitor of LT3972 circuits. However, these capacitors can cause problems if the LT3972 is plugged into a
live supply (see Linear Technology Application Note 88 for
a complete discussion). The low loss ceramic capacitor,
combined with stray inductance in series with the power
source, forms an under damped tank circuit, and the
+
LOW
IMPEDANCE
ENERGIZED
24V SUPPLY
+
+
CLOSING SWITCH
SIMULATES HOT PLUG
STRAY
INDUCTANCE
DUE TO 6 FEET
(2 METERS) OF
TWISTED PAIR
+
22μF
35V
AI.EI.
I
IN
0.7W
V
IN
V
IN
LT3972
4.7μF
20V/DIV
10A/DIV
I
IN
DANGER
RINGING V
MAY EXCEED
IN
ABSOLUTE MAXIMUM RATING
20μs/DIV
(10a)
V
IN
(10b)
20V/DIV
10A/DIV
20V/DIV
10A/DIV
I
IN
20μs/DIV
V
IN
I
IN
LT3972
4.7μF0.1μF
LT3972
4.7μF
(10c)
20μs/DIV
3680 F10
Figure 10. A Well Chosen Input Network Prevents Input Voltage Overshoot and
Ensures Reliable Operation when the LT3972 is Connected to a Live Supply
3972f
17
Page 18
LT3972
APPLICATIONS INFORMATION
voltage at the VIN pin of the LT3972 can ring to twice the
nominal input voltage, possibly exceeding the LT3972’s
rating and damaging the part. If the input supply is poorly
controlled or the user will be plugging the LT3972 into an
energized supply, the input network should be designed
to prevent this overshoot. Figure 10 shows the waveforms
that result when an LT3972 circuit is connected to a 24V
supply through six feet of 24-gauge twisted pair. The
fi rst plot is the response with a 4.7μF ceramic capacitor
at the input. The input voltage rings as high as 50V and
the input current peaks at 26A. A good solution is shown
in Figure 10b. A 0.7Ω resistor is added in series with the
input to eliminate the voltage overshoot (it also reduces
the peak input current). A 0.1μF capacitor improves high
frequency fi ltering. For high input voltages its impact on
effi ciency is minor, reducing effi ciency by 1.5 percent for
a 5V output at full load operating from 24V.
High Temperature Considerations
The PCB must provide heat sinking to keep the LT3972
cool. The Exposed Pad on the bottom of the package must
be soldered to a ground plane. This ground should be tied
to large copper layers below with thermal vias; these layers will spread the heat dissipated by the LT3972. Place
additional vias can reduce thermal resistance further. With
these steps, the thermal resistance from die (or junction)
to ambient can be reduced to JA = 35°C/W or less. With
100 LFPM airfl ow, this resistance can fall by another 25%.
Further increases in airfl ow will lead to lower thermal resistance. Because of the large output current capability of
the LT3972, it is possible to dissipate enough heat to raise
the junction temperature beyond the absolute maximum of
125°C. When operating at high ambient temperatures, the
maximum load current should be derated as the ambient
temperature approaches 125°C.
Power dissipation within the LT3972 can be estimated by
calculating the total power loss from an effi ciency measurement and subtracting the catch diode loss and inductor
loss. The die temperature is calculated by multiplying the
LT3972 power dissipation by the thermal resistance from
junction to ambient.
Other Linear Technology Publications
Application Notes 19, 35 and 44 contain more detailed
descriptions and design information for buck regulators
and other switching regulators. The LT1376 data sheet
has a more extensive discussion of output ripple, loop
compensation and stability testing. Design Note 100
shows how to generate a bipolar output supply using a
buck regulator.
TYPICAL APPLICATIONS
V
6.3V TO 33V
IN
TRANSIENT
TO 62V
10μF
680pF
D: ON SEMI MBRA340
L: NEC MPLC0730L4R7
18
15k
5V Step-Down Converter
V
ON OFF
63.4k
RUN/SS BOOST
V
C
RT
PG
SYNC
f = 600kHz
IN
LT3972
GND
BD
SW
V
OUT
5V
3.5A
0.47μF
D
FB
100k
L
4.7μH
536k
47μF
3680 TA02
3972f
Page 19
TYPICAL APPLICATIONS
V
4.4V TO 33V
IN
TRANSIENT
TO 62V
4.7μF
680pF
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
19k
3.3V Step-Down Converter
V
IN
ON OFF
63.4k
RUN/SS BOOST
V
C
LT3972
RT
PG
SYNC
GND
f = 600kHz
BD
SW
LT3972
V
OUT
3.3V
3.5A
0.47μF
D
FB
100k
L
3.3μH
316k
22μF
3680 TA03
V
4V TO 33V
TRANSIENT
TO 62V
4.7μF
IN
15.4k
680pF
D1: ON SEMI MBRA340
D2: MBR0540
L: NEC MPLC0730L3R3
2.5V Step-Down Converter
V
IN
ON OFF
63.4k
RUN/SS BOOST
V
C
LT3972
RT
PG
SYNC
GND
f = 600kHz
BD
SW
V
OUT
2.5V
D2
1μF
D1
FB
100k
L
3.3μH
215k
3.5A
47μF
3680 TA04
3972f
19
Page 20
LT3972
TYPICAL APPLICATIONS
5V, 2MHz Step-Down Converter
V
8.6V TO 22V
TRANSIENT
TO 62V
4.7μF
15V TO 33V
TRANSIENT
TO 62V
10μF
IN
15k
680pF
D: ON SEMI MBRA340
L: NEC MPLC0730L2R2
V
IN
17.4k
680pF
D: ON SEMI MBRA340
L: NEC MBP107558R2P
ON OFF
12.7k
ON OFF
63.4k
V
IN
RUN/SS BOOST
V
C
RT
PG
SYNC
f = 2MHz
LT3972
GND
BD
SW
FB
12V Step-Down Converter
V
IN
RUN/SS BOOST
V
C
RT
PG
SYNC
f = 600kHz
LT3972
GND
BD
SW
FB
0.47μF
D
100k
0.47μF
D
50k
536k
715k
L
2.2μH
L
8.2μH
3680 TA05
3680 TA06
V
OUT
5V
2.5A
22μF
V
OUT
12V
3.5A
47μF
20
3972f
Page 21
TYPICAL APPLICATIONS
V
IN
3.5V TO 27V
4.7μF
16.9k
680pF
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
1.8V Step-Down Converter
V
IN
ON OFF
78.7k
RUN/SS BOOST
V
C
LT3972
RT
PG
SYNC
GND
f = 500kHz
BD
SW
LT3972
V
OUT
1.8V
3.5A
0.47μF
D
FB
100k
L
3.3μH
127k
47μF
3680 TA08
3972f
21
Page 22
LT3972
PACKAGE DESCRIPTION
DD Package
10-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1699)
0.675 p0.05
3.50 p0.05
1.65 p0.05
(2 SIDES)2.15 p0.05
PACKAGE
OUTLINE
0.25 p 0.05
2.38 p0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
PIN 1
TOP MARK
(SEE NOTE 6)
0.200 REF
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-2).
CHECK THE LTC WEBSITE DATA SHEET FOR CURRENT STATUS OF VARIATION ASSIGNMENT
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE
TOP AND BOTTOM OF PACKAGE
0.50
BSC
(2 SIDES)
3.00 p0.10
(4 SIDES)
0.75 p0.05
1.65 p 0.10
(2 SIDES)
0.00 – 0.05
R = 0.115
TYP
2.38 p0.10
(2 SIDES)
BOTTOM VIEW—EXPOSED PAD
106
15
0.25 p 0.05
0.50 BSC
0.38 p 0.10
(DD) DFN 1103
22
3972f
Page 23
PACKAGE DESCRIPTION
2.794 p 0.102
(.110 p .004)
MSE Package
10-Lead Plastic MSOP, Exposed Die Pad
(Reference LTC DWG # 05-08-1664 Rev B)
BOTTOM VIEW OF
EXPOSED PAD OPTION
0.889 p 0.127
(.035 p .005)
LT3972
2.06 p 0.102
1
(.081 p .004)
1.83 p 0.102
(.072 p .004)
5.23
(.206)
MIN
0.305 p 0.038
(.0120 p .0015)
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
GAUGE PLANE
0.18
(.007)
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
DETAIL “A”
DETAIL “A”
2.083 p 0.102
(.082 p .004)
0.50
(.0197)
BSC
0o – 6o TYP
0.53 p 0.152
(.021 p .006)
3.20 – 3.45
(.126 – .136)
SEATING
PLANE
3.00 p 0.102
(.118 p .004)
(NOTE 3)
4.90 p 0.152
(.193 p .006)
1.10
(.043)
MAX
0.17 –0.27
(.007 – .011)
TYP
10
12
0.50
(.0197)
BSC
0.497 p 0.076
6
45
(.0196 p .003)
REF
3.00 p 0.102
(.118 p .004)
(NOTE 4)
0.86
(.034)
REF
0.1016 p 0.0508
(.004 p .002)
MSOP (MSE) 0307 REV B
8910
7
3
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
3972f
23
Page 24
LT3972
TYPICAL APPLICATION
1.2V Step-Down Converter
V
IN
3.6V TO 27V
ON OFF
4.7μF
17k
78.7k
470pF
D: ON SEMI MBRA340
L: NEC MPLC0730L3R3
V
IN
RUN/SS BOOST
V
C
LT3972
RT
PG
SYNC
GND
f = 500kHz
BD
SW
FB
100k
RELATED PARTS
PART NUMBER DESCRIPTION COMMENTS
LT1933 500mA (I
LT1936 36V, 1.4A (I
Converter
LT1940 Dual 25V, 1.4A (I
Converter
LT1976/LT1967 60V, 1.2A (I
DC/DC Converters with Burst Mode Operation
LT3434/LT3435 60V, 2.4A (I
DC/DC Converters with Burst Mode Operation
LT3437 60V, 400mA (I
Burst Mode Operation
LT3480 36V with Transient Protection to 60V, 2A (I
Effi ciency Step-Down DC/DC Converter with Burst Mode Operation
LT3481 34V with Transient Protection to 36V, 2A (I
Effi ciency Step-Down DC/DC Converter with Burst Mode Operation
LT3493 36V, 1.4A (I
DC/DC Converter
LT3505 36V with Transient Protection to 40V, 1.4A (I
High Effi ciency Step-Down DC/DC Converter
LT3508 36V with Transient Protection to 40V, Dual 1.4A (I
High Effi ciency Step-Down DC/DC Converter
LT3680 36V, 3.5A, 2.4MHz, Low Quiescent Current (<75μA) Step-Down
DC/DC Converter
LT3684 34V with Transient Protection to 36V, 2A (I
High Effi ciency Step-Down DC/DC Converter
LT3685 36V with Transient Protection to 60V, Dual 2A (I
High Effi ciency Step-Down DC/DC Converter
LT3693 36V, 3.5A, 2.4MHz, Step-Down DC/DC Converter V
Linear Technology Corporation
24
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
), 500kHz Step-Down Switching Regulator in SOT-23 VIN: 3.6V to 36V, V
OUT
), 500kHz, High Effi ciency Step-Down DC/DC
OUT
TM
ThinSOT
VIN: 3.6V to 36V, V
MS8E Package
), 1.1MHz, High Effi ciency Step-Down DC/DC
OUT
VIN: 3.6V to 25V, V
TSSOP16E Package
), 200kHz/500kHz, High Effi ciency Step-Down
OUT
VIN: 3.3V to 60V, V
TSSOP16E Package
), 200kHz/500kHz, High Effi ciency Step-Down
OUT
VIN: 3.3V to 60V, V
TSSOP16 Package
), Micropower Step-Down DC/DC Converter with
OUT
), 2.4MHz, High
OUT
), 2.8MHz, High
OUT
), 750kHz High Effi ciency Step-Down
OUT
), 3MHz,
OUT
), 3MHz,
OUT
VIN: 3.3V to 60V, V
3mm
× 3mm DFN10 and TSSOP16E Packages
VIN: 3.6V to 38V, V
3mm
× 3mm DFN10 and MSOP10E Packages
VIN: 3.6V to 34V, V
3mm
× 3mm DFN10 and MSOP10E Packages
VIN: 3.6V to 36V, V
2mm
× 3mm DFN6 Package
VIN: 3.6V to 34V, V
3mm
× 3mm DFN8 and MSOP8E Packages
VIN: 3.7V to 37V, V
4mm
× 4mm QFN24 and TSSOP16E Packages
: 3.6V to 36V, V
V
IN
3mm DFN10, MS10E Package
), 2.8MHz,
OUT
OUT
), 2.4MHz,
VIN: 3.6V to 34V, V
3mm
× 3mm DFN10 and MSOP10E Packages
VIN: 3.6V to 38V, V
3mm
× 3mm DFN10 and MSOP10E Packages
: 3.6V to 36V, V
IN
3mm DFN10, MS10E Package
●
www.linear.com
0.47μF
D
52.3k
Package
V
OUT
1.2V
3.5A
L
3.3μH
100μF
3680 TA09
= 1.2V, IQ = 1.6mA, ISD < 1μA,
OUT(MIN)
= 1.2V, IQ = 1.9mA, ISD < 1μA,
OUT(MIN)
= 1.2V, IQ = 3.8mA, ISD < 30μA,
OUT(MIN)
= 1.2V, IQ = 100μA, ISD < 1μA,
OUT(MIN)
= 1.2V, IQ = 100μA, ISD < 1μA,
OUT(MIN)
= 1.25V, IQ = 100μA, ISD < 1μA,
OUT(MIN)
= 0.78V, IQ = 70μA, ISD < 1μA,
OUT(MIN)
= 1.26V, IQ = 50μA, ISD < 1μA,
OUT(MIN)
= 0.8V, IQ = 1.9mA, ISD < 1μA,
OUT(MIN)
= 0.78V, IQ = 2mA, ISD = 2μA,
OUT(MIN)
= 0.8V, IQ = 4.6mA, ISD = 1μA,
OUT(MIN)
= 0.8V, IQ = 75μA, ISD < 1μA, 3mm ×
OUT(MIN)
= 1.26V, IQ = 850μA, ISD < 1μA,
OUT(MIN)
= 0.78V, IQ = 70μA, ISD < 1μA,
OUT(MIN)
= 0.8V, IQ = 1.3mA, ISD < 1μA, 3mm ×
OUT(MIN)
LT 0908 • PRINTED IN USA
© LINEAR TECHNOLOGY CORPORATION 2008
3972f
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