Datasheet LT1676 Datasheet (Linear Technology)

FEATURES

■

Wide Input Range: 7.4V to 60V

■

700mA Peak Switch Current Rating

■

Adaptive Switch Drive Maintains Efficiency at High
Load Without Pulse Skipping at Light Load

■

True Current Mode Control

■

100kHz Fixed Operating Frequency

■

Synchronizable to 250kHz

■

Low Supply Current in Shutdown: 30µA

■

Available in 8-Pin SO and PDIP Packages

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APPLICATIO S

■

Automotive DC/DC Converters

■

Telecom 48V Step-Down Converters

■

Cellular Phone Battery Charger Accessories

■

IEEE 1394 Step-Down Converters

LT1676

Wide Input Range,

High Efficiency, Step-Down

Switching Regulator

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DESCRIPTIO

The LT®1676 is a wide input range, high efficiency Buck
(step-down) switching regulator. The monolithic die in­cludes all oscillator, control and protection circuitry. The
part can accept input voltages as high as 60V and contains
an output switch rated at 700mA peak current. Current
mode control offers excellent dynamic input supply rejec­tion and short-circuit protection.

The LT1676 contains several features to enhance effi­ciency. The internal control circuitry is normally powered
via the VCC pin, thereby minimizing power drawn directly
from the VIN supply (see Applications Information). The
action of the LT1676 switch circuitry is also load depen­dent. At medium to high loads, the output switch circuitry
maintains high rise time for good efficiency. At light loads,
rise time is deliberately reduced to avoid pulse skipping
behavior.

TYPICAL APPLICATIO

V

IN

8V TO 50V

1

+

39µF
63V

6

SHDN

SYNC

V

IN

LT1676

GND

5

V

CC

V

SW

FB
V

C

4

U

2
3

7
8

2200pF

22k

Figure 1

220µH*

+

MBR160

100pF

*65T #30 ON MAGNETICS
MPP #55030

100µF
10V

The available SO-8 package and 100kHz switching fre­quency allow for minimal PC board area requirements.

, LTC and LT are registered trademarks of Linear Technology Corporation.

36.5k
1%

12.1k
1%

1676 F01

5V
400mA

Efficiency vs VIN and I

90

80

70

60

50

EFFICIENCY (%)

VIN = 12V

40

= 24V

V

IN

30

V

= 36V

IN

= 48V

V

IN

20

1

10 100 1000

I

(mA)

LOAD

LOAD

1676 TA01

1

LT1676

WW

W

ABSOLUTE MAXIMUM RA TIN GS

U

U

W

PACKAGE/ORDER INFORMA TION

(Note 1)

Supply Voltage ........................................................ 60V

Switch Voltage......................................................... 60V

SHDN, SYNC Pin Voltage........................................... 7V

VCC Pin Voltage ....................................................... 30V

FB Pin Voltage ........................................................... 3V

Operating Junction Temperature Range

LT1676C................................................0°C to 125°C

LT1676I............................................ –40°C to 125°C

Storage Temperature Range................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)..................300°C

TOP VIEW

1

SHDN

V

2

CC

V

3

SW

GND

4

N8 PACKAGE
8-LEAD PDIP

T

= 125°C, θJA = 130°C/W (N8)

JMAX

= 125°C, θJA = 110°C/W (S8)

T

JMAX

Consult factory for Military grade parts.

8-LEAD PLASTIC SO

V

8

C

FB

7

SYNC

6

V

5

IN

S8 PACKAGE

ORDER PART

NUMBER

LT1676CN8
LT1676CS8
LT1676IN8
LT1676IS8

S8 PART MARKING

1676
1676I

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.

VIN = 48V, VSW open, VCC = 5V, VC = 1.4V unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Power Supplies

V

IN(MIN)

I

VIN

I

VCC

V

VCC

Feedback Amplifier

V

REF

I

IN

g

m

I

, I

SRC

V

CL

Output Switch

V

ON

I

LIM

Current Amplifier

Minimum Input Voltage 6.7 7.0 V

● 7.4 V

VIN Supply Current VC = 0V 620 800 µA

● 900 µA

VCC Supply Current VC = 0V 3.2 4.0 mA

● 5.0 mA

VCC Dropout Voltage (Note 2) ● 2.8 3.1 V
Shutdown Mode I

Reference Voltage 1.225 1.240 1.255 V

FB Pin Input Bias Current 600 1500 nA
Feedback Amplifier Transconductance ∆lc = ±10µA 400 650 1000 µmho

Feedback Amplifier Source or Sink Current 60 100 170 µA

SNK

Feedback Amplifier Clamp Voltage 2.0 V
Reference Voltage Line Regulation 12V ≤ VIN ≤ 60V ● 0.01 %/V
Voltage Gain 200 600 V/V

Output Switch On Voltage ISW = 0.5A 1.0 1.5 V
Switch Current Limit (Note 3) ● 0.55 0.70 1.0 A

Control Pin Threshold Duty Cycle = 0% 0.9 1.1 1.25 V
Control Voltage to Switch Transconductance 2 A/V

VIN

V

= 0V 30 50 µA

SHDN

● 75 µA

● 1.215 1.265 V

● 200 1500 µmho

● 45 220 µA

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2

LT1676

DUTY CYCLE (%)

0

SWITCH CURRENT LIMIT (mA)

1000

800

600

400

200

0

80

1676 G03

2010 30 50 70 90

40

60

100

TA = 25°C

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C.

VIN = 48V, VSW open, VCC = 5V, VC = 1.4V unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Timing

f Switching Frequency 90 100 110 kHz

● 85 115 kHz

Maximum Switch Duty Cycle ● 85 90 %

t

ON(MIN)

Boost Operation

Sync Function

SHDN Pin Function

V

SHDN

I

SHDN

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Note 2: Control circuitry powered from V

Minimum Switch On Time High dV/dt Mode, RL = 50Ω (Note 4) 300 ns

VC Pin Boost Threshold 1.35 V
dV/dt Below Threshold 0.2 V/ns
dV/dt Above Threshold 1.6 V/ns

Minimum Sync Amplitude ● 1.5 2.2 V
Synchronization Range ● 130 250 kHz
SYNC Pin Input R 40 kΩ

Shutdown Mode Threshold 0.5 V

● 0.2 0.8 V

Upper Lockout Threshold Switching Action On 1.260 V
Lower Lockout Threshold Switching Action Off 1.245 V
Shutdown Pin Current V

= 0V 12 20 µA

SHDN

V

= 1.25V 2.5 10 µA

SHDN

Note 3: Switch current limit is DC trimmed and tested in production.
Inductor dl/dt rate will cause a somewhat higher current limit in actual

CC

.

application.
Note 4: Minimum switch on time is production tested with a 50Ω resistive

load to ground.

TYPICAL PERFORMANCE CHAR ACTERISTICS

Minimum Input Voltage vs
Temperature

7.4

7.2

7.0

6.8

6.6

INPUT VOLTAGE (V)

6.4

6.2

6.0
–50

–25 0

50 100 125

25 75

TEMPERATURE (°C)

LT1676 G01

UW

1.50

1.25

1.00

0.75

0.50

SWITCH VOLTAGE (V)

0.25

0

Switch-On Voltage vs
Switch Current

25°C

400 600 700

300 500

0

–55°C

125°C

100 200

SWITCH CURRENT (mA)

Switch Current Limit vs
Duty Cycle

1676 G02

3

LT1676

UW

TYPICAL PERFORMANCE CHAR ACTERISTICS

SHDN Pin Shutdown Threshold
vs Temperature

900

800

700

600

500

400

SHDN PIN VOLTAGE (mV)

300

200

–50

–25 0

TEMPERATURE (°C)

50 100 125

25 75

Switching Frequency
vs Temperature

106

104

102

LT1676 G04

SHDN Pin Input Current
vs Voltage

5

0

–5

–10

–15

SHDN PIN INPUT CURRENT (µA)

–20

1

0

SHDN PIN VOLTAGE (V)

3

2

Minimum Synchronization Voltage
vs Temperature

2.25

2.00

1.75

25°C
–55°C
125°C

4

1676 G05

SHDN Pin Lockout Thresholds
vs Temperature

1.30

1.28
UPPER THRESHOLD

1.26

LOWER THRESHOLD

1.24

SHDN PIN VOLTAGE (V)

1.22

1.20

–50

5

–25 0

TEMPERATURE (°C)

50 100 125

25 75

LT1676 G06

Switch Minimum On-Time
vs Temperature

600

V

= 48V

IN

= 50Ω

R

L

500

FB =

400

100

98

SWITCHING FREQUENCY (kHz)

96

94

–50 25 75

–25 0

TEMPERATURE (°C)

50 100 125

VC Pin Switching Threshold,
Boost Threshold, Clamp Voltage
vs Temperature

2.2

2.0

1.8

1.6

1.4

PIN VOLTAGE (V)

C

V

1.2

1.0

0.8
–50

–25 0

25 75

TEMPERATURE (°C)

CLAMP

VOLTAGE

THRESHOLD

SWITCHING

THRESHOLD

50 100 125

BOOST

1676 G07

LT1676 G10

1.50

1.25

1.00

MINIMUM SYNCHRONIZATION VOLTAGE (V)

0.75
–50 25 75

–25 0

TEMPERATURE (°C)

50 100 125

Feedback Amplifier Output
Current vs FB Pin Voltage

100

50

0

–50

–100

FEEDBACK AMPLIFIER OUTPUT CURRENT (µA)

–150

1.0

1.1

1.2

FB PIN VOLTAGE (V)

1.3

25°C
–55°C
125°C

1.4

1676 G08

1676 G11

1.5

300

200

100

SWITCH MINIMUM ON-TIME (ns)

0

–50 25 75

–25 0

TEMPERATURE (°C)

50 100 125

Error Amplifier Transconductance
vs Temperature

750

700

650

600

550

500

TRANSCONDUCTANCE (µmho)

450

400

–50

–25 0

TEMPERATURE (°C)

50 100 125

25 75

1676 G09

LT1676 G12

4

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PIN FUNCTIONS

LT1676

SHDN (Pin 1):

When pulled below the shutdown mode
threshold, nominally 0.30V, this pin turns off the regula­tor and reduces VIN input current to a few tens of micro­amperes (shutdown mode).

When this pin is held above the shutdown mode thresh­old, but below the lockout threshold, the part will be
operational with the exception that output switching
action will be inhibited (lockout mode). A user-adjustable
undervoltage lockout can be implemented by driving this
pin from an external resistor divider to VIN. This action is
logically “ANDed” with the internal UVLO, set at nominally

6.7V, such that minimum VIN can be increased above

6.7V, but not decreased (see Applications Information).
If unused, this pin should be left open. However, the high

impedance nature of this pin renders it susceptible to
coupling from the high speed VSW node, so a small
capacitor to ground, typically 100pF or so is recom­mended when the pin is left “open.”

VCC (Pin 2): This pin is used to power the internal control
circuitry off of the switching supply output. Proper use of
this pin enhances overall power supply efficiency. During
start-up conditions, internal control circuitry is powered
directly from VIN. If the output capacitor is located more
than an inch from the VCC pin, a separate 0.1µF bypass
capacitor to ground may be required right at the pin.

VSW (Pin 3): This is the emitter node of the output switch
and has large currents flowing through it. This node
moves at a high dV/dt rate, especially when in “boost”
mode. Keep the traces to the switching components as

short as possible to minimize electromagnetic radiation
and voltage spikes.

GND (Pin 4): This is the device ground pin. The internal
reference and feedback amplifier are referred to it. Keep
the ground path connection to the FB divider and the V

C

compensation capacitor free of large ground currents.
VIN (Pin 5): This is the high voltage supply pin for the

output switch. It also supplies power to the internal control
circuitry during start-up conditions or if the VCC pin is left
open. A high quality bypass capacitor that meets the input
ripple current requirements is needed here. (See Applica­tions Information.)

SYNC (Pin 6): Pin used to synchronize internal oscillator
to the external frequency reference. It is directly logic
compatible and can be driven with any signal between
10% and 90% duty cycle. The sync function is internally
disabled if the FB pin voltage is low enough to cause
oscillator slowdown. If unused, this pin should be grounded.

FB (Pin 7): This is the inverting input to the feedback
amplifier. The noninverting input of this amplifier is inter­nally tied to the 1.24V reference. This pin also slows down
the frequency of the internal oscillator when its voltage is
abnormally low, e.g., 2/3 of normal or less. This feature
helps maintain proper short-circuit protection.

VC (Pin 8): This is the control voltage pin which is the
output of the feedback amplifier and the input of the
current comparator. Frequency compensation of the over­all loop is effected by placing a capacitor, (or in most cases
a series RC combination) between this node and ground.

WUW

TIMING DIAGRAMS

High dV/dt Mode Low dV/dt Mode

V

IN

V

SW

0

SWDR

SWON

BOOST

SWOFF

1676 TD01

V

V

SW

SWDR

SWON

BOOST

SWOFF

IN

0

1676 TD02

5

LT1676

BLOCK DIAGRA

2

V

CC

SHDN

SYNC

GND

1

6

4

8

V

C

7

FB

V

BG

BIAS

gm

FB
AMP

W

V

B

V

BG

BOOST

COMP

V

5

IN

R1 R

Q3

Q2

Q5

SWON

BOOST

I

COMP

Q4

I

I I

SWOFF

SWDR

SWDR

LOGICOSC

V

TH

SWON
BOOST
SWOFF

I

SENSE

Q1

V

3

SW

D1

1676 BD

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OPERATIO

The LT1676 is a current mode switching regulator IC that
has been optimized for high efficiency operation in high
input voltage, low output voltage Buck topologies. The
Block Diagram shows an overall view of the system.
Several of the blocks are straightforward and similar to
those found in traditional designs, including: Internal Bias
Regulator, Oscillator and Feedback Amplifier. The novel
portion includes an elaborate Output Switch section and
Logic Section to provide the control signals required by
the switch section.

The LT1676 operates much the same as traditional
current mode switchers, the major difference being its
specialized output switch section. Due to space con­straints, this discussion will not reiterate the basics of
current mode switcher/controllers and the “Buck” topol­ogy. A good source of information on these topics is
Application Note 19.

Output Switch Theory

One of the classic problems in delivering low output
voltage from high input voltage at good efficiency is that
minimizing AC switching losses requires very fast volt­age (dV/dt) and current (dI/dt) transition at the output
device. This is in spite of the fact that in a bipolar
implementation, slow lateral PNPs must be included in
the switching signal path.

Fast positive-going slew rate action is provided by lateral
PNP Q3 driving the Darlington arrangement of Q1 and Q2.
The extra β available from Q2 greatly reduces the drive
requirements of Q3.

Although desirable for dynamic reasons, this topology
alone will yield a large DC forward voltage drop. A second
lateral PNP, Q4, acts directly on the base of Q1 to reduce
the voltage drop after the slewing phase has taken place.
To achieve the desired high slew rate, PNPs Q3 and Q4 are
“force-fed” packets of charge via the current sources
controlled by the boost signal.

6

OPERATIO

L

V

fI

VV

V

OUT

PK

IN OUT

IN

=





















•

–

LT1676

U

Please refer to the High dV/dt Mode Timing Diagram. A
typical oscillator cycle is as follows: The logic section first
generates an SWDR signal that powers up the current
comparator and allows it time to settle. About 1µs later, the
SWON signal is asserted and the BOOST signal is pulsed
for a few hundred nanoseconds. After a short delay, the
VSW pin slews rapidly to VIN. Later, after the peak switch
current indicated by the control voltage VC has been
reached (current mode control), the SWON and SWDR
signals are turned off, and SWOFF is pulsed for several
hundred nanoseconds. The use of an explicit turn-off
device, i.e., Q5, improves turn-off response time and thus
aids both controllability and efficiency.

The system as previously described handles heavy loads
(continuous mode) at good efficiency, but it is actually
counterproductive for light loads. The method of jam­ming charge into the PNP bases makes it difficult to turn
them off rapidly and achieve the very short switch ON
times required by light loads in discontinuous mode.
Further
adversely affects light load controllability.

The solution is to employ a “boost comparator” whose
inputs are the VC control voltage and a fixed internal

more, the high leading edge dV/dt rate similarly

threshold reference, VTH. (Remember that in a current
mode switching topology, the VC voltage determines the
peak switch current.) When the VC signal is above VTH, the
previously described “high dV/dt” action is performed.
When the VC signal is below VTH, the boost pulses are
absent, as can be seen in the Low dV/dt Mode Timing
Diagram. Now the DC current, activated by the SWON
signal alone, drives Q4 and this transistor drives Q1 by
itself. The absence of a boost pulse, plus the lack of a
second NPN driver, result in a much lower slew rate which
aids light load controllability.

A further aid to overall efficiency is provided by the
specialized bias regulator circuit, which has a pair of
inputs, VIN and VCC. The VCC pin is normally connected to
the switching supply output. During start-up conditions,
the LT1676 powers itself directly from VIN. However, after
the switching supply output voltage reaches about 2.9V,
the bias regulator uses this supply as its input. Previous
generation Buck controller ICs without this provision
typically required hundreds of milliwatts of quiescent
power when operating at high input voltage. This both
degraded efficiency and limited available output current
due to internal heating.

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APPLICATIONS INFORMATION

Selecting a Power Inductor

There are several parameters to consider when selecting
a power inductor. These include inductance value, peak
current rating (to avoid core saturation), DC resistance,
construction type, physical size, and of course, cost.

In a typical application, proper inductance value is dictated
by matching the discontinuous/continuous crossover point
with the LT1676 internal low-to-high dV/dt threshold. This
is the best compromise between maintaining control with
light loads while maintaining good efficiency with heavy
loads. The fixed internal dV/dt threshold has a nominal
value of 1.4V, which referred to the VC pin threshold and
control voltage to switch transconductance, corresponds
to a peak current of about 200mA. Standard Buck con­verter theory yields the following expression for induc­tance at the discontinuous/continuous crossover:

For example, substituting 48V, 5V, 200mA and 100kHz
respectively for VIN, V
220µH. Note that the left half of this expression is indepen-
dent of input voltage while the right half is only a weak
function of VIN when VIN is much greater than V
means that a single inductor value will work well over a
range of “high” input voltage. And although a progres­sively smaller inductor is suggested as VIN begins to
approach V
under these conditions are much more forgiving with
respect to controllability and efficiency issues. Therefore
when a wide input voltage range must be accommodated,
say 10V to 50V for 5V
inductance value based on the maximum input voltage.

, note that the much higher ON duty cycles

OUT

, IPK and f yields a value of about

OUT

OUT

, the user should choose an

OUT

. This

7

LT1676

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APPLICATIONS INFORMATION

Once the inductance value is decided, inductor peak
current rating and resistance need to be considered. Here,
the inductor peak current rating refers to the onset of
saturation in the core material, although manufacturers
sometimes specify a “peak current rating” which is
derived from a worst-case combination of core saturation
and self-heating effects. Inductor winding resistance alone
limits the inductor’s current carrying capability as the I2R
power threatens to overheat the inductor. If applicable,
remember to include the condition of output short circuit.
Although the peak current rating of the inductor can be
exceeded in short-circuit operation, as core saturation per
se is not destructive to the core, excess resistive self­heating is still a potential problem.

The final inductor selection is generally based on cost,
which usually translates into choosing the smallest physi­cal size part that meets the desired inductance value,
resistance and current carrying capability. An additional
factor to consider is that of physical construction. Briefly
stated, “open” inductors built on a rod- or barrel-shaped
core generally offer the smallest physical size and lowest
cost. However their open construction does not contain
the resulting magnetic field, and they may not be accept­able in RFI-sensitive applications. Toroidal style induc­tors, many available in surface mount configuration, offer
improved RFI performance, generally at an increase in
cost and physical size. And although custom design is
always a possibility, most potential LT1676 applications
can be handled by the array of standard, off-the-shelf
inductor products offered by the major suppliers.

Selecting Freewheeling Diode

Highest efficiency operation requires the use of a Schottky
type diode. DC switching losses are minimized due to its
low forward voltage drop, and AC behavior is benign due
to its lack of a significant reverse recovery time. Schottky
diodes are generally available with reverse voltage ratings
of 60V and even 100V, and are price competitive with other
types.

The use of so-called “ultrafast” recovery diodes is gener­ally not recommended. When operating in continuous
mode, the reverse recovery time exhibited by “ultrafast”
diodes will result in a slingshot type effect. The power

internal switch will ramp up VIN current into the diode in an
attempt to get it to recover. Then, when the diode has
finally turned off, some tens of nanoseconds later, the V
node voltage ramps up at an extremely high dV/dt, per­haps 5 to even 10V/ns! With real world lead inductances,
the VSW node can easily overshoot the VIN rail. This can
result in poor RFI behavior and if the overshoot is severe
enough, damage the IC itself.

Selecting Bypass Capacitors

The basic topology as shown in Figure 1 uses two bypass
capacitors, one for the VIN input supply and one for the
V

output supply.

OUT

User selection of an appropriate output capacitor is rela­tively easy, as this capacitor sees only the AC ripple current
in the inductor. As the LT1676 is designed for Buck or
step-down applications, output voltage will nearly always
be compatible with tantalum type capacitors, which are
generally available in ratings up to 35V or so. These
tantalum types offer good volumetric efficiency and many
are available with specified ESR performance. The product
of inductor AC ripple current and output capacitor ESR will
manifest itself as peak-to-peak voltage ripple on the output
node. (Note: If this ripple becomes too large, heavier
control loop compensation, at least at the switching fre­quency, may be required on the VC pin.) The most
demanding applications, requiring very low output ripple,
may be best served not with a single extremely large
output capacitor, but instead by the common technique of
a separate L/C lowpass post filter in series with the output.
(In this case, “Two caps are better than one.”)

The input bypass capacitor is normally a more difficult
choice. In a typical application e.g., 48VIN to 5V
relatively heavy VIN current is drawn by the power switch
for only a small portion of the oscillator period (low ON
duty cycle). The resulting RMS ripple current, for which
the capacitor must be rated, is often several times the DC
average VIN current. Similarly, the “glitch” seen on the V
supply as the power switch turns on and off will be related
to the product of capacitor ESR, and the relatively high
instantaneous current drawn by the switch. To compound
these problems is the fact that most of these applications
will be designed for a relatively high input voltage, for

SW

OUT

,

IN

8

LT1676

FB DIVIDER THEVENIN VOLTAGE (V)

0

0

f

OSC

(kHz)

20

40

60

80

100

120

0.25 0.50 0.75 1.00

1676 F02

1.25

R

TH

LT1676

FB

RTH = 22k

RTH = 4.7kRTH = 10k

U

WUU

APPLICATIONS INFORMATION

which tantalum capacitors are generally unavailable. Rela­tively bulky “high frequency” aluminum electrolytic types,
specifically constructed and rated for switching supply
applications, may be the only choice.

Minimum Load Considerations

As discussed previously, a lightly loaded LT1676 with V
pin control voltage below the boost threshold will operate
in low dV/dt mode. This affords greater controllability at
light loads, as minimum tON requirements are relaxed. In
many applications, it is possible to operate the LT1676
down to zero external load without “pulse skipping”!
In these cases, the LT1676’s modest VCC current
requirement of several milliamperes provides enough of a
load to avoid pulse skipping.

However, some users may be indifferent to pulse skipping
behavior, but instead may be concerned with maintaining
maximum possible efficiency at light loads. This require­ment can be satisfied by forcing the part into Burst Mode
operation. The use of an external comparator whose
output controls the shutdown pin allows high efficiency at
light loads through Burst Mode operation behavior (see
Typical Applications and Figure 8).

Maximum Load/Short-Circuit Considerations

The LT1676 is a current mode controller. It uses the V
node voltage as an input to a current comparator which
turns off the output switch on a cycle-by-cycle basis as
this peak current is reached. The internal clamp on the V
node, nominally 2V, then acts as an output switch peak
current limit. This action becomes the switch current limit
specification. The maximum available output power is
then determined by the switch current limit.

C

TM

C

C

t

ON(MIN)

. When combined with the large ratio of VIN to
(VF + I • R), the diode forward voltage plus inductor I • R
voltage drop, the potential exists for a loss of control.
Expressed mathematically the requirement to maintain
control is:

VIR

•≤+

ft

•

ON

F

V

IN

where:
f = switching frequency
tON = switch ON time
VF = diode forward voltage
VIN = Input voltage
I • R = inductor I • R voltage drop

If this condition is not observed, the current will not be
limited at IPK, but will cycle-by-cycle ratchet up to some
higher value. Using the nominal LT1676 clock frequency
of 100KHz, a VIN of 48V and a (VF + I • R) of say 0.7V, the
maximum tON to maintain control would be approximately
140ns, an unacceptably short time.

The solution to this dilemma is to slow down the oscillator
when the FB pin voltage is abnormally low thereby indicat­ing some sort of short-circuit condition. Figure 2 shows
the typical response of Oscillator Frequency vs FB divider
Thevenin voltage and impedance. Oscillator frequency is
unaffected until FB voltage drops to about 2/3 of its normal
value. Below this point the oscillator frequency decreases
roughly linearly down to a limit of about 25kHz. This lower

A potential controllability problem could occur under
short-circuit conditions. If the power supply output is
short circuited, the feedback amplifier responds to the low
output voltage by raising the control voltage, VC, to its
peak current limit value. Ideally, the output switch would
be turned on, and then turned off as its current exceeded
the value indicated by VC. However, there is finite response
time involved in both the current comparator and turnoff
of the output switch. These result in a minimum on time

Burst Mode is a trademark of Linear Technology Corporation.

Figure 2. Oscillator Frequency vs FB Divider
Thevenin Voltage and Impedance

9

LT1676

U

WUU

APPLICATIONS INFORMATION

oscillator frequency during short-circuit conditions can
then maintain control with the effective minimum ON time.

A further potential problem with short-circuit operation
might occur if the user were operating the part with its
oscillator slaved to an external frequency source via the
SYNC pin. However, the LT1676 has circuitry that auto­matically disables the sync function when the oscillator is
slowed down due to abnormally low FB voltage.

Feedback Divider Considerations

An LT1676 application typically includes a resistive divider
between V
the FB pin to the reference voltage V
a fixed ratio between the two resistors, but a second
degree of freedom is offered by the overall impedance
level of the resistor pair. The most obvious effect this has
is one of efficiency—a higher resistance feedback divider
will waste less power and offer somewhat higher effi­ciency, especially at light load.

However, remember that oscillator slowdown to achieve
short-circuit protection (discussed above) is dependent
on FB pin behavior, and this in turn, is sensitive to FB node
external impedance. Figure 2 shows the typical relation­ship between FB divider Thevenin voltage and impedance,
and oscillator frequency. This shows that as feedback
network impedance increases beyond 10k, complete os­cillator slowdown is not achieved, and short-circuit pro­tection may be compromised. And as a practical matter,
the product of FB pin bias current and larger FB network
impedances will cause increasing output voltage error.
(Nominal cancellation for 10k of FB Thevenin impedance
is included internally.)

Thermal Considerations

Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces­sive die temperatures. The packages are rated at 110°C/W
for the 8-pin SO (S8) and 130°C/W for 8-pin PDIP (N8).

Quiescent power is given by:

PQ = I

(This assumes that the VCC pin is connected to V

and ground, the center node of which drives

OUT

. This establishes

REF

• VIN + I

VIN

VCC

• V

OUT

OUT

.)

Power loss internal to the LT1676 related to actual output
current is composed of both DC and AC switching losses.
These can be roughly estimated as follows:

DC switching losses are dominated by output switch “ON
voltage”, i.e.,

PDC = V
VON = Output switch ON voltage, typically 1V at 500mA

I

OUT

DC = ON duty cycle

AC switching losses are typically dominated by power lost
due to the finite rise time and fall time at the VSW node.
Assuming, for simplicity, a linear ramp up of both voltage
and current and a current rise/fall time equal to 15ns,

PAC = 1/2 • V
tr = (VIN/1.6)ns in high dV/dt mode

(VIN/0.16)ns in low dV/dt mode
tf = (VIN/1.6)ns (irrespective of dV/dt mode)
f = switching frequency

Total power dissipation of the die is simply the sum of
quiescent, DC and AC losses previously calculated.

P

D(TOTAL)

Frequency Compensation

Loop frequency compensation is performed by connect­ing a capacitor, or in most cases a series RC, from the
output of the error amplifier (VC pin) to ground. Proper
loop compensation may be obtained by empirical meth­ods as described in detail in Application Note 19. Briefly,
this involves applying a load transient and observing the
dynamic response over the expected range of VIN and
I

values.

LOAD

As a practical matter, a second small capacitor, directly
from the VC pin to ground is generally recommended to
attenuate capacitive coupling from the V
value for this capacitor is 100pF. (See Switch Node Con­siderations).

Switch Node Considerations

For maximum efficiency, switch rise and fall times are
made as short as practical. To prevent radiation and high

• I

OUT

• I

IN

• DC

OUT

ON

= Output current

= PQ + PDC + P

• (tr + tf + 30ns) • f

AC

SW

pin. A typical

10

LT1676

U

WUU

APPLICATIONS INFORMATION

frequency resonance problems, proper layout of the com­ponents connected to the IC is essential, especially the
power path. B field (magnetic) radiation is minimized by
keeping output diode, switch pin and intput bypass
capacitor leads as short as possible. E field radiation is
kept low by minimizing the length and area of all traces
connected to the switch pin (VSW). A ground plane should
always be used under the switcher circuitry to prevent
interplane coupling.

The high speed switching current path is shown schemati­cally in Figure 3. Minimum lead length in these paths is
essential to ensure clean switching and minimal EMI. The
paths containing the input capacitor, output switch and
output diode are the only ones containing nanosecond rise
and fall times. Keep these paths as short as possible.

Additionally, it is possible for the LT1676 to cause EMI
problems by “coupling to itself”. Specifically, this can
occur if the VSW pin is allowed to capacitively couple in an
uncontrolled manner to the part’s high impedance nodes,

i.e., SHDN, SYNC, VC and FB. This can cause erratic
operation such as odd/even cycle behavior, pulse width
“nervousness”, improper output voltage and/or prema­ture current limit action.

As an example, assume that the capacitance between the
VSW node and a high impedance pin node is 0.1pF, and
further assume that the high impedance node in question
exhibits a capacitance of 1pF to ground. Due to the high
dV/dt, large excursion behavior of the VSW node, this will
couple a nearly 5V transient to the high impedance pin,
causing abnormal operation. (This assumes the “typical”
48VIN to 5V

application.) An explicit 100pF capacitor

OUT

added to the node will reduce the amplitude of the distur­bance to more like 50mV (although settling time will
increase).

Specific pin recommendations are as follows:

SHDN: If unused, add a 100pF capacitor to ground.
SYNC: Ground if unused.

V

IN

+

V

IN

Figure 3. High Speed Current Switching Paths

LT1676

C1

V

SW

L1

+

D1

V

OUT

C2

1676 F03

U

TYPICAL APPLICATIONS

Minimum Component Count Application

Figure 4a shows a basic “minimum component count”
application. The circuit produces 5V at up to 500mA I
with input voltages in the range of 12V to 48V. The typical
P

OUT/PIN

efficiency is shown in Figure 4b. No pulse
skipping is observed down to zero external load. As
shown, the SHDN and SYNC pins are unused, however
either (or both) can be optionally driven by external signals
as desired.

OUT

VC: Add a capacitor directly to ground in addition to the
explicit compensation network. A value of one-tenth of
the main compensation capacitor is recommended, up
to a maximum of 100pF.

FB: Assuming the VC pin is handled properly, this pin
usually requires no explicit capacitor of its own, but
keep this node physically small to minimize stray ca­pacitance.

User Programmable Undervoltage Lockout

Figure 5 adds a resistor divider to the basic application.
This is a simple, cost-effective way to add a user-program­mable undervoltage lockout (UVLO) function. Resistor R5
is chosen to have approximately 200µA through it at the
nominal SHDN pin lockout threshold of roughly 1.25V.
The somewhat arbitrary value of 200µA was chosen to be
significantly above the SHDN pin input current to minimize
its error contribution, but significantly below the typical

3.2mA the LT1676 draws in lockout mode. Resistor R4 is
then chosen to yield this same 200µA, less 2.5µA, with the

11

LT1676

I

LOAD

(mA)

1

60

EFFICIENCY (%)

70

80

90

10 100 1000

1676 F04b

50

40

30

20

VIN = 12V
V

IN

= 24V

V

IN

= 36V

V

IN

= 48V

TYPICAL APPLICATIONS

V

IN

12V TO

48V

1

+

C1
39µF
63V

C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO
D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224

C5
100pF

6

SHDN

SYNC

V

IN

LT1676

GND

5

2

V

CC

3

V

SW

7

FB

8

V

C

4

U

D1
MBRS1100

C3
2200pF
X7R

R3
22k
5%

FOR 3.3V V
R1: 24.3K, R2: 14.7k
L1: 150µH, DO3316P-154

: 0mA TO 500mA

I

OUT

VERSION:

OUT

220µH

L1

C4
100pF

V

OUT

+

C2
100µF
10V

R1

36.5k
1%

R2

12.1k
1%

1676 F04a

5V
0mA to 500mA

desired VIN UVLO voltage minus 1.25V applied across it.
(The 2.5µA factor is an allowance to minimize error due to
SHDN pin input current.)

Behavior is as follows: Normal operation is observed at the
nominal input voltage of 48V. As the input voltage is
decreased to roughly 43V, switching action will stop, V
will drop to zero, and the LT1676 will draw its VIN and V
quiescent currents from the VIN supply. At a much lower
input voltage, typically 18V or so at 25°C, the voltage on

12

Figure 4a. Minimum Component Count Application

Figure 4b. P

V

IN

R4
210k
1%

R5

6.19k
1%

+

C1

1

C5

6

SHDN

SYNC

V

IN

LT1676

GND

5

2

V

CC

3

V

SW

7

FB

8

V

C

C3

4

R3

L1

+

D1

C4

C2

R1

R2

1676 F05

V

OUT

OUT/PIN

Efficiency

Figure 5. User Programmable Undervoltage Lockout

the SHDN pin will drop to the shutdown threshold, and the
part will draw its shutdown current only from the VIN rail.
The resistive divider of R4 and R5 will continue to draw
power from VIN. (The user should be aware that while the

OUT

CC

SHDN pin
ing temperature effects, the SHDN pin
old is more coarse, and exhibits considerably more
temperature drift. Nevertheless the shutdown threshold

lockout

threshold is relatively accurate includ-

shutdown

thresh-

will always be well below the lockout threshold.)

U

TYPICAL APPLICATIONS

LT1676

Micropower Undervoltage Lockout

Certain applications may require very low current drain
when in undervoltage lockout mode. This can be accom­plished with the addition of a few more external compo­nents. Figure 6 shows an LTC®1440 micropower
comparator/reference added to control the LT1676 via its
SHDN pin. The extremely low input bias current of the
CMOS comparator allows the impedance of the resistor
divider R4/R5 to be increased, thereby minimizing power
drain. Hysteresis is externally programmable via resistor
divider R6/R7. The LTC1440 output directly controls the
LT1676 via its shutdown pin, driving it to either 5V (ON) or
0V (Full Shutdown). A simple linear voltage regulator to
power the LTC1440 is provided by Q1, Q2 and R7. Just
below the UVLO threshold, nominally 43V, total current
drain is typically 50µA.

Burst Mode Operation Configuration

Figure 4b demonstrates that power supply efficiency de­grades with lower output load current. This is not surpris­ing, as the LT1676 itself represents a fixed power overhead.

A possible way to improve light load efficiency is in Burst
Mode operation.

Figure 7 shows the LT1676 configured for Burst Mode
operation. Output voltage regulation is now provided in a
“bang-bang” digital manner, via comparator U2, an
LTC1440. Resistor divider R3/R4 provides a scaled ver­sion of the output voltage, which is compared against U2’s
internal reference. Intentional hysteresis is set by the R5/
R6 divider. As the output voltage falls below the regulation
range, the LT1676 is turned on. The output voltage rises,
and as it climbs above the regulation range, the LT1676 is
turned off. Efficiency is maximized, as the LT1676 is only
powered up while it is providing heavy output current.
Figure 7b shows that efficiency is typically maintained at
75% or better down to a load current of 10mA. Even at a
load of 1mA, efficiency is still a respectable 59% to 68%,
depending on VIN.

Resistor divider R1/R2 is still present, but does not
directly influence output voltage. It is chosen to ensure
that the LT1676 delivers high output current throughout
the voltage regulation range. Its presence is also required

NC

R8
10M

Q2
2N2369

Q1
PN2484

V

IN

5

V

IN

4

7

+

HYST
GND

V

V

IN
IN

REF

SW

2

CC

3

L1

+

D1
MBRS1100

7

FB

8

V

C

+
–

12

C3

2200pF
R3
22k

3
4

6
5

R6
22k

R7

2.4M

220µH

C4
100pF

C2
100µF
10V

V

IN

C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080

R4

C4, C5: X7R OR COG/NPO

6.8M

D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
L1: COILCRAFT DO3316P-224

R5
240k

1676 F06

R1

36.5
1%

R2

12.1k
1%

V

OUT

5V

6

SYNC

+

C1
39µF
63V

U1

LT1676

1

SHDN

GND

V

8

OUT

U2

LTC1440

–

V

Figure 6. Micropower Undervoltage Lockout

13

LT1676

U

TYPICAL APPLICATIONS

to maintain proper short-circuit protection. Transistors
Q1, Q2 and resistor R7 form a high VIN, low quiescent
current voltage regulator to power U2.

Burst Mode Operation Configuration with UVLO

Figure 7a uses an external comparator to control the
LT1676 via its SHDN pin. As such, the user’s ability to set
an undervoltage lockout (UVLO) threshold with a resistor
divider from VIN to SHDN pin to ground is lost. This ability
is regained in the slightly more complicated circuit shown
in Figure 8.

A dual comparator, the LTC1442, replaces the previous
single comparator. The second comparator monitors a
resistive divider between VIN and ground to provide the
(user-adjustable) UVLO function. The two comparator
outputs are logically combined in a CMOS NOR gate (U3)
to drive the LT1676 SHDN pin.

V

IN

SYNC

SHDN

OUT

LTC1440

V

V

IN

U1

LT1676

GND

+

V

U2

–

5

V

4

7

HYST

GND

V

IN
IN

REF

SW

2

CC

3

7

FB

8

V

C

C3
100pF

3

+

4

–

6
5

12

R5
22k

R6

2.4M

L1

+

D1

C2

+

C1

R7
10M

Q1
PN2484

Q2
2N2369

NC

C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO3316-224

6

1

8

Minimum Size Inductor Application

Figure 4a employs power path parts that are capable of
delivering the full rated output capability of the LT1676.
Potential users with low output current requirements may
be interested in substituting a physically smaller and less
costly power inductor. The circuit shown on the last page
of this data sheet is topologically identical to the basic
application, but specifies a much smaller inductor, and, a
somewhat smaller input electrolytic capacitor. This circuit
is capable of delivering up to 150mA at 5V, or, up to 200mA
at 3.3V. The only disadvantage is that due to the increased
resistance in the inductor, the circuit is no longer capable
of withstanding indefinite short circuits to ground. The
LT1676 will still current limit at its nominal I

value, but

LIM

this will overheat the inductor. Momentary short circuits
of a few seconds or less can still be tolerated.

V

OUT

R3
323k
1%

R4
100k
1%

1676 F07a

5V

90

80

VIN = 12V

70

60

50

EFFICIENCY (%)

40

30

20

1

V

IN

VIN = 48V

VIN = 36V

= 24V

10 100 1000

I

(mA)

LOAD

1676 F07b

R1
39k
5%

R2
10k
5%

14

(a)

Figure 7. Burst Mode Operation Configuration for High Efficiency at Light Load

(b)

U

TYPICAL APPLICATIONS

R7
10M

Q1
PN2484

Q2
2N2369

NC

C1: PANASONIC HFQ 39µF AT 63V
C2: AVX D CASE 100µF 10V
TPSD107M010R0080
D1: MOTOROLA 100V, 1A,
SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO3316-224

Figure 8. Burst Mode Operation Configuration with Micropower UVLO

4

7S02

LT1676

V

IN

5

V

IN

6

+

C1

5

U3

3

SYNC

LT1676

1

SHDN

GND

OUTA

1

2

LTC1442

OUTB

2

V

CC

3

V

SW

U1

7

FB

8

V

C

R5
22k

C3

4

+

V

+

INA

–

INB

U2

REF

HYST

–

V

R6

2.4M

L1

+

D1

V

IN

R8

6.8M

R9
240k

R1

C2

39k

R2
10k

R3
323k
1%

R4
100k
1%

1676 F08

V

OUT

5V

PACKAGE DESCRIPTION

0.300 – 0.325

(7.620 – 8.255)

0.065

(1.651)

0.009 – 0.015

(0.229 – 0.381)

+0.035

0.325

–0.015
+0.889

8.255

()

–0.381

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

×

°

45

0.016 – 0.050

0.406 – 1.270

TYP

(2.540 ± 0.254)

0°– 8° TYP

U

Dimensions in inches (millimeters) unless otherwise noted.

N8 Package

8-Lead PDIP (Narrow 0.300)

(LTC DWG # 05-08-1510)

0.045 – 0.065

(1.143 – 1.651)

0.100 ± 0.010

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

0.130 ± 0.005

(3.302 ± 0.127)

0.125

(3.175)

MIN

0.018 ± 0.003

(0.457 ± 0.076)

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

TYP

0.020

(0.508)

MIN

N8 1197

0.255 ± 0.015*
(6.477 ± 0.381)

0.228 – 0.244

(5.791 – 6.197)

8

1

0.400*

(10.160)

MAX

876

1234

0.189 – 0.197*
(4.801 – 5.004)

7

6

3

2

5

5

0.150 – 0.157**
(3.810 – 3.988)

4

SO8 0996

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

15

LT1676

TYPICAL APPLICATION

V

IN

12V TO

48V

+

C1
12µF
63V

C1: PANASONIC HFQ
C2: AVX D CASE TPSD107M010R0080
C4, C5: X7R OR COG/NPO

U

C5
100pF

Minimum Inductor Size Application

5

V

IN

1

SHDN

LT1676

6

SYNC

GND

2

V

CC

3

V

SW

7

FB

8

V

C

R3

4

22k
5%

D1: MOTOROLA 100V, 1A, SMD SCHOTTKY
MBRS1100 (T3)
L1: COILCRAFT DO1608C-224

D1

C3
2200pF
X7R

L1

220µH

+

C4
100pF

C2
100µF
10V

V

OUT

5V
0mA to 150mA

R1

36.5k
1%

R2

12.1k
1%

1676 TA02

FOR 3.3V V
I

OUT

L1: 150µH, DO1608C-154
R1: 24.3K, R2: 14.7k

VERSION:

OUT

: 0mA TO 200mA

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PART NUMBER DESCRIPTION COMMENTS

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1676f LT/TP 0499 4K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1998

16

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

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●

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