Datasheet LT1578-2.5, LT1578 Datasheet (Linear Technology)

FEATURES

■

1.5A Switch Current

■

High Efficiency—Low Loss 0.2Ω Switch

■

Constant 200kHz Switching Frequency

■

4V to 15V Input VoltageRange

■

Minimum Output: 1.21V

■

Low Supply Current: 1.9mA

■

Low Shutdown Current: 20µA

■

Easily Synchronizable Up to 400kHz

■

Cycle-by-Cycle Current Limit

■

Reduced EMI Generation

■

Low Thermal Resistance SO-8 Package

■

Uses Small Low Value Inductors

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APPLICATIO S

LT1578/LT1578-2.5

1.5A, 200kHz Step-Down
Switching Regulator

U

DESCRIPTIO

The LT®1578 is a 200kHz monolithic buck mode switching
regulator. A 1.5A switch is included on the die along with
all the necessary oscillator, control and logic circuitry. The
topology is current mode for fast transient response and
good loop stability. The LT1578 is a modified version of the
LT1507 that has been optimized for noise sensitive appli­cations. It will operate over a 4V to 15V input range.

In addition, the reference voltage has been lowered to al­low the device to produce output voltages down to 1.2V.
Quiescent current has been reduced by a factor of two.
Switch on resistance has been reduced by 30%. Switch tran­sition times have been slowed to reduce EMI generation.
The oscillator frequency has been reduced to 200kHz to
maintain high efficiency over a wide output current range.

■

Portable Computers

■

Battery-Powered Systems

■

Battery Chargers

■

Distributed Power Systems

TYPICAL APPLICATION

3.3V Buck Converter

INPUT

5V TO 15V

* RIPPLE CURRENT RATING ≥ I

** INCREASE L1 TO 30µH FOR LOAD

CURRENTS ABOVE 0.6A AND TO
60µH ABOVE 1A
SEE APPLICATIONS INFORMATION

C3*

10µF TO

50µF

OPEN = ON

+

V

BOOST

IN

LT1578

SHDN

OUT

/2

GND

V

C

U

0.33µF

V

SW

FB

C

C

100pF

The pinout has been changed to improve PC layout by al­lowing the high current, high frequency switching circuitry
to be easily isolated from low current, noise sensitive con­trol circuitry. The new SO-8 package includes a fused
ground lead that significantly reduces the thermal resistance
of the device to extend the ambient operating temperature
range. Standard surface mount external parts can be used
including the inductor and capacitors.

, LTC and LT are registered trademarks of Linear Technology Corporation.

Efficiency vs Load Current

90

C2

L1**

15µH

D1
1N5818

R2

4.99k

R1

8.66k

+

D2
1N914

OUTPUT**

3.3V, 1.25A

C1
100µF, 10V
SOLID
TANTALUM

1578 TA01

85

80

75

70

65

EFFICIENCY (%)

60

V
V

55

L = 25µH

50

0

0.25

= 3.3V

OUT

= 5V

IN

0.50 0.75 1.00
LOAD CURRENT (A)

1.25 1.50

1578 TA02

1

LT1578/LT1578-2.5

WW

W

ABSOLUTE MAXIMUM RATINGS

(Note 1)

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PACKAGE/ORDER INFORMATION

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ORDER PART

Input Voltage .......................................................... 16V

BOOST Pin Above Input Voltage ............................. 10V

V

SHDN Pin Voltage..................................................... 7V

SENSE Pin Voltage .................................................... 4V

FB Pin Voltage (Adjustable Part)............................ 3.5V

BOOST

GND

FB Pin Current (Adjustable Part)............................ 1mA

SYNC Pin Voltage ..................................................... 7V

Operating Junction Temperature Range

LT1578C............................................... 0°C to 125° C

LT1578I ........................................... – 40°C to 125°C

θJA =80°C/W WITH FUSED GROUND PIN
CONNECTED TO GROUND PLANE OR
LARGE LANDS

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

ELECTRICAL CHARACTERISTICS

ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Feedback Voltage 1.195 1.21 1.225 V

All Conditions ● 1.18 1.24 V

Sense Voltage (Fixed 2.5) 2.46 2.5 2.54 V

All Conditions ● 2.44 2.56 V

Sense Pin Resistance 5.7 9.5 13.7 kΩ
Reference Voltage Line Regulation 4.3V ≤ VIN ≤ 15V ● 0.01 0.03 %/V
Feedback Input Bias Current ● 0.5 2 µA
Error Amplifier Voltage Gain (Notes 2, 10) 200 400
Error Amplifier Transconductance (Note 10) ∆I (V

VC Pin to Switch Current Transconductance 1.5 A/V
Error Amplifier Source Current VFB = 1.1V ● 40 110 190 µA
Error Amplifier Sink Current VFB = 1.4V ● 50 130 200 µA
VC Pin Switching Threshold Duty Cycle = 0 0.8 V
VC Pin High Clamp 2.1 V
Switch Current Limit VC Open, VFB = 1.1V, DC ≤ 50% ● 1.5 2 3.5 A
Slope Compensation (Note 8) DC = 80% 0.3 A
Switch On Resistance (Note 7) ISW = 1.5A 0.2 0.35 Ω

Maximum Switch Duty Cycle VFB = 1.1V 90 94 %

Minimum Switch Duty Cycle (Note 9) 8%
Switch Frequency VC Set to Give 50% Duty Cycle 180 200 220 kHz

Switch Frequency Line Regulation 4.3V ≤ VIN ≤ 15V ● 0 0.15 %/V
Frequency Shifting Threshold on FB Pin ∆f = 10kHz ● 0.4 0.74 1.0 V
Minimum Input Voltage (Note 3) ● 4.0 4.3 V
Minimum Boost Voltage (Note 4) ISW ≤ 1.5A ● 2.3 3.0 V

) = ±10µA 800 1050 1300 µMho

C

The ● denotes specifications which apply over the full operating tempera-

Consult factory for Military grade parts.

TOP VIEW

1

SW

2

V

IN

3

4

S8 PACKAGE

8-LEAD PLASTIC SO

NUMBER

8

SYNC
SHDN

7

FB/SENSE

6

V

5

C

LT1578CS8
LT1578IS8
LT1578CS8-2.5
LT1578IS8-2.5

S8 PART MARKING

1578
1578I

● 400 1700 µMho

● 0.45 Ω

● 86 94 %

● 160 240 kHz

157825
578I25

2

LT1578/LT1578-2.5

JUNCTION TEMPERATURE (°C)

–50

1.23

1.22

1.21

1.20

1.19
100

1576 G03

–25 0 25 50 75 125

FEEDBACK VOLTAGE (V)

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating tempera-

ture range, otherwise specifications are at TJ = 25°C. VIN = 5V, VC = 1.5V, Boost = VIN + 5V, switch open, unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS

Boost Current (Note 5) ISW = 0.5A ● 918 mA

ISW = 1.5A ● 27 50 mA
VIN Supply Current (Note 6) ● 1.9 2.7 mA
Shutdown Supply Current V

Lockout Threshold VC Open ● 2.34 2.42 2.50 V
Shutdown Thresholds VC Open Device Shutting Down ● 0.13 0.37 0.60 V

Synchronization Threshold 1.5 2.2 V
Synchronizing Range 250 400 kHz
SYNC Pin Input Resistance 40 kΩ

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Note 2: Gain is measured with a V

swing equal to 200mV above the

C

switching threshold level to 200mV below the upper clamp level.
Note 3: Minimum input voltage is not measured directly, but is guaranteed

by other tests. It is defined as the voltage where internal bias lines are still
regulated so that the reference voltage and oscillator frequency remain
constant. Actual minimum input voltage to maintain a regulated output will
depend on output voltage and load current. See Applications Information.

Note 4: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the internal power switch.

Note 5: Boost current is the current flowing into the boost pin with the pin
held 5V above input voltage. It flows only during switch on time.

Note 6: Input supply current is the bias current drawn by the input pin
with switching disabled.

= 0V, VIN ≤ 15V, VSW = 0V, VC Open 20 50 µA

SHDN

● 75 µA

Device Starting Up ● 0.25 0.45 0.7 V

Note 7: Switch on resistance is calculated by dividing V

to VSW voltage

IN

by the forced current (1.5A). See Typical Performance Characteristics for
the graph of switch voltage at other currents.

Note 8: Slope compensation is the current subtracted from the switch
current limit at 80% duty cycle. See Maximum Output Load Current in the
Applications Information section for further details.

Note 9: Minimum on-time is 400ns typical. For a 200kHz operating
frequency this means the minimum duty cycle is 8%. In frequency
foldback mode, the effective duty cycle will be less than 8%.

Note 10: Transconductance and voltage gain refer to the internal amplifier
exclusive of the voltage divider. To calculate gain and transconductance
referred to the sense pin on the fixed voltage parts, divide values shown by
the ratio 2.5/1.21.

TYPICAL PERFORMANCE CHARACTERISTICS

Switch Voltage Drop

0.5

0.4

0.3

0.2

SWITCH VOLTAGE (V)

0.1

0

0

0.50 0.75 1.00

0.25
SWITCH CURRENT (A)

25°C

125°C

–20°C

1.25 1.50

1576 G01

UW

2.5

2.0

1.5

1.0

SWITCH PEAK CURRENT (A)

0.5

0

0

MINIMUM

20

DUTY CYCLE (%)

TYPICAL

40

Feedback Pin VoltageSwitch Peak Current Limit

60

80

100

1576 G02

3

LT1578/LT1578-2.5

FEEDBACK VOLTAGE (V)

0

SWITCHING FREQUENCY (kHz)

OR CURRENT (µA)

2.0

1576 G12

0.5

1.0

1.5

250

200

150

100

50

0

FEEDBACK PIN CURRENT

SWITCHING FREQUENCY

W

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TYPICAL PERFORMANCE CHARACTERISTICS

Shutdown Pin Bias Current
(V

= Lockout Threshold)

SHDN

4

AT 2.44V STANDBY THRESHOLD
(CURRENT FLOWS OUT OF PIN)

3

2

1

SHDN PIN CURRENT (µA)

0

–25 0 25 50 75 125

–50

JUNCTION TEMPERATURE (°C)

Standby Thresholds

2.46

2.45

2.44
ON

2.43

2.42

SHUTDOWN PIN VOLTAGE (V)

2.41

STANDBY

100

1576 G04

Shutdown Pin Bias Current
(V

= Shutdown Threshold)

SHDN

180

160

140

120

100

80

60

SHDN PIN CURRENT (µA)

CURRENT REQUIRED TO FORCE

40

SHUTDOWN (FLOWS OUT OF PIN).
AFTER SHUTDOWN, CURRENT

20

DROPS TO A FEW µA

0

–25 0 25 50 75 125

–50

JUNCTION TEMPERATURE (°C)

Shutdown Supply Current

25

20

15

10

5

INPUT SUPPLY CURRENT (µA)

V

SHDN

100

1576 G05

= 0V

Shutdown Thresholds

0.8

0.7

0.6

0.5

0.4

0.3

0.2

SHUTDOWN PIN VOLTAGE (V)

0.1

0

–50

SHUTDOWN

050

–25 25 75 125

JUNCTION TEMPERATURE (°C)

Shutdown Supply Current

30

25

20

15

10

INPUT SUPPLY CURRENT (µA)

5

START-UP

100

1576 G06

VIN = 10V

2.40

–50

–25 0

Error Amplifier Transconductance

2000

1500

1000

500

V

GAIN (µMho)

–500

4

1 × 10

FB

0

ERROR AMPLIFIER EQUIVALENT CIRCUIT

R

LOAD

10 1k 10k 1M

50 100 125

25 75

JUNCTION TEMPERATURE (°C)

PHASE

GAIN

R

OUT

570k

–3

)(

= 50Ω

100 100k

FREQUENCY (Hz)

C

OUT

2.4pF

V

1576 G09

1576 G07

C

200

150

100

50

0

–50

0

0

1600

1400

1200

PHASE (DEG)

1000

800

600

400

TRANSCONDUCTANCE (µMho)

200

5

INPUT VOLTAGE (V)

10

15

1576 G08

0

0

0.1 0.2 0.3 0.4
SHUTDOWN VOLTAGE (V)

Error Amplifier Transconductance Frequency Foldback

0

–25 25 75 125

–50

050

JUNCTION TEMPERATURE (°C)

100

1576 G11

1576 G010

W

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TYPICAL PERFORMANCE CHARACTERISTICS

Switching Frequency

240

220

Switch Current Limit Foldback

3.0

2.5

2.0

LT1578/LT1578-2.5

Minimum Input Voltage to Start
with 3.3V Output

4.50

4.25

200

FREQUENCY (kHz)

180

160

–25 0 25 50 75 125

–50

JUNCTION TEMPERATURE (°C)

Maximum Output Current
at V

= 5V

OUT

1.6

1.4

1.2

1.0

0.8

0.6

OUTPUT CURRENT (A)

0.4

0.2

0

6

912

INPUT VOLTAGE (V)

L = 60µH

L = 30µH

L = 15µH

100

1576 G13

1578 G15

1.5

1.0

SWITCH CURRENT LIMIT (A)

0.5

0

0

0.4 0.6 0.8

0.2
FEEDBACK PIN VOLTAGE (V)

1.0 1.2

1578 G19

Maximum Output Current
at V

= 3.3V

OUT

1.6

1.4

1.2

1.0

0.8

0.6

OUTPUT CURRENT (A)

0.4

0.2

15

0

6 8 10 12

4

L = 60µH

L = 30µH
L = 15µH

14

INPUT VOLTAGE (V)

1578 G16

4.00

INPUT VOLTAGE (V)

3.75

3.50
1

10 100 1000

LOAD CURRENT (mA)

Maximum Output Current
at V

= 2.5V

OUT

1.6

1.4

1.2

1.0

0.8

0.6

OUTPUT CURRENT (A)

0.4

0.2

0

6 8 10 12

4

INPUT VOLTAGE (V)

1576 G14

L = 60µH
L = 30µH

L = 15µH

14

1578 G17

BOOST Pin Current

30

25

20

15

10

BOOST PIN CURRENT (mA)

5

0

0

Kool Mµ is a registered trademark of Magnetics, Inc.
Metglas is a registered trademark of AlliedSignal, Inc.

0.50 0.75 1.00

0.25
SWITCH CURRENT (A)

1.25 1.50

1576 G20

VC Pin Shutdown Threshold

1.0

0.8

0.6

0.4

THRESHOLD VOLTAGE (V)

0.2

0

–25 0 25 50 75 125

–50

JUNCTION TEMPERATURE (°C)

100

1576 G21

5

LT1578/LT1578-2.5

UUU

PIN FUNCTIONS

VSW (Pin 1): The switch pin is the emitter of the on-chip
power NPN switch. This pin is driven up to the input pin
voltage during switch on time. Inductor current drives the
switch pin negative during switch off time. Negative volt­age is clamped with the external catch diode. Maximum
negative switch voltage allowed is –0.8V.

VIN (Pin 2): This is the collector of the on-chip power NPN
switch. This pin powers the internal circuitry and internal
regulator. At NPN switch on and off, high dI/dt edges occur
through this pin. Keep the external bypass and catch diode
close to this pin. Trace inductance in this path will create
a voltage spike at switch off, adding to the VCE voltage
across the internal NPN.

BOOST (Pin 3): The BOOST pin is used to provide a drive
voltage, higher than the input voltage, to the internal
bipolar NPN power switch. Without this added voltage, the
typical switch voltage loss would be about 1.5V. The
additional boost voltage allows the switch to saturate with
its voltage drop approximating that of a 0.2Ω FET struc­ture. Efficiency improves from 75% for conventional bipo­lar designs to >88% for the LT1578.

pin should be attached to a large copper area to improve
thermal resistance.

VC (Pin 5): The VC pin is the output of the error amplifier
and the input to the peak switch current comparator. It is
normally used for frequency compensation, but can do
double duty as a current clamp or control loop override.
This pin sits at about 1V for very light loads and 2V at
maximum load. It can be driven to ground to shut off the
regulator, but if driven high, current must be limited to
4mA.

FB/SENSE (Pin 6): The feedback pin is used to set output
voltage using an external voltage divider that generates

1.21V at the pin with the desired output voltage. The fixed
voltage (2.5V) parts have the divider included on the chip
and the FB pin is used as a sense pin, connected directly
to the 2.5V output. Three additional functions are per­formed by the FB pin. When the pin voltage drops below

0.7V, the switch current limit and the switching frequency
are reduced and the external sync function is disabled. See
Feedback Pin Function section in Applications Information
for details.

GND (Pin 4): The GND pin connection needs consideration
for two reasons. First, it acts as the reference for the
regulated output, so load regulation will suffer if the
“ground” end of the load is not at the same voltage as the
GND pin of the IC. This condition will occur when load
current or other currents flow through metal paths be­tween the GND pin and the load ground point. Keep the
ground path short between the GND pin and the load and
use a ground plane when possible. The second consider­ation is EMI caused by GND pin current spikes. Internal
capacitance between the VSW pin and the GND pin creates
very narrow (<10ns) current spikes in the GND pin. If the
GND pin is connected to system ground with a long metal
trace, this trace may radiate EMI. Keep the path between
the input bypass and the GND pin short. The GND pin of the
SO-8 package is directly attached to the internal tab. This

SHDN (Pin 7): The shutdown pin is used to turn off the
regulator and to reduce input drain current to a few
microamperes. Actually, this pin has two separate thresh­olds, one at 2.42V to disable switching, and a second at

0.4V to force complete micropower shutdown. The 2.42V
threshold functions as an accurate undervoltage lockout
(UVLO). This can be used to prevent the regulator from
operating until the input voltage has reached a predeter­mined level.

SYNC (Pin 8): The SYNC pin is used to synchronize the
internal oscillator to an external signal. It is directly logic
compatible and can be driven with any signal between
10% and 90% duty cycle. The synchronizing range is
equal to
not used, this pin should be grounded. See Synchronizing
section in Applications Information for details.

initial

operating frequency, up to 400kHz. When

6

BLOCK DIAGRAM

LT1578/LT1578-2.5

W

The LT1578 is a constant frequency, current mode buck
converter. This means that there is an internal clock and
two feedback loops that control the duty cycle of the power
switch. In addition to the normal error amplifier, there is a
current sense amplifier that monitors switch current on a
cycle-by-cycle basis. A switch cycle starts with an oscilla­tor pulse which sets the RS flip-flop to turn the switch on.
When switch current reaches a level set by the inverting
input of the comparator, the flip-flop is reset and the
switch turns off. Output voltage control is obtained by
using the output of the error amplifier to set the switch
current trip point. This technique means that the error
amplifier commands current to be delivered to the output
rather than voltage. A voltage fed system will have low
phase shift up to the resonant frequency of the inductor

INPUT

2.9V BIAS

REGULATOR

INTERNAL
V

CC

0.025Ω

+

–

CURRENT SENSE
AMPLIFIER DC
VOLTAGE GAIN = 35

and output capacitor, then an abrupt 180° shift will occur.
The current fed system will have 90° phase shift at a much
lower frequency, but will not have the additional 90° shift
until well beyond the LC resonant frequency. This makes
it much easier to frequency compensate the feedback loop
and also gives much quicker transient response.

High switch efficiency is attained by using the BOOST pin
to provide a voltage to the switch driver which is higher
than the input voltage, allowing the switch to saturate. This
boosted voltage is generated with an external capacitor
and diode. Two comparators are connected to the shut­down pin. One has a 2.42V threshold for undervoltage
lockout and the second has a 0.4V threshold for complete
shutdown.

SYNC

SHDN

SHUTDOWN

COMPARATOR

+

3.5µA

0.4V

+

–

–

LOCKOUT
COMPARATOR

SLOPE COMP

200kHz

OSCILLATOR

Σ

0.8V

CURRENT
COMPARATOR

+

–

FOLDBACK

CURRENT

LIMIT

CLAMP

V

C

Q2

Figure 1. Block Diagram

S

R

S

FLIP-FLOP

R

FREQUENCY

SHIFT CIRCUIT

ERROR

AMPLIFIER

g

= 1000µMho

m

DRIVER

CIRCUITRY

–

+

BOOST

Q1
POWER
SWITCH

V

SW

FB

1.21V2.42V
GND

1578 BD

7

LT1578/LT1578-2.5

U

WUU

APPLICATIONS INFORMATION

FEEDBACK PIN FUNCTIONS

The feedback (FB) pin on the LT1578 is used to set output
voltage and provide several overload protection features.
The first part of this section deals with selecting resistors
to set output voltage and the remaining part talks about
foldback frequency and current limiting created by the FB
pin. Please read both parts before committing to a final
design. The fixed 2.5V LT1578-2.5 has internal divider
resistors and the FB pin, renamed SENSE, is connected
directly to the 2.5V output.

The suggested value for the output divider resistor (see
Figure 2) from FB to ground (R2) is 5k or less, and a
formula for R1 is shown below. The output voltage error
caused by ignoring the input bias current on the FB pin is
less than 0.25% with R2 = 5k. Please read the following
if divider resistors are increased above the suggested
values.

RV

2121

R

1

=

−

121

.

.

()

OUT

More Than Just Voltage Feedback

The feedback pin is used for more than just output voltage
sensing. It also reduces switching frequency and current
limit when output voltage is very low (see the Frequency
Foldback graph in Typical Performance Characteristics).
This is done to control power dissipation in both the IC and
the external diode and inductor during short-circuit con­ditions. A shorted output requires the switching regulator
to operate at very low duty cycles, and the average current
through the diode and inductor is equal to the short-circuit
current limit of the switch (typically 2A for the LT1578,
folding back to less than 0.77A). Minimum switch on time
limitations would prevent the switcher from attaining a
sufficiently low duty cycle if switching frequency were
maintained at 200kHz, so frequency is reduced by about
5:1 when the feedback pin voltage drops below 0.7V (see
Frequency Foldback graph). This does not affect operation
with normal load conditions; one simply sees a gear shift
in switching frequency during start-up as the output
voltage rises.

LT1578

VCGND

Q2

TO FREQUENCY

SHIFTING

1.4V

ERROR

AMPLIFIER

+

–

R5
5k

TO SYNC CIRCUIT

Figure 2. Frequency and Current Limit Foldback

1.21V

Q1

R3

1k

R4

1k

V

SW

R1

FB

R2
5k

OUTPUT
5V

+

1578 F02

8

LT1578/LT1578-2.5

U

WUU

APPLICATIONS INFORMATION

In addition to lower switching frequency, the LT1578 also
operates at lower switch current limit when the feedback
pin voltage drops below 0.7V. Q2 in Figure 2 performs this
function by clamping the VC pin to a voltage less than its
normal 2.1V upper clamp level. This
greatly reduces power dissipation in the IC, diode and
inductor during short-circuit conditions. External synchro­nization is also disabled to prevent interference with
foldback operation. Again, it is nearly transparent to the
user under normal load conditions. The only loads that may
be affected are current sources, such as lamps and mo­tors, that maintain high load current with output voltage
less than 50% of final value. In these rare situations the
feedback pin can be clamped above 0.7V to defeat foldback
current limit.
that frequency shifting will also be defeated, so a combina­tion of high input voltage and dead shorted output may
cause the LT1578 to lose control of current limit.

The internal circuitry which forces reduced switching
frequency also causes current to flow out of the feedback
pin when output voltage is low. The equivalent circuitry is
shown in Figure 2. Q1 is completely off during normal
operation. If the FB pin falls below 0.7V, Q1 begins to
conduct current and reduces frequency at the rate of
approximately 1kHz/µA. To ensure adequate frequency
foldback (under worst-case short-circuit conditions), the
external divider Thevinin resistance must be low enough
to pull 35µA out of the FB pin with 0.5V on the pin (R

14.3k).

current limit are affected by output voltage divider imped­ance. Although divider impedance is not critical, caution
should be used if resistors are increased beyond the
suggested values and short-circuit conditions will occur
with high input voltage

increase and the protection accorded by frequency and
current foldback will decrease.

MAXIMUM OUTPUT LOAD CURRENT

Caution:

The net result is that reductions in frequency and

clamping the feedback pin means

. High frequency pickup will

foldback current limit

≤

DIV

graphically in Typical Performance Characteristics and as
shown in the formula below:

IP = 1.5A for DC ≤ 50%

IP = 1.67 – 0.18 (DC) – 0.32(DC)2 for 50% < DC < 90%
DC = Duty cycle = V
Example: with V

I

SW(MAX)

Current rating decreases with duty cycle because the
LT1578 has internal slope compensation to prevent cur­rent mode subharmonic switching. For more details, read
Application Note 19. The LT1578 is a little unusual in this
regard because it has nonlinear slope compensation which
gives better compensation with less reduction in current
limit.

Maximum load current would be equal to maximum
switch current
finite inductor size, maximum load current is reduced by
one-half peak-to-peak inductor current. The following
formula assumes continuous mode operation, implying
that the term on the right is less than one-half of IP.

I

OUT(MAX)

Continuous Mode

For the conditions above and L = 15µH,

I

OUT MAX

At VIN = 15V, duty cycle is 33%, so IP is just equal to a fixed

1.5A, and I

= 1.67 – 0.18 (0.625) – 0.32(0.625)2 = 1.43A

=

()

OUT(MAX)

OUT/VIN

= 5V, VIN = 8V; DC = 5/8 = 0.625, and;

OUT

for an infinitely large inductor

VVV

()

OUT IN OUT

I

−

P

=−

143

.

2 15 10 200 10 8

=−=

143 031 112

...

is equal to:

2



••



−

()

LfV

()()( )

()

−

IN

58 5

−

()

63





A

, but with



()



Maximum load current for a buck converter is limited by
the maximum switch current rating (IP) of the LT1578.
This current rating is 1.5A up to 50% duty cycle (DC),
decreasing to 1.3A at 80% duty cycle. This is shown

515 5

()

15

.

−


2 15 10 200 10 15

••



15 056 094

.. .

=− =−A

()

63





−


()



9

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

Note that there is less load current available at the higher
input voltage because inductor ripple current increases.
This is not always the case. Certain combinations of
inductor value and input voltage range may yield lower
available load current at the lowest input voltage due to
reduced peak switch current at high duty cycles. If load
current is close to the maximum available, please check
maximum available current at both input voltage
extremes. To calculate actual peak switch current with a
given set of conditions, use:

VVV

II

SW PEAK OUT

For lighter loads where discontinuous operation can be
used, maximum load current is equal to:

I

OUT(MAX)

Discontinuous mode

Example: with L = 5µH, V

(

=+

)

=

()

IA

OUT MAX

=

()

OUT IN OUT

2



1 5 200 10 5 10 15

.••



−

()

LfV

2

()()( )

OUT

2 5 15 5

IN

2

IfLV

()()()( )

PIN

2

VVV

()

OUT IN OUT

= 5V, and V

36





−

()

()

−

()

) = 15V,

IN(MAX

−



()



=

034

.

physical size of the inductor. Higher values allow more
output current because they reduce peak current seen by
the LT1578 switch, which has a 1.5A limit. Higher values
also reduce output ripple voltage, and reduce core loss.
Graphs in the Typical Performance Characteristics section
show maximum output load current versus inductor size
and input voltage.

When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable
component height, output voltage ripple, EMI, fault cur­rent in the inductor, saturation, and of course, cost. The
following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.

1. Choose a value in microhenries from the graphs of
maximum load current and core loss. Choosing a small
inductor may result in discontinuous mode operation
at lighter loads, but the LT1578 is designed to work
well in either mode. Keep in mind that lower core loss
means higher cost, at least for closed core geometries
like toroids.

Assume that the average inductor current is equal to
load current and decide whether or not the inductor
must withstand continuous fault conditions. If maxi­mum load current is 0.5A, for instance, a 0.5A inductor
may not survive a continuous 1.5A overload condition.
Dead shorts will actually be more gentle on the induc­tor because the LT1578 has foldback current limiting.

The main reason for using such a tiny inductor is that it is
physically very small, but keep in mind that peak-to-peak
inductor current will be very high. This will increase output
ripple voltage. If the output capacitor has to be made larger
to reduce ripple voltage, the overall circuit could actually
wind up larger.

CHOOSING THE INDUCTOR AND OUTPUT CAPACITOR

For most applications the output inductor will fall in the
range of 15µH to 60µH. Lower values are chosen to reduce

10

2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe­cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving
because they saturate softly, whereas ferrite cores
saturate abruptly. Other core materials fall somewhere
in between. The following formula assumes continu­ous mode of operation, but it errs only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

VVV

II

=+

PEAK OUT

OUT IN OUT

VIN = Maximum input voltage
f = Switching frequency, 200kHz

3. Decide if the design can tolerate an “open” core geom­etry like a rod or barrel, with high magnetic field
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media, for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small and there are no helpful
guidelines to calculate when the magnetic field radia­tion will be a problem.

4. Start shopping for an inductor (see representative
surface mount units in Table 1) which meets the
requirements of core shape, peak current (to avoid
saturation), average current (to limit heating), and fault
current (if the inductor gets too hot, wire insulation will
melt and cause turn-to-turn shorts). Keep in mind that
all good things like high efficiency, low profile, and high
temperature operation will increase cost, sometimes
dramatically. Get a quote on the cheapest unit first to
calibrate yourself on price, then ask for what you really
want.

5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology’s applica­tions department if you feel uncertain about the final
choice. They have experience with a wide range of
inductor types and can tell you about the latest devel­opments in low profile, surface mounting, etc.

−

()

fLV

2

()()( )

IN

Table 1

SERIES CORE
VENDOR/ VALUE DC CORE RESIS- MATER- HEIGHT
PART NO. (µH) (Amps) TYPE TANCE(Ω) IAL (mm)

Coiltronics

CTX15-2 15 1.7 Tor 0.059 KMµ 6.0
CTX33-2 33 1.4 Tor 0.106 KMµ 6.0
CTX68-4 68 1.2 Tor 0.158 KMµ 6.4
CTX15-1P 15 1.4 Tor 0.087 52 4.2
CTX33-2P 33 1.3 Tor 0.126 52 6.0
CTX68-4P 68 1.1 Tor 0.238 52 6.4

Sumida

CDRH74-150 15 1.47 SC 0.081 Fer 4.5
CDH115-330 33 1.68 SC 0.082 Fer 5.2
CDRH125-680 68 1.5 SC 0.12 Fer 6
CDH74-330 33 1.45 SC 0.17 Fer 5.2

Coilcraft

DO3308P-153 15 2 SC 0.12 Fer 3
DO3316P-333 33 2 SC 0.1 Fer 5.21
DO3316P-683 68 1.4 SC 0.18 Fer 5.21

Pulse

PE-53602 35 1.4 Tor 0.166 Fer 9.1
PE-53604 73 1.3 Tor 0.290 Fer 9.1
PE-53632 22 2.7 Tor 0.063 Fer 9.1
PE-53633 40 2.7 Tor 0.085 Fer 10

Gowanda

SMP3316-152K 15 3.5 SC 0.041 Fer 6
SMP3316-332K 33 2.3 SC 0.092 Fer 6
SMP3316-682K 68 1.7 SC 0.178 Fer 6
Tor = Toroid

SC = Semi-closed geometry
Fer = Ferrite core material
52 = Type 52 powdered iron core material
KMµ = Kool Mµ

11

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APPLICATIONS INFORMATION

Output Capacitor

The output capacitor is normally chosen by its Effective
Series Resistance (ESR), because this is what determines
output ripple voltage. To get low ESR takes
physically smaller capacitors have high ESR. The ESR
range for typical LT1578 applications is 0.05Ω to 0.2Ω. A
typical output capacitor is an AVX type TPS, 100µF at 10V,
with a guaranteed ESR less than 0.1Ω. This is a “D” size
surface mount solid tantalum capacitor. TPS capacitors
are specially constructed and tested for low ESR, so they
give the lowest ESR for a given volume. The value in
microfarads is not particularly critical, and values from
22µF to greater than 500µF work well, but you cannot
cheat mother nature on ESR. If you find a tiny 22µF solid
tantalum capacitor, it will have high ESR, and output ripple
voltage will be terrible. Table 2 shows some typical solid
tantalum surface mount capacitors.

Table 2. Surface Mount Solid Tantalum Capacitor ESR
and Ripple Current

E Case Size ESR (Max., Ω) Ripple Current (A)

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
AVX TAJ 0.7 to 0.9 0.4

D Case Size

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1

C Case Size

AVX TPS 0.2 (typ) 0.5 (typ)

Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true, and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the
tantalum capacitors fail during very high
which do not occur at the output of regulators. High

discharge

dead shorted, do not harm the capacitors.
Unlike the input capacitor, RMS ripple current in the

output capacitor is normally low enough that ripple cur­rent rating is not an issue. The current waveform is
triangular with a typical value of 200mA
to calculate this is:

surges, such as when the regulator output is

output

volume

capacitor. Solid

turn-on

. The formula

RMS

, so

surges,

Output Capacitor Ripple Current (RMS):

VVV

029.

()

I

RIPPLE RMS

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempt­ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor’s ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumen­tal in giving acceptable loop phase margin. Ceramic
capacitors remain capacitive to beyond 300kHz and usu­ally resonate with their ESL before their ESR provides any
damping. They are appropriate for input bypassing be­cause of their high ripple current ratings and tolerance of
turn-on surges.

OUTPUT RIPPLE VOLTAGE

Figure 3 shows a typical output ripple voltage waveform
for the LT1578. Ripple voltage is determined by the high
frequency impedance of the output capacitor, and ripple
current through the inductor. Peak-to-peak ripple current
through the inductor into the output capacitor is:

I

=

P

-P

For high frequency switchers, the sum of ripple current
slew rates may also be relevant and can be calculated
from:

dIdtV

Σ

=

=

(

)

VVV

()

()

OUT IN OUT

VLf

()()()

IN

IN

L

OUT IN OUT

LfV

()()( )

−

−

()

IN

12

LT1578/LT1578-2.5

I

IVV

V

D AVG

OUT IN OUT

IN

(

)

=

−

()

U

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APPLICATIONS INFORMATION

Peak-to-peak output ripple voltage is the sum of a
created by peak-to-peak ripple current times ESR, and a

square

wave created by parasitic inductance (ESL) and
ripple current slew rate. Capacitive reactance is assumed
to be small compared to ESR or ESL.

V I ESR ESL

RIPPLE

=

()( )

P-P

Example: with VIN =10V, V

+

()

OUT

dI

Σ

dt

= 5V, L = 30µH, ESR = 0.1Ω,

ESL = 10nH:

−

510 5

()

IA

=

P-P

dI

Σ

dt

VA

RIPPLE

..

0 042 0 003 45

=+=

20mV/DIV

200mA/DIV

20mV/DIV

200mA/DIV



10 30 10 200 10

()



10

==

•

30 10

=

()()

Figure 3. LT1578 Ripple Voltage Waveform

()

=

−

63



••

−

6

.. • .•

0 42 0 1 10 10 0 33 10



+

6




P-P

.•

033 10

mV

2µs/DIV 1578 F03

042




96

−





CATCH DIODE

The suggested catch diode (D1) is a 1N5818 Schottky, or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch off time. Peak reverse voltage is equal to
regulator input voltage. Average forward current in normal
operation can be calculated from:

.

triwave

V

AT

OUT

= 1A

I

OUT

INDUCTOR
CURRENT
AT I

= 1A

OUT

AT

V

OUT

= 50mA

I

OUT

INDUCTOR
CURRENT

= 50mA

AT I

OUT

This formula will not yield values higher than 1A with
maximum load current of 1.25A unless the ratio of input to
output voltage exceeds 5:1. The only reason to consider a
larger diode is the worst-case condition of a high input
voltage and

overloaded

circuit conditions, foldback current limit will reduce diode
current to less than 1A, but if the output is overloaded and
does not fall to less than 1/3 of nominal output voltage,
foldback will not take effect. With the overloaded condi­tion, output current will increase to a typical value of 1.8A,
determined by peak switch current limit of 2A. With
VIN = 15V, V

IA



D AVG

()



= 4V (5V overloaded) and I

OUT

1 8 15 4

−

()

=

15

This is safe for short periods of time, but it would be
prudent to check with the diode manufacturer if continu­ous operation under these conditions must be tolerated.

BOOST␣ PIN␣ CONSIDERATIONS

For most applications, the boost components are a 0.33µF
capacitor and a 1N914 or 1N4148 diode. The anode is
connected to the regulated output voltage and this gener­ates a voltage across the boost capacitor nearly identical
to the regulated output. In certain applications, the anode
may instead be connected to the unregulated input volt­age. This could be necessary if the regulated output
voltage is very low (< 3V) or if the input voltage is less than
6V. Efficiency is not affected by the capacitor value, but the
capacitor should have an ESR of less than 1Ω to ensure
that it can be recharged fully under the worst-case condi­tion of minimum input voltage. Almost any type of film or
ceramic capacitor will work fine.

WARNING!

Peak voltage on the BOOST pin is the sum of

unregulated input voltage plus the voltage across the

(not shorted) output. Under short-

= 1.8A:

OUT

=

132..

13

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

boost capacitor. This normally means that peak BOOST
pin voltage is equal to input voltage plus output voltage,
but

when the boost diode is connected to the regulator
input, peak BOOST pin voltage is equal to twice the input
voltage. Be sure that BOOST pin voltage does not exceed
its maximum rating

For nearly all applications, a 0.33µF boost capacitor works
just fine, but for the curious, more details are provided
here. The size of the boost capacitor is determined by
switch drive current requirements. During switch on time,
drain current on the capacitor is approximately I
peak load current of 1.25A, this gives a total drain of 25mA.
Capacitor ripple voltage is equal to the product of on time
and drain current divided by capacitor value;
∆V = (tON)(25mA/C). To keep capacitor ripple voltage to
less than 0.5V (a slightly arbitrary number) at the worst­case condition of tON = 4.7µs, the capacitor needs to be

0.24µF. Boost capacitor ripple voltage is not a critical
parameter, but if the minimum voltage across the capaci­tor drops to less than 3V, the power switch may not
saturate fully and efficiency will drop. An
formula for absolute minimum capacitor value is:

.

/ 50. At

OUT

approximate

IVV

//50

()( )

C

MIN

OUT OUT IN

=

fV V

()

()

OUT

−

3

f = Switching frequency
V

= Regulated output voltage

OUT

VIN = Minimum input voltage
This formula can yield capacitor values substantially less

than 0.24µF, but it should be used with caution since it
does not take into account secondary factors such as
capacitor series resistance, capacitance shift with tem­perature and output overload.

SHUTDOWN FUNCTION AND
UNDERVOLTAGE LOCKOUT

Figure 4 shows how to add undervoltage lockout (UVLO)
to the LT1578. Typically, UVLO is used in situations where
the input supply is

current limited

, or has a relatively high
source resistance. It is particularly useful for input sup­plies with foldback current limiting. A switching regulator
draws constant power from the source, so source current
increases as source voltage drops. This looks like a
negative resistance load to the source and can cause the
source to current limit and latch under low source voltage

INPUT

R

FB

LT1578

V

IN

2.42V

R

HI

SHDN

R

C1

LO

3.5µA

0.4V

Figure 4. Undervoltage Lockout

–

STANDBY

+

+

TOTAL
SHUTDOWN

–

GND

SW

OUTPUT

+

1578 F04

14

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

conditions. UVLO helps prevent the regulator from oper­ating at source voltages where these problems might
occur.

Threshold voltage for lockout is about 2.42V. A 3.5µA bias
current flows
generated current is used to force a default high state on
the shutdown pin if the pin is left open. When low shut­down current is not an issue, the error due to this current
can be minimized by making RLO 10k or less. If shutdown
current is an issue, RLO can be raised to 100k, but the error
due to initial bias current and changes with temperature
should be considered.

Rk

LO

R

HI

VIN = Minimum input voltage
Keep the connections from the resistors to the shutdown

pin short and make sure that interplane or surface capaci­tance to the switching nodes are minimized. If high resis­tor values are used, the shutdown pin should be bypassed
with a 1000pF capacitor to prevent coupling problems
from the switch node. If hysteresis is desired in the
undervoltage lockout point, a resistor RFB can be added to
the output node. Resistor values can be calculated from:

R

=

HI

=

RRV V

FB HI OUT

25k suggested for R
VIN =

Input voltage at which switching stops as input
voltage descends to trip level

∆V = Hysteresis in input voltage level
Example: output voltage is 5V, switching is to stop if input

voltage drops below 12V and should not restart unless

out

of the pin at threshold. This internally

=

10

to 100k 25k suggested

RV V

LO IN

=

()

VR A

242 35

..µ

RV VV V

LO IN OUT

[]

()( )

()

−

242

.

−

LO

()

−+

2

./

∆∆

42

()

R

−

242 35

..

LO

LO

()

/

∆

+

1

µ

A

input rises back to 13.5V. ∆V is therefore 1.5V and
VIN = 12V. Let RLO = 25k.

k

25 12 24215 5 1 15

R

=

HI

SWITCH NODE CONSIDERATIONS

For maximum efficiency, switch rise and fall times are
made as short as possible. To prevent radiated EMI and
high frequency resonance problems, proper layout of the
components connected to the switch node is essential. B
field (magnetic) radiation is minimized by keeping catch
diode, switch pin, and input bypass capacitor leads as
short as possible. E field radiation is kept low by minimiz­ing the length and area of all traces connected to the switch
pin and BOOST pin. A ground plane should always be used
under the switcher circuitry to prevent interplane cou­pling. A suggested layout for the critical components is
shown in Figure 5. Note that the feedback resistors and
compensation components are kept as far as possible
from the switch node. Also note that the high current
ground path of the catch diode and input capacitor are kept
very short and separate from the analog ground line.

The high speed switching current path is shown schemati­cally in Figure 6. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, catch diode, and input capacitor is
the only one containing nanosecond rise and fall times. If
you follow this path on the PC layout, you will see that it is
irreducibly short. If you move the diode or input capacitor
away from the LT1578, get your resumé in order. The
other paths contain only some combination of DC and
200kHz triwave, so are much less critical.

25 10 35

=

Rk k

=

111 5 1 5 370

FB

−+

../ .

[]

242 25 35

k

.

()

233

.

()

()

kA

−

..

/.

()

k

=

111

=

+

µ

15

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

TAKE OUTPUT DIRECTLY FROM END

CAPACITOR DIRECTLY

TO HEAVY GROUND

MINIMIZE AREA
OF CONNECTIONS
TO SWITCH NODE

AND BOOST NODE

KEEP INPUT

CAPACITOR
AND CATCH

DIODE CLOSE

TO REGULATOR

AND TERMINATE

THEM TO THE

SAME POINT

CONNECT OUTPUT

C1

L1

SW

D1

C3

V

IN

GND

OF OUTPUT CAPACITOR TO AVOID
PARASITIC RESISTANCE AND
INDUCTANCE (KELVIN CONNECTION)

V

OUT

D2

C2

BOOST

SYNC

SHDN

R1

FB

V

C

GND

MINIMIZE SIZE

OF FEEDBACK PIN

CONNECTIONS

TO AVOID PICKUP

R2

C

C

TERMINATE
FEEDBACK
RESISTORS AND
COMPENSATION
COMPONENTS
DIRECTLY TO

R

C

SWITCHER
GROUND PIN

GROUND RING NEED NOT BE AS SHOWN

(NORMALLY EXISTS AS INTERNAL PLANE)

Figure 5. Suggested Layout for LT1578

SWITCH NODE

HIGH

V

IN

FREQUENCY

CIRCULATING

PATH

L1

Figure 6. High Speed Switching Path

1578 F05

5V

LOAD

1578 F06

16

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PARASITIC RESONANCE

Resonance or “ringing” may sometimes be seen on the
switch node (see Figure 7). Very high frequency ringing
following the switch voltage rise time is caused by switch/
diode/input capacitance lead inductance and diode ca­pacitance. Schottky diodes have very high “Q” junction
capacitance that can ring for many cycles when excited at
high frequency. If total lead length for the input capacitor,
diode and switch path is 1 inch, the inductance will be
approximately 25nH. At switch off, this will produce a
spike across the NPN output device in addition to the input
voltage. At higher currents this spike can be in the order of
10V to 20V or higher with a poor layout, potentially
exceeding the absolute max switch voltage. The path
around switch, catch diode and input capacitor must be
kept as short as possible to ensure reliable operation.

When looking at this, a >100MHz oscilloscope must be
used, and waveforms should be observed on the leads of
the package. This switch off spike will also cause the SW
node to go below ground. The LT1578 has special circuitry
inside which mitigates this problem, but negative voltages
over 1V lasting longer than 10ns should be avoided. Note
that 100MHz oscilloscopes are barely fast enough to see
the details of the falling edge overshoot in Figure 7.

A second, much lower frequency ringing is seen during
switch off time if load current is low enough to allow the
inductor current to fall to zero during part of the switch off
time (see Figure 8). Switch and diode capacitance reso­nate with the inductor to form damped ringing at 1MHz to
10 MHz. This ringing is not harmful to the regulator and it
has not been shown to contribute significantly to EMI. Any
attempt to damp it with an RC snubber will slightly degrade
efficiency.

5V/DIV

5V/DIV

50mA/DIV

50ns/DIV 1578 F07

Figure 7. Switch Node Response

1µs/DIV 1578 F08

Figure 8. Discontinuous Mode Ringing

RISE AND FALL
WAVEFORMS ARE
SUPERIMPOSED
(PULSE WIDTH IS

NOT

350ns)

SWITCH NODE
VOLTAGE

INDUCTOR
CURRENT

INPUT BYPASSING AND VOLTAGE RANGE

Input Bypass Capacitor

Step-down converters draw current from the input supply
in pulses. The average height of these pulses is equal to
load current, and the duty cycle is equal to V

OUT/VIN

. Rise
and fall times of the current are very fast. A local bypass
capacitor across the input supply is necessary to ensure
proper operation of the regulator and minimize the ripple
current fed back into the input supply.

The capacitor also
forces switching current to flow in a tight local loop,
minimizing EMI

.

Do not cheat on the ripple current rating of the input
bypass capacitor, but also do not be overly concerned with
the value in microfarads

. The input capacitor is intended
to absorb all the switching current ripple, which can have
an RMS value as high as one half of the load current. Ripple
current ratings on the capacitor must be observed to
ensure reliable operation. In many cases it is necessary to
parallel two capacitors to obtain the required ripple rating.
Both capacitors must be of the same value and manufac­turer to guarantee power sharing. The actual value of the
capacitor in microfarads is not particularly important

17

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

because at 200kHz, any value above 15µF is essentially
resistive. RMS ripple current rating is the critical param­eter. Actual RMS current can be calculated from:

IIVVVV

RIPPLE RMS OUT OUT IN OUT IN

The term inside the radical has a maximum value of 0.5
when input voltage is twice output, and stays near 0.5 for
a relatively wide range of input voltages. It is common
practice therefore to simply use the worst-case value and
assume that RMS ripple current is one half of load current.
At maximum output current of 1.5A for the LT1578, the
input bypass capacitor should be rated at 0.75A ripple
current. Note however, that there are many secondary
considerations in choosing the final ripple current rating.
These include ambient temperature, average versus peak
load current, equipment operating schedule, and required
product lifetime. For more details, see Application Notes
19 and 46, and Design Note 95.

=−

()

()

2

/

series for instance, see Table 3), but even these units may
fail if the input voltage surge approaches the maximum
voltage rating of the capacitor. AVX recommends derating
capacitor voltage by 2:1 for high surge applications. The
highest voltage rating is 50V, so 25V may be a practical
input voltage upper limit when using solid tantalum ca­pacitors for input bypassing.

Larger capacitors may be necessary when the input volt­age is very close to the minimum specified on the data
sheet. Small voltage dips during switch on time are not
normally a problem, but at very low input voltage they may
cause erratic operation because the input voltage drops
below the minimum specification. Problems can also
occur if the input-to-output voltage differential is near
minimum. The amplitude of these dips is normally a
function of capacitor ESR and ESL because the capacitive
reactance is small compared to these terms. ESR tends to
be the dominate term and is inversely related to physical
capacitor size within a given capacitor type.

Input Capacitor Type

Some caution must be used when selecting the type of
capacitor used at the input to regulators. Aluminum
electrolytics are lowest cost, but are physically large to
achieve adequate ripple current rating, and size con­straints (especially height) may preclude their use.
Ceramic capacitors are now available in larger values, and
their high ripple current and voltage rating make them
ideal for input bypassing. Cost is fairly high and footprint
may also be somewhat large. Solid tantalum capacitors
would be a good choice, except that they have a history of
occasional spectacular failures when they are subjected to
large current surges during power-up. The capacitors can
short and then burn with a brilliant white light and lots of
nasty smoke. This phenomenon occurs in only a small
percentage of units, but it has led some OEMs to forbid
their use in high surge applications. The input bypass
capacitors of regulators can see these high surges when
a battery or high capacitance source is connected. Several
manufacturers have developed a line of solid tantalum
capacitors specially tested for surge capability (AVX TPS

SYNCHRONIZING

The SYNC pin is used to synchronize the internal oscillator
to an external signal. The SYNC input must pass from a
logic level low, through the maximum synchronization
threshold with a duty cycle between 10% and 90%. The
input can be driven directly from a logic level output. The
synchronizing range is equal to
up to 400kHz. This means that
frequency is equal to the worst-case
frequency (250kHz), not the typical operating frequency of
200kHz. Caution should be used when synchronizing
above 280kHz because at higher sync frequencies the
amplitude of the internal slope compensation used to
prevent subharmonic switching is reduced. This type of
subharmonic switching only occurs at input voltages less
than twice output voltage. Higher inductor values will tend
to eliminate this problem. See Frequency Compensation
section for a discussion of an entirely different cause of
subharmonic switching before assuming that the cause is
insufficient slope compensation. Application Note 19 has
more details on the theory of slope compensation.

initial

operating frequency

minimum

practical sync

high

self-oscillating

18

LT1578/LT1578-2.5

P

W

PW

P

SW

BOOST

Q

=

()()()

+







()( )







=+ =

=

()( )

=

=







+







+

()( )

=

−

−−

02 1 5

10

60 10 1 10 200 10

01 012 022

5150

10

005

10 0 55 10 5 1 6 10

5 0 004

10

0

2

93

2

33

2

.

••

.. .

/

.

.• .•

.

.. 02W

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APPLICATIONS INFORMATION

At power-up, when VC is being clamped by the FB pin (see
Figure 2, Q2), the sync function is disabled. This allows the
frequency foldback to operate in the shorted output con­dition. During normal operation, switching frequency is
controlled by the internal oscillator until the FB pin reaches

0.7V, after which the SYNC pin becomes operational. If no
synchronization is required, this pin should be connected
to ground.

THERMAL CALCULATIONS

Power dissipation in the LT1578 chip comes from four
sources: switch DC loss, switch AC loss, boost circuit
current, and input quiescent current. The following formu­las show how to calculate each of these losses. These
formulas assume continuous mode operation, so they
should not be used for calculating efficiency at light load
currents.

Switch loss:

RI V

P

SW

Boost current loss:

P

BOOST

Quiescent current loss:

SW OUT OUT

=

VI

=

2

()( )

ns I V f

+

60

()()()

V

IN

2

()

OUT OUT

50/

V

IN

OUT IN

Total power dissipation is 0.22 + 0.05 + 0.02 = 0.29W.
Thermal resistance for LT1578 package is influenced by

the presence of internal or backside planes. With a full
plane under the SO package, thermal resistance will be
about 80°C/W. No plane will increase resistance to about
120°C/W. To calculate die temperature, add in worst-case
ambient temperature:

TJ = TA + θJA (P

With the SO-8 package (θJA = 80°C/W), at an ambient
temperature of 50°C,

TJ = 50 + 80 (0.29) = 73.2°C

Die temperature is highest at low input voltage, so use
lowest continuous input operating voltage for thermal
calculations.

TOT

)

−−

PV V

Q IN OUT

RSW = Switch resistance (≈0.2Ω)
60ns = Equivalent switch current/voltage overlap time
f = Switch frequency
Example: with VIN = 10V, V



=

055 10 16 10

.• .•



2



V



OUT



+

33



+





0 004

.

()





V

IN




= 5V and I

OUT




= 1A:

OUT

FREQUENCY COMPENSATION

Loop frequency compensation of switching regulators
can be a rather complicated problem because the reactive
components used to achieve high efficiency also intro­duce multiple poles into the feedback loop. The inductor
and output capacitor on a conventional step-down con­verter actually form a resonant tank circuit that can exhibit
peaking and a rapid 180° phase shift at the resonant
frequency. By contrast, the LT1578 uses a “current mode”
architecture to help alleviate the phase shift created by the
inductor. The basic connections are shown in Figure 9.
Figure 10 shows a Bode plot of the phase and gain of the
power section of the LT1578, measured from the VC pin to

19

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APPLICATIONS INFORMATION

the output. Gain is set by the 1.5A/V transconductance of
the LT1578 power section and the effective complex
impedance from output to ground. Gain rolls off smoothly
above the 160Hz pole frequency set by the 100µF output
capacitor. Phase drop is limited to about 85°. Phase
recovers and gain levels off at the zero frequency (≈16kHz)
set by capacitor ESR (0.1Ω).

Error amplifier transconductance phase and gain are shown
in Figure 11. The error amplifier can be modeled as a
transconductance of 1000µMho, with an output imped-
ance of 570kΩ in parallel with 2.4pF. In all practical
applications, the compensation network from the VC pin to
ground has a much lower impedance than the output
impedance of the amplifier at frequencies above 200Hz.

LT1578

GND

CURRENT MODE

POWER STAGE

= 1.5A/V

g

m

V

C

R

C

C

F

C

C

AMPLIFIER

ERROR

–

+

V

1.21V

SW

OUTPUT

R1

FB

ESR

+

C1

R2

1578 F09

This means that the error amplifier characteristics them­selves do not contribute excess phase shift to the loop, and
the phase/gain characteristics of the error amplifier sec­tion are completely controlled by the external compensa­tion network.

In Figure 12, full loop phase/gain characteristics are
shown with a compensation capacitor of 100pF, giving the
error amplifier a pole at 2.8kHz, with phase rolling off to
90° and staying there. The overall loop has a gain of 66dB
at low frequency, rolling off to unity-gain at 58kHz. The
phase plot shows a two-pole characteristic until the ESR
of the output capacitor brings it back to single pole above
16kHz. Phase margin is about 77° at unity-gain.

2000

1500

1000

500

V

GAIN (µMho)

FB

0

ERROR AMPLIFIER EQUIVALENT CIRCUIT

R

LOAD

–500

10 1k 10k 1M

PHASE

GAIN

–3

1 × 10

)(

= 50Ω

100 100k

R

OUT

570k

FREQUENCY (Hz)

C

2.4pF

OUT

V

C

1578 F11

200

150

PHASE (DEG)

100

50

0

–50

20

Figure 9. Model for Loop Response Figure 11. Error Amplifier Gain and Phase

40

20

0

GAIN (dB)

–20

–40

10

PHASE

100 1k

FREQUENCY (Hz)

GAIN

VIN = 10V

= 5V

V

OUT

= 500mA

I

OUT

10k 100k

40

0

–40

–80

–120

1578 F07

Figure 10. Response from VC Pin to Output

PHASE (DEG)

80

60

40

VIN = 10V

20

LOOP GAIN (dB)

–20

= 5V

V

OUT

= 500mA

I

OUT

= 100µF

C

OUT

0

10V, AVX TPS

= 100pF

C

C

L = 30µH

100 100k

10 1k 10k 1M

FREQUENCY (Hz)

PHASE

GAIN

1578 F12

Figure 12. Overall Loop Characteristics

180

135

LOOP PHASE (DEG)

90

45

0

–45

LT1578/LT1578-2.5

V

k

V

C RIPPLE

()

−

−

=

()

()

−

()()()

()

()()

=

15 1 10 10 5 01 121

10 30 10 200 10

0 151

3

63

•..

••

.

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APPLICATIONS INFORMATION

Analog experts will note that around 7kHz, phase dips
close to the zero phase margin line. This is typical of
switching regulators, especially those that operate over a
wide range of loads. This region of low phase is not a
problem as long as it does not occur near unity-gain. In
practice, the variability of output capacitor ESR tends to
dominate all other effects with respect to loop response.
Variations in ESR
but at the same time phase moves with it so that adequate
phase margin is maintained over a very wide range of ESR
(≥ ±3:1).

What About a Resistor in the Compensation Network?

It is common practice in switching regulator design to add
a “zero” to the error amplifier compensation to increase
loop phase margin. This zero is created in the external
network in the form of a resistor (RC) in series with the
compensation capacitor. Increasing the size of this resis­tor generally creates better and better loop stability, but
there are two limitations on its value. First, the combina­tion of output capacitor ESR and a large value for RC may
cause loop gain to stop rolling off altogether, creating a
gain margin problem. An approximate formula for R
where gain margin falls to zero is:

will

cause unity-gain to move around,

C

evidenced by alternating pulse widths seen at the switch
node. In more severe cases, the regulator squeals or
hisses audibly even though the output voltage is still
roughly correct. None of this will show on a Bode plot
since this is an amplitude insensitive measurement.

have shown that if ripple voltage on the VC is held to less
than 100mV

The formula below will give an estimate of VC ripple
voltage when RC is added to the loop, assuming that RC is
large compared to the reactance of CC at 200kHz.

V

C RIPPLE

()

GMA = Error amplifier transconductance (1000µMho)
If a series compensation resistor of 15k gave the best

overall loop response, with adequate gain margin, the
resulting VC pin ripple voltage with VIN = 10V, V
ESR = 0.1Ω, L = 30µH, would be:

, the LT1578 will generally be well behaved

P-P

R G V V ESR

()( )

C MA IN OUT

=

−

()()()

VLf

()()()

IN

121.

OUT

Tests

.

= 5V,

V

R Loop

C

GMP = Transconductance of power stage = 1.5A/V
GMA = Error amplifier transconductance = 1(10–3)
ESR = Output capacitor ESR

1.21 = Reference voltage
With V

would yield zero gain margin, so this represents an upper
limit. There is a second limitation however which has
nothing to do with theoretical small signal dynamics. This
resistor sets high frequency gain of the error amplifier,
including the gain at the switching frequency. If the
switching frequency gain is high enough, an excessive
amout of output ripple voltage will appear at the VC pin
resulting in improper operation of the regulator. In a
marginal case,

Gain =1

()

= 5V and ESR = 0.1Ω, a value of 27.5k for R

OUT

=

G G ESR

()()()()

MP MA

subharmonic

OUT

121.

C

switching occurs, as

This ripple voltage is high enough to possibly create
subharmonic switching. In most situations a compromise
value (<10k in this case) for the resistor gives acceptable
phase margin and no subharmonic problems. In other
cases, the resistor may have to be larger to get acceptable
phase response, and some means must be used to control
ripple voltage at the VC pin. The suggested way to do this
is to add a capacitor (CF) in parallel with the RC/CC network
on the VC pin. The pole frequency for this capacitor is
typically set at one-fifth of the switching frequency so that
it provides significant attenuation of the switching ripple,
but does not add unacceptable phase shift at the loop
unity-gain frequency. With RC = 15k,

C

=

F

5

π

fR

2

()()()

=

π •

2 200 10 15

C

5

3

k

()

()

=

265

pF

21

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

How Do I Test Loop Stability?

The “standard” compensation for LT1578 is a 100pF
capacitor for CC, with RC = 0. While this compensation will
work for most applications, the “optimum” value for loop
compensation components depends, to various extents,
on parameters which are not well controlled. These in­clude

inductor value

load current and ripple current variations),

tance

(±20% to ±50% due to production tolerance,

temperature, aging and changes at the load),

capacitor ESR

temperature and aging), and finally,

output load current

designer to check out the final design to ensure that it is
“robust” and tolerant of all these variations.

(±30% due to production tolerance,

output capaci-

output

(±200% due to production tolerance,

DC input voltage and

. This makes it important for the

SWITCHING

REGULATOR

+

100µF TO
1000µF

One way to check switching regulator loop stability is by
pulse loading the regulator output while observing the
transient response at the output, using the circuit shown
in Figure 13. The regulator loop is “hit” with a small
transient AC load current at a relatively low frequency,
50Hz to 1kHz. This causes the output to jump a few
millivolts, then settle back to the original value, as shown
in Figure 14. A well behaved loop will settle back cleanly,
whereas a loop with poor phase or gain margin will “ring”
as it settles. The
stability, and the
approximate unity-gain frequency of the loop.

number

frequency

of rings indicates the degree of

of the ringing shows the

Amplitude

of the signal is not particularly important, as long as the
amplitude is not so high that the loop behaves nonlinearly.

RIPPLE FILTER

470Ω

3300pF 330pF

4.7k

TO X1
OSCILLOSCOPE
PROBE

ADJUSTABLE

INPUT SUPPLY

10mV/DIV

5A/DIV

ADJUSTABLE

DC LOAD

50Ω

TO
OSCILLOSCOPE
SYNC

100Hz TO 1kHz
100mV TO 1V

P-P

Figure 13. Loop Stability Test Circuit

0.2ms/DIV 1578 F14

Figure 14. Loop Stability Check

1578 F13

V

AT

OUT

= 500mA

I

OUT

BEFORE FILTER

V

AT

OUT

= 500mA

I

OUT

AFTER FILTER

V

AT

OUT

= 50mA

I

OUT

AFTER FILTER
LOAD PULSE

THROUGH 50Ω
f ≈ 780Hz

22

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APPLICATIONS INFORMATION

The output of the regulator contains both the desired low
frequency transient information and a reasonable amount
of high frequency (200kHz) ripple. The ripple makes it
difficult to observe the small transient, so a two-pole,
100kHz filter has been added. This filter is not particularly
critical; even if it attenuated the transient signal slightly,
this wouldn’t matter because amplitude is not critical.

After verifying that the setup is working correctly, start
varying load current and input voltage to see if you can find
any combination that makes the transient response look
suspiciously “ringy.” This procedure may lead to an ad­justment for best loop stability or faster loop transient
response. Nearly always you will find that loop response
looks better if you add in several kΩ for RC. Do this only
if necessary, because as explained before, RC above 1k
may require the addition of CF to control VC pin ripple.
If everything looks OK, use a heat gun and cold spray on
the circuit (especially the output capacitor) to bring out
any temperature-dependent characteristics.

Keep in mind that this procedure does not take initial
component tolerance into account. You should see fairly
clean response under all load and line conditions to ensure
that component variations will not cause problems. One
note here: according to Murphy, the component most
likely to be changed in production is the output capacitor,
because that is the component most likely to have manu­facturer variations (in ESR) large enough to cause prob­lems. It would be a wise move to lock down the sources of
the output capacitor in production. Also, try varying com­ponent values by a factor of 2 and see if the behavior is still
acceptable. Double and halve the values of RC and CC and
output capacitors. If the regulator still works correctly, it
will likely be good in production.

A possible exception to the “clean response” rule is at very
light loads, as evidenced in Figure 14 with I
Switching regulators tend to have dramatic shifts in loop
response at very light loads, mostly because the inductor
current becomes discontinuous. One common result is very
slow but stable characteristics. A second possibility is low
phase margin, as evidenced by ringing at the output with
transients. The good news is that the low phase margin at

LOAD

= 50mA.

light loads is not particularly sensitive to component varia­tion, so if it looks reasonable under a transient test, it will
probably not be a problem in production. Note that

quency

of the light load ringing may vary with component

fre-

tolerance but phase margin generally hangs in there.

POSITIVE-TO-NEGATIVE CONVERTER

The circuit in Figure 15 is a classic positive-to-negative
topology using a grounded inductor. It differs from the
standard approach in the way the IC chip derives its
feedback signal. Because the LT1578 accepts only posi­tive feedback signals, the ground pin must be tied to the
regulated negative output. A resistor divider to ground or,
in this case, the sense pin, then provides the proper
feedback voltage for the chip.

D1

1N4148

C2

L1*

0.33µF

4.99k

D2
1N5818

15µH

R1

15.8k

C1

R2

+

100µF
10V TANT
×2

OUTPUT**
–5V, 0.5A

1578 F15

INPUT

5.5V TO

15V

+

C3

10µF TO

50µF

* INCREASE L1 TO 30µH OR 60µH FOR HIGHER CURRENT APPLICATIONS.

SEE APPLICATIONS INFORMATION

** MAXIMUM LOAD CURRENT DEPENDS ON MINIMUM INPUT VOLTAGE

AND INDUCTOR SIZE. SEE APPLICATIONS INFORMATION

Figure 15. Positive-to-Negative Converter

BOOST

V

IN

LT1578

GND

V

SW

FB

V

C

C

C

R

C

Inverting regulators differ from buck regulators in the
basic switching network. Current is delivered to the output
as

square waves with a peak-to-peak amplitude much

greater than load current

. This means that

maximum load
current will be significantly less than the LT1578’s 1.5A
maximum switch current, even with large inductor values

.
The buck converter in comparison, delivers current to the
output as a triangular wave superimposed on a DC level
equal to load current, and load current can approach 1.5A

23

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

with large inductors. Output ripple voltage for the positive­to-negative converter will be much higher than a buck
converter. Ripple current in the output capacitor will also
be much higher. The following equations can be used to
calculate operating conditions for the positive-to-negative
converter.

Maximum load current:




VV

()

OUT IN



−

035

()



035..

()

OUT

+

= 5V, L = 30µH,

P

I

MAX




I

−

P




=

VV

()( )

IN OUT

VVfL

2

()()()

+

OUT IN

VV VV

+−

()

OUT IN OUT F

IP = Maximum rated switch current
VIN = Minimum input voltage
V

= Output voltage

OUT

VF = Catch diode forward voltage

0.35 = Switch voltage drop at 1.5A
Example: with V

VF = 0.5V, IP = 1.5A: I

IN(MIN)

MAX

= 5.5V, V

= 0.6A. Note that this equation
does not take into account that maximum rated switch
current (IP) on the LT1578 is reduced slightly for duty
cycles above 50%. If duty cycle is expected to exceed 50%
(input voltage less than output voltage), use the actual I
value from the Electrical Characteristics table.

Operating duty cycle:

This duty cycle is close enough to 50% that IP can be
assumed to be 1.5A.

OUTPUT DIVIDER

If the adjustable part is used, the resistor connected to
V

(R2) should be set to approximately 5k. R1 is

OUT

calculated from:

RV

2121

R

1

=

−

121

.

.

()

OUT

INDUCTOR VALUE

Unlike buck converters, positive-to-negative converters
cannot use large inductor values to reduce output ripple
voltage. At 200kHz, values larger than 75µH make almost
no change in output ripple. The graph in Figure 16 shows
peak-to-peak output ripple voltage for a 5V to –5V con­verter versus inductor value. The criteria for choosing the

150

)

P-P

120

90

60

5V TO –5V CONVERTER
OUTPUT CAPACITOR’S
ESR = 0.1Ω

DISCONTINUOUS
I

= 0.1A

LOAD

DISCONTINUOUS
I

LOAD

= 0.25A

VV

+

DC

=

VVV

OUT F

−+ +03.

IN OUT F

(This formula uses an average value for switch loss, so it
may be several percent in error.)

With the conditions above:

+

505

DC =

−++

55 03 5 05

.. .

.

=

51

%

24

30

OUTPUT RIPPLE VOLTAGE (mV

0

0

Figure 16. Ripple Voltage on Positive-to-Negative Converter

CONTINUOUS
I

LOAD

15

30

INDUCTOR SIZE (µH)

> 0.38A

45

60

75

1578 F16

LT1578/LT1578-2.5

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APPLICATIONS INFORMATION

inductor is therefore typically based on ensuring that peak
switch current rating is not exceeded. This gives the
lowest value of inductance that can be used, but in some
cases (lower output load currents) it may give a value that
creates unnecessarily high output ripple voltage. A com­promise value is often chosen that reduces output ripple.
As you can see from the graph,
give arbitrarily low ripple, but
high ripple.

The difficulty in calculating the minimum inductor size
needed is that you must first know whether the switcher
will be in continuous or discontinuous mode at the critical
point where switch current is 1.5A. The first step is to use
the following formula to calculate the load current where
the switcher must use continuous mode. If your load
current is less than this, use the discontinuous mode
formula to calculate the minimum inductor value needed.
If the load current is higher, use the continuous mode
formula.

Output current where continuous mode is needed:

VI

()()

I

=

CONT

Minimum inductor discontinuous mode:

L

=

MIN

Minimum inductor continuous mode:

VV VV V

4

()

IN OUT IN OUT F

VI

2

()()

OUT OUT

fI

()( )

P

IN P

+

2

large

inductors will not

small

inductors can give

22

++

()

For the example above, with maximum load current of

0.25A:

22

55 15

..

IA

()()

=

CONT

This says that discontinuous mode can be used and the
minimum inductor needed is found from:

LH

MIN

In practice, the inductor should be increased by about 30%
over the calculated minimum to handle losses and varia­tions in value. This suggests a minimum inductor of 7.3µH
for this application, but looking at the ripple voltage chart
shows that output ripple voltage could be reduced by a fac­tor of two by using a 30µH inductor. There is no rule of thumb
here to make a final decision. If modest ripple is needed and
the larger inductor does the trick, this is probably the best
solution. If ripple is noncritical use the smaller inductor. If
ripple is extremely critical, a second stage filter may have
to be added in any case, and the lower value of inductance
can be used. Keep in mind that the output capacitor is the
other critical factor in determining output ripple voltage.
Ripple shown on the graph (Figure 16) is with a capacitor’s
ESR of 0.1Ω. This is
“E” size surface mount solid tantalum capacitors, but the
final capacitor chosen must be looked at carefully for ESR
characteristics.

455555505

.. .

+

()

25 025

.

=

()( )



200 10 1 5



3

•.

reasonable for AVX type TPS “D” or

++

()

=

2



()



56

.µ

=

038

.

L

VV

()( )

=

MIN

21

fV V I I

+

()

()

IN OUT P OUT

IN OUT








−+






VV

+

()

OUT

V

IN





F













25

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APPLICATIONS INFORMATION

Ripple Current in the Input and Output Capacitors

Positive-to-negative converters have high ripple current in
both the input and output capacitors. For long capacitor
lifetime, the RMS value of this current must be less than
the high frequency ripple current rating of the capacitor.
The following formula will give an
RMS ripple current.

conduction mode and a large inductor value

tors will give somewhat higher ripple current, especially in
discontinuous mode. The exact formulas are very com­plex and appear in Application Note 44, pages 30 and 31.
For our purposes here, a simple fudge factor (ff) is added.
The value for ff is about 1.2 for load currents above 0.38A
(in continuous conduction mode) and L ≥10µH. It in-
creases to about 2.0 for smaller inductors at lower load
currents (in discontinuous conduction mode).

Capacitor ff I

ff = Fudge factor (1.2 to 2.0)

I

RMS

This formula assumes continuous

=

()( )

OUT

approximate

V

OUT

V

IN

value for

. Small induc-

Diode Current

Average

current will be considerably higher.
Peak diode current:

Keep in mind that during start-up and output overloads,
the average diode current may be much higher than with
normal loads. Care should be used if diodes rated less than
1A are used, especially if continuous overload conditions
must be tolerated.

diode current is equal to load current.

Continuous

()

I

OUT

Discontinuous

Mode

VV

+

IN OUT

V

IN

Mode =

=

VV

()( )

+

IN OUT

2

LfV V

()()

()

IN OUT

2I

()( )

OUT

()()

+

V

Lf

OUT

Peak

diode

26

PACKAGE DESCRIPTION

LT1578/LT1578-2.5

U

Dimensions in inches (millimeters) unless otherwise noted.

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.189 – 0.197*
(4.801 – 5.004)

7

8

5

6

0.228 – 0.244

(5.791 – 6.197)

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

× 45°

0°– 8° TYP

0.016 – 0.050

(0.406 – 1.270)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.150 – 0.157**
(3.810 – 3.988)

1

3

2

4

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

BSC

SO8 1298

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

27

LT1578/LT1578-2.5

TYPICAL APPLICATION

U

Dual Output SEPIC␣ Converter

The circuit in Figure 17 generates both positive and
negative 5V outputs with a single piece of magnetics. The
inductor L1 is a 33µH surface mount inductor from
Coiltronics. It is manufactured with two identical windings
that can be connected in series or parallel. The topology for
the 5V output is a standard buck converter. The –5V
topology would be a simple flyback winding coupled to the
buck converter if C4 were not present. C4 creates the
SEPIC (Single-Ended Primary Inductance Converter) to­pology which improves regulation and reduces ripple
current in L1. Without C4, the voltage swing on L1B
compared to L1A would vary due to relative loading and

INPUT

6V

TO 15V

+

C3
22µF

GND

* L1 IS A SINGLE CORE WITH TWO WINDINGS

COILTRONICS CTX33-2

** AVX TSPD107M010

†

IF LOAD CAN GO TO ZERO, AN OPTIONAL
PRELOAD OF 1k TO 5k MAY BE USED TO
IMPROVE LOAD REGULATION

35V TANT

V

IN

SHDN

GND

BOOST

LT1578

V

SW

FB

V

C

C

C

100pF

C4**

100µF

coupling losses. C4 provides a low impedance path to
maintain an equal voltage swing in L1B, improving regu­lation. In a flyback converter, during switch on time, all the
converter’s energy is stored in L1A only, since no current
flows in L1B. At switch off, energy is transferred by
magnetic coupling into L1B, powering the –5V rail. C4
pulls L1B positive during switch on time, causing current
to flow, and energy to build in L1B and C4. At switch off,
the energy stored in both L1B and C4 supply the –5V rail.
This reduces the current in L1A and changes L1B current
waveform from square to triangular. For details on this
circuit see Design Note 100.

C2

0.33µF

+

L1A*

D1
1N5818

L1B*

D2

1N914

33µH

1N5818

D3

15.8k

R2

4.99k

OUTPUT
5V

R1

+

C1**
100µF
10V TANT

+

C5**
100µF
10V TANT

OUTPUT

†

–5V

1578 F17

Figure 17. Dual Output SEPIC Converter

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Burst Mode is a trademark of Linear Technology Corporation.

1578f LT/TP 0100 4K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1999

28

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

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