Datasheet LT1534-1, LT1534 Datasheet (Linear Technology)

FEATURES

■

Greatly Reduced Conducted and Radiated EMI

■

Low Switching Harmonic Content

■

Independent Control of Switch Voltage and
Current Slew Rates

■

2A Current Limited Power Switch

■

Regulates Positive and Negative Voltages

■

20kHz to 250kHz Oscillator Frequency

■

Easily Synchronized to External Clock

■

Wide Input Voltage Range: 2.7V to 23V

■

Low Shutdown Current: 12µA Typical

■

Easier Layout than with Conventional Switchers

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APPLICATIO S

■

Precision Instrumentation Systems

■

Isolated Supplies for Industrial Automation

■

Medical Instruments

■

Wireless Communications

■

Single Board Data Acquisition Systems

LT1534/LT1534-1
Ultralow Noise

2A Switching Regulators

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DESCRIPTIO

The LT®1534/LT1534-1 are a new class of switching regula­tor designed to reduce conducted and radiated electromag­netic interference (EMI). Ultralow noise and EMI are achieved
by providing user control of the output switch slew rates.
Voltage and current slew rates can be independently pro­grammed to optimize switcher harmonic content versus
efficiency. The LT1534/LT1534-1 can reduce high frequency
harmonic power by as much as 40dB with only minor losses
in efficiency.

The LT1534/LT1534-1 utilize a current mode architecture
optimized for low noise boost topologies. The ICs include a
2A power switch along with all necessary oscillator, control
and protection circuitry. Unique error amp circuitry can
regulate both positive and negative voltages. The internal
oscillator may be synchronized to an external clock for more
accurate placement of switching harmonics. Protection fea­tures include cycle-by-cycle current limit protection, under­voltage lockout and thermal shutdown.

Low minimum supply voltage and low supply current during
shutdown make the LT1534/LT1534-1 well suited for por­table applications. The LT1534/LT1534-1 are available in the
16-pin narrow SO package.

, LTC and LT are registered trademarks of Linear Technology Corporation.

TYPICAL APPLICATION

L1

3.3V

50µH

+

220pF

C
33µF

6.3V

3300pF

IN

16.9k

6.8k

15nF

12

SHDN

4

SYNC

5

C

6

R

11

V

15

V

T

T

C

GND

IN

LT1534-1

9,16

PGND

NFB

R

R

101,8,

COL

VSL

CSL

2

3

14

13

7

FB

2.49k

Figure 1. Low Noise 3.3V to 5V Boost Converter

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1N5817

10Ω

L2

24k

24k

A

C1

+

47µF

6.3V
×2

1nF

7.5k

CIN: MATSUSHITA ECGCOJB330
C1, C2: MATSUSHITA ECGCOJB47O
L1, L3: COILTRONICS CTX50-4
L2: COILCRAFT B08T (28nH) OR PC TRACE

1534 TA01

L3

50µH

OPTIONAL

B

+

C2
47µF

5V
650mA

50mV/DIV

2mV/DIV

5V Output Noise (BW = 100MHz)

A

B

10µs/DIV

1534 TA02

1

LT1534/LT1534-1

WW

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ABSOLUTE MAXIMUM RATINGS

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(Note 1)

Input Voltage (VIN) .................................................. 30V

Switch Voltage (COL) .............................................. 35V

SHDN Pin Voltage.................................................... 30V

Feedback Pin Current (FB) .................................... 10mA

Negative Feedback Pin Current (NFB) .................. ±10mA

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PACKAGE/ORDER INFORMATION

ORDER PART

NUMBER

LT1534CS
LT1534IS

1

NC

2

COL

3

*NC

4

SYNC

5

C

T

6

R

T

7

FB

8

NFB

16-LEAD PLASTIC SO

*DO NOT CONNECT

T

JMAX

TOP VIEW

16
15
14
13
12
11
10

9

S PACKAGE

= 125°C, θJA = 100°C/W

PGND
COL
V

IN

R

VSL

R

CSL

SHDN
V

C

GND

Operating Junction Temperature Range

LT1534C................................................ 0°C to 125°C

LT1534I ............................................ –40°C to 125°C

Maximum Junction Temperature .......................... 125°C

Storage Temperature Range ................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec.)................. 300°C

TOP VIEW

1

GND

2

COL

3

PGND

4

SYNC

5

C

T

6

R

T

7

FB

8

GND

S PACKAGE

**FOUR CORNER PINS ARE FUSED TO INTERNAL

CONNECT THESE FOUR PINS TO EXPANDED

16-LEAD PLASTIC SO

DIE ATTACH PADDLE FOR HEAT SINKING.

PC LANDS FOR PROPER HEAT SINKING.

T

= 125°C, θJA = 50°C/W

JMAX

16

GND

15

V

IN

14

R

VSL

13

R

CSL

12

SHDN

11

V

C

10

NFB

9

GND

ORDER PART

NUMBER

LT1534CS-1
LT1534IS-1

Consult factory for Military grade parts.

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VC = 0.9V, VFB = V

The ● denotes the specifications which apply over the full operating

. COL, SHDN, NFB, all other pins open

REF

unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Supply and Protection

V

IN

V

IN(MIN)

I

VIN

I

VIN(OFF)

V

SHDN

I

SHDN

Error Amplifiers

V

REF

I

FB

FB

REG

Recommended Operating Range ● 2.7 23 V
Minimum Input Voltage ● 2.55 2.7 V
Operating Supply Current 2.7V ≤ VIN ≤ 23V, R
Shutdown Supply Current 2.7V ≤ VIN ≤ 23V, V

2.7V ≤ V

≤ 23V, V

IN

, R

VSL

SHDN
SHDN

, RT = 17k ● 12 30 mA

CSL

= 0V ● 12 50 µA

= 0V 12 30 µA
Shutdown Threshold 2.7V ≤ VIN ≤ 23V ● 0.4 0.8 1.2 V
Shutdown Input Current –2 µA

Reference Voltage Measured at Feedback Pin 1.235 1.250 1.265 V

● 1.215 1.250 1.275 V

Feedback Input Current VFB = V

REF

● 250 900 nA

Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 23V ● 0.003 0.03 %/V

2

LT1534/LT1534-1

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, VC = 0.9V, VFB = V

The ● denotes the specifications which apply over the full operating

. COL, SHDN, NFB, all other pins open

REF

unless otherwise noted.

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS
Error Amplifiers

V

NFR

I

NFR

NFB

REG

g

m

I

ESK

I

ESRC

V

CLH

V

CLL

A

V

Oscillator and Sync

f

MAX

f

SYNC

R

SYNC

V

FBfs

Output Switches

DC

MAX

t

IBL

BV

COL

R

ON

I

LIM

∆IIN/∆I

Slew Control

V

SLEWR

V

SLEWF

I

SLEWR

I

SLEWF

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Negative Feedback Reference Voltage Measured at Negative Feedback Pin with ● –2.550 –2.500 –2.420 V

Feedback Pin Open

Negative Feedback Input Current V

NFB

= V

NFR

● –37 –25 µA

Negative Feedback Reference Voltage 2.7V ≤ VIN ≤ 23V ● 0.002 0.05 %/V
Line Regulation

Error Amplifier Transconductance ∆IC = ±25µA 1100 1500 1900 µmho

● 700 2300 µmho

Error Amplifier Sink Current VFB = V
Error Amplifier Source Current VFB = V

+ 150mV, VC = 0.9V, V

REF

– 150mV, VC = 0.9V, V

REF

= 1V ● 120 200 350 µA

SHDN

= 1V ● 120 200 350 µA

SHDN

Error Amplifier Clamp Voltage High Clamp, VFB = 1V 1.33 V
Error Amplifier Clamp Voltage Low Clamp, VFB = 1.5V 0.1 V
Error Amplifier Voltage Gain 180 250 V/V

Maximum Switch Frequency 250 kHz
Synchronization Frequency Range f

= 250kHz ● 375 kHz

OSC

SYNC Pin Input Resistance 40 kΩ
FB Pin Threshold for Frequency Shift 5% Reduction from Nominal 0.4 V

Maximum Switch Duty Cycle R

VSL

= R

CSL

= 4.9k, f

= 25kHz ● 88 91 %

OSC

Switch Current Limit Blanking Time 200 ns
Output Switch Breakdown Voltage 2.7V ≤ VIN ≤ 23V ● 35 V
Output Switch-On Resistance I

= 1.5A, Both COL Pins Tied Together ● 0.25 0.43 Ω

COL

Switch Current Limit Duty Cycle = 30% 2 A

Duty Cycle = 80% 1.6 A

Supply Current Increase During 16 mA/A

SW

Switch-On Time

Output Voltage Slew Rising Edge R
Output Voltage Slew Falling Edge R
Output Current Slew Rising Edge R
Output Current Slew Falling Edge R

VSL

VSL

VSL

VSL

, R

= 17k 11 V/µs

CSL

, R

= 17k 14.5 V/µs

CSL

, R

= 17k 1.3 A/µs

CSL

, R

= 17k 1.3 A/µs

CSL

3

LT1534/LT1534-1

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TYPICAL PERFORMANCE CHARACTERISTICS

Minimum Input Voltage (VIN)
vs Temperature

2.70

2.65

2.60

2.55

INPUT VOLTAGE (V)

2.50

2.45
–50

–25 25

0

JUNCTION TEMPERATURE (°C)

50

Feedback Voltage and
Input Current vs Temperature

1.30

1.29

1.28

1.27

1.26

1.25

1.24

1.23

FEEDBACK VOLTAGE (V)

1.22

1.21

1.20
–50

0

–25 25

125

100

75

1534 G01

V

FB

I

FB

50

TEMPERATURE (°C)

75

150

100

125

1533 G04

Change in Maximum Switch

–100

–200

(mA)

–300

LIM

∆I

–400

–500

–600

150

Current (I

0

0

2.0

1.8

FEEDBACK INPUT CURRENT (µA)

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2
0

) vs Duty Cycle

LIM

125°C

20 40 60

DUTY CYCLE (%)

85°C

25°C

80

100

1533 G02

Negative Feedback Voltage and
Input Current vs Temperature

–2.30

–2.35

–2.40

–2.45

–2.50

–2.55

–2.60

NEGATIVE FB VOLTAGE (V)

–2.65

–2.70

–25 25 75

–50

Output Switch Saturation Voltage
vs Switch Current

0.8

0.7

0.6

0.5

0.4

0.3

SWITCH VOLTAGE (V)

0.2

0.1

0

0

0

TEMPERATURE (°C)

0.5
SWITCH CURRENT (A)

50

25°C

100

V

I

NFB

125

1.0

NFB

1533 G05

85°C

150

150°C

35

30

25

20

15

NFB IINPUT CURRENT (µA)

1.5

–40°C

2.0

1534 G03

Error Amplifier Transconductance
vs Temperature

2000

gm = ∆IVC/∆V

1900
1800
1700
1600
1500
1400
1300
1200

TRANSCONDUCTANCE (mho)

1100
1000

–50 25 75 150

–25 0

FB

50

TEMPERATURE (°C)

4

100 125

1534 G08

Switching Frequency vs
Feedback Pin Voltage

100

TA = 25°C

90
80
70
60
50
40
30
20

SWITCHING FREQUENCY (% TYPICAL)

10

0

0

0.1
FEEDBACK PIN VOLTAGE (V)

0.2

0.3 0.4

0.5

1533 G06

0.6

Error Amplifier Output Current (VC)

500
400
300
200
100

0
–100
–200
–300

ERROR AMPLIFIER OUTPUT (µA)

–400
–500

–400 –300 –200 –100

FEEDBACK PIN VOLTAGE FROM NOMINAL (mV)

–40°C

0 100 300

200

125°C

25°C

400

1534 G07

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TYPICAL PERFORMANCE CHARACTERISTICS

VC Pin Threshold and Clamp
Voltage vs Temperature

1.6

1.4

1.2

1.0

0.8

0.6

PIN VOLTAGE (V)

C

V

0.4

0.2

0

–50 25 75

–25 0

VC PIN CLAMP

THRESHOLD

TEMPERATURE (°C)

VOLTAGE

VC PIN

50

100 125

1534 G09

0.5A/DIV

100mV/DIV

LT1534/LT1534-1

Load Transient Response

I

LOAD

V

OUT

VIN = 3.3V 500µs/DIV 1534 G10
V

= 5V (NODE A)

OUT

= 0.1A TO 0.5A

I

LOAD

FIGURE 1 CIRCUIT

Typical Output Switch Voltage
and Current Slew Rates vs Slew
Setting Resistance (R

40

I

35

30

25

20

15

VOLTAGE SLEW (V/µs)

10

5

0

0 1/20k 1/10k 1/6.7k 1/5k

I
V
V

RISE
FALL

RISE
FALL

1/R (mmho)

VSL

= R

CSL

TA = 25°C

(%)

OUT

∆V

–0.5

)

4

CURRENT SLEW (A/µs)

3

2

1

0

1534 G11

Load Regulation

2.0

1.5

1.0

0.5

0

FIGURE 1 CIRCUIT

Percent Change in Oscillator
Frequency vs Temperature (NPO
Cap and Metal Film R)

2.0

1.5

1.0

0.5

0

–0.5

–1.0

CHANGE IN FREQUENCY (%)

–1.5

–2.0

–50

–25 25

0

TEMPERATURE (°C)

50

125

100

75

1534 G12

–1.0

–1.5

0

100 300

200

LOAD CURRENT (mA)

400

500

600

700

1534-1 G13

5

LT1534/LT1534-1

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PIN FUNCTIONS

(LT1534/LT1534-1)

COL (Pins 2, 15/Pin 2): These two pins should be con-

nected together externally to create the collector of the
power switch. The emitter returns to PGND through a
sense resistor. Large currents flow into these pins so it is
desirable to keep external trace lengths short to minimize
radiation.

SYNC (Pin 4): The SYNC pin can be used to synchronize
the oscillator to an external clock (see Oscillator Sync in
Applications Information section for more details). The
SYNC pin may either be floated or tied to ground if not
used.

CT (Pin 5): The oscillator capacitor pin is used in
conjunction with RT to set the oscillator frequency.
For RT = 16.9k,

C

= 129/f

T(NF)

RT (Pin 6): The oscillator resistor pin is used to set the
charge and discharge currents of the oscillator capacitor.
The nominal value is 16.9k. It is possible to adjust this
resistance ±25% to get a more accurate oscillator fre­quency.

FB (Pin 7): The feedback pin is used for positive voltage
sensing and oscillator frequency shifting during start-up
and short-circuit conditions. It is the inverting input to the
error amplifier. The noninverting input of this amplifier
connects internally to a 1.25V reference. This pin should
be left open if not used.

NFB (Pin 8/Pin 10): The negative voltage feedback pin is
used for sensing a negative output voltage. The pin is
connected to the inverting input of the negative feedback
amplifier through a 100k source resistor. The negative
feedback amplifier provides a gain of – 0.5 to the feedback
amplifier; therefore, the nominal regulation point is –2.5V
on NFB. This pin should be left open if not used.

GND (Pin 9/Pins 1, 8, 9, 16): Signal Ground. The internal
error amplifier, negative feedback amplifier, oscillator,
slew control circuitry and the bandgap reference are

OSC(kHz)

referred to this ground. Keep the connection to the feed­back divider and VC compensation network free of large
ground currents.

VC (Pin 10/Pin 11): The compensation pin is used for
frequency compensation and current limiting. It is the
output of the error amplifier and the input of the current
comparator. Loop frequency compensation can be per­formed with an RC network connected from the VC pin to
ground.

SHDN (Pin 11/Pin 12): The shutdown pin is used for
disabling the switcher. Grounding this pin will disable all
internal circuitry. Normally this output can be tied high (to
VIN) or may be left floating.

R

(Pin 12/Pin 13): A resistor to ground sets the current

CSL

slew rate for the power switch. The minimum resistor
value is 3.9k and the maximum value is 68k. Current slew
will be approximately:

I

SLEW(A/µs)

R

(Pin 13/Pin 14): A resistor to ground sets the voltage

VSL

slew rate for the power switch collector. The minimum
resistor value is 3.9k and the maximum value is 68k.
Voltage slew will be approximately:

V

SLEW(V/µs)

VIN (Pin 14/Pin 15): Input Supply Pin. Bypass this pin with
a ≥ 4.7µF low ESR capacitor. When VIN is below 2.55V the
part will go into undervoltage lockout where it will stop
output switching and pull the VC pin low.

PGND (Pin 16/Pin 3): Power Switch Ground. This ground
comes from the emitters of the power switches. In normal
operation this pin should have approximately 25nH induc­tance to ground. This can be done by trace inductance
(approximately 1") or with wire or a specific inductive
component (e.g., small ferrite bead). This inductance
ensures stability in the current slew control loop during
turn-off. Too much inductance (>50nH) may produce
oscillation on the output voltage slew edges.

= 33/R

= 220/R

CSL(kΩ)

VSL(kΩ)

6

BLOCK DIAGRA

V

C

+

NEGATIVE
FEEDBACK

AMP

–

NFB

SYNC

FB

R

T

C

T

100k 50k

W

1.25V

LT1534/LT1534-1

SHDN V

LDO REGULATOR

–

g

m

ERROR

AMP

+

+

OSCILLATOR

INTERNAL V

–

COMP

+

GND

IN

CC

PGND COL COL

+

–

SQ

FF

R

OUTPUT

DRIVER

SLEW CONTROL

1534 BD

R

VSL

R

CSL

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OPERATIO

In noise sensitive applications, switching regulators tend
to be ruled out as a power supply option due to their
propensity for generating unwanted noise. When switch­ing supplies are required due to efficiency or input/output
voltage constraints, great pains must be taken to work
around the noise generated by a typical supply. These
steps may include precise synchronization of the power
supply oscillator to an external clock, synchronizing the
rest of the circuit to the power supply oscillator, or halting
power supply switching during noise sensitive operations.
The LT1534 greatly simplifies the task of eliminating
supply noise by enabling the design of an inherently low
noise switching regulator power supply.

The LT1534 is a fixed frequency, current mode switching
regulator with unique circuitry to control the voltage and
current slew rates of the output switch. Slew control
capability provides much greater control over power sup­ply components that can create conducted and radiated
electromagnetic interference. The current mode control
provides excellent AC and DC line regulation and simplifies
loop compensation.

Current Mode Control

A switching cycle begins with an oscillator discharge pulse
which resets the RS flip-flop, turning on the output driver

(refer to Block Diagram). The switch current is sensed
across an internal resistor and the resulting voltage is
amplified and compared to the output of the error amplifier
(VC pin). The driver is turned off once the output of the
current sense amplifier exceeds the voltage on the VC pin.
Internal slope compensation is provided to ensure stabil­ity under high duty cycle conditions.

Output regulation is obtained using the error amp to set
the switch current trip point. The error amp is a transcon­ductance amplifier that integrates the difference between
the feedback output voltage and an internal 1.25V refer­ence. The output of the error amp adjusts the switch
current trip point to provide the required load current at
the desired regulated output voltage. This method of
controlling current rather than voltage provides faster
input transient response, cycle by cycle current limiting
for better output switch protection and greater ease in
compensating the feedback loop.

The VC pin serves three different purposes. It is used for
loop compensation, current limit adjustment and soft
starting. During normal operation the VC voltage will be
between 0.2V and 1.33V. An external clamp may be used
for lowering the current limit. A capacitor coupled to an
external clamp can be used for soft starting.

7

LT1534/LT1534-1

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OPERATIO

The negative feedback amplifier allows for direct regula­tion of negative output voltages. The voltage on the NFB
pin gets amplified by a gain of – 0.5 and driven onto the FB
input, i.e., the NFB pin regulates to –2.5V while the
amplifier output internally drives the FB pin to 1.25V as in
normal operation. The negative feedback amplifier input
impedance is 100k (typ) referred to ground.

Slew Control

Control of output voltage and current slew rates is done via
two feedback loops. One loop controls the output switch
collector voltage dV/dt and the other loop controls the
emitter current dI/dt. Output slew control is achieved by
comparing the currents generated by these two slewing
events to currents created by external resistors R
R

. The two control loops are combined internally to

CSL

provide a smooth transition from current slew control to
voltage slew control.

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WUU

VSL

and

APPLICATIONS INFORMATION

Internal Regulator

Most of the control circuitry operates from an internal 2.4V
low dropout regulator that is powered from VIN. The
internal low dropout design allows VIN to vary from 2.7V
to 23V with virtually no change in device performance.
When the part is put into shutdown, the internal regulator
is turned off, leaving only a small (12µA typ) current drain
from VIN.

Protection Features

There are three modes of protection in the LT1534. The
first is overcurrent limit. This is achieved via the clamping
action of the VC pin. The second is thermal shutdown that
disables both output drivers and pulls the VC pin low in the
event of excessive chip temperature. The third is under­voltage lockout that also disables both outputs
and pulls the VC pin low whenever VIN drops below 2.5V.

Reducing EMI from switching power supplies has tradi­tionally invoked fear in designers. Many switchers are
designed solely on efficiency and as such produce wave­forms filled with high frequency harmonics that then
propagate through the rest of the power supply.

The LT1534 provides control over two of the more impor­tant variables for controlling EMI with switching inductive
loads: switch voltage slew rate and switch current slew
rate. The use of this part will reduce noise and EMI over
conventional switch mode controllers. Because these
variables are under control, a supply built with this part will
exhibit far less tendency to create EMI and less chance of
wandering into problems during production.

It is beyond the scope of this data sheet to get into EMI
fundamentals. AN70 contains much information concern­ing noise in switching regulators and should be consulted.

Oscillator Frequency

The oscillator determines the switching frequency and
therefore the fundamental positioning of all harmonics.
The use of good quality external components is important
to ensure oscillator frequency stability. The oscillator is a

sawtooth design. A current defined by external resistor R
is used to charge and discharge the capacitor CT. The
discharge rate is approximately ten times the charge rate.

By allowing the user to have control over both compo­nents, trimming of oscillator frequency can be more easily
achieved.

The external capacitance CT is chosen by:

C

= 2180/[f

T(nF)

where f
For RT equal to 16.9k, this simplifies to:

C

T(nF)

(e.g., CT = 1.29nF for f

A good quality temperature stable capacitor should be
chosen.

Nominally RT should be 16.9k. Since it sets up current, its
temperature coefficient should be selected to compliment
the capacitor. Ideally, both should have low temperature
coefficients.

is the desired oscillator frequency in kHz.

OSC

= 129/f

OSC(kHz)

OSC(kHz)

• R

T(kΩ)

= 100kHz)

OSC

]

T

8

LT1534/LT1534-1

U

WUU

APPLICATIONS INFORMATION

If the FB pin is below 0.4V the oscillator discharge time will
increase, causing the oscillation frequency to decrease by
approximately 6:1. This feature helps minimize power
dissipation during start-up and short-circuit conditions.

Oscillator frequency is important for noise reduction in
two ways: 1) the lower the oscillator frequency the lower
the harmonics of waveforms are, making it easier to filter
them, 2) the oscillator will control the placement of output
frequency harmonics which can aid in specific problems
where you might be trying to avoid a certain frequency
bandwidth that is used for detection elsewhere.

Oscillator Sync

If a more precise frequency is desired (e.g., to accurately
place harmonics) the oscillator can be synchronized to an
external clock. Set the RC timing components for an
oscillator frequency 10% lower than the desired sync
frequency.

Drive the SYNC pin with a square wave (with greater than

1.4V amplitude). The rising edge of the sync square wave
will initiate clock discharge. The sync pulse should have a
minimum of 0.5µs pulse width.

Be careful in synchronizing to frequencies much different
from the part since the internal oscillator charge slope
determines slope compensation. It would be possible to
get into subharmonic oscillation if the sync doesn’t allow
for the charge cycle of the capacitor to initiate slope
compensation. In general, this will not be a problem until
the sync frequency is greater than 1.5 times the oscillator
free-run frequency.

Slew Rate Setting

Setting the voltage and current slew rates is easy. External
resistors to ground on the R
the slew rates. Determining what slew rate to use is more
difficult. There are several ways to approach the problem.

First start by putting a 50k resistor pot with a 3.9k series
resistance on each pin. In general, the next step will be to
monitor the noise that you are concerned with. Be careful
in measurement technique (consult AN70). Keep probe
ground leads very short.

VSL

and R

pins determine

CSL

Usually it will be desirable to keep the voltage and current
slew resistors approximately the same. There are circum­stances where a better optimization can be found by
adjusting each separately, but as these values are sepa­rated further, a loss of independence of control will occur.

Starting from the lowest resistor setting adjust the pots
until the noise level meets your guidelines. Note that
slower slewing waveforms will dissipate more power so
that efficiency will drop. You can also monitor this as you
make your slew adjustment.

It is possible to use a single slew setting resistor. In this
case the R

VSL

and R

pins are tied together. A resistor

CSL

with a value of 2k to 34k (one half the individual resistors)
can then be tied from these pins to ground.

Emitter Inductance

A small inductance in the power ground minimizes a
potential dip in the output current falling edge that can
occur under fast slewing, 25nH is usually sufficient. Greater
than 50nH may produce unwanted oscillations in the
voltage output. The inductance can be created by wire or
board trace with the equivalent of one inch of straight
length. A spiral board trace will require less length.

Positive Output Voltage Setting

Sensing of a positive output voltage is usually done using
a resistor divider from the output to the FB pin. The
positive input to the error amp is connected internally to a

1.25V bandgap reference. The FB pin will regulate to this
voltage.

R1

FB PIN

R2

Figure 2

V

OUT

1534 F01

Referring to Figure 2, R1 is determined by:

RR



12





V

OUT

125

.



1=−





The FB bias current represents a small error and can

usually be ignored for values of R1||R2 up to 10k.

9

LT1534/LT1534-1

U

WUU

APPLICATIONS INFORMATION

One word of caution. Sometimes a feedback zero is added
to the control loop by placing a capacitor across R1 above.
If the feedback zero capacitively pulls the FB pin above the
internal regulator voltage (2.4V typ), output regulation
may be disrupted. A series resistance with the feedback
pin can eliminate this potential problem.

Negative Output Voltage Setting

Negative output voltage can be sensed using the NFB pin.
In this case regulation will occur when the NFB pin is at
–2.5V. The input bias current for the NFB pin is –25µA
(I

) and must be accounted for when selecting divider

NFB

resistor values.

R1

NFB PIN

I

NFB

R2

Figure 3

Referring to Figure 3, R1 is chosen such that:

–V

1534 F02

OUT

Thermal Considerations

Computing power dissipation for this IC requires careful
attention to detail. Reduced output slewing causes the part
to dissipate more power than would occur with fast edges.
However, much improvement in noise can be produced
with modest decrease in supply efficiency.

Power dissipation is a function of topology, input voltage,
switch current and slew rates. It is impractical to come up
with an all-encompassing formula. It is therefore recom­mended that package temperature be measured in each
application. The part has an internal thermal shutdown to
prevent device destruction, but this should not replace
careful thermal design.

1. Dissipation due to input current:

PVmA



=+

VIN IN





11

60



I





where I is the average switch current.

2. Dissipation due to the driver saturation:

.

−

=

RR

12

V

OUT

•
.•

+µ

25 2 25

25

RA

A suggested value for R2 is 2.5k. The NFB pin is normally
left open if the FB pin is being used.

Dual Polarity Output Voltage Sensing

Certain applications may benefit from sensing both posi­tive and negative output voltages. When doing this each
output voltage resistor divider is individually set as previ­ously described. When both FB and NFB pins are used, the
LT1534 will act to prevent either output from going
beyond its set output voltage. The highest output (lightest
load) will dominate control of the regulator. This technique
would prevent either output from going unregulated high
at no load. However, this technique will also compromise
output load regulation.

Shutdown

If the shutdown pin is pulled low, the regulator will turn off.
The supply current will be reduced to less than 20µA.

P

= (V

VSAT

where V
approximately 0.1 + (0.2)(I), DC

)(I)(DC

SAT

is the output saturation voltage which is

SAT

MAX

)

is the maximum

MAX

duty cycle.

3. Dissipation due to output slew using approximations
for slew rates:







Rf

()

()

VSL OSC









P

=

SLEW

Note if V







VI



IN

()










33 10

()








and ∆I are small with respect to VIN and I,

SAT

2

+





2

I

∆





4



R

()

CSL

9






IV

IN

()




+

220 10

()

2

V

−



SAT

4

9

2









then:



R

()

VSL



fVI

()()()

OSC IN











P

SLEW



IR

()( )

CSL



=



33 10 220 10



()



+

99





V

()

IN

()

10

LT1534/LT1534-1

VC PIN

1534 F03

R

VC

2k

C

VC

0.01µF

C

VC2

4.7nF

U

WUU

APPLICATIONS INFORMATION

where ∆I is the ripple current in the switch, R
R

are the slew resistors and f

VSL

is the oscillator

OSC

frequency.

Power dissipation PD is the sum of these three terms. Die
junction temperature is then computed as:

TJ = T

where T

+ (PD)(θJA)

AMB

is ambient temperature and θJA is the package

AMB

thermal resistance. For the 16-pin SO with fused leads the
θJA is 50°C/W.

For example, with f

= 40kHz, 0.4A average current and

OSC

0.1A of ripple, the maximum duty cycle is 88%. Assume
slew resistors are both 17k and V

is 0.26V, then:

SAT

PD = 0.176W + 0.094W + 0.158W = 0.429W

In an S16 fused lead package the die junction temperature
would be 21°C above ambient.

Frequency Compensation

Loop frequency compensation is accomplished by way of
a series RC network on the output of the error amplifier (V
pin). Referring to Figure 4, the main pole is formed by
capacitor CVC and the output impedance of the error
amplifier (approximately 400kΩ). The series resistor R
creates a “zero” which improves loop stability and tran­sient response. A second capacitor C

, typically one-

VC2

tenth the size of the main compensation capacitor, is
sometimes used to reduce the switching frequency ripple
on the VC pin. VC pin ripple is caused by output voltage
ripple attenuated by the output divider and multiplied by
the error amplifier. Without the second capacitor, VC pin
ripple is:

VgR

125.

V

CPIN RIPPLE

where V

()( )()()

RIPPLE

=

= Output ripple (V

RIPPLE m VC

V

OUT

)

P-P

gm = Error amplifier transconductance
RVC = Series resistor on VC pin
V

= DC output voltage

OUT

To prevent irregular switching, VC pin ripple should be
kept below 50mV
maximum output load current and will also be increased if

. Worst-case VC pin ripple occurs at

P-P

CSL

and

VC

poor quality (high ESR) output capacitors are used. The
addition of a 0.0047µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RVC will also reduce VC pin ripple, but loop phase
margin may be inadequate.

Figure 4

Capacitors

While the IC reduces the source of switcher noise, it is
essential for the lowest noise, that the filter capacitors
should have low parasitic impedance. Sanyo OS-CON,
Panasonic Specialty Polymer and tantalum capacitors are
the preferred types. Aluminum electrolytics are not suit­able for this application. In general, ESR is more critical
than capacitance. At higher frequencies, ESL can also be

C

important. Paralleling capacitors can reduce both ESR and
ESL.

Design Note 95 offers more information about capacitor
selection. The following is a brief summary:

Solid tantalum capacitors have small size and low
impedance. Typically they are available for voltages
below 50V. They may have a problem with surge
currents (AVX TPS line addresses this issue).

OS-CON capacitors have very low impedance but are
only available for 25V or less. Form factor may be a
problem. Sometimes their very low ESR can cause loop
stability problems.

Ceramic capacitors are generally used for high fre­quency and high voltage bypass. They too can have
such a low ESR as to cause loop stability problems.
Often they can resonate with their ESL before ESR
becomes effective.

Specialty Polymer Aluminum: Panasonic has come out
with their series CD capacitors. While they are only
available for voltages below 16V, they have very low
ESR and good surge capability.

11

LT1534/LT1534-1

U

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APPLICATIONS INFORMATION

Input Capacitor

The ESR of this capacitor acts with high frequency current
components to produce much of the conducted noise of
the switcher. Values of 1µF to 47µF are typical with ESR
less than 0.3Ω. Place the capacitor close to the IC and
inductor.

The input capacitor can see a high surge current when a
battery of high capacitance source is connected “live.”
Some solid tantalum capacitors can fail under this con­dition. Several manufacturers have developed a line of
solid tantalum capacitors specially tested for surge capa­bility (e.g., AVX TPS series). However, even these units
may fail if the input voltage approaches the maximum
voltage rating of the capacitor. AVX recommends derat­ing capacitor voltage by 2:1 for high surge applications.

Output Filter Capacitor

Output capacitors are usually chosen on the basis of ESR
since this will determine output ripple. However, low ESR
is also needed for low output noise and this will typically
be the tougher requirement. Typically required ESR will be
less than 0.2Ω . Typical capacitance values are in the 47µF
to 500µF range. Again keep connection length as short as
possible. Table 1 shows some typical surface mount
capacitors.

Table 1

SIZE CAPACITOR ESR (MAX Ω)

E CASE AVX TPS, Sprague 593D 0.1 to 0.3

AVX TAJ 0.7 to 0.9

D CASE AVX TPS, Sprague 593D 0.1 to 0.3

AVX TAJ 0.9 to 2.0
Panasonic CD 0.05 to 0.18

C CASE AVX TPS 0.2 (Typ)

AVX TAJ 1.8 to 3.0

B CASE AVX TAJ 2.5 to 10

Fast Voltage Slew Edges

A very fast voltage slew under certain operating conditions
may produce ringing on the COL voltage waveform. While
there is small harmonic energy in this, it can be eliminated
by placing an RC network of 10Ω in series with 1000pF
from the COL pin to ground.

Switching Diodes

In general, switching diodes should be Schottky diodes
such as 1N5817-19 or MBR320-330.

Choosing the Inductor

For a boost converter, inductor selection involves trade­offs of size, maximum output power, transient response
and filtering characteristics. Higher inductor values pro­vide more output power and lower input ripple. However,
they are physically larger and can impede transient re­sponse. Low inductor values have high magnetizing cur­rent, which can reduce maximum power and increase
input current ripple.

The following procedure can be used to handle these
trade-offs:

1. Assume that the average inductor current for a boost
converter is equal to load current times V
decide whether the inductor must withstand continu­ous overload conditions. If average inductor current at
maximum load current is 0.5A, for instance, a 0.5A
inductor may not survive a continuous 1.5A overload
condition. Also be aware that boost converters are not
short-circuit protected, and under output short condi­tions, only the available current of the input supply
limits inductor current.

OUT/VIN

and

12

LT1534/LT1534-1

U

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APPLICATIONS INFORMATION

2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe­cially with smaller inductors and lighter loads, so don’t
omit this step. Powdered iron cores are forgiving be­cause they saturate softly, whereas ferrite cores satu­rate abruptly. Other core material falls in between. The
following formula assumes continuous mode opera­tion but it errors only slightly on the high side for dis­continuous mode, so it can be used for all conditions.



V

II

=+

PEAK OUT

OUT




V

IN



L = inductance value
VIN = supply voltage
V

= output voltage

OUT

I = output current
f = oscillator frequency

VV V

()

IN OUT IN

•••2

LfV

–

OUT








3. Choose a core geometry. For low EMI problems a
closed structure should be used such as a pot core, ER
core, E core or toroid (see AN70 appendix I).

4. Select an inductor that can handle peak current, aver­age current (heating effects) and fault current.

5. Finally, double check output voltage ripple. The experts
in the Linear Technology Applications department have
experience with a wide range of inductor types and can
assist you in making a good choice.

Further Help

AN70 has more information on noise in switching regula­tors and its measurement. AN19 has general information
on switcher design. The Linear Technology applications
group is always ready to lend a helping hand.

U

TYPICAL APPLICATIO S

Low Noise ±12V Dual Output Flyback Converter with Dual Polarity Output Voltage Sensing

2.7V TO 10V

10µF

P6KE-20A

1N4148

15

V

2700pF

16.9k

10k

0.047µF

IN

SHDN

4

SYNC

5

C

LT1534-1

T

6

R

T

11

V

C

GND

9,16

NFB

COL

PGND

R

VSL

R

CSL

101,8,

2.49k
1%

212

3

14

13

7

FB

MBR330

T1

17

10

4

8

2

9

3

NOTE 1

9.31k
1%

MBR330

10Ω

1nF

NOTE 1: 25nH TRACE INDUCTANCE
OR COILCRAFT B10T
L1, L2: COILTRONICS CTX50-4

L1

50µH

+

C1
47µF

+

C2
47µF

21.5k
1%

2.49k
1%

+

47µF

+

47µF

L2

50µH

OPTIONAL

T1: PHILIPS EFD-15-3F3 CORE

GAP FOR PRIMARY

L = 100µH

12V
50mA

–12V
50mA

PIN 1 TO TO 10, 16 TURNS 24AWG
PIN 3 TO 8, 45 TURNS 38AWG
PIN 4 TO 7, 45 TURNS 36AWG
PIN 2 TO 9, 16 TURNS 24AWG

1534 TA04

13

LT1534/LT1534-1

U

TYPICAL APPLICATIO S

Ultralow Noise Regulator for a Thermo-Electric Cooler,
Maintaining Sensitive Electronics at Low Temperatures

12V

110k

10k

D1: MOTOROLA MBR5320T3
L1: MIDCOM 38440
L2: COILCRAFT B07T
L3, L4: SUMIDA CD43-220
U2A, U2B: LT1490

100Ω

0.1µF

U2B

L3

22µH

L4

22µH

+
–

3.3M

0.1µF

+

+

100k

C3
22µF
16V

C4
22µF
16V

499k

R6
25k

10k

1534 TA05

0A TO 1.5A
THERMOELECTRIC
COOLER

499k

R

T

10k
NTC

R1

50mΩ

R2
10k

R3
690k

10k

0.01µF

3.3k

5.6V

C2

+

47µF

16V

1000pF

L2

22nF

PGND

V

IN

FB

LT1534-1

C

T

2200pF

D1

10Ω

COL

U1

R

T

R

R

GND

18k 10k

C1

47µF

16V

L1

100µH

1 TO 5 6 TO 10

1k

V

C

CSL

VSL

2N3904

R4

5.1k

Q5

0.1µF

+

–

U2A

+

2N3904

24k

R5

1.6k

INPUT VOLTAGE

4V TO 12V

1300pF

14

16.9k

2k

11

4

5

6

33nF

SHDN
SYNC

C

T

R

T

V

C

4.7µF

LT1534-1

GND

9,16

Ultralow Noise 5V to –3V Cuk Converter

–3V

A Cuk converter is a natural topology for a low noise

0.7A

4.7µF

1534 TA06

C1

47µF

L1B
20µH

MBR320

15

V

IN

212

COL

3

PGND

14

R

VSL

13

R

CSL

7

FB

NFB

101,8,

4.99k
1%

L1A

20µH

NOTE 1

1k

1%

NOTE 1: 25nH TRACE INDUCTANCE
OR COILCRAFT B10T
L1: COILTRONICS CTX20-4

converter. The Cuk converter is a dual of a buck boost
converter. C1 is the primary means of storing and trans­ferring energy. Like a buck boost, the DC transfer function
is approximately V

OUT/VIN

= DC/(1 – DC). The output
voltage, though negative, can be higher or lower in mag­nitude from the input. The two inductors can be separate
however, by placing them on the same winding input and
output current ripple can be greatly reduced. The addi­tional slew control provided by the LT1534 will reduce the
high frequency content even further.

PACKAGE DESCRIPTION

LT1534/LT1534-1

U

Dimensions in inches (millimeters) unless otherwise noted.

S Package

16-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.386 – 0.394*

(9.804 – 10.008)

13

16

14

15

12

11

10

9

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

×

°

45

0.016 – 0.050

(0.406 – 1.270)

0° – 8° TYP

0.228 – 0.244

(5.791 – 6.197)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.150 – 0.157**
(3.810 – 3.988)

4

5

0.050

(1.270)

BSC

3

2

1

7

6

8

0.004 – 0.010

(0.101 – 0.254)

S16 1098

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

15

LT1534/LT1534-1

TYPICAL APPLICATION

V

IN

3V TO 12V

4

SYNC

12

SHDN

10

NFB

11

V

C

R

T

6.8k

0.01µF

220pF

16.9k

U

Low Noise Wide Input Range ±5V Supply

+

C1
10µF
16V

15

V

IN

LT1534-1

GND

9,16

R

VSL

141,8

4k TO
25k

C

T

5

6

1500pF

COL

PGND

R

CSL

FB

13

2

3

7

4k TO
25k

28nH

2.49k

C2

10µF

16V

+

1

4

2

5

+

10Ω

V

OUT1

1nF

C3

10µF

16V

L2

7.5k

*TOTAL OUTPUT CURRENT ≤ 300mA
C1, C2, C3: MATSUSHITA ECGCICB6R8
C4, C5: MATSUSHITA ECGC0JB470
L2: COILCRAFT B08T OR PC TRACE
T1: COILTRONICS VP2-0216

1N5817

T1

6

3

12

9

11

1N5817
T1
8

10
T1
7

V

OUT1

5V
150mA*

+

C4
47µF

6.3V

+

C5
47µF

6.3V

V

OUT2

–5V
150mA*

1534 TA03

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RMS

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16

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com

1534fa LT/TP 0300 2K REV A • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1998