Datasheet LT1511, LT1511ISW, LT1511CSW Datasheet (Linear Technology)

FEATURES

LT1511

Constant-Current/

Constant-Voltage 3A Battery

Charger with Input Current Limiting

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DESCRIPTIO

■

Simple Design to Charge NiCd, NiMH and Lithium
Rechargeable Batteries—Charging Current
Programmed by Resistors or DAC

■

Adapter Current Loop Allows Maximum Possible
Charging Current During Computer Use

■

Precision 0.5% Accuracy for Voltage Mode Charging

■

High Efficiency Current Mode PWM with 4A Internal
Switch

■

5% Charging Current Accuracy

■

Adjustable Undervoltage Lockout

■

Automatic Shutdown When AC Adapter is Removed

■

Low Reverse Battery Drain Current: 3µA

■

Current Sensing Can Be at Either Terminal of the Battery

■

Charging Current Soft-Start

■

Shutdown Control

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APPLICATIO S

■

Chargers for NiCd, NiMH, Lead-Acid, Lithium
Rechargeable Batteries

■

Switching Regulators with Precision Current Limit

, LTC and LT are registered trademarks of Linear Technology Corporation.

*See LT1510 for 1.5A Charger

The LT®1511 current mode PWM battery charger is the
simplest, most efficient solution to fast charge modern
rechargeable batteries including lithium-ion (Li-Ion), nickel­metal-hydride (NiMH) and nickel-cadmium (NiCd) that
require constant-current and/or constant-voltage charg­ing. The internal switch is capable of delivering 3A* DC
current (4A peak current). Full-charging current can be
programmed by resistors or a DAC to within 5%. With 0.5%
reference voltage accuracy, the LT1511 meets the critical
constant-voltage charging requirement for Li-Ion cells.

A third control loop is provided to regulate the current
drawn from the AC adapter. This allows simultaneous
operation of the equipment and battery charging without
overloading the adapter. Charging current is reduced to
keep the adapter current within specified levels.

The LT1511 can charge batteries ranging from 1V to 20V.
Ground sensing of current is not required and the battery’s
negative terminal can be tied directly to ground. A saturat­ing switch running at 200kHz gives high charging effi­ciency and small inductor size. A blocking diode is not
required between the chip and the battery because the
chip goes into sleep mode and drains only 3µA when the
wall adapter is unplugged.

TYPICAL APPLICATIO

D1

MBRD340

NOTE: COMPLETE LITHIUM-ION CHARGER,
NO TERMINATION REQUIRED. R
AND C1 ARE OPTIONAL FOR I
*TOKIN OR UNITED CHEMI-CON/MARCON
CERAMIC SURFACE MOUNT
**20µH COILTRONICS CTX20-4

†

SEE APPLICATIONS INFORMATION FOR

INPUT CURRENT LIMIT AND UNDERVOLTAGE LOCKOUT

L1**
20µH

MBR0540T

IN

, R7

S4

LIMITING

U

GND

C2

0.47µF

D2

SW

BOOST

COMP1

200pF

SPIN

OVP SENSE BAT

Figure 1. 3A Lithium-Ion Battery Charger

CLP

CLN

V

CC

LT1511

UV

PROG

V

C

R

R

S3

200Ω

200Ω

1%

1%

R

S1

0.033Ω

BATTERY CURRENT

SENSE

50pF

S2

+ +

1k

R3
390k

0.25%
BATTERY
VOLTAGE SENSE

R4
162k

0.25%

10µF

0.33µF

CIN*
10µF

+

500Ω

C1
1µF

R7

300Ω

C
22µF
TANT

C
1µF

OUT

†

PROG

D3

MBRD340

†

R

S4

ADAPTER
CURRENT SENSE

R

PROG

4.93k
1%

+

4.2V

+

4.2V

V

IN

11V TO 28V

TO MAIN
SYSTEM POWER

R5†
UNDERVOLTAGE
LOCKOUT

R6
5k

V

BAT

2 Li-Ion

1511 • F01

(ADAPTER INPUT)

1

LT1511

WW

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ABSOLUTE MAXIMUM RATINGS

(Note 1)

Supply Voltage

(V

, CLP and CLN Pin Voltage) ...................... 30V

MAX

Switch Voltage with Respect to GND...................... –3V

Boost Pin Voltage with Respect to VCC................... 25V

Boost Pin Voltage with Respect to GND ................. 57V

Boost Pin Voltage with Respect to SW Pin .............. 30V

VC, PROG, OVP Pin Voltage...................................... 8V

I

(Average)........................................................... 3A

BAT

Switch Current (Peak) .............................................. 4A

Operating Junction Temperature Range

Commercial ...........................................0°C to 125°C

Industrial ......................................... –40°C to 125°C

Operating Ambient Temperature

Commercial ............................................ 0°C to 70°C

Industrial ........................................... –40°C to 85°C

Storage Temperature Range................. –65°C to 150°C

Lead Temperature (Soldering, 10 sec)..................300°C

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PACKAGE/ORDER INFORMATION

TOP VIEW

1

GND**

2

SW

3

BOOST

4

GND**

5

GND**

6

UV

7

GND**

8

OVP

9

CLP

10

CLN

11

COMP1

12

SENSE

24-LEAD PLASTIC SO WIDE

T

= 125°C, θJA = 30°C/W**

JMAX

SW PACKAGE

24

GND**

23

GND**

22

V

CC1

21

V

CC2

20

V

CC3

19

PROG

18

V

C

17

UV

OUT

16

GND**

15

COMP2

14

BAT

13

SPIN

Consult factory for Military grade parts.

*
*
*

ORDER PART

NUMBER

LT1511CSW
LT1511ISW

*ALL V

**ALL GND PINS ARE

PINS SHOULD

CC

BE CONNECTED
TOGETHER CLOSE TO
THE PINS

FUSED TO INTERNAL DIE
ATTACH PADDLE FOR
HEAT SINKING. CONNECT
THESE PINS TO
EXPANDED PC LANDS
FOR PROPER HEAT
SINKING. 30°C/W
THERMAL RESISTANCE
ASSUMES AN INTERNAL
GROUND PLANE
DOUBLING AS A HEAT
SPREADER

UU

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, V
RS2 = RS3 = 200Ω (see Block Diagram), V

= VCC. No load on any outputs unless otherwise noted.

CLN

The ● denotes specifications which apply over the full operating

BAT

= 8V, V

(maximum operating VCC) = 28V,

MAX

PARAMETER CONDITIONS MIN TYP MAX UNITS
Overall

Supply Current V

Sense Amplifier CA1 Gain and Input Offset Voltage 8V ≤ V
(With R
(Measured across R

= 200Ω, RS3 = 200Ω)R

S2

)(Note 2) R

S1

= 2.7V, VCC ≤ 20V ● 4.5 6.8 mA

PROG

V

= 2.7V, 20V < VCC ≤ 25V ● 4.6 7.0 mA

PROG

≤ 25V , 0V ≤ V

CC

= 4.93k ● 95 100 105 mV

PROG

= 49.3k ● 81012 mV

PROG

< 0°C 7 13 mV

T

J

= 28V, V

V

CC

R

PROG

R

PROG

T

< 0°C 6 14 mV

J

= 20V

BAT

= 4.93k ● 90 110 mV
= 49.3k ● 713mV

BAT

≤ 20V

VCC Undervoltage Lockout (Switch OFF) Threshold Measured at UV Pin ● 678 V
UV Pin Input Current 0.2V ≤ VUV ≤ 8V ● 0.1 5 µA
UV Output Voltage at UV
UV Output Leakage Current at UV
Reverse Current from Battery (When V

Pin In Undervoltage State, I

OUT

Pin 8V ≤ VUV, V

OUT

Is V

CC

BAT

UVOUT

≤ 20V, VUV ≤ 0.4V 3 15 µA

= 70µA ● 0.1 0.5 V

UVOUT

= 5V ● 0.1 3 µA

Not Connected, VSW Is Floating)

2

LT1511

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, V
RS2 = RS3 = 200Ω (see Block Diagram), V

PARAMETER CONDITIONS MIN TYP MAX UNITS
Overall

Boost Pin Current V

Switch

Switch ON Resistance 8V ≤ V

∆I

/∆ISW During Switch ON V

BOOST

Switch OFF Leakage Current V

Minimum I
Minimum I
Maximum V

Current Sense Amplifier CA1 Inputs (Sense, BAT)

Input Bias Current ● –50 –125 µA
Input Common Mode Low ● –0.25 V
Input Common Mode High ● VCC – 2 V
SPIN Input Current –100 –200 µA

Reference

Reference Voltage (Note 3) R

Reference Voltage All Conditions of V

Oscillator

Switching Frequency 180 200 220 kHz
Switching Frequency All Conditions of V

Maximum Duty Cycle

Current Amplifier CA2

Transconductance VC = 1V, IVC = ±1µA 150 250 550 µmho
Maximum VC for Switch OFF ● 0.6 V
I

Current (Out of Pin) VC ≥ 0.6V 100 µA

VC

for Switch ON ● 2420 µA

PROG

for Switch OFF at V

PROG

for Switch ON ● VCC – 2 V

BAT

≤ 1V ● 1 2.4 mA

PROG

= VCC. No load on any outputs unless otherwise noted.

CLN

CC

V

CC

2V ≤ V
8V ≤ V

V

BOOST
BOOST
SW

20V < VCC ≤ 28V ● 4 200 µA

PROG

VA Supplying I

TA = 25°C9093%

VC < 0.45V 3 mA

The ● denotes specifications which apply over the full operating

= 8V, V

BAT

= 20V, V
= 28V, V

BOOST
BOOST

≤ V

CC

– VSW ≥ 2V ● 0.15 0.25 Ω
= 24V, ISW ≤ 3A 25 35 mA/A

= 0V, VCC ≤ 20V ● 2 100 µA

= 4.93k, Measured at OVP with

= 0V 0.1 10 µA

BOOST

= 0V 0.25 20 µA

BOOST

– VCC < 8V (Switch ON) 6 9 mA
– VCC ≤ 25V (Switch ON) 8 12 mA

, ISW = 3A,

MAX

and Switch OFF 2.453 2.465 2.477 V

PROG

> 0°C ● 2.441 2.489 V

CC,TJ

TJ < 0°C (Note 4) ● 2.43 2.489 V

> 0°C ● 170 200 230 kHz

CC,TJ

TJ < 0°C ● 160 230 kHz

(maximum operating VCC) = 28V,

MAX

● 85 %

3

LT1511

DUTY CYCLE (%)

010305070

I

CC

(mA)

80

1511 • TPC03

20

40

60

8
7
6
5
4
3
2
1
0

125°C

0°C

25°C

V

CC

= 16V

ELECTRICAL CHARACTERISTICS

temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, V

The ● denotes specifications which apply over the full operating

BAT

= 8V, V

(maximum operating VCC) = 28V.

MAX

No load on any outputs unless otherwise noted.

PARAMETER CONDITIONS MIN TYP MAX UNITS
Voltage Amplifier VA

Transconductance (Note 3) Output Current from 50µA to 500µA 0.25 0.6 1.3 mho
Output Source Current V

OVP

= V

+ 10mV, V

REF

PROG

= V

+ 10mV 1.1 mA

REF

OVP Input Bias Current At 0.75mA VA Output Current ±3 ±10 nA

At 0.75mA VA Output Current, TJ > 90°C –15 25 nA

Current Limit Amplifier CL1, 8V ≤ Input Common Mode

Turn-On Threshold 0.75mA Output Current 93 100 107 mV
Transconductance Output Current from 50µA to 500µA 0.5 1 2 mho
CLP Input Current 0.75mA Output Current, VUV ≥ 0.4V 0.3 1 µA
CLN Input Current 0.75mA Output Current VUV ≥ 0.4V 0.8 2 mA

Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.

Note 2: Tested with Test Circuit 1.

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Note 3: Tested with Test Circuit 2.
Note 4: A linear interpolation can be used for reference voltage

specification between 0°C and –40°C.

TYPICAL PERFORMANCE CHARACTERISTICS

Thermally Limited Maximum
Charging Current Efficiency of Figure 1 Circuit

3.0

2.8

2.6

2.4

(θJA=30°C/W)

2.2
T

MAXIMUM CHARGING CURRENT (A)

4

AMAX

T

JMAX

2.0

5

NOTE: FOR 4.2V AND 8.4V BATTERIES MAXIMUM
CHARGING CURRENT IS 3A FOR V

8.4V BATTERY
≥ 11V

V

IN

4.2V BATTERY
≥ 8V

V

IN

=60°C
=125°C

10

15

INPUT VOLTAGE (V)

12.6V BATTERY

16.8V BATTERY

20

25

– V

IN

1511 • TPC01

≥ 3V

BAT

ICC vs Duty Cycle

100

VIN = 16.5

98
96
94
92
90
88

EFFICIENCY (%)

86
84
82

30

80

V

BAT

0.2

0.6 1.4

= 8.4V

INCLUDES LOSS

IN DIODE D3

1.0
I

BAT

CHARGER EFFICIENCY

1.8 3.02.62.2

(A)

1511 • TPC02

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VC (V)

0 0.2 0.6 1.0 1.4 1.8

I

VC

(mA)

–1.20
–1.08
–0.96
–0.84
–0.72
–0.60
–0.48
–0.36
–0.24
–0.12

0

0.12

1.6

1511 • TPC09

0.4

0.8

1.2

2.0

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TYPICAL PERFORMANCE CHARACTERISTICS

Switching Frequency vs
Temperature

210

205

200

195

190

FREQUENCY (kHz)

185

ICC vs V

7.0
MAXIMUM DUTY CYCLE

6.5

6.0

(mA)

CC

I

5.5

5.0

CC

0°C
25°C

125°C

0.003

0.002

0.001

(V)

REF

∆V

–0.001

–0.002

V

Line Regulation

REF

0

LT1511

ALL TEMPERATURES

180

–20

200 40 80 12060

IVA vs ∆V

4

3

(mV)

2

OVP

∆V

1

0

0.20.1 0.3 0.5 0.7 0.9

0

PROG Pin Characteristics

6

(mA)

0

PROG

I

–6

0123

TEMPERATURE (°C)

(Voltage Amplifier)

OVP

0.4
IVA (mA)

V

PROG

100

125°C

25°C

0.6

125°C

(V)

0.8

1511 • TPC04

1511• TPC07

25°C

1511 • TPC10

140

1.0

4.5
0

10 15 20

5

VCC (V)

25 30

1511 • TPC05

–0.003

0

10 15 20

5

VCC (V)

25 30

1511 • TPC06

VC Pin CharacteristicsMaximum Duty Cycle

98
97
96
95
94
93

DUTY CYCLE (%)

92
91
90

0

Switch Current vs Boost Current
vs Boost Voltage

50
45
40
35
30
25
20
15

BOOST CURRENT (mA)

10

5
0

54

0 0.2 0.4 0.6 0.8 1.0 1.2

40 80

20 60 100 140

TEMPERATURE (°C)

VCC = 16V

V

BOOST

SWITCH CURRENT (A)

120

1511 • TPC08

= 38V

28V
18V

1.4 1.8 2.01.6

1511 • TPC11

Reference Voltage
vs Temperature

2.470

2.468

2.466

2.464

2.462

REFERENCE VOLTAGE (V)

2.460

2.458
0

50 75 100

25

TEMPERATURE

125 150

LT1511 • TPC12

5

LT1511

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PIN FUNCTIONS

GND (Pins 1, 4, 5, 7, 16, 23, 24): Ground Pin.
SW (Pin 2): Switch Output. The Schottky catch diode must

be placed with very short lead length in close proximity to
SW pin and GND.

BOOST (Pin 3): This pin is used to bootstrap and drive the
switch power NPN transistor to a low on-voltage for low
power dissipation. In normal operation, V
V

when switch is on. Maximum allowable V

BAT

55V.
UV (Pin 6): Undervoltage Lockout Input. The rising thresh-

old is at 6.7V with a hysteresis of 0.5V. Switching stops in
undervoltage lockout. When the supply (normally the wall
adapter output) to the chip is removed, the UV pin has to
be pulled down to below 0.7V (a 5k resistor from adapter
output to GND is required) otherwise the reverse battery
current drained by the chip will be approximately 200µA
instead of 3µA. Do not leave UV pin floating. If it is
connected to VIN with no resistor divider, the built-in 6.7V
undervoltage lockout will be effective.

OVP (Pin 8): This is the input to the amplifier VA with a
threshold of 2.465V. Typical input current is about 3nA out
of pin. For charging lithium-ion batteries, VA monitors the
battery voltage and reduces charging when battery voltage
reaches the preset value. If it is not used, the OVP pin
should be grounded.

CLP (Pin 9): This is the positive input to the supply current
limit amplifier CL1. The threshold is set at 100mV. When
used to limit supply current, a filter is needed to filter out
the 200kHz switching noise.

CLN (Pin 10): This is the negative input to the amplifier
CL1.

COMP1 (Pin 11): This is the compensation node for the
amplifier CL1. A 200pF capacitor is required from this pin
to GND if input current amplifier CL1 is used. At input
adapter current limit, this node rises to 1V. By forcing
COMP1 low with an external transistor, amplifier CL1 will
be defeated (no adapter current limit). COMP1 can source
200µ A.

BOOST

= VCC +

is

BOOST

SENSE (Pin 12): Current Amplifier CA1 Input. Sensing can
be at either terminal of the battery.

SPIN (Pin 13): This pin is for the internal amplifier CA1
bias. It has to be connected to RS1 as shown in the 3A
Lithium Battery Charger (Figure 1).

BAT (Pin 14): Current Amplifier CA1 Input.
COMP2 (Pin 15): This is also a compensation node for the

amplifier CL1. It gets up to 2.8V at input adapter current
limit and/or at constant-voltage charging.

UV

undervoltage lockout status. It stays low in undervoltage
state. With an external pull-up resistor , it goes high at valid
VCC. Note that the base drive of the open collector NPN
comes from CLN pin. UV
higher than 2V. Pull-up current should be kept under
100µA.

VC (Pin 18): This is the control signal of the inner loop of
the current mode PWM. Switching starts at 0.7V. Higher
VC corresponds to higher charging current in normal
operation. A capacitor of at least 0.33µ F to GND filters out
noise and controls the rate of soft-start. To shut down
switching, pull this pin low. Typical output current is 30µA.

PROG (Pin 19): This pin is for programming the charging
current and for system loop compensation. During normal
operation, V
GND the switching will stop. When a microprocessor
controlled DAC is used to program charging current, it
must be capable of sinking current at a compliance up to

2.465V.

V

good bypass, a low ESR capacitor of 20µF or higher is
required, with the lead length kept to a minimum. V
should be between 8V and 28V and at least 3V higher than
V
when VCC goes below 7V. Note that there is a parasitic
diode inside from SW pin to VCC pin. Do not force V
below SW by more than 0.7V with battery present. All three
VCC pins should be shorted together close to the pins.

(Pin 17): This is an open collector output for

OUT

stays low only when CLN is

OUT

stays close to 2.465V. If it is shorted to

PROG

(Pins 20, 21, 22): This is the supply of the chip. For

CC

. Undervoltage lockout starts and switching stops

BAT

CC

CC

6

BLOCK DIAGRAM

LT1511

W

UV

GND

–

UV

OUT

+

+

6.7V
200kHz

B1

OSCILLATOR

–

+

V

CC

S

R
R
R

V

CC

Q

SW

+

CA1

–

R1
1k

I

PROG

BOOST

SW

SPIN

SENSE

BAT

R

S3

R

S2

I

BAT

R

S1

BAT

+

0.7V

+

V

SW

V

CC

–

–

SHUTDOWN

+

+

1.5V

V

BAT

PWM

C1

+

–

SLOPE COMPENSATION

R2

R3

+

0VP

CLP

CLN

COMP1
COMP2

1511 BD

–

V

C

75k

CA2

+

V

REF

gm = 0.64

+

VA

Ω

V

REF

–

2.465V

100mV

+

CL1

–

C

PROG

PROG

I

PROG

R

PROG

(I

)(RS2)

PROG

I

=

BAT

R

S1

R

2.465V

=

(())

R

(R

PROG
S3

= RS2)

S2

R

S1

7

LT1511

TEST CIRCUITS

0.047µF

1k

+

≈ 0.65V

Test Circuit 1

LT1511

–

V

C

60k

CA2

+

V

REF

300Ω

1µF

1k

PROG

R

PROG

+

CA1

–

SPIN

SENSE

BAT

R

S3

200Ω

R

S2

200Ω

R

S1

10Ω

+

V

BAT

+

LT1006

1511 • TC01

–

20k

Test Circuit 2

LT1511

PROG

I

PROG

10k

0.47µF

R

PROG

+

2.465V

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OPERATION

The LT1511 is a current mode PWM step-down (buck)
switcher. The battery DC charging current is programmed
by a resistor R
pin (see Block Diagram). Amplifier CA1 converts the
charging current through RS1 to a much lower current
I

fed into the PROG pin. Amplifier CA2 compares the

PROG

output of CA1 with the programmed current and drives the
PWM loop to force them to be equal. High DC accuracy is
achieved with averaging capacitor C
has both AC and DC components. I
and generates a ramp signal that is fed to the PWM control
comparator C1 through buffer B1 and level shift resistors

(or a DAC output current) at the PROG

PROG

. Note that I

PROG

goes through R1

PROG

PROG

+

VA

OVP

–

V

REF

10k

–

LT1013

+

1511 • TC02

R2 and R3, forming the current mode inner loop. The
Boost pin drives the switch NPN QSW into saturation and
reduces power loss. For batteries like lithium-ion that
require both constant-current and constant-voltage charg­ing, the 0.5%, 2.465V reference and the amplifier VA
reduce the charging current when battery voltage reaches
the preset level. For NiMH and NiCd, VA can be used for
overvoltage protection. When input voltage is not present,
the charger goes into low current (3µA typically) sleep
mode as input drops down to 0.7V below battery voltage.
To shut down the charger, simply pull the VC pin low with
a transistor.

8

LT1511

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APPLICATIONS INFORMATION

Input and Output Capacitors

In the 3A Lithium Battery Charger (Figure 1), the input
capacitor (CIN) is assumed to absorb all input switching
ripple current in the converter, so it must have adequate
ripple current rating. Worst-case RMS ripple current will
be equal to one half of output charging current. Actual
capacitance value is not critical. Solid tantalum capacitors
such as the AVX TPS and Sprague 593D series have high
ripple current rating in a relatively small surface mount
package, but

tors are used for input bypass

can be created when the adapter is hot-plugged to the
charger and solid tantalum capacitors have a known
failure mechanism when subjected to very high turn-on
surge currents. Highest possible voltage rating on the
capacitor will minimize problems. Consult with the manu­facturer before use. Alternatives include new high capacity
ceramic (5µF to 20µF) from Tokin or United Chemi-Con/
Marcon, et al., and the old standby, aluminum electrolytic,
which will require more microfarads to achieve adequate
ripple rating. Sanyo OS-CON can also be used.

The output capacitor (C
output switching current ripple. The general formula for
capacitor current is:

I

RMS

For example, VCC = 16V, V
and f = 200kHz, I

EMI considerations usually make it desirable to minimize
ripple current in the battery leads, and beads or inductors
may be added to increase battery impedance at the 200kHz
switching frequency. Switching ripple current splits be­tween the battery and the output capacitor depending on
the ESR of the output capacitor and the battery imped­ance. If the ESR of C
is rased to 4Ω with a bead or inductor, only 5% of the
current ripple will flow in the battery.

caution must be used when tantalum capaci-

. High input surge currents

) is also assumed to absorb

OUT

V

BAT

()

V

CC

= 8.4V, L1 = 20µH,

BAT

=

0.29 (V
(L1)(f)

RMS

) 1 –

BAT

= 0.3A.

is 0.2Ω and the battery impedance

OUT

Soft-Start

The LT1511 is soft started by the 0.33µ F capacitor on the
VC pin. On start-up, VC pin voltage will rise quickly to 0.5V,
then ramp at a rate set by the internal 45µ A pull-up current
and the external capacitor. Battery charging current starts
ramping up when VC voltage reaches 0.7V and full current
is achieved with VC at 1.1V. With a 0.33µ F capacitor, time
to reach full charge current is about 10ms and it is
assumed that input voltage to the charger will reach full
value in less than 10ms. The capacitor can be increased up
to 1µF if longer input start-up times are needed.

In any switching regulator, conventional timer-based soft
starting can be defeated if the input voltage rises much
slower than the time out period. This happens because the
switching regulators in the battery charger and the com­puter power supply are typically supplying a fixed amount
of power to the load. If input voltage comes up slowly
compared to the soft start time, the regulators will try to
deliver full power to the load when the input voltage is still
well below its final value. If the adapter is current limited,
it cannot deliver full power at reduced output voltages and
the possibility exists for a quasi “latch” state where the
adapter output stays in a current limited state at reduced
output voltage. For instance, if maximum charger plus
computer load power is 30W, a 15V adapter might be
current limited at 2.5A. If adapter voltage is less than
(30W/2.5A = 12V) when full power is drawn, the adapter
voltage will be sucked down by the constant 30W load until
it reaches a lower stable state where the switching regu­lators can no longer supply full load. This situation can be
prevented by utilizing
than the minimum adapter voltage where full power can be
achieved.

A fixed undervoltage lockout of 7V is built into the VCC pin,
but an additional adjustable lockout is also available on the
UV pin. Internal lockout is performed by clamping the V
pin low. The VC pin is released from its clamped state when
the UV pin rises above 6.7V and is pulled low when the UV
pin drops below 6.2V (0.5V hysteresis). At the same time
UV
signal can be used to alert the system that charging is
about to start. The charger will start delivering current
about 4ms after VC is released, as set by the 0.33µF

goes high with an external pull-up resistor. This

OUT

undervoltage lockout

, set higher

C

9

LT1511

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APPLICATIONS INFORMATION

capacitor. A resistor divider is used to set the desired V
lockout voltage as shown in Figure 2. A typical value for R6
is 5k and R5 is found from:

R6(V –V )

R5=

UV

IN

V

UV

VUV = Rising lockout threshold on the UV pin
VIN = Charger input voltage that will sustain full load power
Example: With R6 = 5k, VUV = 6.7V and setting VIN at 12V;
R5 = 5k (12V – 6.7V)/6.7V = 4k
The resistor divider should be connected directly to the

adapter output as shown, not to the VCC pin to prevent
battery drain with no adapter voltage. If the UV pin is not
used, connect it to the adapter output (not VCC) and
connect a resistor no greater than 5k to ground. Floating
the pin will cause reverse battery current to increase from
3µ A to 200µA.

If connecting the unused UV pin to the adapter output is
not possible for some reason, it can be grounded. Al­though it would seem that grounding the pin creates a
permanent lockout state, the UV circuitry is arranged for
phase reversal with low voltages on the UV pin to allow the
grounding technique to work.

100mV

+

CL1

–

LT1511

Figure 2. Adapter Current Limiting

CLP

+

1µF

CLN

V

CC

+

UV

*R

=

S4

ADAPTER CURRENT LIMIT

500Ω

RS4*

100mV

1511 • F02

CC

AC ADAPTER

OUTPUT

V

R5

R6

IN

being charged without complex load management algo­rithms. Additionally, batteries will automatically be charged at
the maximum possible rate of which the adapter is capable.

This feature is created by sensing total adapter output
current and adjusting charging current downward if a
preset adapter current limit is exceeded. True analog
control is used, with closed loop feedback ensuring that
adapter load current remains within limits. Amplifier CL1
in Figure 2 senses the voltage across RS4, connected
between the CLP and CLN pins. When this voltage exceeds
100mV, the amplifier will override programmed charging
current to limit adapter current to 100mV/RS4. A lowpass
filter formed by 500Ω and 1µF is required to eliminate
switching noise. If the current limit is not used, both CLP
and CLN pins should be connected to VCC.

Charging Current Programming

The basic formula for charging current is (see Block
Diagram):

I

BAT

where R

R

= I

PROG

is the total resistance from PROG pin to ground.

PROG

S2

()

R

S1

2.465V

=

()()

R

PROG

R

S2

R

S1

For the sense amplifier CA1 biasing purpose, RS3 should
have the same value as RS2 and SPIN should be connected
directly to the sense resistor (RS1) as shown in the Block
Diagram.

For example, 3A charging current is needed. To have low
power dissipation on R

and enough signal to drive the

S1

amplifier CA1, let RS1 = 100mV/3A = 0.033Ω. This limits
RS1 power to 0.3W. Let R

)(R

(I

RS2 = RS3 =

BAT

PROG

PROG

= 5k, then:

)(RS1)

2.465V

(3A)(5k)(0.033)

= = 200Ω

2.465V

Adapter Limiting

An important feature of the LT1511 is the ability to
automatically adjust charging current to a level which
avoids overloading the wall adapter. This allows the
product to operate at the same time that batteries are

10

Charging current can also be programmed by pulse width
modulating I

with a switch Q1 to R

PROG

at a frequency

PROG

higher than a few kHz (Figure 3). Charging current will be
proportional to the duty cycle of the switch with full current
at 100% duty cycle.

LT1511

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APPLICATION S INFORMATION

LT1511

PROG

300Ω

R

PROG

4.7k

5V
0V

PWM

I

= (DC)(3A)

BAT

Figure 3. PWM Current Programming

Q1
VN2222

Lithium-Ion Charging

The 3A Lithium Battery Charger (Figure 1) charges lithium­ion batteries at a constant 3A until battery voltage reaches
a limit set by R3 and R4. The charger will then automati­cally go into a constant-voltage mode with current de­creasing to zero over time as the battery reaches full
charge. This is the normal regimen for lithium-ion charg­ing, with the charger holding the battery at “float” voltage
indefinitely. In this case no external sensing of full charge
is needed.

Battery Voltage Sense Resistors Selection

C

PROG

1µF

1511 • F03

When power is on, there is about 200µ A of current flowing
out of the BAT and Sense pins. If the battery is removed
during charging, and total load including R3 and R4 is less
than the 200µ A, V
loop has turned switching off. To keep V

could float up to VCC even though the

BAT

regulated to

BAT

the battery voltage in this condition, R3 and R4 can be
chosen to draw 0.5mA and Q3 can be added to disconnect
them when power is off (Figure 4). R5 isolates the OVP pin
from any high frequency noise on VIN. An alternative way is
to use a Zener diode with a breakdown voltage two or three
volts higher than battery voltage to clamp the V

R3
12k

0.25%

LT1511

OVP

Figure 4. Disconnecting Voltage Divider

VN2222

Q3

R4

4.99k

0.25%

R5

220k

V

IN

+

4.2V

–
+

4.2V

–

LT1511 • F04

voltage.

BAT

V

BAT

To minimize battery drain when the charger is off, current
through the R3/R4 divider is set at 15µ A. The input current
to the OVP pin is 3nA and the error can be neglected.

With divider current set at 15µ A, R4 = 2.465/15µ A = 162k
and,

R4 V 2.465

()

R3

=

390k

=

−

()

BAT

2.465

162 k 8.4 2.465

=

−

()

2.465

Li-Ion batteries typically require float voltage accuracy of
1% to 2%. Accuracy of the LT1511 OVP voltage is ±0.5%
at 25°C and ±1% over full temperature. This leads to the
possibility that very accurate (0.1%) resistors might be
needed for R3 and R4. Actually, the temperature of the
LT1511 will rarely exceed 50°C in float mode because
charging currents have tapered off to a low level, so 0.25%
resistors will normally provide the required level of overall
accuracy.

Some battery manufacturers recommend termination of
constant-voltage float mode after charging current has
dropped below a specified level (typically around 10% of
the full current)

and

a further time out period of 30 minutes
to 90 minutes has elapsed. This may extend the life of the
battery, so check with manufacturers for details. The
circuit in Figure 5 will detect when charging current has
dropped below 400mA. This logic signal is used to initiate
a timeout period, after which the LT1511 can be shut down
by pulling the VC pin low with an open collector or drain.
Some external means must be used to detect the need for
additional charging or the charger may be turned on
periodically to complete a short float-voltage cycle.

Current trip level is determined by the battery voltage, R1
through R3 and the sense resistor (RS1). D2 generates
hysteresis in the trip level to avoid multiple comparator
transitions.

11

LT1511

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APPLICATIONS INFORMATION

I

BAT

R

S3

200Ω

R

S1

0.033Ω

V

BAT

C1

R1*

BAT

1.6k

R2
560k

R3
430k

0.1µF

D2

1N4148

200Ω

3

2

R

S2

ADAPTER

–

LT1011

+

OUTPUT

Figure 5. Current Comparator for Initiating Float Time Out

Nickel-Cadmium and Nickel-Metal-Hydride Charging

The circuit in the 3A Lithium Battery Charger (Figure 1) can
be modified to charge NiCd or NiMH batteries. For ex­ample, 2-level charging is needed; 2A when Q1 is on and
200mA when Q1 is off.

LT1511

1k

0.33µF

SENSE

LT1511

BAT

3.3V OR 5V

D1

1

1N4148

8

PROG

R1

49.3k

4

R4
470k

7

* TRIP CURRENT =

(1.6k)(8.4V)

= ≈ 400mA

(560k + 430k)(0.033Ω)

R2

5.49k

Q1

NEGATIVE EDGE
TO TIMER

R1(V

BAT

(R2 + R3)(R

1511 • F04

)

)

S1

2.465 4000

()()

R1

=

I

LOW HI LOW

R2

2.465 4000

()()

=

−

II

All battery chargers with fast charge rates require some
means to detect full charge state in the battery to terminate
the high charging current. NiCd batteries are typically
charged at high current until temperature rise or battery
voltage decrease is detected as an indication of near full
charge. The charging current is then reduced to a much
lower value and maintained as a constant trickle charge.
An intermediate “top off” current may be used for a fixed
time period to reduce 100% charge time.

NiMH batteries are similar in chemistry to NiCd but have
two differences related to charging. First, the inflection
characteristic in battery voltage as full charge is ap­proached is not nearly as pronounced. This makes it more
difficult to use dV/dt as an indicator of full charge, and
change of temperature is more often used with a tempera­ture sensor in the battery pack. Secondly, constant trickle
charge may not be recommended. Instead, a moderate
level of current is used on a pulse basis (≈ 1% to 5% duty
cycle) with the time-averaged value substituting for a
constant low trickle. Please contact the Linear Technology
Applications Department about charge termination cir­cuits.

If overvoltage protection is needed, R3 and R4 should be
calculated according to the procedure described in Lithium­Ion Charging section. The OVP pin should be grounded if
not used.

When a microprocessor DAC output is used to control
charging current, it must be capable of sinking current at a
compliance up to 2.5V if connected directly to the PROG pin.

1511 • F05

Figure 6. 2-Level Charging

For 2A full current, the current sense resistor (RS1) should
be increased to 0.05Ω so that enough signal (10mV) will
be across RS1 at 0.2A trickle charge to keep charging
current accurate.

For a 2-level charger, R1 and R2 are found from;

12

Thermal Calculations

If the LT1511 is used for charging currents above 1.5A, a
thermal calculation should be done to ensure that junction
temperature will not exceed 125°C. Power dissipation in
the IC is caused by bias and driver current, switch resis­tance and switch transition losses. The SO wide package,
with a thermal resistance of 30°C/W, can provide a full 3A
charging current in many situations. A graph is shown in
the Typical Performance Characteristics section.

LT1511

U

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APPLICATIONS INFORMATION

P 3.5mA V 1.5mA V

=

BIAS IN BAT

P

DRIVER

P

SW

RSW = Switch ON resistance ≈ 0.16Ω
tOL = Effective switch overlap time ≈ 10ns
f = 200kHz

Example: VIN = 15V, V

P 3.5mA 15 1.5mA 8.4

BIAS

+

P

DRIVER

P

SW

0.81 0.09 0.9W

Total Power in the IC is: 0.27 + 0.33 + 0.9 = 1.5W

()()

V

()

BAT

+

V

IN

IV

()( )

BAT BAT

=

2

IRV

()()( )

BAT

=

=

()()

2

8.4

()

7.5mA 0.012 3 0.27W

[]

15

3 8.4

()( )

=

2

3 0.16 8.4

()( )( )

=

=+=

15

+

2

7.5mA 0.012 I

[]

2

55 V

()

SW BAT

V

IN

= 8.4V, I

BAT

+

+

()()



2

+

1





5515

()

+

()

+

()()

V



BAT

+

1





30

IN

tVI f

+

()()( )()

BAT

BAT






OL IN BAT

= 3A;

()

=

8430.






0.33W

=

−

9

10 15 3 200kHz

()()( )

SW

LT1511

C2

L1

V

X

AVV

384331

()()()

P

DRIVER

The average IVX required is:

P

DRIVER

Fused-lead packages conduct most of their heat out the
leads. This makes it very important to provide as much PC
board copper around the leads as is practical. Total
thermal resistance of the package-board combination is
dominated by the characteristics of the board in the
immediate area of the package. This means both lateral
thermal resistance across the board and vertical thermal
resistance through the board to other copper layers. Each
layer acts as a thermal heat spreader that increases the
heat sinking effectiveness of extended areas of the board.

=

011

.

==

V

X

33

+

I

VX

Figure 7. Lower V

..

55 15

W

.

34

V

BOOST

D2

SPIN

1511 • F07

10µF

BOOST

33

.



+





30

V

()

mA

V






=

011

.

W

Temperature rise will be (1.5W)(30°C/W) = 45°C. This
assumes that the LT1511 is properly heat sunk by con­necting the seven fused ground pins to expanded traces
and that the PC board has a backside or internal plane for
heat spreading.

The P
diode D2 (see Figure 1) to a lower system voltage (lower
than V

Then P

For example, VX = 3.3V then:

term can be reduced by connecting the boost

DRIVER

) instead of V

BAT

DRIVER

=

.

BAT

V



IN

+

1





30

IVV

()( )()

BAT BAT X

V

55

()

X






Total board area becomes an important factor when the
area of the board drops below about 20 square inches. The
graph in Figure 8 shows thermal resistance vs board area
for 2-layer and 4-layer boards with continuous copper
planes. Note that 4-layer boards have significantly lower
thermal resistance, but both types show a rapid increase
for reduced board areas. Figure 9 shows actual measured
lead temperatures for chargers operating at full current.
Battery voltage and input voltage will affect device power
dissipation, so the data sheet power calculations must be
used to extrapolate these readings to other situations.

Vias should be used to connect board layers together.
Planes under the charger area can be cut away from the
rest of the board and connected with vias to form both a

13

LT1511

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APPLICATIONS INFORMATION

low thermal resistance system and to act as a ground
plane for reduced EMI.

Glue-on, chip-mounted heat sinks are effective only in
moderate power applications where the PC board copper
cannot be used, or where the board size is small. They
offer very little improvement in a properly laid out multi­layer board of reasonable size.

Higher Duty Cycle for the LT1511 Battery Charger

Maximum duty cycle for the LT1511 is typically 90%, but
this may be too low for some applications. For example, if
an 18V ±3% adapter is used to charge ten NiMH cells, the
charger must put out 15V maximum. A total of 1.6V is lost
in the input diode, switch resistance, inductor resistance
and parasitics, so the required duty cycle is 15/16.4 =

91.4%. As it turn out, duty cycle can be extended to 93%

45

STANDARD CONNECTION HIGH DUTY CYCLE CONNECTION

by restricting boost voltage to 5V instead of using V

BAT

as
is normally done. This lower boost voltage also reduces
power dissipation in the LT1511, so it is a win-win deci­sion. Connect an external source of 3V to 6V at VX node in
Figure 10 with a 10µF CX bypass capacitor.

Even Lower Dropout

For even lower dropout and/or reducing heat on the board,
the input diode D3 should be replaced with a FET (see
Figure 11). It is pretty straightforward to connect a
P-channel FET across the input diode and connect its gate
to the battery so that the FET commutates off when the
input goes low. The problem is that the gate must be
pumped low so that the FET is fully turned on even when
the input is only a volt or two above the battery voltage.
Also there is a turn-off speed issue. The FET should turn

40

35

30

25

20

MEASURED FROM AIR AMBIENT

THERMAL RESISTANCE (°C/W)

15

TO DIE USING COPPER LANDS
AS SHOWN ON DATA SHEET

10

0

510

2-LAYER BOARD

4-LAYER BOARD

20 30 35

15 25

BOARD AREA (IN2)

LT1511 • F08

Figure 8. LT1511 Thermal Resistance

110

NOTE: PEAK DIE TEMPERATURE WILL BE
ABOUT 10°C HIGHER THAN LEAD TEMPER-

100

ATURE AT 3A CHARGING CURRENT

90

80

70

60

VIN = 16V

LEAD TEMPERATURE (°C)

50

40

V

BAT

I

CHRG

= 25°C

T

A

0

= 8.4V

= 3A

510

2-LAYER BOARD

4-LAYER BOARD

4-LAYER BOARD

WITH V

BOOST

20 30 35

15 25

BOARD AREA (IN2)

= 3.3V

LT1511 • F09

Figure 9. LT1511 Lead Temperature

0.47µF

C3

D2

SW

BOOST

SPIN
SENSE BAT

V

IN

Q1 = Si4435DY
Q2 = TP0610L

C3

0.47µF

LT1511

3V TO 6V

V

BAT

D2

V

X

C

X

10µF

+ +

Figure 10. High Duty Cycle

HIGH DUTY CYCLE CONNECTION

R
50k

Q1

Q2

X

D1

3V TO 6V

0.47µF

V

X

10µF

+

C2

D2

C

X

Figure 11. Replacing the Input Diode

SW

BOOST

LT1511

SPIN
SENSE BAT

V

CC

SW

BOOST

LT1511

SPIN
SENSE BAT

+

V

1511 F10

V

BAT

1511 F11

BAT

14

LT1511

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APPLICATIONS INFORMATION

off instantly when the input is dead shorted to avoid large
current surges from the battery back through the charger
into the FET. Gate capacitance slows turn-off, so a small
P-channel (Q2) is to discharge the gate capacitance quickly
in the event of an input short. The body diode of Q2 creates
the necessary pumping action to keep the gate of Q1 low
during normal operation. Note that Q1 and Q2 have a V
spec limit of 20V. This restricts VIN to a maximum of 20V.
For low dropout operation with VIN > 20V consult factory.

Optional Connection of Input Diode and
Current Sense Resistor

The typical application shown in Figure 1 on the first page
of this data sheet shows a single diode to isolate the V
pin from the adapter input. This simple connection may be
unacceptable in situations where the main system power
must be disconnected from both the battery
adapter under some conditons. In particular, if the adapter
is disconnected or turned off and it is desired to also

L1

LT1511

SW

PARASITIC

INTERNAL

DIODE

R

S1

CLP

CLN
V

CC

R7

500Ω

+

C1
1µF

+

C

IN

D3

ADAPTER
IN

R

S4

TO
SYSTEM
POWER

1511 F12a

and

GS

CC

the

disconnect the system load from the battery, the system
will remain powered through the parasitic diode from the
SW pin to the VCC pin.

The circuit in Figure 12b allows system power to go to 0V
without drawing battery current by adding an additional
diode, D4. To ensure proper operation, the LT1511 current
sense amplifier inputs (CLP and CLN) were designed to
work above VCC and not to draw current from VCC when the
inputs are pulled to ground by a powered-down adapter.

Layout Considerations

Switch rise and fall times are under 10ns for maximum
efficiency. To prevent radiation, the catch diode, SW pin
and input bypass capacitor leads should be kept as short
as possible. A ground plane should be used under the
switching circuitry to prevent interplane coupling and to
act as a thermal spreading path. All ground pins should be
connected to expanded traces for low thermal resistance.
The fast-switching high current ground path, including the
switch, catch diode and input capacitor, should be kept
very short. Catch diode and input capacitor should be
close to the chip and terminated to the same point. This
path contains nanosecond rise and fall times with several
amps of current. The other paths contain only DC and/or
200kHz tri-wave and are less critical. Figure 13 indicates
the high speed, high current switching path. Figure 14
shows critical path layout. Contact Linear Technology for
an actual LT1511 circuit PCB layout or Gerber file.

Figure 12a. Standard Connection

R7

L1

LT1511

SW

PARASITIC

INTERNAL

DIODE

R

S1

CLP

CLN
V

500Ω

+

C1
1µF

CC

+

D3

C

IN

ADAPTER
IN

R

S4

TO
SYSTEM
POWER

D4

1511 F12b

Figure 12b. Modified Input Diode Connection

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

SWITCH NODE

HIGH

FREQUENCY

V

C

IN

IN

CIRCULATING

PATH

L1

D1

C

OUT

BAT

LT1511 • F13

V

BAT

Figure 13. High Speed Switching Path

15

LT1511

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APPLICATIONS INFORMATION

L1

NOTE: CONNECT ALL GND PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING

Figure 14. Critical Electrical and Thermal Path Layout

U

PACKAGE DESCRIPTION

Dimensions in inches (millimeters) unless otherwise noted.

D1

TO

GND

GND
SW

GND
GND

GND

GND

GND

GND
GND

V

GND

C

CC1

TO

GND

R

S1

C

IN

IN

C

OUT

LT1511 • F14

SW Package

24-Lead Plastic Small Outline (Wide 0.300)

0.291 – 0.299**

(LTC DWG # 05-08-1620)

(7.391 – 7.595)

0.010 – 0.029

(0.254 – 0.737)

0.009 – 0.013

(0.229 – 0.330)

NOTE:

1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS.
THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

*

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

**

NOTE 1

× 45°

0.016 – 0.050

(0.406 – 1.270)

(2.362 – 2.642)

° – 8° TYP

0

0.093 – 0.104

0.050

(1.270)

TYP

0.014 – 0.019

(0.356 – 0.482)

TYP

0.037 – 0.045

(0.940 – 1.143)

NOTE 1

0.004 – 0.012

(0.102 – 0.305)

2324

2345678

1

0.598 – 0.614*

(15.190 – 15.600)

22 21 20 19 181716 15

910

11 12

1314

0.394 – 0.419

(10.007 – 10.643)

S24 (WIDE) 0996

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16

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

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www.linear-tech.com

1511fb LT/TP 0399 REV B 2K • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1995