Datasheet LT1373 Datasheet (Linear Technology)

FEATURES

LT1373

250kHz Low Supply Current

High Efficiency

1.5A Switching Regulator

U

DESCRIPTIO

■

1mA IQ at 250kHz

■

Uses Small Inductors: 15µH

■

All Surface Mount Components

■

Only 0.6 Square Inch of Board Space

■

Low Minimum Supply Voltage: 2.7V

■

Constant Frequency Current Mode

■

Current Limited Power Switch: 1.5A

■

Regulates Positive or Negative Outputs

■

Shutdown Supply Current: 12µA Typ

■

Easy External Synchronization

■

8-Pin SO or PDIP Packages

U

APPLICATIO S

■

Boost Regulators

■

CCFL Backlight Driver

■

Laptop Computer Supplies

■

Multiple Output Flyback Supplies

■

Inverting Supplies

The LT®1373 is a low supply current high frequency
current mode switching regulator. It can be operated in all
standard switching configurations including boost, buck,
flyback, forward, inverting and “Cuk.” A 1.5A high effi­ciency switch is included on the die, along with all oscilla­tor, control and protection circuitry. All functions of the
LT1373 are integrated into 8-pin SO/PDIP packages.

Compared to the 500kHz LT1372, which draws 4mA of
quiescent current, the LT1373 switches at 250kHz, typi­cally consumes only 1mA and has higher efficiency. High
frequency switching allows for small inductors to be used.
All surface mount components consume less than 0.6
square inch of board space.

New design techniques increase flexibility and maintain
ease of use. Switching is easily synchronized to an exter­nal logic level source. A logic low on the shutdown pin
reduces supply current to 12µA. Unique error amplifier
circuitry can regulate positive or negative output voltage
while maintaining simple frequency compensation tech­niques. Nonlinear error amplifier transconductance re­duces output overshoot on start-up or overload recovery.
Oscillator frequency shifting protects external compo­nents during overload conditions.

, LTC and LT are registered trademarks of Linear Technology Corporation.

TYPICAL APPLICATIO

5V-to-12V Boost Converter

5V

L1*

MBRS120T3

22µH

5

V

4

ON

OFF

+

C1**
22µF

IN

S/S

LT1373

GND

6, 7

8

V

SW

2

FB

V

C

1

C2

0.01µF

R3
5k

U

12V Output Efficiency

100

D1

†

V

OUT

R1
215k
1%

R2

24.9k
1%

LT1373 • TA01

12V

+

C4**
22µF

†

MAX I

OUT

L1

I

OUT

0.3A

15µH

0.35A

22µH

SUMIDA CD75-220KC (22µH) OR

*

COILCRAFT D03316-153 (15µH)

**

AVX TPSD226M025R0200

VIN = 5V
f = 250kHz

90

80

70

EFFICIENCY (%)

60

50

1

10 100 1000

OUTPUT CURRENT (mA)

LT1373 • TA02

1

LT1373

PACKAGE/ORDER I FOR ATIO

UU

W

WWWU

ABSOLUTE AXI U RATI GS

(Note 1)

Supply Voltage ....................................................... 30V

Switch Voltage

LT1373 ............................................................... 35V

LT1373HV .......................................................... 42V

S/S Pin Voltage....................................................... 30V

Feedback Pin Voltage (Transient, 10ms) .............. ±10V

Feedback Pin Current........................................... 10mA

Negative Feedback Pin Voltage

(Transient, 10ms)............................................. ±10V

Operating Junction Temperature Range

Commercial ........................................ 0°C to 125°C*

Industrial ......................................... – 40°C to 125°C

Short Circuit ......................................... 0°C to 150°C

Storage Temperature Range ................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec)................. 300°C

ORDER PART

TOP VIEW

V

1

C

FB

2

NFB

3

S/S

4

N8 PACKAGE
8-LEAD PDIP

T

JMAX

T

JMAX

Consult factory for Military grade parts.
*Units shipped prior to Date Code 9552 are rated at 100°C maximum

operating temperature.

= 125°C, θ
= 125°C, θ

8-LEAD PLASTIC SO

JA
JA

V

8

SW

GND

7

GND S

6

V

5

IN

S8 PACKAGE

= 100°C/ W (N8)
= 120°C/ W (S8)

LT1373CN8
LT1373HVCN8
LT1373CS8
LT1373HVCS8

NUMBER

LT1373IN8
LT1373HVIN8
LT1373IS8
LT1373HVIS8

S8 PART MARKING

1373
1373I

1373H
1373HI

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

V

REF

I

FB

V

NFB

I

NFB

g

m

A

V

f Switching Frequency 2.7V ≤ VIN ≤ 25V 225 250 275 kHz

Reference Voltage Measured at Feedback Pin 1.230 1.245 1.260 V

Feedback Input Current VFB = V

Reference Voltage Line Regulation 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● 0.01 0.03 %/V
Negative Feedback Reference Voltage Measured at Negative Feedback Pin –2.51 –2.45 –2.39 V

Negative Feedback Input Current V
Negative Feedback Reference Voltage 2.7V ≤ VIN ≤ 25V, VC = 0.8V ● 0.01 0.05 %/V

Line Regulation
Error Amplifier Transconductance ∆IC = ±5µA 250 375 500 µmho

Error Amplifier Source Current VFB = V
Error Amplifier Sink Current VFB = V
Error Amplifier Clamp Voltage High Clamp, VFB = 1V 1.70 1.95 2.30 V

Error Amplifier Voltage Gain 250 V/V
VC Pin Threshold Duty Cycle = 0% 0.8 1 1.25 V

Maximum Switch Duty Cycle ● 90 95 %
Switch Current Limit Blanking Time 340 500 ns

, VSW, S/S and NFB pins open, unless otherwise noted.

REF

V

= 0.8V ● 1.225 1.245 1.265 V

C

REF

Feedback Pin Open, V

= V

NFB

NFR

– 150mV, VC = 1.5V ● 25 50 90 µA

REF

+ 150mV, VC = 1.5V ● 850 1500 µA

REF

Low Clamp, V

0°C ≤ T
–40°C ≤ T

FB

≤ 125°C ● 210 250 290 kHz

J

≤ 0°C (I Grade) 200 290 kHz

J

= 0.8V ● –2.55 –2.45 –2.35 V

C

= 1.5V 0.25 0.40 0.52 V

50 150 nA

● 275 nA

● –12 –7 –2 µA

● 150 600 µmho

2

LT1373

TEMPERATURE (°C)

–50

1.8

INPUT VOLTAGE (V)

2.0

2.2

2.4

2.6

050

100

150

LT1373 • G03

2.8

3.0

–25 25

75

125

ELECTRICAL CHARACTERISTICS

The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C.
VIN = 5V, VC = 0.6V, VFB = V

SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS

BV Output Switch Breakdown Voltage LT1373 ● 35 47 V

V

SAT

Output Switch “On” Resistance ISW = 1A ● 0.5 0.85 Ω

, VSW, S/S and NFB pins open, unless otherwise noted.

REF

LT1373HV
0°C ≤ T
–40°C ≤ T

≤ 125°C ● 42 47 V

J

≤ 0°C (I Grade) 40 V

J

I

LIM

Switch Current Limit Duty Cycle = 50% ● 1.5 1.9 2.7 A

Duty Cycle = 80% (Note 2)

∆I
∆I

IN

SW

Supply Current Increase During Switch On-Time 10 20 mA/A

Control Voltage to Switch Current 2A/V
Transconductance

Minimum Input Voltage ● 2.4 2.7 V

I

Q

Supply Current 2.7V ≤ VIN ≤ 25V ● 1 1.5 mA
Shutdown Supply Current 2.7V ≤ VIN ≤ 25V, V

0°C ≤ T
–40°C ≤ T

≤ 125°C ● 12 30 µA

J

≤ 0°C (I Grade) 50 µA

J

S/S

≤ 0.6V

Shutdown Threshold 2.7V ≤ VIN ≤ 25V ● 0.6 1.3 2 V
Shutdown Delay ● 5 12 100 µs
S/S Pin Input Current 0V ≤ V

≤ 5V ● –10 15 µA

S/S

Synchronization Frequency Range ● 300 340 kHz

Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.

Note 2: For duty cycles (DC) between 50% and 90%, minimum
guaranteed switch current is given by I

UW

TYPICAL PERFOR A CE CHARACTERISTICS

● 1.3 1.7 2.5 A

= 0.667 (2.75 – DC).

LIM

SWITCH SATURATION VOLTAGE (V)

Switch Saturation Voltage
vs Switch Current

1.0

0.9

0.8

0.7

0.6

0.5

0.4

0.3

0.2

0.1
0

0.2

0.4

0

0.6

SWITCH CURRENT (A)

0.8

100°C

1.0

1.2

150°C

–55°C

1.4

1.6

25°C

1.8

LT1373 • G01

2.0

Switch Current Limit
vs Duty Cycle

3.0

2.5

2.0

1.5

1.0

SWITCH CURRENT LIMIT (A)

0.5

0

20 40 60 80

DUTY CYCLE (%)

–55°C

25°C AND
125°C

LT1373 • G02

Minimum Input Voltage
vs Temperature

10010030 50 70 90

3

LT1373

TEMPERATURE (°C)

–50

0

TRANSCONDUCTANCE (µmho)

200

500

0

50

75

LT1373 • G09

100

400

300

–25 25

100

125

150

gm =

∆I (V

C

)

∆V (FB)

TEMPERATURE (°C)

–50

–20

NEGATIVE FEEDBACK INPUT CURRENT (µA)

–12
–14

0

0

50

75

LT1373 • G12

–16
–18

–4
–6

–2

–8

–10

–25 25

100

125

150

V

NFB

= V

NFR

UW

TYPICAL PERFOR A CE CHARACTERISTICS

Shutdown Delay and Threshold
vs Temperature

20
18

SHUTDOWN DELAY (µs)

16
14
12
10

8
6
4
2
0

–50

SHUTDOWN

–25 25

SHUTDOWN
THRESHOLD

DELAY

0

50

TEMPERATURE (°C)

75

S/S Pin Input Current
vs Voltage

5

VIN = 5V

4
3
2
1

0
–1
–2
–3

S/S PIN INPUT CURRENT (µA)

–4
–5

1

2

08

–1

S/S PIN VOLTAGE (V)

5

3

4

Minimum Synchronization
Voltage vs Temperature

)

3.0
f

0

SYNC

–50

= 330kHz

050

–25 25

TEMPERATURE (°C)

P-P

SHUTDOWN THRESHOLD (V)

2.5

2.0

1.5

1.0

0.5

MINIMUM SYNCHRONIZATION VOLTAGE (V

100

75

125

LT1373 • G05

150

100

125

LT1373 • G04

150

2.0

1.8

1.6

1.4

1.2

1.0

0.8

0.6

0.4

0.2
0

Switching Frequency
vs Feedback Pin Voltage

110
100

90
80
70
60
50
40
30
20

SWITCHING FREQUENCY (% OF TYPICAL)

6

7

9

LT1373 • G07

10

0.2

0.3

0.1 0.9

0

FEEDBACK PIN VOLTAGE (V)

0.4

0.5

0.6

0.7

0.8

1.0

LT1373 • G08

Error Amplifier Output Current
vs Feedback Pin Voltage

100

–25

–50

ERROR AMPLIFIER OUTPUT CURRENT (µA)

–75

75

50

25

0

125°C

–0.3 –0.2

FEEDBACK PIN VOLTAGE (V)

25°C
–55°C

V

–0.1 0.1

REF

Error Amplifier Transconductance
vs Temperature

LT1373 • G06

2.4

2.2

2.0

1.8

1.6

1.4

1.2

PIN VOLTAGE (V)

C

1.0

V

0.8

0.6

0.4

4

VC Pin Threshold and High
Clamp Voltage vs Temperature

VC HIGH CLAMP

VC THRESHOLD

–50

–25 25

0

TEMPERATURE (°C)

50

75

100

125

LT1373 • G10

150

Feedback Input Current
vs Temperature

400

VFB = V

–25

REF

0

50

25

TEMPERATURE (°C)

350

300

250

200

150

100

FEEDBACK INPUT CURRENT (nA)

50

0

–50

Negative Feedback Input
Current vs Temperature

75

100

125

LT1373 • G11

150

LT1373

U

UU

PI FU CTIO S

VC (Pin 1): Compensation Pin. The VC pin is used for
frequency compensation, current limiting and soft start. It
is the output of the error amplifier and the input of the
current comparator. Loop frequency compensation can be
performed with an RC network connected from the V
to ground.

FB (Pin 2): T

he feedback pin is used for positive output
voltage sensing and oscillator frequency shifting. It is the
inverting input to the error amplifier. The noninverting
input of this amplifier is internally tied to a 1.245V
reference. Load on the FB pin should not exceed 100µA
when the NFB pin is used. See Applications Information.

NFB (Pin 3): The negative feedback pin is used for negative
output voltage sensing. It is connected to the inverting
input of the negative feedback amplifier through a 400k
source resistor.

S/S (Pin 4): Shutdown and Synchronization Pin. The S/S
pin is logic level compatible. Shutdown is active low and
the shutdown threshold is typically 1.3V. For normal
operation, pull the S/S pin high, tie it to V

or leave it

IN

C

pin

floating. To synchronize switching, drive the S/S pin be­tween 300kHz and 340kHz.

V

(Pin 5): Input Supply Pin. Bypass VIN with 10µF or

IN

more. The part goes into undervoltage lockout when V

IN

drops below 2.5V. Undervoltage lockout stops switching
and pulls the V

pin low.

C

GND S (Pin 6): The ground sense pin is a “clean” ground.
The internal reference, error amplifier and negative feed­back amplifier are referred to the ground sense pin. Con­nect it to ground. Keep the ground path connection to the
output resistor divider and the V

compensation network

C

free of large ground currents.
GND (Pin 7): The ground pin is the emitter connection of

the power switch and has large currents flowing through it.
It should be connected directly to a good quality ground
plane.

V

(Pin 8): The switch pin is the collector of the power

SW

switch and has large currents flowing through it. Keep the
traces to the switching components as short as possible to
minimize radiation and voltage spikes.

BLOCK DIAGRA

S/S

FEEDBACK

NFB

FB

400k

200k

1.245V
REF

GND SENSE

W

SHUTDOWN

DELAY AND RESET

SYNC

+

NEGATIVE

AMP

–

–

ERROR

AMP

+

250kHz

OSC

5:1 FREQUENCY

SHIFT

V

IN

LOW DROPOUT

2.3V REG

LOGIC DRIVER

COMP

CURRENT

AMP

V

C

AV ≈ 6

SW

ANTI-SAT

SWITCH

+

0.08Ω

–

GND LT1373 • BD

5

LT1373

OPERATIO

U

The LT1373 is a current mode switcher. This means that
switch duty cycle is directly controlled by switch current
rather than by output voltage. Referring to the Block
Diagram, the switch is turned “On” at the start of each
oscillator cycle. It is turned “Off” when switch current
reaches a predetermined level. Control of output voltage
is obtained by using the output of a voltage sensing error
amplifier to set current trip level. This technique has
several advantages. First, it has immediate response to
input voltage variations, unlike voltage mode switchers
which have notoriously poor line transient response.
Second, it reduces the 90° phase shift at mid-frequencies
in the energy storage inductor. This greatly simplifies
closed-loop frequency compensation under widely vary­ing input voltage or output load conditions. Finally, it
allows simple pulse-by-pulse current limiting to provide
maximum switch protection under output overload or
short conditions. A low dropout internal regulator pro­vides a 2.3V supply for all internal circuitry. This low
dropout design allows input voltage to vary from 2.7V to
25V with virtually no change in device performance. A
250kHz oscillator is the basic clock for all internal timing.
It turns “On” the output switch via the logic and driver
circuitry. Special adaptive anti-sat circuitry detects onset
of saturation in the power switch and adjusts driver
current instantaneously to limit switch saturation. This
minimizes driver dissipation and provides very rapid
turn-off of the switch.

A 1.245V bandgap reference biases the positive input of
the error amplifier. The negative input of the amplifier is
brought out for positive output voltage sensing. The error
amplifier has nonlinear transconductance to reduce out-

put overshoot on start-up or overload recovery. When
the feedback voltage exceeds the reference by 40mV,
error amplifier transconductance increases ten times,
which reduces output overshoot. The feedback input also
invokes oscillator frequency shifting, which helps pro­tect components during overload conditions. When the
feedback voltage drops below 0.6V, the oscillator fre­quency is reduced 5:1. Lower switching frequency allows
full control of switch current limit by reducing minimum
switch duty cycle.

Unique error amplifier circuitry allows the LT1373 to
directly regulate negative output voltages. The negative
feedback amplifier’s 400k source resistor is brought out
for negative output voltage sensing. The NFB pin regulates
at –2.45V while the amplifier output internally drives the
FB pin to 1.245V. This architecture, which uses the same
main error amplifier, prevents duplicating functions and
maintains ease of use. (Consult Linear Technology Mar­keting for units that can regulate down to –1.25V.)

The error signal developed at the amplifier output is
brought out externally. This pin (VC) has three different
functions. It is used for frequency compensation, current
limit adjustment and soft starting. During normal regula­tor operation this pin sits at a voltage between 1V (low
output current) and 1.9V (high output current). The error
amplifier is a current output (gm) type, so this voltage can
be externally clamped for lowering current limit. Like­wise, a capacitor coupled external clamp will provide soft
start. Switch duty cycle goes to zero if the VC pin is pulled
below the control pin threshold, placing the LT1373 in an
idle mode.

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APPLICATIO S I FOR ATIO

Positive Output Voltage Setting

The LT1373 develops a 1.245V reference (V

) from the

REF

FB pin to ground. Output voltage is set by connecting the
FB pin to an output resistor divider (Figure 1). The FB pin
bias current represents a small error and can usually be
ignored for values of R2 up to 25k. The suggested value for
R2 is 24.9k. The NFB pin is normally left open for positive
output applications.

6

V

OUT

= V

V

R1

OUT

FB

PIN

V

REF

Figure 1. Positive Output Resistor Divider

R1 = R2

R2

R1

1 +

REF

()

R2

V

OUT

– 1

()

1.245

LT1373 • F01

WUUU

APPLICATIO S I FOR ATIO

LT1373

Negative Output Voltage Setting

The LT1373 develops a –2.45V reference (V

) from the

NFR

NFB pin to ground. Output voltage is set by connecting the
NFB pin to an output resistor divider (Figure 2). The –7µA
NFB pin bias current (I

) can cause output voltage errors

NFB

and should not be ignored. This has been accounted for in
the formula in Figure 2. The suggested value for R2 is

2.49k. The FB pin is normally left open for negative output
applications. See Dual Polarity Output Voltage Sensing for
limitations of FB pin loading when using the NFB pin.

–V

OUT

= V

–V

R1

I

NFB

PIN

NFB

V

NFR

Figure 2. Negative Output Resistor Divider

OUT

V

R1 =

R2

2.45

()

R2

R1

+ I

NFB

()

R2

 – 2.45

OUT

+ (7 • 10–6)

(R1)1 +

NFB

LT1373 • F02

Dual Polarity Output Voltage Sensing

Certain applications benefit from sensing both positive
and negative output voltages. One example is the Dual
Output Flyback Converter with Overvoltage Protection
circuit shown in the Typical Applications section. Each
output voltage resistor divider is individually set as de­scribed above. When both the FB and NFB pins are used,
the LT1373 acts to prevent either output from going
beyond its set output voltage. For example in this applica­tion, if the positive output were more heavily loaded than
the negative, the negative output would be greater and
would regulate at the desired set-point voltage. The posi­tive output would sag slightly below its set-point voltage.
This technique prevents either output from going unregu­lated high at no load. Please note that the load on the FB
pin should not exceed 100µA when the NFB pin is used.
This situation occurs when the resistor dividers are used
at both FB

and

NFB. True load on FB is not the full divider
current unless the positive output is shorted to ground.
See Dual Output Flyback Converter application.

Shutdown and Synchronization

The dual function S/S pin provides easy shutdown and
synchronization. It is logic level compatible and can be
pulled high, tied to VIN or left floating for normal operation.

A logic low on the S/S pin activates shutdown, reducing
the part’s supply current to 12µA. Typical synchronization
range is from 1.05 and 1.8 times the part’s natural switch­ing frequency, but is only guaranteed between 300kHz and
340kHz. A 12µs resetable shutdown delay network guar-
antees the part will not go into shutdown while receiving
a synchronization signal.

Caution should be used when synchronizing above
330kHz because at higher sync frequencies the ampli­tude of the internal slope compensation used to prevent
subharmonic switching is reduced. This type of
subharmonic switching only occurs when the duty cycle
of the switch is above 50%. Higher inductor values will
tend to eliminate problems.

Thermal Considerations

Care should be taken to ensure that the worst-case input
voltage and load current conditions do not cause exces­sive die temperatures. The packages are rated at 120°C/W
for SO (S8) and 130°C/W for PDIP (N8).

Average supply current (including driver current) is:

IIN = 1mA + DC (ISW/60 + ISW • 0.004)
ISW = switch current
DC = switch duty cycle

Switch power dissipation is given by:

PSW = (ISW)2 • RSW • DC
RSW = output switch “On” resistance

Total power dissipation of the die is the sum of supply
current times supply voltage plus switch power:

P

D(TOTAL)

= (IIN • VIN) + P

SW

Choosing the Inductor

For most applications the inductor will fall in the range of
10µH to 50µH. Lower values are chosen to reduce physical
size of the inductor. Higher values allow more output
current because they reduce peak current seen by the
power switch which has a 1.5A limit. Higher values also
reduce input ripple voltage, and reduce core loss.

When choosing an inductor you might have to consider
maximum load current, core and copper losses, allowable

7

LT1373

WUUU

APPLICATIO S I FOR ATIO

component height, output voltage ripple, EMI, fault cur­rent in the inductor, saturation, and of course, cost. The
following procedure is suggested as a way of handling
these somewhat complicated and conflicting requirements.

1. Assume that the average inductor current (for a boost

converter) is equal to load current times V

OUT/VIN

and
decide whether or not the inductor must withstand
continuous overload conditions. If average inductor
current at maximum load current is 0.5A, for instance,
a 0.5A inductor may not survive a continuous 1.5A
overload condition. Also, be aware that boost convert­ers are not short-circuit protected, and that under
output short conditions, inductor current is limited only
by the available current of the input supply.

2. Calculate peak inductor current at full load current to
ensure that the inductor will not saturate. Peak current
can be significantly higher than output current, espe­cially with smaller inductors and lighter loads, so don’t
omit this step. Powered iron cores are forgiving be­cause they saturate softly, whereas ferrite cores satu­rate abruptly. Other core materials fall in between
somewhere. The following formula assumes continu­ous mode operation, but it errors only slightly on the
high side for discontinuous mode, so it can be used for
all conditions.

V

I

= I

PEAK

V

IN

OUT

= minimum input voltage

OUT

•
V

IN

V

IN (VOUT – VIN

+

2(f)(L)(V

OUT

)

)

f = 250kHz switching frequency

inductor gets too hot, wire insulation will melt and cause
turn-to-turn shorts). Keep in mind that all good things
like high efficiency, low profile and high temperature
operation will increase cost, sometimes dramatically.

5. After making an initial choice, consider the secondary
things like output voltage ripple, second sourcing, etc.
Use the experts in the Linear Technology application
department if you feel uncertain about the final choice.
They have experience with a wide range of inductor
types and can tell you about the latest developments in
low profile, surface mounting, etc.

Output Capacitor

The output capacitor is normally chosen by its effective
series resistance (ESR), because this is what determines
output ripple voltage. At 500kHz, any polarized capacitor
is essentially resistive. To get low ESR takes

volume

, so

physically smaller capacitors have high ESR. The ESR
range for typical LT1373 applications is 0.05Ω to 0.5Ω. A
typical output capacitor is an AVX type TPS, 22µF at 25V,
with a guaranteed ESR less than 0.2Ω. This is a “D” size
surface mount solid tantalum capacitor. TPS capacitors
are specially constructed and tested for low ESR, so they
give the lowest ESR for a given volume. To further reduce
ESR, multiple output capacitors can be used in parallel.
The value in microfarads is not particularly critical and
values from 22µF to greater than 500µF work well, but you
cannot cheat mother nature on ESR. If you find a tiny 22µF
solid tantalum capacitor, it will have high ESR and output
ripple voltage will be terrible. Table 1 shows some typical
solid tantalum surface mount capacitors.

3. Decide if the design can tolerate an “open” core geom­etry like a rod or barrel, which have high magnetic field
radiation, or whether it needs a closed core like a toroid
to prevent EMI problems. One would not want an open
core next to a magnetic storage media for instance!
This is a tough decision because the rods or barrels are
temptingly cheap and small, and there are no helpful
guidelines to calculate when the magnetic field radia­tion will be a problem.

4. Start shopping for an inductor which meets the require­ments of core shape, peak current (to avoid saturation),
average current (to limit heating), and fault current, (if the

8

Table 1. Surface Mount Solid Tantalum Capacitor
ESR and Ripple Current

E CASE SIZE ESR (MAX Ω) RIPPLE CURRENT (A)

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
AVX TAJ 0.7 to 0.9 0.4

D CASE SIZE

AVX TPS, Sprague 593D 0.1 to 0.3 0.7 to 1.1
AVX TAJ 0.9 to 2.0 0.36 to 0.24

C CASE SIZE

AVX TPS 0.2 (Typ) 0.5 (Typ)
AVX TAJ 1.8 to 3.0 0.22 to 0.17

B CASE SIZE

AVX TAJ 2.5 to 10 0.16 to 0.08

WUUU

APPLICATIO S I FOR ATIO

LT1373

Many engineers have heard that solid tantalum capacitors
are prone to failure if they undergo high surge currents.
This is historically true and type TPS capacitors are
specially tested for surge capability, but surge ruggedness
is not a critical issue with the
tantalum capacitors fail during very high
which do not occur at the output of regulators. High

discharge

dead shorted, do not harm the capacitors.
Single inductor boost regulators have large RMS ripple

current in the output capacitor, which must be rated to
handle the current. The formula to calculate this is:

Output Capacitor Ripple Current (RMS)

I

Input Capacitors

The input capacitor of a boost converter is less critical due
to the fact that the input current waveform is triangular,
and does not contain large squarewave currents as is
found in the output capacitor. Capacitors in the range of
10µF to 100µF with an ESR (effective series resistance) of

0.3Ω or less work well up to a full 1.5A switch current.

Higher ESR capacitors may be acceptable at low switch
currents. Input capacitor ripple current for boost con­verter is:

I

f = 250kHz switching frequency

The input capacitor can see a very high surge current when
a battery or high capacitance source is connected “live”,
and solid tantalum capacitors can fail under this condition.
Several manufacturers have developed a line of solid
tantalum capacitors specially tested for surge capability
(AVX TPS series, for instance), but even these units may
fail if the input voltage approaches the maximum voltage
rating of the capacitor. AVX recommends derating capaci­tor voltage by 2:1 for high surge applications. Ceramic and

surges, such as when the regulator output is

RIPPLE

RIPPLE

(RMS) = I

= I

0.3(V

=

IN

(f)(L)(V

OUT

OUT

)(V

OUT

OUT

output

DC

1 – DC

V

OUT – VIN

V

– VIN)

)

capacitor. Solid

turn-on

IN

surges,

aluminum electrolytic capacitors may also be used and
have a high tolerance to turn-on surges.

Ceramic Capacitors

Higher value, lower cost ceramic capacitors are now
becoming available in smaller case sizes. These are tempt­ing for switching regulator use because of their very low
ESR. Unfortunately, the ESR is so low that it can cause
loop stability problems. Solid tantalum capacitor ESR
generates a loop “zero” at 5kHz to 50kHz that is instrumen­tal in giving acceptable loop phase margin. Ceramic ca­pacitors remain capacitive to beyond 300kHz and usually
resonate with their ESL before ESR becomes effective.
They are appropriate for input bypassing because of their
high ripple current ratings and tolerance of turn-on surges.
Linear Technology plans to issue a Design Note on the use
of ceramic capacitors in the near future.

Output Diode

The suggested output diode (D1) is a 1N5818 Schottky or
its Motorola equivalent, MBR130. It is rated at 1A average
forward current and 30V reverse voltage. Typical forward
voltage is 0.42V at 1A. The diode conducts current only
during switch-off time. Peak reverse voltage for boost
converters is equal to regulator output voltage. Average
forward current in normal operation is equal to output
current.

Frequency Compensation

Loop frequency compensation is performed on the output
of the error amplifier (VC pin) with a series RC network. The
main pole is formed by the series capacitor and the output
impedance (≈1MΩ) of the error amplifier. The pole falls in
the range of 5Hz to 30Hz. The series resistor creates a
“zero” at 2kHz to 10kHz, which improves loop stability and
transient response. A second capacitor, typically one tenth
the size of the main compensation capacitor, is sometimes
used to reduce the switching frequency ripple on the V
pin. VC pin ripple is caused by output voltage ripple
attenuated by the output divider and multiplied by the error
amplifier. Without the second capacitor, VC pin ripple is:

VC Pin Ripple =

1.245(V

RIPPLE

V

OUT

)(gm)(RC)

C

9

LT1373

WUUU

APPLICATIO S I FOR ATIO

V

= output ripple (V

RIPPLE

P-P

)
gm = error amplifier transconductance (≈375µmho)
RC = series resistor on VC pin
V

= DC output voltage

OUT

To prevent irregular switching, VC pin ripple should be
kept below 50mV

. Worst-case VC pin ripple occurs at

P-P

maximum output load current and will also be increased if
poor quality (high ESR) output capacitors are used. The
addition of a 0.001µF capacitor on the VC pin reduces
switching frequency ripple to only a few millivolts. A low
value for RC will also reduce VC pin ripple, but loop phase
margin may be inadequate.

Switch Node Considerations

For maximum efficiency, switch rise and fall time are made
as short as possible. To prevent radiation and high fre­quency resonance problems, proper layout of the compo­nents connected to the switch node is essential. B field
(magnetic) radiation is minimized by keeping output di­ode, switch pin and output bypass capacitor leads as short
as possible. E field radiation is kept low by minimizing the
length and area of all traces connected to the switch pin.
A ground plane should always be used under the switcher
circuitry to prevent interplane coupling.

The high speed switching current path is shown schemati­cally in Figure 3. Minimum lead length in this path is
essential to ensure clean switching and low EMI. The path
including the switch, output diode and output capacitor is
the only one containing nanosecond rise and fall times.
Keep this path as short as possible.

SWITCH

L1

NODE

V

IN

HIGH

FREQUENCY

CIRCULATING

PATH

LOAD

V

OUT

LT1373 • F03

Figure 3

More Help

For more detailed information on switching regulator
circuits, please see AN19. Linear Technology also offers a
computer software program, SwitcherCADTM, to assist in
designing switching converters. In addition, our applica­tions department is always ready to lend a helping hand.

SwitcherCAD is a trademark of Linear Technology Corporation.

U

TYPICAL APPLICATIONS N

Positive-to-Negative Converter with Direct Feedback

V

IN

OFF

+

ON

4

C1
22µF

2.7V TO 16V

S/S

LT1373

V

C

1

R1
5k

5

V

IN

C2

0.01µF

D2
P6KE-15A
D3
1N4148

8

V

SW

3

NFB

GND

6, 7

*COILTRONICS CTX20-2 [phone]

†

214

•

MBRS130LT3

MAX I

OUT

V

I

IN

OUT

3V

0.3A
5V

0.5A
9V

0.75A

T1*

•

3

D1

Dual Output Flyback Converter with Overvoltage Protection

R2

275k

1%

+

C3
47µF

+

†

R2

2.55k
1%

R3

2.49k
1%

–V

OUT

–5V

LT1373 • TA03

OFF

ON

C1
100µF

4

S/S

FB

V

C

1

V

IN

4.75V TO 13V

52

V

IN

V

SW

LT1373

NFB

GND

6, 7

C2

0.01µF

R3
5k

*DALE LPE-4841-100MB [phone]

R1

302.6k
1%

MBRS140T3

T1*

2, 3

5

+

4
8

+

1

6, 7

•

•

•

MBRS140T3

P6KE-20A

1N4148

8

3

C3
47µF

C4
47µF

R4

12.4k
1%

R5

2.49k
1%

V

OUT

15V

–V

OUT

–15V

LT1373 • TA04

10

TYPICAL APPLICATIO S

V

IN

5V

C1

22µF

10V

*

**

U

Low Ripple 5V to –3V “Cuk”† Converter

L1*

2

••

C2

47µF

5

V

4

7

6

IN

S/S

GND

GND S

LT1373

+

SUMIDA CLS62-100L
MOTOROLA MBR0520LT3

†

PATENTS MAY APPLY

V

NFB

SW

V

C

16V

8

+

3

1

D1**

R4
5k

C4

0.01µF

LT1373

V

OUT

C3
47µF
16V

–3V
250mA

C6

0.1µF

LT1373 • TA05

3

41

R1
1k
1%

+

R2

5.49k
1%

PACKAGE DESCRIPTION

0.300 – 0.325

(7.620 – 8.255)

0.065

(1.651)

0.009 – 0.015

(0.229 – 0.381)

+0.035

0.325

–0.015

+0.889

8.255

()

–0.381

*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)

0.010 – 0.020

(0.254 – 0.508)

0.008 – 0.010

(0.203 – 0.254)

*

DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE

**

DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE

× 45°

0.016 – 0.050

(0.406 – 1.270)

TYP

0°– 8° TYP

U

Dimensions in inches (millimeters) unless otherwise noted.

N8 Package

8-Lead PDIP (Narrow 0.300)

(LTC DWG # 05-08-1510)

0.045 – 0.065

(1.143 – 1.651)

0.100
(2.54)

BSC

0.130 ± 0.005

(3.302 ± 0.127)

0.125

(3.175)

MIN

0.018 ± 0.003

(0.457 ± 0.076)

0.020

(0.508)

MIN

0.255 ± 0.015*
(6.477 ± 0.381)

S8 Package

8-Lead Plastic Small Outline (Narrow 0.150)

(LTC DWG # 05-08-1610)

0.053 – 0.069

(1.346 – 1.752)

0.014 – 0.019

(0.355 – 0.483)

TYP

0.004 – 0.010

(0.101 – 0.254)

0.050

(1.270)

BSC

0.228 – 0.244

(5.791 – 6.197)

0.400*
(10.160)

MAX

876

1234

0.189 – 0.197*
(4.801 – 5.004)

7

8

1

2

6

3

5

N8 1098

5

0.150 – 0.157**
(3.810 – 3.988)

4

SO8 1298

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen­tation that the interconnection of circuits as described herein will not infringe on existing patent rights.

11

LT1373

TYPICAL APPLICATIO S

U

90% Efficient CCFL Supply

C2

27pF

V

IN

4.5V

+

TO 30V

2.7V TO

5.5V

OFF

C1 = WIMA MKP-20

L1 = COILCRAFT D03316-104

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

T1 = COILTRONICS CTX 110609
* = 1% FILM RESISTOR

DO NOT SUBSTITUTE COMPONENTS

COILTRONICS [phone]
COILCRAFT [phone]

ON

10µF

330Ω

1N5818

+

2.2µF

S/S

GND

5

V

IN

LT1373

V

SW

V

FB

V

C

16, 7

+

5mA MAX

LAMP

10

T1

12345

C1

0.1µF

Q1

84

2

2µF

OPTIONAL REMOTE

DIMMING

CCFL BACKLIGHT APPLICATION CIRCUITS
CONTAINED IN THIS DATA SHEET ARE
COVERED BY U.S. PATENT NUMBER 5408162
AND OTHER PATENTS PENDING

L1
100µH

22k

1N4148

Q2

562Ω*

20k
DIMMING

D2
1N4148

10k

0.1µF

LT1372 • TA06

D1
1N4148

Two Li-Ion Cells to 5V SEPIC Conveter

V

IN

4V TO 9V

FB

1

R1
5k

8

2

C4

0.01µF

•

L1A
33µH

C2

2.2µF

MBRS130LT3

•

L1B
33µH

†

MAX I

OUT

I

OUT

0.45A

0.55A

0.65A

0.72A

D1

†

V

OUT

+

5V

C3
100µF
10V

LT1373 • TA07

R2
75k
1%

R3

24.9k
1%

V

IN

4V
5V
7V
9V

5

V

S/S

GND

LT1373

6, 7

IN

V

SW

V

C

4

ON

OFF

C1

+

33µF
20V

C1 = AVX TPSD 336M020R0200
C2 = TOKIN 1E225ZY5U-C203-F
C3 = AVX TPSD 107M010R0100
L1 = COILTRONICS CTX33-2, SINGLE
INDUCTOR WITH TWO WINDINGS

RELATED PARTS

PART NUMBER DESCRIPTION COMMENTS

LT1172 100kHz 1.25A Boost Switching Regulator Also for Flyback, Buck and Inverting Configurations
LTC®1265 13V 1.2A Monolithic Buck Converter Includes PMOS Switch On-Chip
LT1302 Micropower 2A Boost Converter Converts 2V to 5V at 600mA
LT1308A/LT1308B 600kHz 2A Switch DC/DC Converter 5V at 1A from a Single Li-Ion Cell
LT1370 500kHz High Efficiency 6A Boost Converter 6A, 0.065Ω Internal Switch
LT1372 500kHz 1.5A Boost Switching Regulator Also Regulates Negative Flyback Outputs
LT1374 4.5A, 500kHz Step-Down Converter 4.5A, 0.07Ω Internal Switch
LT1376 500kHz 1.5A Buck Switching Regulator Handles Up to 25V Inputs
LT1377 1MHz 1.5A Boost Switching Regulator Only 1MHz Integrated Switching Regulator Available
LT1613 1.4MHz Switching Regulator in 5-Lead SOT-23 5V at 200mA from 4.4V Input
LT1615 Micropower Step-Up DC/DC in 5-Lead SOT-23 20µA IQ, 36V, 350mA Switch
LT1949 600kHz, 1A Switch PWM DC/DC Converter 1.1A, 0.5Ω, 30V Internal Switch, VIN as Low as 1.5V

1373fb LT/TP 0200 2K REV B • PRINTED IN THE USA

 LINEAR TECHNOLOGY CORPORATION 1995

12

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7417

[phone] ● FAX: [phone]

●

www.linear-tech.com