LMD18245
3A, 55V DMOS Full-Bridge Motor Driver
LMD18245 3A, 55V DMOS Full-Bridge Motor Driver
April 1998
General Description
The LMD18245 full-bridge power amplifier incorporates all
the circuit blocks required to drive and control current in a
brushed type DC motor or one phase of a bipolar stepper
motor.The multi-technology process used tobuild thedevice
combines bipolar and CMOS control and protection circuitry
with DMOS power switches on the same monolithic structure. The LMD18245 controls the motor current via a fixed
off-time chopper technique.
An all DMOS H-bridge power stage delivers continuous output currents up to3A (6A peak)at supplyvoltages up to 55V.
The DMOS power switches feature low R
ficiency, and a diode intrinsic to the DMOS body structure
eliminates the discrete diodes typically required to clamp bipolar power stages.
An innovative current sensing method eliminates the power
loss associated witha senseresistor in serieswith themotor.
Afour-bit digital-to-analog converter (DAC) provides a digital
path for controllingthe motorcurrent, and, by extension, simplifies implementationof full, half and microstep stepper motor drives. For higher resolution applications, an external
DAC can be used.
DS(ON)
for high ef-
Features
n DMOS power stage rated at 55V and 3A continuous
n Low R
n Internal clamp diodes
n Low-loss current sensing method
n Digital or analog control of motor current
n TTL and CMOS compatible inputs
n Thermal shutdown (outputs off) at T
n Overcurrent protection
n No shoot-through currents
n 15-lead TO-220 molded power package
of typically 0.3Ω per power switch
DS(ON)
=
155˚C
J
Applications
n Full, half and microstep stepper motor drives
n Stepper motor and brushed DC motor servo drives
n Automated factory, medical and office equipment
Functional Block and Connection Diagram (15-Lead TO-220 Molded Power Package (T) )
DS011878-1
Order Number LMD18245T
See NS Package Number TA15A
© 1998 National Semiconductor Corporation DS011878 www.national.com
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
DC Voltage at:
OUT 1, V
COMP OUT, RC, M4, M3, M2, M1, BRAKE, +12V
DIRECTION, CS OUT, and DAC REF
DC Voltage PGND to SGND
Continuous Load Current 3A
Peak Load Current (Note 2) 6A
Junction Temperature (T
, and OUT 2 +60V
CC
±
) +150˚C
J(max)
400mV
Power Dissipation (Note 3) :
TO-220 (T
TO-220 (T
=
25˚C, Infinite Heatsink) 25W
A
=
25˚C, Free Air) 3.5W
A
ESD Susceptibility (Note 4) 1500V
Storage Temperature Range (T
) −40˚C to +150˚C
S
Lead Temperature (Soldering, 10 seconds) 300˚C
Operating Conditions (Note 1)
Temperature Range (T
Supply Voltage Range (V
CS OUT Voltage Range 0V to +5V
DAC REF Voltage Range 0V to +5V
) (Note 3) −40˚C to +125˚C
J
) +12V to +55V
CC
MONOSTABLE Pulse Range 10 µs to 100 ms
Electrical Characteristics (Note 2)
The following specifications apply for V
perature range, −40˚C ≤ T
≤ +125˚C. All other limits apply for T
J
Symbol Parameter Conditions Typical Limit Units
I
CC
Quiescent Supply Current DAC REF=0V, V
POWER OUTPUT STAGE
R
V
T
Q
t
D(ON)
DS(ON)
DIODE
rr
rr
Switch ON Resistance I
Body Diode Forward Voltage I
Diode Reverse Recovery Time I
Diode Reverse Recovery Charge I
Output Turn ON Delay Time
Sourcing Outputs I
Sinking Outputs I
t
D(OFF)
Output Turn OFF Delay Time
Sourcing Outputs I
Sinking Outputs I
t
ON
Output Turn ON Switching Time
Sourcing Outputs I
Sinking Outputs I
t
OFF
Output Turn OFF Switching Time
Sourcing Outputs I
Sinking Outputs I
t
pw
t
DB
Minimum Input Pulse Width Pins 10 and 11 2 µs
Minimum Dead Band (Note 6) 40 ns
CURRENT SENSE AMPLIFIER
Current Sense Output I
Current Sense Linearity Error 0.5A ≤ I
Current Sense Offset I
=
+42V, unless otherwise stated. Boldface limits apply over the operating tem-
CC
=
=
T
25˚C.
A
J
(Note 5) (Note 5) (Limits)
=
+20V 8 mA
CC
15 mA (max)
=
3A 0.3 0.4 Ω (max)
LOAD
0.6 Ω (max)
=
I
6A 0.3 0.4 Ω (max)
LOAD
0.6 Ω (max)
=
3A 1.0 V
DIODE
1.5 V(max)
=
1A 80 ns
DIODE
=
1A 40 nC
DIODE
=
3A 5 µs
LOAD
=
3A 900 ns
LOAD
=
3A 600 ns
LOAD
=
3A 400 ns
LOAD
=
3A 40 µs
LOAD
=
3A 1 µs
LOAD
=
3A 200 ns
LOAD
=
3A 80 ns
LOAD
=
1A (Note 7) 200 µA (min)
LOAD
250 175 µA (min)
300 µA (max)
325 µA (max)
≤ 3A (Note 7)
LOAD
=
0A 5 µA
LOAD
±
6
±
9
20 µA (max)
%
(max)
%
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Electrical Characteristics (Note 2) (Continued)
The following specifications apply for V
perature range, −40˚C ≤ T
≤ +125˚C. All other limits apply for T
J
Symbol Parameter Conditions Typical Limit Units
DIGITAL-TO-ANALOG CONVERTER (DAC)
Resolution 4 Bits (min)
Monotonicity 4 Bits (min)
Total Unadjusted Error 0.125 0.25 LSB (max)
Propagation Delay 50 ns
I
REF
DAC REF Input Current DAC REF=+5V −0.5 µA
COMPARATOR AND MONOSTABLE
Comparator High Output Level 6.27 V
Comparator Low Output Level 88 mV
Comparator Output Current
Source 0.2 mA
Sink 3.2 mA
t
DELAY
Monostable Turn OFF Delay (Note 8) 1.2 µs
PROTECTION AND PACKAGE THERMAL RESISTANCES
Undervoltage Lockout, V
T
JSD
Shutdown Temperature, T
Package Thermal Resistances
θ
JC
θ
JA
Junction-to-Case, TO-220 1.5 ˚C/W
Junction-to-Ambient, TO-220 35 ˚C/W
LOGIC INPUTS
V
IL
V
IH
I
IN
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the devicemay occur. Electrical specifications do notapply when operating the device
outside the rated Operating Conditions.
Note 2: Unless otherwise stated, load currents are pulses with widths less than 2 ms and duty cycles less than 5%.
Note 3: The maximum allowable powerdissipation at any ambient temperature is P
eration, T
ing T
junction-to-case thermal resistance of the package, θ
Note 4: ESDrating is based on thehuman body model of 100 pFdischarged through a 1.5 kΩ resistor.M1,M2,M3 and M4, pins 8,7, 6 and 4 are protectedto 800V.
Note 5: All limits are 100%production tested at 25˚C. Temperature extreme limits are guaranteed via correlation using accepted SQC (Statistical Quality Control)
methods. All limits are used to calculate AOQL (Average Outgoing Quality Level). Typicals are at T
Note 6: Asymmetric turn OFF and ON delay times and switching times ensure a switch turns OFF before the other switch in the same half H-bridge begins to turn
ON (preventing momentary short circuits between the power supply and ground). The transitional period during which both switches are OFF is commonly referred
to as the dead band.
Note 7: (I
The current sense linearity is specified as the slope of the line between the 0.5A and 1Adata points minus the slope of the line between the 2A and 3Adata points
all divided by the slope of the line between the 0.5A and 1A data points.
Note 8: Turn OFF delay, t
DMOS switch beginning to turn OFF. With V
5V at 1.2V/µs, and t
Low Level Input Voltage −0.1 V (min)
High Level Input Voltage 2 V (min)
Input Current V
is the ambient temperaturein ˚C, and θJAis the junction-to-ambient thermalresistance in ˚C/W. Exceeding P
A
above 125˚C. If the junction temperature exceeds 155˚C, internal circuitry disables the power bridge. When a heatsink is used, θJAis the sum of the
J
LOAD,ISENSE
) data points are taken for load currents of 0.5A, 1A, 2A and 3A. The current sense gain is specified as I
, is defined as the time from the voltage at the output of the current sense amplifier reaching the DAC output voltage to the lower
DELAY
is measured as the time from the voltage at RC reaching 2V to the time the voltage at OUT 1 reaches 3V.
DELAY
=
+42V, unless otherwise stated. Boldface limits apply over the operating tem-
CC
CC
J
=
0V or 12V
IN
, and the case-to-ambient thermal resistance of the heatsink.
JC
=
32V, DIRECTION high, and 200Ω connected between OUT1 and V
CC
A
Max
=
=
T
25˚C.
J
=
)/θJA, where 125˚Cis the maximum junction temperature for op-
(125 − T
A
=
J
(Note 5) (Note 5) (Limits)
0.5 LSB (max)
±
10 µA (max)
2.0 µs (max)
5 V (min)
8 V (max)
155 ˚C
0.8 V (max)
12 V (max)
±
10 µA (max)
voids the Electrical Specificationsby forc-
max
25˚C and represent the most likely parametric norm.
SENSE/ILOAD
, the voltage at RC is increased from 0V to
CC
for the 1A data point.
3 www.national.com
Typical Performance Characteristics
RDS(ON) vs Temperature
RDS(ON) vs
Supply Voltage
Supply Current vs
Supply Voltage
DS011878-29
DS011878-31
RDS(ON) vs Load Current
DS011878-30
Current Sense Output
vs Load Current
DS011878-32
Supply Current vs
Temperature
DS011878-33
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DS011878-34
Connection Diagram
DS011878-2
15-Lead TO-220 Molded Power Package
See NS Package Number TA15A
Pinout Descriptions
and Connection Diagrams)
Pin 1, OUT 1: Output node of the first half H-bridge.
Pin 2, COMP OUT: Output of the comparator. If the voltage
at CS OUT exceeds that providedby theDAC, the comparator triggers the monostable.
Pin 3, RC: Monostable timing node. A parallel resistorcapacitor network connected between this node and ground
sets the monostable timing pulse at about 1.1 RC seconds.
Pin 5, PGND: Ground return node ofthe powerbridge. Bond
wires (internaI) connect PGND to the tab of the TO-220
package.
Pins 4 and 6 through 8, M4 through M1: Digital inputs of
the DAC. These inputs make up a four-bit binary number
with M4 as the most significant bit or MSB. The DAC provides an analog voltage directly proportional to the binary
number applied at M4 through M1.
Pin 9, V
Pin 10, BRAKE: Brake logic input. Pulling the BRAKE input
logic-high activates both sourcing switches of the power
bridge—effectively shorting the load. See
the load in this manner forces the load current to recirculate
and decay to zero.
Pin 11, DIRECTION: Direction logic input. The logic level at
this input dictates the direction of current flow in the load.
See
Pin 12, SGND: Ground returnnode of allsignal levelcircuits.
: Power supply node.
CC
Table 1
.
Top View
Order Number LMD18245T
(See Functional Block
Table 1
. Shorting
Pin 13, CS OUT: Output of the current sense amplifier. The
current sense amplifier sources 250 µA(typical) per ampere
of totalforward current conducted by the upper two switches
of the power bridge.
Pin 14, DAC REF: Voltage reference input of the DAC. The
DAC provides an analog voltage equal to V
where D is the decimal equivalent (0–15) of the binary num-
DAC REF
x D/16,
ber applied at M4 through M1.
Pin 15, OUT 2: Output node of the second half H-bridge.
TABLE 1. Switch Control Logic Truth Table
BRAKE DIRECTION MONO Active Switches
H X X Source 1, Source 2
L H L Source 2
L H H Source 2, Sink 1
L L L Source 1
L L H Source 1, Sink 2
X=don’t care
MONO is the output of the monostable.
Functional Descriptions
TYPICAL OPERATION OF A CHOPPER AMPLIFIER
Chopper amplifiers employ feedback driven switching of a
power bridge to control and limit current in the winding of a
motor (
Figure 1
power switches and four diodes connected in an H configuration. Control circuitry (not shown) monitors the winding
current and compares it to a threshold. While the winding
current remainsless than the threshold, a source switch and
a sink switch in opposite halves of the bridge force the supply voltage across the winding, and the winding current increases rapidly towards V
As the winding current surpasses the threshold, the control
circuitry turns OFF the sink switch for a fixed period or
off-time.
site upper diode short the winding, and the winding current
recirculates and decays slowly towards zero (
Figure 1e
turns back ON the sink switch, and the winding current again
increases rapidly towards V
again). The above sequence repeats to provide a current
chopping action that limits the winding current to the threshold (
Figure 1g
reaches the threshold. During a change in the direction of
the winding current, the diodes provide a decay path for the
initial winding current (
bridge shorts the winding fora fixed period, this typeof chopper amplifier is commonly referred to as a
chopper.
). The bridge consists of four solid state
/R (
CC
Figure 1a
and
Figure 1d
During the off-time, the source switch and the oppo-
Figure 1b
and
). At the end of the off-time, the control circuitry
/R (
CC
Figure 1a
and
Figure 1d
). Chopping only occurs if the winding current
Figure 1c
and
Figure 1f
). Since the
fixed off-time
).
5 www.national.com
Functional Descriptions (Continued)
(a)
(c)
(e)
DS011878-3
DS011878-5
(b)
DS011878-4
(d)
DS011878-6
(f)
DS011878-7
DS011878-8
(g)
DS011878-9
FIGURE 1. Chopper Amplifier Chopping States: Full VCCApplied Across the Winding (a) and (d), Shorted Winding (b)
and (e), Winding Current Decays During a Change in the Direction of the Winding Current (c) and (f), and the
Chopped Winding Current (g)
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Functional Descriptions (Continued)
THE LMD18245 CHOPPER AMPLIFIER
The LMD18245 incorporates all the circuit blocks needed to
implement a fixed off-time chopper amplifier. These blocks
include: an all DMOS, full H-bridge with clamp diodes, an
amplifier for sensing the load current, a comparator, a
monostable, and a DAC for digital control of the chopping
threshold.Also incorporated are logic, levelshifting anddrive
blocks for digital control of the direction of the load current
and braking.
THE H-BRIDGE
The power stage consists of fourDMOS powerswitches and
associated bodydiodes connected in an H-bridge configuration (
Figure 2
). Turning ON a source switch and a sink
switch in opposite halves of the bridge forces the full supply
voltage less the switch drops across the motor winding.
While the bridge remainsin thisstate, the winding current increases exponentially towards a limit dictated by the supply
voltage, the switch drops, and the winding resistance. Subsequently turning OFF the sink switchcauses avoltage transient that forward biases the body diode of the other source
switch. The diode clamps the transient at one diode drop
above the supply voltageand providesan alternative current
path. While the bridge remains in this state, it essentially
shorts the winding and the winding current recirculates and
decays exponentially towards zero. During a change in the
direction of the winding current, both the switches and the
body diodes provide a decay path for the initial winding current (
Figure 3
).
DS011878-10 DS011878-11
FIGURE 2. The DMOS H-Bridge
DS011878-12 DS011878-13
FIGURE 3. Decay Paths for Initial Winding Current During a Change in the Direction of the Winding Current
7 www.national.com
Functional Descriptions (Continued)
THE CURRENT SENSE AMPLIFIER
Many transistor cells in parallel make up the DMOS power
switches. The current sense amplifier (
small fractionof the cells of both upper switches to providea
unique, low-loss means for sensing theload current.In practice, each upper switch functions as a 1x sense device in
parallel with a 4000x power device. Thecurrent sense amplifier forces the voltage at the source of the sense device to
equal that at the source of the power device; thus, the devices share the totaldrain current in proportion to the 1:4000
cell ratio. Only the current flowing from drain to source, the
forward current, registers at the output of the current sense
amplifier. The current sense amplifier, therefore, sources
250 µA per ampereof total forward current conducted by the
upper two switches of the power bridge.
The sensecurrent develops a potential across R
portional to the load current; for example,per ampereof load
current, the sense current develops one volt across a 4 kΩ
resistor (the product of 250 µA per ampere and 4 kΩ). Since
chopping of the load current occurs as the voltage at CS
OUT surpasses the threshold (the DAC output voltage), R
sets the gain of the chopper amplifier; for example, a 2 kΩ
resistor sets the gain at two amperes of load current pervolt
of the threshold (the reciprocal of the product of 250 µA per
ampere and 2 kΩ). A quarter watt resistor suffices. A low
value capacitor connected in parallel with R
fects of switching noise from the current sense signal.
While the specified maximum DC voltage compliance at CS
OUT is 12V, the specified operating voltage range at CS
OUT is 0V to 5V.
Figure 4
filters the ef-
S
) uses a
that ispro-
S
THE DIGITAL-TO-ANALOG CONVERTER (DAC)
The DAC sets the threshold voltage for chopping at
V
of the binary number appliedat M4 through M1, thedigital in-
x D/16, where D is the decimal equivalent (0–15)
DAC REF
puts of the DAC. M4 is the MSB or most significant bit. For
applications that require higher resolution, an external DAC
can drive the DAC REF input. While the specified maximum
DC voltage compliance at DACREF is12V,the specified operating voltage range at DAC REF is 0V to 5V.
THE COMPARATOR, MONOSTABLE AND WINDING
CURRENT THRESHOLD FOR CHOPPING
As the voltage at CS OUT surpasses that at the output of the
DAC, the comparator triggers the monostable, and the
monostable, once triggered, provides a timing pulse to the
control logic. During the timing pulse, the power bridge
shorts the motor winding, causing current in the winding to
recirculate and decay slowly towards zero (
Figure 1e
nected between RC (pin
again). A parallel resistor-capacitor network con-
#
3) and ground sets the timing
pulse or off-time at about 1.1 RC seconds.
Chopping ofthe winding current occurs as the voltage at CS
S
OUT exceeds that at the output of the DAC; so chopping occurs at a winding current threshold of about
(V
x D/16)÷((250 x 10−6)xRS)) amperes.
DAC REF
Figure 1b
and
FIGURE 4. The Source Switches of the Power Bridge and the Current Sense Amplifier
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DS011878-14
Applications Information
POWER SUPPLY BYPASSING
Step changes in current drawn from the power supply occur
repeatedly during normal operation and may cause large
voltage spikes across inductance in the power supply line.
Care must betaken tolimit voltage spikesat V
the 60VAbsolute Maximum Rating. At a change in the direction of the load current, the initial load current tends to raise
the voltageat the power supply rail(
Figure 3
transients caused by the reverse recovery of the clamp diodes tend to pull down the voltage at the power supply rail.
Bypassing the power supply line at V
the device and minimize the adverse effects of normal op-
CC
eration on the power supply rail. Using botha1µFhigh frequency ceramic capacitor and a large-value aluminum electrolytic capacitor is highly recommended. A value of 100 µF
per ampere of load current usually suffices for the aluminum
electrolytic capacitor. Both capacitors should have short
leads and be located within one half inch of V
OVERCURRENT PROTECTION
If the forward current in either source switch exceeds a 12A
threshold, internal circuitry disables both source switches,
forcing a rapid decay of the fault current (
mately 3 µs after the fault current reaches zero, the device
restarts.Automatic restartallows animmediate returnto normal operation once the fault condition has been removed. If
the faultpersists, the device will begin cycling into and outof
thermal shutdown. Switching large fault currents may cause
potentially destructive voltage spikes across inductance in
the power supply line; therefore, the power supply line must
be properly bypassed at V
an extended overcurrent fault.
for the motor driver to survive
CC
to less than
CC
) again.Current
is required to protect
.
CC
Figure 5
). Approxi-
In the case of a locked rotor, the inductance of the winding
tends to limit the rate of change of the fault current to a value
easily handled by the protection circuitry. In the case of a low
inductance short from either output to ground or between
outputs, thefault current could surge past the 12A shutdown
threshold, forcing the device to dissipate a substantial
amount of power for the brief period required to disable the
source switches. Because the fault power must be dissipated by only one source switch, a short from output to
ground represents the worstcase fault.Any overcurrent fault
is potentially destructive, especially while operating with high
supply voltages (≥30V), so precautions are in order. Sinking
V
for heat with 1 square inch of 1 ounce copper on the
CC
printed circuit board is highly recommended. The sink
switches are not internally protected against shorts to V
CC
THERMAL SHUTDOWN
Internal circuitry senses the junction temperature near the
power bridge and disables the bridge if the junction temperature exceeds about 155˚C. When the junction temperature
cools past the shutdown threshold (lowered by a slight hysteresis), the device automatically restarts.
UNDERVOLTAGE LOCKOUT
Internal circuitry disables the power bridge if the power supply voltage drops below a rough threshold between 8V and
5V.Should the powersupply voltagethen exceed the threshold, the device automatically restarts.
.
Trace: Fault Current at 5A/div
Horizontal: 20 µs/div
FIGURE 5. Fault Current with V
=
30V, OUT 1 Shorted to OUT 2, and CS OUT Grounded
CC
The Typical Application
Figure 6
shows the typical application, the power stage of a
chopper drive for bipolar stepper motors.The 20 kΩ resistor
and 2.2 nF capacitorconnected between RC and ground set
the off-time at about 48 µs, and the 20 kΩ resistor connected
between CSOUT and ground sets the gain at about 200 mA
DS011878-15
per volt of the threshold for chopping. Digital signals control
the thresholds for chopping, the directions of the winding
currents, and, by extension, the drive type (full step, half
step, etc.). A µprocessor or µcontroller usually provides the
digital control signals.
9 www.national.com
The Typical Application (Continued)
FIGURE 6. Typical Application Circuit for Driving Bipolar Stepper Motors
ONE-PHASE-ON FULL STEP DRIVE (WAVE DRIVE)
To make the motor take full steps, windings A and B can be
energized in the sequence
→
→A→
*
*
B
A→B→A
…,
where Arepresents winding A energized with current in one
*
direction and A
represents winding Aenergized with current
in the opposite direction. The motor takes one full step each
time one winding isde-energized and the other is energized.
To make the motor step in theopposite direction,the orderof
the above sequence must be reversed.
Figure 7
shows the
winding currents and digital control signals for a wave drive
application of the typical application circuit.
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DS011878-16
TWO-PHASE-ON FULL STEP DRIVE
To makethe motortake fullsteps, windingsAand Bcan also
be energized in the sequence
→
AB→A
*
B→A*B
→AB→
*
*
AB
…,
and because both windings are energized at all times, this
sequence produces more torque than that produced with
wave drive. The motor takes one full step at each change of
direction of either winding current.
Figure 8
shows the winding currents and digital control signals for this application of
the typical application circuit, and
Figure 9
shows, for a
single phase, thewinding currentand voltage atthe outputof
the associated current sense amplifier.
The Typical Application (Continued)
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase B Winding Current at 1A/div
Horizontal: 1 ms/div
*500 steps/second
BRAKE A=BRAKE B=0
FIGURE 7. Winding Currents and Digital Control Signals for One-Phase-On Drive (Wave Drive)
DS011878-17
DS011878-18
11 www.national.com
The Typical Application (Continued)
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase B Winding Current at 1A/div
Horizontal: 1 ms/div
*500 steps/second
M4 A through M1 A=M4 B through M1 B=1
BRAKE A=BRAKE B=0
FIGURE 8. Winding Currents and Digital Control Signals for Two-Phase-On Drive
DS011878-19
DS011878-20
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase A Sense Voltage at 5V/div
Horizontal: 1 ms/div
*500 steps/second
FIGURE 9. Winding Current and Voltage at the Output of the Associated Current Sense Amplifier
HALF STEP DRIVE WITHOUT TORQUE
COMPENSATION
To make the motor take half steps, windings A and B can be
energized in the sequence
→
*
*
A→AB→B→A
→
*B*
A
B→A
→
→A→
*
*
B
AB
…
The motor takes one half step each time the number of energized windings changes. It is important to note that al-
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DS011878-21
though half stepping doubles the step resolution, changing
the number of energized windings from two to one decreases (one to two increases) torque by about 40%, resulting in significant torque ripple and possibly noisy operation.
Figure 10
shows the winding currents and digital control signals for this half step application of the typical application
circuit.
The Typical Application (Continued)
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase B Winding Current at 1A/div
Horizontal: 1 ms/div
*500 steps/second
BRAKE A=BRAKE B=0
DS011878-22
FIGURE 10. Winding Currents and Digital Control Signals for Half Step Drive without Torque Compensation
HALF STEP DRIVE WITH TORQUE COMPENSATION
To make the motor take half steps, the windingscan also be
Figure 11
energized with sinusoidal currents (
). Controlling
the winding currents in the fashion shown doubles the step
resolution without the significant torque ripple of the prior
drive technique. The motor takes one half step each time the
level of either winding current changes. Half step drive with
torque compensation is microstepping drive. Along with the
obvious advantage of increased step resolution, microstepping reduces both full step oscillations and resonances that
occur as the motor andload combination is driven atits natural resonant frequency or subharmonics thereof. Both of
these advantages are obtained by replacing full steps with
bursts of microsteps. When compared to full step drive, the
motor runs smoother and quieter.
Figure 12
shows the lookup table for this application of the
typical application circuit. Dividing 90˚electrical per full step
by two microsteps per full step yields 45˚ electrical per microstep. α, therefore, increases from0 to 315˚ in increments
of 45˚. Each full 360˚ cycle comprises eight half steps.
Rounding |cosα| to four bits gives D A, the decimal equivalent of the binary number applied at M4 A through M1 A. DIRECTION A controls the polarity of the current in winding A.
Figure 11
shows the sinusoidal winding currents.
DS011878-23
13 www.national.com
The Typical Application (Continued)
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase B Winding Current at 1A/div
Horizontal: 2 ms/div
*500 steps/second
BRAKE A=BRAKE B=0
DS011878-24
DS011878-25
90˚ ELECTRICAL/FULL STEP÷2 MICROSTEPS/FULL STEP=45˚ ELECTRICAL/MICROSTEP
FIGURE 11. Winding Currents and Digital Control Signals for Half Step Drive with Torque Compensation
|
FORWARD
↓
α |cos(α)| D A DIRECTION A |sin(α)| D B DIRECTlON B
01151 001
45 0.707 11 1 0.707 11 1
90 0 0 0 1 15 1
135 0.707 11 0 0.707 11 1
↑
180 1 15 0 0 0 0
REVERSE 225 0.707 11 0 0.707 11 0
| 270 0 0 1 1 15 0
315 0.707 11 1 0.707 11 0
REPEAT
FIGURE 12. Lookup Table for Half Step Drive with Torque Compensation
www.national.com 14
The Typical Application (Continued)
QUARTER STEP DRIVE WITH TORQUE
COMPENSATION
Figure 13
quarter step drive (four microsteps per full step) with torque
compensation.
shows the winding currents and lookup table for a
TopTrace: Phase AWinding Current at 1A/div
Bottom Trace: Phase B Winding Current at 1A/div
Horizontal: 2ms/div
*250 steps/second
DS011878-26
90˚ ELECTRICAL/FULL STEP÷4 MICROSTEPS/FULL STEP=22.5˚ ELECTRICAL/MICROSTEP
α |cos(α)| D A DIRECTION A |sin(α)| D B DIRECTION B
01151 001
22.5 0.924 14 1 0.383 6 1
| 45 0.707 11 1 0.707 11 1
FORWARD 67.5 0.383 6 1 0.924 14 1
↓
90 0 0 0 1 15 1
112.5 0.383 6 0 0.924 14 1
↑
135 0.707 11 0 0.707 11 1
REVERSE 157.5 0.924 14 0 0.383 6 1
| 180 1 15 0 0 0 0
202.5 0.924 14 0 0.383 6 0
225 0.707 11 0 0.707 11 0
247.5 0.383 6 0 0.924 14 0
270 0 0 1 1 15 0
292.5 0.383 6 1 0.924 14 0
315 0.707 11 1 0.707 11 0
337.5 0.924 14 1 0.383 6 0
REPEAT
BRAKE A=BRAKE B=0
FIGURE 13. Winding Currents and Lookup Table for Quarter Step Drive with Torque Compensation
15 www.national.com
Test Circuit and Switching Time Definitions
www.national.com 16
DS011878-28
17
Physical Dimensions inches (millimeters) unless otherwise noted
15-Lead TO-220 Power Package (T)
Order Number LMD18245T
NS Package Number TA15A
LMD18245 3A, 55V DMOS Full-Bridge Motor Driver
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