September 1998
LM4766 Overture
Dual 40W Audio Power Amplifier with Mute
LM4766
Overture
™
Audio Power Amplifier Series Dual 40W Audio Power
Amplifier with Mute
General Description
The LM4766 is a stereo audio amplifier capable of delivering
typically 40W per channel of continuous average output
power into an 8Ω load with less than 0.1%(THD + N).
The performance of the LM4766, utilizing its Self Peak Instantaneous Temperature (˚Ke) (SPiKe
cuitry, places it in a class above discrete and hybrid amplifiers by providing an inherently, dynamically protected Safe
Operating Area (SOA). SPiKe Protection means that these
parts are safeguarded at the output against overvoltage, undervoltage, overloads, including thermal runaway and instantaneous temperature peaks.
Each amplifier within the LM4766 has an independent
smooth transition fade-in/out mute that minimizes output
pops. The IC’s extremely low noise floor at 2 µV and its extremely low THD + N value of 0.06%at the rated power
make the LM4766 optimum for high-end stereo TVs or minicomponent systems.
™
) Protection Cir-
Typical Application
Key Specifications
j
THD+N at 1 kHz at 2 x 30W continuous average
output power into 8Ω: 0.1%(max)
j
THD+N at 1 kHz at continuous average
output power of 2 x 30W into 8Ω: 0.009%(typ)
Features
n SPiKe Protection
n Minimal amount of external components necessary
n Quiet fade-in/out mute mode
n Non-Isolated 15-lead TO-220 package
Applications
n High-end stereo TVs
n Component stereo
n Compact stereo
Connection Diagram
Plastic Package
™
Audio Power Amplifier Series
DS100928-2
Top View
Non-Isolated Package
Order Number LM4766T
See NS Package Number TA15A
DS100928-1
FIGURE 1. Typical Audio Amplifier Application Circuit
Note: Numbers in parentheses represent pinout for amplifier B.
*
Optional component dependent upon specific design requirements.
SPiKe™Protection and Overture™are trademarks of National Semiconductor Corporation.
© 1999 National Semiconductor Corporation DS100928 www.national.com
Absolute Maximum Ratings (Notes 4, 5)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |V
(No Input) 78V
Supply Voltage |V
(with Input) 74V
Common Mode Input Voltage (V
Differential Input Voltage 60V
Output Current Internally Limited
Power Dissipation (Note 6) 62.5W
ESD Susceptability (Note 7) 3000V
|+|VEE|
CC
|+|VEE|
CC
or VEE) and
CC
|+|VEE| ≤ 60V
|V
CC
Junction Temperature (Note 8) 150˚C
Thermal Resistance
Non-Isolated T-Package
θ
JC
1˚C/W
Soldering Information
T Package 260˚C
Storage Temperature −40˚C to +150˚C
Operating Ratings (Notes 4, 5)
Temperature Range
≤ TA≤ T
T
MIN
Supply Voltage |V
MAX
|+|VEE| (Note 1) 20V to 60V
CC
−20˚C ≤ TA≤ +85˚C
Electrical Characteristics (Notes 4, 5)
The following specifications apply for V
Limits apply for T
=
25˚C.
A
CC
=
+30V, V
Symbol Parameter Conditions LM4766 Units
| + Power Supply Voltage GND − VEE≥ 9V 18 20 V (min)
|V
CC
|V
| (Note 11) 60 V (max)
EE
P
O
Output Power THD + N=0.1%(max),
(Note 3) (Continuous Average) f=1 kHz, f=20 kHz 40 30 W/ch (min)
THD + N Total Harmonic Distortion 30 W/ch, R
Plus Noise A
X
SR
talk
Channel Separation f=1 kHz, V
Slew Rate V
(Note 3)
I
total
Total Quiescent Power Both Amplifiers V
(Note 2) Supply Current V
V
OS
(Note 2)
I
B
I
OS
I
O
V
OD
Input Offset Voltage V
Input Bias Current V
Input Offset Current V
Output Current Limit |VCC|=|VEE|=10V, t
Output Dropout Voltage |VCC–VO|, V
(Note 2) (Note 12) |V
PSRR Power Supply Rejection Ratio V
(Note 2) V
CMRR Common Mode Rejection Ratio V
(Note 2) V
A
VOL
(Note 2)
Open Loop Voltage Gain R
GBWP Gain Bandwidth Product f
e
IN
Input Noise IHF—A Weighting Filter 2.0 8 µV (max)
(Note 3) R
=
−30V, I
EE
L
=
26 dB
V
O
=
1.2 Vrms, t
IN
=
=
0V, I
O
O
=
0V, I
CM
CM
CM
V
O
CC
CM
V
CC
V
CM
CC
CM
L
O
IN
O–VEE
O
=
0V, I
O
=
0V, I
O
=
0V
CC
|, V
EE
=
30V to 10V, V
=
0V, I
O
=
30V, V
=
0V, I
O
=
50V to 10V, V
=
20V to −20V, I
=
2kΩ,∆V
=
100 kHz, V
=
600Ω (Input Referred)
MUTE
=
−0.5 mA with R
=
8Ω unless otherwise specified.
L
Typical Limit
(Note 9) (Note 10)
=
8Ω,20Hz≤f≤20 kHz 0.06
=
10.9 Vrms 60 dB
=
2 ns 9 5 V/µs (min)
rise
=
0V, 48 100 mA (max)
CM
0mA
=
0 mA 1 10 mV (max)
=
0 mA 0.2 1 µA (max)
=
0 mA 0.01 0.2 µA (max)
=
10 ms, 4 3 Apk (min)
ON
=
EE
=
=
=
0mA
0mA
O
IN
=
20V, I
−20V, I
=
=
+100 mA 1.5 4 V (max)
O
=
−100 mA 2.5 4 V (max)
O
=
−30V, 125 85 dB (min)
EE
−30V to −10V 110 85 dB (min)
=
−10V to −50V, 110 75 dB (min)
EE
=
0mA
O
40V 115 80 dB (min)
=
50 mVrms 8 2 MHz (min)
(Limits)
%
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Electrical Characteristics (Notes 4, 5) (Continued)
The following specifications apply for V
Limits apply for T
=
25˚C.
A
CC
=
+30V, V
Symbol Parameter Conditions LM4766 Units
SNR Signal-to-Noise Ratio P
A
M
Note 1: Operation is guaranteed up to 60V, however,distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations are not taken into
account. Refer to the Application Information section for a complete explanation.
Note 2: DC Electrical Test; Refer to Test Circuit
Note 3: AC Electrical Test; Refer to Test Circuit
Note 4: All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
Note 5: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is func-
tional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular testconditionswhichguarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is
given, however, the typical value is a good indication of device performance.
Note 6: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of
θ
JC
Note 7: Human body model, 100 pF discharged through a 1.5 kΩ resistor.
Note 8: The operating junction temperature maximum is 150˚C, however, the instantaneous Safe Operating Area temperature is 250˚C.
Note 9: Typicals are measured at 25˚C and represent the parametric norm.
Note 10: Limits are guarantees that all parts are tested in production to meet the stated values.
Note 11: V
ferential between V
Note 12: The output dropout voltage, V
formance Characteristics section.
Mute Attenuation Pin 6,11 at 2.5V 115 80 dB (min)
#
1.
#
2.
=
1˚C/W (junction to case) for the T package. Refer to the section Determining the Correct Heat Sink in the Application Information section.
must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In addition, the voltage dif-
EE
and VEEmust be greater than 14V.
CC
, is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltagegraph in the Typical Per-
OD
=
−30V, I
EE
MUTE
=
−0.5 mA with R
=
8Ω unless otherwise specified.
L
Typical Limit
(Note 9) (Note 10)
=
1W, A — Weighted, 98 dB
O
Measured at 1 kHz, R
=
P
25W, A — Weighted 112 dB
O
Measured at 1 kHz, R
=
25Ω
S
=
25Ω
S
(Limits)
Test Circuit#1 (Note 2) (DC Electrical Test Circuit)
DS100928-3
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Test Circuit#2 (Note 3) (AC Electrical Test Circuit)
Bridged Amplifier Application Circuit
DS100928-4
FIGURE 2. Bridged Amplifier Application Circuit
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DS100928-5
Single Supply Application Circuit
FIGURE 3. Single Supply Amplifier Application Circuit
Note:*Optional components dependent upon specific design requirements.
DS100928-6
Auxiliary Amplifier Application Circuit
FIGURE 4. Special Audio Amplifier Application Circuit
DS100928-7
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Equivalent Schematic (excluding active protection circuitry)
LM4766 (One Channel Only)
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DS100928-8
External Components Description
Components Functional Description
1R
B
2R
i
3R
f
4C
i
(Note 13)
5C
S
6R
V
(Note 13)
7R
IN
(Note 13)
8C
IN
(Note 13)
9R
SN
(Note 13)
10 C
SN
(Note 13)
11 L (Note 13) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce
12 R (Note 13)
13 R
A
14 C
A
15 R
INP
(Note 13)
16 R
BI
17 R
E
18 R
M
19 C
M
20 S
1
Note 13: Optional components dependent upon specific design requirements.
Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the
load upon power down of the system due to the low input impedance of the circuitry when the
undervoltage circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.
Inverting input resistance to provide AC gain in conjunction with Rf.
Feedback resistance to provide AC gain in conjunction with Ri.
Feedback capacitor which ensures unity gain at DC. Also creates a highpass filter with R
).
1/(2πR
iCi
=
at f
i
C
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for
proper placement and selection of bypass capacitors.
Acts as a volume control by setting the input voltage level.
Sets the amplifier’s input terminals DC bias point when C
create a highpass filter at f
=
1/(2πR
C
INCIN
). Refer to
is present in the circuit. Also works with CINto
IN
Figure 4
.
Input capacitor which blocks the input signal’s DC offsets from being passed onto the amplifier’s inputs.
Works with C
Works with R
The pole is set at f
to stabilize the output stage by creating a pole that reduces high frequency instabilities.
SN
to stabilize the output stage by creating a pole that reduces high frequency instabilities.
SN
=
1/(2πR
C
SNCSN
). Refer to
Figure 4
.
the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and
Figure 4
pass audio signals to the load. Refer to
.
Provides DC voltage biasing for the transistor Q1 in single supply operation.
Provides bias filtering for single supply operation.
Limits the voltage difference between the amplifier’s inputs for single supply operation. Refer to the Clicks
and Pops application section for a more detailed explanation of the function of R
.
INP
Provides input bias current for single supply operation. Refer to the Clicks and Pops application section
for a more detailed explanation of the function of R
.
BI
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the
half-supply point along with C
.
A
Mute resistance set up to allow 0.5 mA to be drawn from pin 6 or 11 to turn the muting function off.
→
is calculated using: RM≤ (|VEE| − 2.6V)/l where l ≥ 0.5 mA. Refer to the Mute Attenuation vs Mute
R
M
Current curves in the Typical Performance Characteristics section.
Mute capacitance set up to create a large time constant for turn-on and turn-off muting.
Mute switch that mutes the music going into the amplifier when opened.
Typical Performance Characteristics
THD+NvsFrequency
DS100928-55
THD+NvsFrequency
DS100928-56
THD+NvsOutput Power
DS100928-58
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Typical Performance Characteristics (Continued)
THD+NvsOutput Power
Channel Separation vs
Frequency
Output Power vs
Supply Voltage
DS100928-57
DS100928-10
THD+NvsDistribution
Clipping Voltage vs
Supply Voltage
Power Dissipation vs
Output Power
DS100928-72
DS100928-68
THD+NvsDistribution
DS100928-73
Output Power vs
Load Resustance
DS100928-74
Power Dissipation vs
Output Power
DS100928-78
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DS100928-76
DS100928-77
Typical Performance Characteristics (Continued)
Max Heatsink Thermal Resistance (˚C/W)
at the Specified Ambient Temperature (˚C)
Note: The maximum heatsink thermal resistance values, θ
thermal compound.
, in the table above were calculated using a θ
SA
DS100928-75
=
0.2˚C/W due to
CS
Safe Area
Pulse Power Limit
DS100928-59
DS100928-64
SPiKe Protection
Response
Pulse Response
DS100928-60
DS100928-66
Pulse Power Limit
DS100928-63
Large Signal Response
DS100928-87
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Typical Performance Characteristics (Continued)
Power Supply
Rejection Ratio
Supply Current vs
Case Temperature
DS100928-65
Mute Attenuation vs
Mute Current (per Amplifier)
DS100928-88
Common-Mode
Rejection Ratio
Input Bias Current vs
Case Temperature
DS100928-89
DS100928-67
Open Loop
Frequency Response
DS100928-90
Mute Attenuation vs
Mute Current (per Amplifier)
DS100928-85
DS100928-86
Application Information
MUTE MODE
The muting function of the LM4766 allows the user to mute
the music going into the amplifier by drawing more than
0.5 mA out of each mute pin on the device. This is accomplished as shown in the TypicalApplication Circuit where the
resistor R
voltage and is used in conjunction with a switch. The switch
when opened cuts off the current flow from pin 6 or 11 to
−V
MuteAttenuation vs Mute Current curves in the Typical Per-
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is chosen with reference to your negative supply
M
, thus placing the LM4766 into mute mode. Refer to the
EE
formance Characteristics section for values of attenuation
per current out of pins 6 or 11. The resistance R
lated by the following equation:
R
≤ (|−VEE| − 2.6V)/Ipin6
M
is calcu-
M
where Ipin6 = Ipin11 ≥ 0.5 mA.
Both pins 6 and 11 can be tied together so that only one re-
sistor and capacitor are required for the mute function. The
mute resistance must be chosen such that greater than 1 mA
is pulled through the resistor R
so that each amplifier is fully
M
Application Information (Continued)
pulled out of mute mode. Takinginto account supply line fluctuations, it is a good idea to pull out 1 mA per mute pin or
2 mA total if both pins are tied together.
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection circuitry allows the power supplies and their corresponding capacitors to come up close to their full values before turning
on the LM4766 such that no DC output spikes occur. Upon
turn-off, the output of the LM4766 is brought to ground before the power supplies such that no transients occur at
power-down.
OVER-VOLTAGE PROTECTION
The LM4766 contains over-voltage protection circuitry that
limits the output current to approximately 4.0 Apk while also
providing voltage clamping, though not through internal
clamping diodes. The clamping effect is quite the same,
however, the output transistors are designed to work alternately by sinking large current spikes.
SPiKe PROTECTION
The LM4766 is protected from instantaneous peaktemperature stressing of the power transistor array.The Safe
Operating graph in the Typical Performance Characteris-
tics section shows the area of device operation where
SPiKe Protection Circuitry is not enabled. The waveform to
the right of the SOA graph exemplifies how the dynamic protection will cause waveform distortion when enabled. Please
refer to AN-898 for more detailed information.
THERMAL PROTECTION
The LM4766 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die reaches 165˚C, the LM4766 shuts
down. It starts operating again when the die temperature
drops to about 155˚C, but if the temperature again begins to
rise, shutdown will occur again at 165˚C. Therefore, the device is allowed to heat up to a relatively high temperature if
the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion between the thermal shutdown temperature limits of 165˚C and
155˚C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen such that
thermal shutdown will not be reached during normal operation. Using the best heat sink possible within the cost and
space constraints of the system will improve the long-term
reliability of any power semiconductor device, as discussed
in the Determining the Correct Heat Sink Section.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understanding if optimum power output is to be obtained. An incorrect
maximum power dissipation calculation may result in inadequate heat sinking causing thermal shutdown and thus limiting the output power.
Equation (1)
sipation point of each amplifier where V
voltage.
exemplifies the theoretical maximum power dis-
is the total supply
CC
2
=
P
DMAX
/2π2R
V
CC
L
(1)
Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calculated. The package dissipation is twice the number which results from
Equation (1)
since there are two amplifiers in each
LM4766. Refer to the graphs of Power Dissipation versus
Output Power in the Typical Performance Characteristics
section which show the actual full range of power dissipation
not just the maximum theoretical point that results from
Equation (1)
.
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry does not operate under
normal circumstances.
The thermal resistance from the die (junction) to the outside
air (ambient) is a combination of three thermal resistances,
θ
, θCS, and θSA. In addition, the thermal resistance, θ
JC
(junction to case), of the LM4766T is 1˚C/W. Using Thermalloy Thermacote thermal compound, the thermal resistance,
θ
(case to sink), is about 0.2˚C/W. Since convection heat
CS
flow (power dissipation) is analogous to current flow, thermal
resistance is analogous to electrical resistance, and temperature drops are analogous to voltage drops, the power
dissipation out of the LM4766 is equal to the following:
=
where T
ture and θ
JMAX
JA
P
=
150˚C, T
=
θ
JC
(T
DMAX
AMB
+ θCS+ θSA.
JMAX−TAMB
is the system ambient tempera-
)/θ
DS100928-52
JA
(2)
Once the maximum package power dissipation has been
calculated using
tance, θ
be calculated. This calculation is made using
Equation (1)
, (heat sink to ambient) in ˚C/W for a heat sink can
SA
which is derived by solving for θSAin
=
θ
[(T
SA
JMAX−TAMB
Again it must be noted that the value of θ
upon the system designer’s amplifier requirements. If the
, the maximum thermal resis-
Equation (3)
)−P
DMAX(θJC+θCS
Equation (2)
.
)]/P
DMAX
is dependent
SA
(3)
ambient temperature that the audio amplifier is to be working
under is higher than 25˚C, then the thermal resistance for the
heat sink, given all other things are equal, will need to be
smaller.
SUPPLY BYPASSING
The LM4766 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM4766 should have its supply leads bypassed with
low-inductance capacitors having short leads that are located close to the package terminals. Inadequate power
supply bypassing will manifest itself by a low frequency oscillation known as “motorboating” or by high frequency instabilities. These instabilities can be eliminated through multiple
bypassing utilizing a large tantalum or electrolytic capacitor
(10 µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1 µF) to prevent
any high frequency feedback through the power supply lines.
www.national.com11
JC
Application Information (Continued)
If adequate bypassing is not provided, the current in the supply leads which is a rectified component of the load current
may be fed back into internal circuitry. This signal causes
distortion at high frequencies requiring that the supplies be
bypassed at the package terminals with an electrolytic capacitor of 470 µF or more.
BRIDGED AMPLIFIER APPLICATION
The LM4766 has two operational amplifiers internally, allowing for a few different amplifier configurations. One of these
configurations is referred to as “bridged mode” and involves
driving the load differentially through the LM4766’s outputs.
This configuration is shown in
eration is different from the classical single-ended amplifier
configuration where one side of its load is connected to
ground.
A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Consequently, theoretically four times the output
power is possible as compared to a single-ended amplifier
under the same conditions. This increase in attainable output
power assumes that the amplifier is not current limited or
clipped.
A direct consequence of the increased power delivered to
the load by a bridge amplifier is an increase in internal power
dissipation. For each operational amplifier in a bridge configuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Thus, for an
audio power amplifier such as the LM4766, which has two
operational amplifiers in one package, the package dissipation will increase by a factor of four. To calculate the
LM4766’s maximum power dissipation point for a bridged
load, multiply
This value of P
heat sink for a bridged amplifier application. Since the inter-
Equation (1)
can be used to calculate the correct size
DMAX
nal dissipation for a given power supply and load is increased by using bridged-mode, the heatsink’s θ
to decrease accordingly as shown by
the section, Determining the Correct Heat Sink, for a more
detailed discussion of proper heat sinking for a given application.
SINGLE-SUPPLYAMPLIFIER APPLICATION
The typical application of the LM4766 is a split supply amplifier. But as shown in
Figure 3
in a single power supply configuration. This involves using
some external components to create a half-supply bias
which is used as the reference for the inputs and outputs.
Thus, the signal will swing around half-supply much like it
swings around ground in a split-supply application. Along
with proper circuit biasing, a few other considerations must
be accounted for to take advantage of all of the LM4766
functions, like the mute function.
CLICKS AND POPS
In the typical application of the LM4766 as a split-supply audio power amplifier, the IC exhibits excellent “click” and “pop”
performance when utilizing the mute and standby modes. In
addition, the device employs Under-Voltage Protection,
which eliminates unwanted power-up and power-down transients. The basis for these functions are a stable and constant half-supply potential. In a split-supply application,
ground is the stable half-supply potential. But in a
Figure 2
. Bridged mode op-
by a factor of four.
will have
Equation (3)
SA
. Refer to
, the LM4766 can also be used
single-supply application, the half-supply needs to charge up
just like the supply rail, V
a clickless and popless turn-on more challenging. Any un-
. This makes the task of attaining
CC
even charging of the amplifier inputs will result in output
clicks and pops due to the differential input topology of the
LM4766.
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4766. In
serves to keep the inputs at the same potential by limiting the
Figure 3
, the resistor R
INP
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal performance.
As shown in
Figure 3
, the resistors labeled RBIhelp bias up
the LM4766 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R
sistors bring up the inputs at the same rate resulting in a pop-
, namely 10 kΩ and 200 kΩ. These re-
BI
less turn-on. Adjusting these resistors values slightly may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
AUDIO POWER AMPLlFIER DESIGN
Design a 30W/8Ω Audio Amplifier
Given:
Power Output 30 Wrms
Load Impedance 8Ω
Input Level 1 Vrms(max)
Input Impedance 47 kΩ
Bandwidth 20 Hz−20 kHz
±
0.25 dB
A designer must first determine the power supply requirements in terms of both voltage and current needed to obtain
the specified output power. V
Equation (4)
and I
OPEAK
from
can be determined from
OPEAK
Equation (5)
.
(4)
(5)
To determine the maximum supply voltage the following conditions must be considered. Add the dropout voltage to the
peak output swing V
of I
. The regulation of the supply determines the un-
OPEAK
loaded voltage which is usually about 15%higher. The sup-
, to get the supply rail at a current
OPEAK
ply voltage will also rise 10%during high line conditions.
Therefore the maximum supply voltage is obtained from the
following equation.
Max supplies ≈
±
(V
OPEAK+VOD
) (1 + regulation) (1.1)
For 30W of output power into an 8Ω load, the required
V
is 21.91V. A minimum supply rail of 25.4V results
OPEAK
from adding V
supplies are
Equation (5)
into an 8Ω load the I
2.74 Apk or 5.48 Apk. At this point it is a good idea to check
and VOD. With regulation, the maximum
OPEAK
±
32V and the required I
. It should be noted that for a dual 30W amplifier
drawn from the supplies is twice
OPEAK
OPEAK
is 2.74A from
the Power Output vs Supply Voltage to ensure that the required output power is obtainable from the device while
maintaining low THD+N. In addition, the designer should
www.national.com 12
Application Information (Continued)
verify that with the required power supply voltage and load
impedance, that the required heatsink value θ
given system cost and size constraints. Once the heatsink
issues have been addressed, the required gain can be determined from
From
Equation (6)
Equation (6)
.
, the minimum AVis: AV≥ 15.5.
By selecting a gain of 21, and with a feedback resistor,R
20 kΩ, the value of R
Thus with R
Since the desired input impedance was 47 kΩ, a value of
i
47 kΩ was selected for R
follows from
i
=
R
=
i
1kΩa non-inverting gain of 21 will result.
IN
Equation (7)
− 1) (7)
R
f(AV
. The final design step is to ad-
is feasible
SA
.
(6)
f
dress the bandwidth requirements which must be stated as a
pair of −3 dB frequency points. Five times away from a −3 dB
point is 0.17 dB down from passband response which is better than the required
±
0.25 dB specified. This fact results in
a low and high frequency pole of 4 Hz and 100 kHz respectively.As stated in the External Components section, R
conjunction with C
≥ 1/(2π*1kΩ*4 Hz)=39.8 µF; use 39 µF.
C
i
create a high-pass filter.
i
The high frequency pole is determined by the product of the
desired high frequency pole, f
=
A
21 and f
=
V
which is less than the guaranteed minimum GBWP of the
=
100 kHz, the resulting GBWP is 2.1 MHz,
H
, and the gain, AV. With a
H
LM4766 of 8 MHz. This will ensure that the high frequency
response of the amplifier will be no worse than 0.17 dB down
at 20 kHz which is well within the bandwidth requirements of
the design.
in
i
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Audio Power Amplifier Series
™
Dual 40W Audio Power Amplifier with Mute
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