Datasheet LM3478MWC, LM3478MMX, LM3478MM, LM3478MDC, LM3478EVAL Datasheet (NSC)

LM3478
High Efficiency Low-Side N-Channel Controller for
Switching Regulator

General Description

The LM3478 is a versatile Low-Side N-FET switching regu­lator controller. It is suitable for use in topologies requiring
low side FET, such as boost, flyback, SEPIC, etc. Moreover,
the LM3478 can be operated at extremely high switching
frequency in order to reduce the overall solution size. The
switching frequency of LM3478 can be adjusted to any value
between 100kHz and 1MHz by using a single external resis­tor. Current mode control provides superior bandwidth and
transient response, besides cycle-by-cycle current limiting.
Output current can be programmed with a single external
resistor.

The LM3478 has built in features such as thermal shutdown,
short-circuit protection, over voltage protection, etc. Power
saving shutdown mode reduces the total supply current to
5µA and allows power supply sequencing. Internal soft-start
limits the inrush current at start-up.

Key Specifications

n Wide supply voltage range of 2.97V to 40V
n 100kHz to 1MHz Adjustable clock frequency

n

±

2.5% (over temperature) internal reference

n 10µA shutdown current (over temperature)

Features

n 8-lead Mini-SO8 (MSOP-8) package
n Internal push-pull driver with 1A peak current capability
n Current limit and thermal shutdown
n Frequency compensation optimized with a capacitor and

a resistor

n Internal softstart
n Current Mode Operation
n Undervoltage Lockout with hysteresis

Applications

n Distributed Power Systems
n Battery Chargers
n Offline Power Supplies
n Telecom Power Supplies
n Automotive Power Systems

Typical Application Circuit

10135501

Typical High Efficiency Step-Up (Boost) Converter

May 2003

LM3478 High Efficiency Low-Side N-Channel Controller for Switching Regulator

© 2003 National Semiconductor Corporation DS101355 www.national.com

Connection Diagram

10135502

8 Lead Mini SO8 Package (MSOP-8 Package)

Package Marking and Ordering Information

Order Number Package Type Package Marking Supplied As:

LM3478MM MSOP-8 S14B 1000 units on Tape and

Reel

LM3478MMX MSOP-8 S14B 3500 units on Tape and

Reel

Pin Description

Pin Name Pin Number Description

I

SEN

1 Current sense input pin. Voltage generated across an external

sense resistor is fed into this pin.

COMP 2 Compensation pin. A resistor, capacitor combination connected to

this pin provides compensation for the control loop.

FB 3 Feedback pin. The output voltage should be adjusted using a

resistor divider to provide 1.26V at this pin.

AGND 4 Analog ground pin.

PGND 5 Power ground pin.

DR 6 Drive pin of the IC. The gate of the external MOSFET should be

connected to this pin.

FA/SD 7 Frequency adjust and Shutdown pin. A resistor connected to this

pin sets the oscillator frequency. A high level on this pin for ≥ 30µs
will turn the device off. The device will then draw less than 10µA
from the supply.

V

IN

8 Power Supply Input pin.

LM3478

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Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.

Input Voltage 45V

FB Pin Voltage -0.4V

<

V

FB

<

7V

FA/SD Pin Voltage

-0.4V<V

FA/SD

<

7V

Peak Driver Output Current (

<

10µs) 1.0A

Power Dissipation Internally Limited

Storage Temperature Range −65˚C to +150˚C

Junction Temperature +150˚C

ESD Susceptibilty

Human Body Model (Note 2) 2kV

Lead Temperature

MM Package
Vapor Phase (60 sec.)
Infared (15 sec.)

215˚C
220˚C

DR Pin Voltage −0.4V ≤ VDR ≤ 8V

I

SEN

Pin Voltage 500mV

Operating Ratings (Note 1)

Supply Voltage 2.97V ≤ V

IN

≤ 40V

Junction
Temperature Range −40˚C ≤ TJ≤ +125˚C

Switching Frequency 100kHz ≤ F

SW

≤ 1MHz

Electrical Characteristics

Specifications in Standard type face are for TJ= 25˚C, and in bold type face apply over the full Operating Temperature
Range. Unless otherwise specified, V

IN

= 12V, RFA= 40kΩ

Symbol Parameter Conditions Typical Limit Units

V

FB

Feedback Voltage V

COMP

= 1.4V,

2.97 ≤ V

IN

≤ 40V

1.26

1.2416/1.228

1.2843/1.292

V

V(min)

V(max)

∆V

LINE

Feedback Voltage
Line Regulation

2.97 ≤ VIN≤ 40V 0.001 %/V

∆V

LOAD

Output Voltage Load
Regulation

I

EAO

Source/Sink

±

0.5 %/V (max)

V

UVLO

Input Undervoltage
Lock-out

2.85

2.97

V

V(max)

V

UV(HYS)

Input Undervoltage
Lock-out Hysteresis

170

130
210

mV

mV (min)

mV (max)

F

nom

Nominal Switching
Frequency

RFA= 40KΩ 400

360
430

kHz

kHz(min)

kHz(max)

R

DS1 (ON)

Driver Switch On
Resistance (top)

IDR= 0.2A, VIN=5V 16 Ω

R

DS2 (ON)

Driver Switch On
Resistance (bottom)

IDR= 0.2A 4.5 Ω

V

DR (max)

Maximum Drive
Voltage Swing(Note 6)

V

IN

<

7.2V V

IN

V

V

IN

≥ 7.2V 7.2

D

max

Maximum Duty
Cycle(Note 7)

100 %

T

min

(on) Minimum On Time 325

210
600

nsec

nsec(min)

nsec(max)

I

SUPPLY

Supply Current
(switching)

(Note 9)

2.0

3.0

mA

mA (max)

I

Q

Quiescent Current in
Shutdown Mode

V

FA/SYNC/SD

= 5V (Note

10), V

IN

=5V

5

10

µA

µA (max)

V

SENSE

Current Sense
Threshold Voltage

VIN= 5V 165

140/ 135
195/ 200

mV

mV (min)

mV (max)

LM3478

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Electrical Characteristics (Continued)

Specifications in Standard type face are for TJ= 25˚C, and in bold type face apply over the full Operating Temperature
Range. Unless otherwise specified, V

IN

= 12V, RFA= 40kΩ

Symbol Parameter Conditions Typical Limit Units

V

SC

Short-Circuit Current
Limit Sense Voltage

VIN= 5V 325

235
395

mV

mV (min)

mV (max)

V

SL

Internal Compensation
Ramp Voltage

VIN=5V 92

52

132

mV

mV(min)

mV(max)

V

OVP

Output Over-voltage
Protection (with
respect to feedback
voltage) (Note 8)

V

COMP

= 1.4V 50

32/ 25
78/ 85

mV

mV(min)

mV(max)

V

OVP(HYS)

Output Over-Voltage
Protection
Hysteresis(Note 8)

V

COMP

= 1.4V 60

20

110

mV

mV(min)

mV(max)

Gm Error Ampifier

Transconductance

V

COMP

= 1.4V

I

EAO

= 100µA

(Source/Sink)

800

600/ 365

1000/ 1265

µmho

µmho (min)

µmho (max)

A

VOL

Error Amplifier Voltage
Gain

V

COMP

= 1.4V

I

EAO

= 100µA

(Source/Sink)

38

26
44

V/V

V/V (min)

V/V (max)

I

EAO

Error Amplifier Output
Current (Source/ Sink)

Source, V

COMP

= 1.4V,

V

FB

=0V

110

80/ 50

140/ 180

µA

µA (min)

µA (max)

Sink, V

COMP

= 1.4V, V

FB

= 1.4V

−140

−100/ −85

−180/ −185

µA

µA (min)

µA (max)

V

EAO

Error Amplifier Output
Voltage Swing

Upper Limit
V

FB

=0V

COMP Pin = Floating

2.2

1.8

2.4

V

V(min)

V(max)

Lower Limit
V

FB

= 1.4V

0.56

0.2

1.0

V

V(min)

V(max)

T

SS

Internal Soft-Start
Delay

VFB= 1.2V, V

COMP

=

Floating

4 msec

T

r

Drive Pin Rise Time Cgs = 3000pf, VDR=0to

3V

25 ns

T

f

Drive Pin Fall Time Cgs = 3000pf, VDR=0to

3V

25 ns

VSD Shutdown threshold

(Note 5)

Output = High 1.27

1.35

V

V (max)

Output = Low 0.65

0.35

V

V (min)

I

SD

Shutdown Pin Current VSD=5V −1 µA

V

SD

=0V +1

TSD Thermal Shutdown 165 ˚C

T

sh

Thermal Shutdown
Hysteresis

10 ˚C

θ

JA

Thermal Resistance MM Package 200 ˚C/W

LM3478

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Electrical Characteristics (Continued)

Note 1: Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which operation of the device

is intended to be functional. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 3: All limits are guaranteed at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature limits are 100%

tested. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. All limits are used to calculate
Average Outgoing Quality Level (AOQL).

Note 4: Typical numbers are at 25˚C and represent the most likely norm.

Note 5: The FA/SD pin should be pulled to V

IN

through a resistor to turn the regulator off. The voltage on the FA/SD pin must be above the maximum limit for Output

= High to keep the regulator off and must be below the limit for Output = Low to keep the regulator on.

Note 6: The voltage on the drive pin, V

DR

is equal to the input voltage when input voltage is less than 7.2V. VDRis equal to 7.2V when the input voltage is greater

than or equal to 7.2V.

Note 7: The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle operation.

Note 8: The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the feedback voltage. The

overvoltage protection threshold is given by adding the feedback voltage, V

FB

to the over-voltage protection specification.

Note 9: For this test, the FA/SD pin is pulled to ground using a 40K resistor.

Note 10: For this test, the FA/SD pin is pulled to 5V using a 40K resistor.

Typical Performance Characteristics Unless otherwise specified, V

IN

= 12V, TJ= 25˚C.

I

Q

vs Input Voltage (Shutdown) I

Supply

vs Input Voltage (Non-Switching)

10135503

10135534

I

Supply

vs VIN(Switching) Switching Frequency vs R

FA

10135535

10135504

LM3478

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Typical Performance Characteristics Unless otherwise specified, V

IN

= 12V, TJ= 25˚C. (Continued)

Frequency vs Temperature Drive Voltage vs Input Voltage

10135554

10135505

Current Sense Threshold vs Input Voltage COMP Pin Voltage vs Load Current

10135545

10135562

Efficiency vs Load Current (3.3V In and 12V Out) Efficiency vs Load Current (5V In and 12V Out)

10135559 10135558

LM3478

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Typical Performance Characteristics Unless otherwise specified, V

IN

= 12V, TJ= 25˚C. (Continued)

Efficiency vs Load Current (9V In and 12V Out) Efficiency vs Load Current (3.3V In and 5V Out

10135560

10135553

Error Amplifier Gain Error Amplifier Phase

10135555 10135556

COMP Pin Source Current vs Temperature Short Circuit Sense Voltage vs Input Voltage

10135536

10135557

LM3478

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Typical Performance Characteristics Unless otherwise specified, V

IN

= 12V, TJ= 25˚C. (Continued)

Compensation Ramp vs Compensation Resistor Shutdown Threshold Hysteresis vs Temperature

10135551

10135546

Duty Cycle vs Current Sense Voltage

10135552

LM3478

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Functional Block Diagram

10135506

Functional Description

The LM3478 uses a fixed frequency, Pulse Width Modulated
(PWM), current mode control architecture. In a typical appli­cation circuit, the peak current through the external MOS­FET is sensed through an external sense resistor. The volt­age across this resistor is fed into the I

SEN

pin. This voltage
is then level shifted and fed into the positive input of the
PWM comparator. The output voltage is also sensed through
an external feedback resistor divider network and fed into
the error amplifier negative input (feedback pin, FB). The
output of the error amplifier (COMP pin) is added to the slope
compensation ramp and fed into the negative input of the
PWM comparator.

At the start of any switching cycle, the oscillator sets the RS
latch using the SET/Blank-out and switch logic blocks. This
forces a high signal on the DR pin (gate of the external
MOSFET) and the external MOSFET turns on. When the
voltage on the positive input of the PWM comparator ex­ceeds the negative input, the RS latch is reset and the
external MOSFET turns off.

The voltage sensed across the sense resistor generally
contains spurious noise spikes, as shown in Figure 1. These
spikes can force the PWM comparator to reset the RS latch
prematurely. To prevent these spikes from resetting the
latch, a blank-out circuit inside the IC prevents the PWM
comparator from resetting the latch for a short duration after
the latch is set. This duration is about 150ns and is called the
blank-out time.

Under extremely light load or no-load conditions, the energy
delivered to the output capacitor when the external MOSFET
is on during the blank-out time is more than what is delivered
to the load. An over-voltage comparator inside the LM3478
prevents the output voltage from rising under these condi­tions. The over-voltage comparator senses the feedback (FB
pin) voltage and resets the RS latch under these conditions.
The latch remains in reset state till the output decays to the
nominal value.

LM3478

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Functional Description (Continued)

SLOPE COMPENSATION RAMP

The LM3478 uses a current mode control scheme. The main
advantages of current mode control are inherent cycle-by­cycle current limit for the switch, and simpler control loop
characteristics. It is also easy to parallel power stages using
current mode control since current sharing is automatic.

Current mode control has an inherent instability for duty
cycles greater than 50%, as shown in Figure 2.InFigure 2,
a small increase in the load current causes the switch cur­rent to increase by ∆I

O

. The effect of this load change, ∆I1,is

:

From the above equation, when D>0.5, ∆I1will be greater
than ∆I

O

. In other words, the disturbance is divergent. So a
very small perturbation in the load will cause the disturbance
to increase.

To prevent the sub-harmonic oscillations, a compensation
ramp is added to the control signal, as shown in Figure 3.

With the compensation ramp,

10135507

FIGURE 1. Basic Operation of the PWM comparator

10135509

FIGURE 2. Sub-Harmonic Oscillation for D>0.5

LM3478

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Functional Description (Continued)

The compensation ramp has been added internally in
LM3478. The slope of this compensation ramp has been
selected to satisfy most of the applications. The slope of the
internal compensation ramp depends on the frequency. This
slope can be calculated using the formula:

M

C=VSL.FS

Volts/second

In the above equation, V

SL

is the amplitude of the internal

compensation ramp. Limits for V

SL

have been specified in

the electrical characteristics.
In order to provide the user additional flexibility, a patented

scheme has been implemented inside the IC to increase the
slope of the compensation ramp externally, if the need
arises. Adding a single external resistor, R

SL

(as shown in

Figure 4) increases the slope of the compensation ramp, M

C

by :

In this equation, ∆VSLis equal to 40.10-6RSL. Hence,

∆VSLversus RSLhas been plotted in Figure 5 for different
frequencies.

10135511

FIGURE 3. Compensation Ramp Avoids Sub-Harmonic Oscillation

LM3478

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Functional Description (Continued)

FREQUENCY ADJUST/SHUTDOWN

The switching frequency of LM3478 can be adjusted be­tween 100kHz and 1MHz using a single external resistor.
This resistor must be connected between FA/SD pin and
ground, as shown in Figure 6. Please refer to the typical
performance characteristics to determine the value of the
resistor required for a desired switching frequency.

The FA/SD pin also functions as a shutdown pin. If a high
signal (refer to the electrical characteristics for definition of
high signal) appears on the FA/SD pin, the LM3478 stops

switching and goes into a low current mode. The total supply
current of the IC reduces to less than 10 µA under these
conditions.

Figure 7 shows implementation of shutdown function when
operating in frequency adjust mode. In frequency adjust
mode, connecting the FA/SD pin to ground forces the clock
to run at a certain frequency. Pulling this pin high shuts down
the IC. In frequency adjust mode, a high signal for more than
30µs shuts down the IC.

10135513

FIGURE 4. Increasing the Slope of the Compensation Ramp

10135551

FIGURE 5. ∆VSLvs R

SL

LM3478

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Functional Description (Continued)

SHORT-CIRCUIT PROTECTION

When the voltage across the sense resistor (measured on
I

SEN

Pin) exceeds 350mV, short-circuit current limit gets
activated. A comparator inside LM3478 reduces the switch­ing frequency by a factor of 5 and maintains this condition till
the short is removed.

Typical Applications

The LM3478 may be operated in either continuous or dis­continuous conduction mode. The following applications are
designed for continuous conduction operation. This mode of
operation has higher efficiency and lower EMI characteristics
than the discontinuous mode.

BOOST CONVERTER

The most common topology for LM3478 is the boost or
step-up topology. The boost converter converts a low input
voltage into a higher output voltage. The basic configuration
for a boost regulator is shown in Figure 8. In continuous
conduction mode (when the inductor current never reaches
zero at steady state), the boost regulator operates in two
cycles. In the first cycle of operation, MOSFET Q is turned
on and energy is stored in the inductor. During this cycle,

diode D is reverse biased and load current is supplied by the
output capacitor, C

OUT

.

In the second cycle, MOSFET Q is off and the diode is
forward biased. The energy stored in the inductor is trans­ferred to the load and output capacitor. The ratio of these two
cycles determines the output voltage. The output voltage is
defined as:

(ignoring the drop across the MOSFET and the diode), or

where D is the duty cycle of the switch, VDis the forward
voltage drop of the diode, and V

Q

is the drop across the
MOSFET when it is on. The following sections describe
selection of components for a boost converter

10135514

FIGURE 6. Frequency Adjust

10135516

FIGURE 7. Shutdown Operation in Frequency Adjust Mode

LM3478

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Typical Applications (Continued)

POWER INDUCTOR SELECTION

The inductor is one of the two energy storage elements in a
boost converter. Figure 9 shows how the inductor current
varies during a switching cycle. The current through an
inductor is quantified as:

10135522

FIGURE 8. Simplified Boost Converter Diagram (a) First cycle of operation. (b) Second cycle of operation

10135524

FIGURE 9. A. Inductor current B. Diode current

LM3478

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Typical Applications (Continued)

If V

L

(t) is constant, diL(t)/dt must be constant. Hence, for a
given input voltage and output voltage, the current in the
inductor changes at a constant rate.

The important quantities in determining a proper inductance
value are I

L

(the average inductor current) and ∆iL(the
inductor current ripple). If ∆iLis larger than IL, the inductor
current will drop to zero for a portion of the cycle and the
converter will operate in discontinuous conduction mode. If
∆i

L

is smaller than IL, the inductor current will stay above
zero and the converter will operate in continuous conduction
mode. All the analysis in this datasheet assumes operation
in continuous conduction mode. To operate in continuous
conduction mode, the following conditions must be met:

I

L

>

∆i

L

Choose the minimum I

OUT

to determine the minimum L. A

common choice is to set ∆i

L

to 30% of IL. Choosing an
appropriate core size for the inductor involves calculating the
average and peak currents expected through the inductor. In
a boost converter,

and I

L_peak=IL

(max) + ∆iL(max),

where

A core size with ratings higher than these values should be
chosen. If the core is not properly rated, saturation will
dramatically reduce overall efficiency.

The LM3478 can be set to switch at very high frequencies.
When the switching frequency is high, the converter can be
operated with very small inductor values. With a small induc­tor value, the peak inductor current can be higher than the
output currents, especially under light load conditions.

The LM3478 senses the peak current through the switch.
The peak current through the switch is the same as the peak
current calculated above.

PROGRAMMING THE OUTPUT VOLTAGE AND OUTPUT
CURRENT

The output voltage can be programmed using a resistor
divider between the output and the feedback pins, as shown
in Figure 10. The resistors are selected such that the voltage
at the feedback pin is 1.26V. R

F1

and RF2can be selected

using the equation,

A 100pF capacitor may be connected between the feedback
and ground pins to reduce noise.

The maximum amount of current that can be delivered at the
output can be controlled by the sense resistor, R

SEN

. Current
limit occurs when the voltage that is generated across the
sense resistor equals the current sense threshold voltage,
V

SENSE

. Limits for V

SENSE

have been specified in the elec-

trical characteristics. This can be expressed as:

I

sw(peak)

*

R

SEN=VSENSE

V

SENSE

represents the maximum value of the control signal
as shown in Figure 2. This control signal, however, is not a
constant value and changes over the course of a period as a
result of the internal compensation ramp (see Figure 3).
Therefore the current limit will also change as a result of the
internal compensation ramp. The actual command signal,
V

CS

, can be better expressed as a function of the sense

voltage and the internal compensation ramp:

V

CS=VSENSE

−(D*VSL)

V

SL

is defined as the internal compensation ramp voltage,

limits are specified in the electrical characteristics.
The peak current through the switch is equal to the peak

inductor current.

I

sw(peak)=IL

+ ∆i

L

Therefore for a boost converter

Combining the three equation yields an expression for R

SEN

LM3478

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Typical Applications (Continued)

CURRENT LIMIT WITH ADDITIONAL SLOPE
COMPENSATION

If an external slope compensation resistor is used (see
Figure 4) the internal control signal will be modified and this
will have an effect on the current limit. The control signal is
given by:

V

CS=VSENSE

−(D*VSL)

Where V

SENSE

and VSLare defined parameters in the elec-

trical characteristics section. If R

SL

is used, then this will add
to the existing slope compensation. The command voltage
will then be given by:

V

CS=VSENSE

−(D*(VSL+ ∆VSL))

Where ∆V

SL

is the additional slope compensation generated
and can be calculated by use of Figure 5 or is equal to 40 x
10

−6

*

RSL. This changes the equation for R

SEN

to:

Therefore RSLcan be used to provide an additional method
for setting the current limit.

POWER DIODE SELECTION

Observation of the boost converter circuit shows that the
average current through the diode is the average load cur­rent, and the peak current through the diode is the peak
current through the inductor. The diode should be rated to
handle more than its peak current. The peak diode current
can be calculated using the formula:

I

D(Peak)=IOUT

/ (1−D) + ∆I

L

In the above equation, I

OUT

is the output current and ∆ILhas

been defined in Figure 9.
The peak reverse voltage for boost converter is equal to the

regulator output voltage. The diode must be capable of
handling this voltage. To improve efficiency, a low forward
drop schottky diode is recommended.

POWER MOSFET SELECTION

The drive pin of LM3478 must be connected to the gate of an
external MOSFET. In a boost topology, the drain of the
external N-Channel MOSFET is connected to the inductor
and the source is connected to the ground. The drive pin
(DR) voltage depends on the input voltage (see typical per­formance characteristics). In most applications, a logic level
MOSFET can be used. For very low input voltages, a sub­logic level MOSFET should be used.

The selected MOSFET directly controls the efficiency. The
critical parameters for selection of a MOSFET are:

1. Minimum threshold voltage, V

TH

(MIN)

2. On-resistance, R

DS

(ON)

3. Total gate charge, Q

g

4. Reverse transfer capacitance, C

RSS

5. Maximum drain to source voltage, V

DS(MAX)

The off-state voltage of the MOSFET is approximately equal
to the output voltage. V

DS(MAX)

of the MOSFET must be
greater than the output voltage. The power losses in the
MOSFET can be categorized into conduction losses and ac
switching or transition losses. R

DS(ON)

is needed to estimate

the conduction losses. The conduction loss, P

COND

,isthe

I

2

R loss across the MOSFET. The maximum conduction loss

is given by:

10135520

FIGURE 10. Adjusting the Output Voltage

LM3478

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Typical Applications (Continued)

where D

MAX

is the maximum duty cycle.

The turn-on and turn-off transitions of a MOSFET require
times of tens of nano-seconds. C

RSS

and Qgare needed to
estimate the large instantaneous power loss that occurs
during these transitions.

The amount of gate current required to turn the MOSFET on
can be calculated using the formula:

I

G=Qg.FS

The required gate drive power to turn the MOSFET on is
equal to the switching frequency times the energy required
to deliver the charge to bring the gate charge voltage to V

DR

(see electrical characteristics and typical performance char­acteristics for the drive voltage specification).

P

Drive=FS.Qg.VDR

INPUT CAPACITOR SELECTION

Due to the presence of an inductor at the input of a boost
converter, the input current waveform is continuous and
triangular, as shown in Figure 9. The inductor ensures that
the input capacitor sees fairly low ripple currents. However,
as the input capacitor gets smaller, the input ripple goes up.
The rms current in the input capacitor is given by:

The input capacitor should be capable of handling the rms
current. Although the input capacitor is not as critical in a
boost application, low values can cause impedance interac­tions. Therefore a good quality capacitor should be chosen
in the range of 100µF to 200µF. If a value lower than 100µF
is used than problems with impedance interactions or
switching noise can affect the LM3478. To improve perfor­mance, especially with V

IN

below 8 volts, it is recommended
to use a 20Ω resistor at the input to provide a RC filter. The
resistor is placed in series with the V

IN

pin with only a bypass

capacitor attached to the V

IN

pin directly (see Figure 11). A

0.1µF or 1µF ceramic capacitor is necessary in this configu­ration. The bulk input capacitor and inductor will connect on
the other side of the resistor with the input power supply.

OUTPUT CAPACITOR SELECTION

The output capacitor in a boost converter provides all the
output current when the inductor is charging. As a result it
sees very large ripple currents. The output capacitor should
be capable of handling the maximum rms current. The rms
current in the output capacitor is:

Where

and D, the duty cycle is equal to (V

OUT−VIN

)/V

OUT

.

The ESR and ESL of the output capacitor directly control the
output ripple. Use capacitors with low ESR and ESL at the
output for high efficiency and low ripple voltage. Surface
Mount tantalums, surface mount polymer electrolytic and
polymer tantalum, Sanyo- OSCON, or multi-layer ceramic
capacitors are recommended at the output.

Designing SEPIC Using LM3478

Since the LM3478 controls a low-side N-Channel MOSFET,
it can also be used in SEPIC (Single Ended Primary Induc­tance Converter) applications. An example of SEPIC using
LM3478 is shown in Figure 12. As shown in Figure 12, the
output voltage can be higher or lower than the input voltage.
The SEPIC uses two inductors to step-up or step-down the
input voltage. The inductors L1 and L2 can be two discrete
inductors or two windings of a coupled transformer since
equal voltages are applied across the inductor throughout
the switching cycle. Using two discrete inductors allows use
of catalog magnetics, as opposed to a custom transformer.
The input ripple can be reduced along with size by using the
coupled windings of transformer for L1 and L2.

Due to the presence of the inductor L1 at the input, the
SEPIC inherits all the benefits of a boost converter. One
main advantage of SEPIC over boost converter is the inher­ent input to output isolation. The capacitor CS isolates the
input from the output and provides protection against
shorted or malfunctioning load. Hence, the A SEPIC is useful
for replacing boost circuits when true shutdown is required.
This means that the output voltage falls to 0V when the
switch is turned off. In a boost converter, the output can only
fall to the input voltage minus a diode drop.

10135593

FIGURE 11. Reducing IC Input Noise

LM3478

www.national.com17

Designing SEPIC Using LM3478

(Continued)

The duty cycle of a SEPIC is given by:

In the above equation, VQis the on-state voltage of the
MOSFET, Q, and V

DIODE

is the forward voltage drop of the

diode.

POWER MOSFET SELECTION

As in boost converter, the parameters governing the selec­tion of the MOSFET are the minimum threshold voltage,
V

TH(MIN)

, the on-resistance, R

DS(ON)

, the total gate charge,

Q

g

, the reverse transfer capacitance, C

RSS

, and the maxi-

mum drain to source voltage, V

DS(MAX)

. The peak switch

voltage in a SEPIC is given by:

V

SW(PEAK)

=VIN+V

OUT+VDIODE

The selected MOSFET should satisfy the condition:

V

DS(MAX)

>

V

SW(PEAK)

The peak switch current is given by:

The rms current through the switch is given by:

POWER DIODE SELECTION

The Power diode must be selected to handle the peak
current and the peak reverse voltage. In a SEPIC, the diode
peak current is the same as the switch peak current. The
off-state voltage or peak reverse voltage of the diode is V

IN

+V

OUT

. Similar to the boost converter, the average diode
current is equal to the output current. Schottky diodes are
recommended.

SELECTION OF INDUCTORS L1 AND L2

Proper selection of the inductors L1 and L2 to maintain
constant current mode requires calculations of the following
parameters.

Average current in the inductors:

I

L2AVE=IOUT

Peak to peak ripple current, to calculate core loss if neces­sary:

Maintaining the condition I

L

>

∆iLto ensure constant current

mode yields:

10135544

FIGURE 12. Typical SEPIC Converter

LM3478

www.national.com 18

Designing SEPIC Using LM3478

(Continued)

Peak current in the inductor, to ensure the inductor does not
saturate:

I

L1PK

must be lower than the maximum current rating set by

the current sense resistor.
The value of L1 can be increased above the minimum rec-

ommended to reduce input ripple and output ripple. How­ever, once D

IL1

is less than 20% of I

L1AVE

, the benefit to

output ripple is minimal.
By increasing the value of L2 above the minimum recom-

mended, ∆

IL2

can be reduced, which in turn will reduce the

output ripple voltage:

where ESR is the effective series resistance of the output
capacitor.

If L1 and L2 are wound on the same core, then L1 = L2 = L.
All the equations above will hold true if the inductance is
replaced by 2L. A good choice for transformer with equal
turns is Coiltronics CTX series Octopack.

SENSE RESISTOR SELECTION

The peak current through the switch, I

SW(PEAK)

can be ad-

justed using the current sense resistor, R

SEN

, to provide a

certain output current. Resistor R

SEN

can be selected using

the formula:

Sepic Capacitor Selection

The selection of SEPIC capacitor, CS, depends on the rms
current. The rms current of the SEPIC capacitor is given by:

The SEPIC capacitor must be rated for a large ACrms cur­rent relative to the output power. This property makes the
SEPIC much better suited to lower power applications where
the rms current through the capacitor is relatively small
(relative to capacitor technology). The voltage rating of the
SEPIC capacitor must be greater than the maximum input
voltage. Tantalum capacitors are the best choice for SMT,

having high rms current ratings relative to size. Ceramic
capacitors could be used, but the low C values will tend to
cause larger changes in voltage across the capacitor due to
the large currents. High C value ceramics are expensive.
Electrolytics work well for through hole applications where
the size required to meet the rms current rating can be
accommodated. There is an energy balance between CS
and L1, which can be used to determine the value of the
capacitor. The basic energy balance equation is:

Where

is the ripple voltage across the SEPIC capacitor, and

is the ripple current through the inductor L1. The energy
balance equation can be solved to provide a minimum value
for C

S

:

Input Capacitor Selection

Similar to a boost converter, the SEPIC has an inductor at
the input. Hence, the input current waveform is continuous
and triangular. The inductor ensures that the input capacitor
sees fairly low ripple currents. However, as the input capaci­tor gets smaller, the input ripple goes up. The rms current in
the input capacitor is given by:

The input capacitor should be capable of handling the rms
current. Although the input capacitor is not as critical in a
boost application, low values can cause impedance interac­tions. Therefore a good quality capacitor should be chosen
in the range of 100µF to 200µF. If a value lower than 100µF
is used than problems with impedance interactions or
switching noise can affect the LM3478. To improve perfor­mance, especially with V

IN

below 8 volts, it is recommended
to use a 20Ω resistor at the input to provide a RC filter. The
resistor is placed in series with the V

IN

pin with only a bypass

capacitor attached to the V

IN

pin directly (see Figure 11). A

0.1µF or 1µF ceramic capacitor is necessary in this configu­ration. The bulk input capacitor and inductor will connect on
the other side of the resistor with the input power supply.

Output Capacitor Selection

The ESR and ESL of the output capacitor directly control the
output ripple. Use low capacitors with low ESR and ESL at

LM3478

www.national.com19

Output Capacitor Selection

(Continued)

the output for high efficiency and low ripple voltage. Surface
mount tantalums, surface mount polymer electrolytic and
polymer tantalum, Sanyo- OSCON, or multi-layer ceramic
capacitors are recommended at the output.

The output capacitor of the SEPIC sees very large ripple
currents (similar to the output capacitor of a boost converter.
The rms current through the output capacitor is given by:

The ESR and ESL of the output capacitor directly control the
output ripple. Use low capacitors with low ESR and ESL at
the output for high efficiency and low ripple voltage. Surface

mount tantalums, surface mount polymer electrolytic and
polymer tantalum, Sanyo- OSCON, or multi-layer ceramic
capacitors are recommended at the output for low ripple.

Other Application Circuit

10135543

FIGURE 13. Typical Flyback Circuit

LM3478

www.national.com 20

Physical Dimensions inches (millimeters)

unless otherwise noted

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www.national.com

LM3478 High Efficiency Low-Side N-Channel Controller for Switching Regulator

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.