Datasheet LM3402HV, LM3402 Datasheet (NSC)

June 2007

LM3402/LM3402HV

0.5A Constant Current Buck Regulator for Driving High
Power LEDs

LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs

General Description

The LM3402/02HV are monolithic switching regulators de­signed to deliver constant currents to high power LEDs. Ideal
for automotive, industrial, and general lighting applications,
they contain a high-side N-channel MOSFET switch with a
current limit of 735 mA (typical) for step-down (Buck) regula­tors. Hysteretic control with controlled on-time coupled with
an external resistor allow the converter output voltage to ad­just as needed to deliver a constant current to series and
series - parallel connected arrays of LEDs of varying number
and type, LED dimming by pulse width modulation (PWM),
broken/open LED protection, low-power shutdown and ther­mal shutdown complete the feature set.

Typical Application

Features

Integrated 0.5A N-channel MOSFET

■

VIN Range from 6V to 42V (LM3402)

■

VIN Range from 6V to 75V (LM3402HV)

■

500 mA Output Current Over Temperature

■

Cycle-by-Cycle Current Limit

■

No Control Loop Compensation Required

■

Separate PWM Dimming and Low Power Shutdown

■

Supports all-ceramic output capacitors and capacitor-less

■

outputs
Thermal shutdown protection

■

MSOP-8, PSOP-8 Packages

■

Applications

LED Driver

■

Constant Current Source

■

Automotive Lighting

■

General Illumination

■

Industrial Lighting

■

20192101

© 2007 National Semiconductor Corporation 201921 www.national.com

Connection Diagrams

LM3402/LM3402HV

20192145

8-Lead Plastic MSOP-8 Package

NS Package Number MUA08A

20192102

8-Lead Plastic PSOP-8 Package

NS Package Number MRA08B

Ordering Information

Order Number Package Type NSC Package Drawing Supplied As

LM3402MM

LM3402MMX 3500 units on tape and reel

LM3402HVMM 1000 units on tape and reel

LM3402HVMMX 3500 units on tape and reel

LM3402MR

LM3402MRX 2500 units on tape and reel

LM3402HVMR 95 units in anti-static rails

LM3402HVMRX 2500 units on tape and reel

MSOP-8 MUA08A

PSOP-8 MRA08B

1000 units on tape and reel

95 units in anti-static rails

Pin Descriptions

Pin(s) Name Description Application Information

1 SW Switch pin Connect this pin to the output inductor and Schottky diode.

2 BOOT MOSFET drive bootstrap pin Connect a 10 nF ceramic capacitor from this pin to SW.

3 DIM Input for PWM dimming Connect a logic-level PWM signal to this pin to enable/disable the

power FET and reduce the average light output of the LED array.

4 GND Ground pin Connect this pin to system ground.

5 CS Current sense feedback pin Set the current through the LED array by connecting a resistor from

this pin to ground.

6 RON On-time control pin A resistor connected from this pin to VIN sets the regulator controlled

on-time.

7 VCC Output of the internal 7V linear

regulator

8 VIN Input voltage pin Nominal operating input range is 6V to 42V (LM3402) or 6V to 75V

DAP GND Thermal Pad Connect to ground. Place 4 to 6 vias from DAP to bottom layer ground

Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor
with X5R or X7R dielectric.

(LM3402HV).

plane.

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Absolute Maximum Ratings (LM3402) (Note 1)

If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.

VIN to GND -0.3V to 45V
BOOT to GND -0.3V to 59V
SW to GND -1.5V
BOOT to VCC -0.3V to 45V
BOOT to SW -0.3V to 14V
VCC to GND -0.3V to 14V
DIM to GND -0.3V to 7V
CS to GND -0.3V to 7V
RON to GND -0.3V to 7V
Junction Temperature 150°C
Storage Temp. Range -65°C to 125°C
ESD Rating (Note 2) 2kV
Soldering Information
Lead Temperature (Soldering,

10sec) 260°C
Infrared/Convection Reflow (15sec) 235°C

Operating Ratings (LM3402) (Note 1)

V

IN

Junction Temperature Range −40°C to +125°C

Thermal Resistance θJA (MSOP-8 Package)
(Note 3) 200°C/W

Thermal Resistance θJA (PSOP-8 Package)
(Note 5) 50°C/W

6V to 42V

LM3402/LM3402HV

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Absolute Maximum Ratings (LM3402HV) (Note 1)

If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.

VIN to GND

LM3402/LM3402HV

BOOT to GND -0.3V to 90V
SW to GND -1.5V
BOOT to VCC -0.3V to 76V
BOOT to SW -0.3V to 14V
VCC to GND -0.3V to 14V
DIM to GND -0.3V to 7V
CS to GND -0.3V to 7V
RON to GND -0.3V to 7V
Junction Temperature 150°C

-0.3V to 76V

Storage Temp. Range -65°C to 125°C
ESD Rating (Note 2) 2kV
Soldering Information
Lead Temperature (Soldering,

10sec) 260°C
Infrared/Convection Reflow (15sec) 235°C

Operating Ratings (LM3402HV) (Note 1)

V

IN

Junction Temperature Range −40°C to +125°C

Thermal Resistance θJA (MSOP-8 Package)
(Note 3) 200°C/W

Thermal Resistance θJA (PSOP-8 Package)
(Note 5) 50°C/W

6V to 75V

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LM3402/LM3402HV

Electrical Characteristics V

= 24V unless otherwise indicated. Typicals and limits appearing in plain type apply

IN

for TA = TJ = +25°C. (Note 4) Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/max
specification limits are guaranteed by design, test, or statistical analysis.

LM3402

Symbol Parameter Conditions Min Typ Max Units

SYSTEM PARAMETERS

t

ON-1

t

ON-2

On-time 1

On-time 2

VIN = 10V, RON = 200 kΩ

VIN = 40V, RON = 200 kΩ

2.1 2.75 3.4 µs

490 650 810 ns

LM3402HV

Symbol Parameter Conditions Min Typ Max Units

SYSTEM PARAMETERS

t

ON-1

t

ON-2

On-time 1

On-time 2

VIN = 10V, RON = 200 kΩ

VIN = 70V, RON = 200 kΩ

2.1 2.75 3.4 µs

290 380 470 ns

LM3402/LM3402HV

Symbol Parameter Conditions Min Typ Max Units

REGULATION AND OVER-VOLTAGE COMPARATORS

V

REF-REG

V

REF-0V

I

CS

SHUTDOWN

V

SD-TH

V

SD-HYS

OFF TIMER

t

OFF-MIN

INTERNAL REGULATOR

V

CC-REG

V

IN-DO

V

CC-BP-TH

V

CC-BP-HYS

V

CC-Z-6

V

CC-Z-8

V

CC-Z-24

V

CC-LIM

V

CC-UV-TH

V

CC-UV-HYS

V

CC-UV-DLY

I

IN-OP

I

IN-SD

CURRENT LIMIT

I

LIM

DIM COMPARATOR

V

IH

V

IL

CS Regulation Threshold CS Decreasing, SW turns on 194 200 206 mV

CS Over-voltage Threshold CS Increasing, SW turns off 300 mV

CS Bias Current CS = 0V 0.1 µA

Shutdown Threshold RON / SD Increasing 0.3 0.7 1.05 V

Shutdown Hysteresis RON / SD Decreasing 40 mV

Minimum Off-time CS = 0V 300 ns

VCC Regulated Output 6.6 7 7.4 V

VIN - VCC Dropout ICC = 5 mA, 6.0V < VIN < 8.0V 300 mV

VCC Bypass Threshold VIN Increasing 8.8 V

VCC Bypass Hysteresis VIN Decreasing 225 mV

VCC Output Impedance
(0 mA < ICC < 5 mA)

VIN = 6V 55

VIN = 8V 50

Ω

VIN = 24V 0.4

VCC Current Limit (Note 3) VIN = 24V, VCC = 0V 16 mA

VCC Under-voltage Lock-out

VCC Increasing 5.25 V

Threshold

VCC Under-voltage Lock-out

VCC Decreasing 150 mV

Hysteresis

VCC Under-voltage Lock-out

100 mV Overdrive 3 µs

Filter Delay

I

Operating Current Non-switching, CS = 0V 600 900 µA

IN

IIN Shutdown Current RON / SD = 0V 90 180 µA

Current Limit Threshold 530 735 940 mA

Logic High DIM Increasing 2.2 V

Logic Low DIM Decreasing 0.8 V

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Symbol Parameter Conditions Min Typ Max Units

I

DIM-PU

DIM Pull-up Current DIM = 1.5V 75 µA

N-MOSFET AND DRIVER

R

DS-ON

V

DR-UVLO

Buck Switch On Resistance ISW = 200mA, BOOT-SW = 6.3V 0.7 1.5

BOOT Under-voltage Lock-out

BOOT–SW Increasing 1.7 3 4 V

Threshold

LM3402/LM3402HV

V

DR-HYS

BOOT Under-voltage Lock-out

BOOT–SW Decreasing 400 mV

Hysteresis

THERMAL SHUTDOWN

T

SD

T

SD-HYS

THERMAL RESISTANCE

θ

JA

Thermal Shutdown Threshold 165 °C

Thermal Shutdown Hysteresis 25 °C

Junction to Ambient MSOP-8 Package 200 °C/W

PSOP-8 Package 50

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.

Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.

Note 3: VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.

Note 4: Typical specifications represent the most likely parametric norm at 25°C operation.

Note 5: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1 oz. copper on the top or bottom PCB layer.

Ω

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Typical Performance Characteristics

V

vs Temperature (VIN = 24V)

REF

V

vs VIN, LM3402 (TA = 25°C)

REF

LM3402/LM3402HV

20192129

V

vs VIN, LM3402HV (TA = 25°C)

REF

20192131

Current Limit vs VIN, LM3402 (TA = 25°C)

20192130

Current Limit vs Temperature (VIN = 24V)

20192132

Current Limit vs VIN, LM3402HV (TA = 25°C)

20192133

20192134

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LM3402/LM3402HV

TON vs VIN,

RON = 100 kΩ (TA = 25°C)

TON vs VIN,

(TA = 25°C)

TON vs VIN,
(TA = 25°C)

TON vs RON, LM3402HV

(TA = 25°C)

20192135

20192137

TON vs RON, LM3402

(TA = 25°C)

VCC vs V

(TA = 25°C)

IN

20192136

20192144

20192138

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20192139

V

vs fSW, LM3402

O-MAX

(TA = 25°C)

V

vs fSW, LM3402

O-MIN

(TA = 25°C)

LM3402/LM3402HV

V

vs fSW, LM3402HV

O-MAX

(TA = 25°C)

20192140

20192142

V

vs fSW, LM3402HV

O-MIN

(TA = 25°C)

20192141

20192143

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Block Diagram

LM3402/LM3402HV

Application Information

THEORY OF OPERATION

The LM3402 and LM3402HV are buck regulators with a wide
input voltage range, low voltage reference, and a fast output
enable/disable function. These features combine to make
them ideal for use as a constant current source for LEDs with
forward currents as high as 500 mA. The controlled on-time
(COT) architecture is a combination of hysteretic mode con­trol and a one-shot on-timer that varies inversely with input
voltage. Hysteretic operation eliminates the need for small­signal control loop compensation. When the converter runs in
continuous conduction mode (CCM) the controlled on-time
maintains a constant switching frequency over the range of
input voltage. Fast transient response, PWM dimming, a low
power shutdown mode, and simple output overvoltage pro­tection round out the functions of the LM3402/02HV.

CONTROLLED ON-TIME OVERVIEW

Figure 1 shows the feedback system used to control the cur­rent through an array of LEDs. A voltage signal, V

SNS

, is

20192103

created as the LED current flows through the current setting
resistor, R
where it is compared against a 200 mV reference, V
on-comparator turns on the power MOSFET when V
below V
on-time, tON, set by an external resistor, RON, and by the input

, to ground. V

SNS

. The power MOSFET conducts for a controlled

REF

is fed back to the CS pin,

SNS

REF

SNS

. The

falls

voltage, VIN. On-time is governed by the following equation:

At the conclusion of tON the power MOSFET turns off for a
minimum off-time, t
plete the CS comparator compares V
waiting to begin the next cycle.

, of 300 ns. Once t

OFF-MIN

SNS

OFF-MIN

and V

is com-

again,

REF

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FIGURE 1. Comparator and One-Shot

The LM3402/02HV regulators should be operated in contin­uous conduction mode (CCM), where inductor current stays
positive throughout the switching cycle. During steady-state
operationin the CCM, the converter maintains a constant
switching frequency, which can be selected using the follow­ing equation:

20192105

The maximum number of LEDs, n
a single series string is governed by V
mum forward voltage of the LEDs used, V

, that can be placed in

MAX

O(MAX)

expression:

and the maxi-

, using the

F(MAX)

LM3402/LM3402HV

VF = forward voltage of each LED, n = number of LEDs in

series

AVERAGE LED CURRENT ACCURACY

The COT architecture regulates the valley of ΔV
portion of V
is also the average inductor current) the valley inductor cur-

. To determine the average LED current (which

SNS

SNS

, the AC

rent is calculated using the following expression:

In this equation t
CS comparator, and is approximately 220 ns. The average
inductor/LED current is equal to I
ductor current ripple, ΔiL:

represents the propagation delay of the

SNS

plus one-half of the in-

L-MIN

IF = IL = I

L-MIN

+ ΔiL / 2

Detailed information for the calculation of ΔiL is given in the
Design Considerations section.

MAXIMUM OUTPUT VOLTAGE

The 300 ns minimum off-time limits on the maximum duty cy­cle of the converter, D
voltage V

is determined by the following equations:

O(MAX)

, and in turn ,the maximum output

MAX

At low switching frequency the maximum duty cycle and out­put voltage are higher, allowing the LM3402/02HV to regulate
output voltages that are nearly equal to input voltage. The
following equation relates switching frequency to maximum
output voltage.

MINIMUM OUTPUT VOLTAGE

The minimum recommended on-time for the LM3402/02HV is
300 ns. This lower limit for tON determines the minimum duty
cycle and output voltage that can be regulated based on input
voltage and switching frequency. The relationship is deter­mined by the following equation:

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HIGH VOLTAGE BIAS REGULATOR

The LM3402/02HV contains an internal linear regulator with
a 7V output, connected between the VIN and the VCC pins.
The VCC pin should be bypassed to the GND pin with a 0.1
µF ceramic capacitor connected as close as possible to the
pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical)
and then regulates at 7V as VIN increases. Operation begins
when VCC crosses 5.25V.

LM3402/LM3402HV

INTERNAL MOSFET AND DRIVER

The LM3402/02HV features an internal power MOSFET as
well as a floating driver connected from the SW pin to the
BOOT pin. Both rise time and fall time are 20 ns each (typical)
and the approximate gate charge is 3 nC. The high-side rail
for the driver circuitry uses a bootstrap circuit consisting of an
internal high-voltage diode and an external 10 nF capacitor,
CB. VCC charges CB through the internal diode while the power
MOSFET is off. When the MOSFET turns on, the internal
diode reverse biases. This creates a floating supply equal to
the VCC voltage minus the diode drop to drive the MOSFET
when its source voltage is equal to VIN.

FAST SHUTDOWN FOR PWM DIMMING

The DIM pin of the LM3402/02HV is a TTL logic compatible
input for low frequency PWM dimming of the LED. A logic low
(below 0.8V) at DIM will disable the internal MOSFET and
shut off the current flow to the LED array. While the DIM pin
is in a logic low state the support circuitry (driver, bandgap,
VCC) remains active in order to minimize the time needed to
turn the LED array back on when the DIM pin sees a logic
high (above 2.2V). A 75 µA (typical) pull-up current ensures
that the LM3402/02HV is on when DIM pin is open circuited,
eliminating the need for a pull-up resistor. Dimming frequen­cy, f

, and duty cycle, D

DIM

rise time and fall time and the delay from activation of the DIM

, are limited by the LED current

DIM

pin to the response of the internal power MOSFET. In general,
f

should be at least one order of magnitude lower than the

DIM

steady state switching frequency in order to prevent aliasing.

PEAK CURRENT LIMIT

The current limit comparator of the LM3402/02HV will engage
whenever the power MOSFET current (equal to the inductor

current while the MOSFET is on) exceeds 735 mA (typical).
The power MOSFET is disabled for a cool-down time that is
10x the steady-state on-time. At the conclusion of this cool­down time the system re-starts. If the current limit condition
persists the cycle of cool-down time and restarting will con­tinue, creating a low-power hiccup mode, minimizing thermal
stress on the LM3402/02HV and the external circuit compo­nents.

OVER-VOLTAGE/OVER-CURRENT COMPARATOR

The CS pin includes an output over-voltage/over-current
comparator that will disable the power MOSFET whenever
V

exceeds 300 mV. This threshold provides a hard limit

SNS

for the output current. Output current overshoot is limited to
300 mV / R

by this comparator during transients.

SNS

The OVP/OCP comparator can also be used to prevent the
output voltage from rising to V
open-circuit. This is the most common failure mode for LEDs,

in the event of an output

O(MAX)

due to breaking of the bond wires. In a current regulator an
output open circuit causes V
maximum duty cycle. Figure 2 shows a method using a zener

to fall to zero, commanding

SNS

diode, Z1, and zener limiting resistor, RZ, to limit output volt­age to the reverse breakdown voltage of Z1 plus 200 mV. The
zener diode reverse breakdown voltage, VZ, must be greater
than the maximum combined VF of all LEDs in the array. The
maximum recommended value for RZ is 1 kΩ.

As discussed in the Maximum Output Voltage section, there
is a limit to how high VO can rise during an output open-circuit
that is always less than VIN. If no output capacitor is used, the
output stage of the LM3402/02HV is capable of withstanding
V

indefinitely, however the voltage at the output end of

O(MAX)

the inductor will oscillate and can go above VIN or below 0V.
A small (typically 10 nF) capacitor across the LED array
dampens this oscillation. For circuits that use an output ca­pacitor, the system can still withstand V
long as CO is rated to handle VIN. The high current paths are

indefinitely as

O(MAX)

blocked in output open-circuit and the risk of thermal stress is
minimal, hence the user may opt to allow the output voltage
to rise in the case of an open-circuit LED failure.

FIGURE 2. Output Open Circuit Protection

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20192112

LM3402/LM3402HV

LOW POWER SHUTDOWN

The LM3402/02HV can be switched to a low power state (I

= 90 µA) by grounding the RON pin with a signal-level

SD

MOSFET as shown in Figure 3. Low power MOSFETs like the

IN-

2N7000, 2N3904, or equivalent are recommended devices
for putting the LM3402/02HV into low power shutdown. Logic
gates can also be used to shut down the LM3402/02HV as

FIGURE 3. Low Power Shutdown

long as the logic low voltage is below the over temperature
minimum threshold of 0.3V. Noise filter circuitry on the RON
pin can cause a few pulses with a longer on-time than normal
after RON is grounded or released. In these cases the OVP/
OCP comparator will ensure that the peak inductor or LED
current does not exceed 300 mV / R

SNS

20192113

.

THERMAL SHUTDOWN

Internal thermal shutdown circuitry is provided to protect the
IC in the event that the maximum junction temperature is ex-

ceeded. The threshold for thermal shutdown is 165°C with a
25°C hysteresis (both values typical). During thermal shut­down the MOSFET and driver are disabled.

13 www.national.com

Design Considerations

SWITCHING FREQUENCY

Switching frequency is selected based on the tradeoffs be­tween efficiency (better at low frequency), solution size/cost
(smaller at high frequency), and the range of output voltage
that can be regulated (wider at lower frequency.) Many appli­cations place limits on switching frequency due to EMI sen-

LM3402/LM3402HV

sitivity. The on-time of the LM3402/02HV can be programmed
for switching frequencies ranging from the 10’s of kHz to over
1 MHz. The maximum switching frequency is limited only by
the minimum on-time requirement.

LED RIPPLE CURRENT

Selection of the ripple current, ΔiF, through the LED array is
analogous to the selection of output ripple voltage in a stan­dard voltage regulator. Where the output ripple in a voltage
regulator is commonly ±1% to ±5% of the DC output voltage,
LED manufacturers generally recommend values for Δi
ranging from ±5% to ±20% of IF. Higher LED ripple current
allows the use of smaller inductors, smaller output capacitors,
or no output capacitors at all. The advantages of higher ripple
current are reduction in the solution size and cost. Lower rip­ple current requires more output inductance, higher switching
frequency, or additional output capacitance. The advantages
of lower ripple current are a reduction in heating in the LED
itself and greater range of the average LED current before the
current limit of the LED or the driving circuitry is reached.

ered, making the magnetics smaller and less expensive.
Alternatively, the circuit could be run at lower frequency but
keep the same inductor value, improving the efficiency and
expanding the range of output voltage that can be regulated.
Both the peak current limit and the OVP/OCP comparator still
monitor peak inductor current, placing a limit on how large
ΔiL can be even if ΔiF is made very small. A parallel output
capacitor is also useful in applications where the inductor or
input voltage tolerance is poor. Adding a capacitor that re­duces ΔiF to well below the target provides headroom for
changes in inductance or VIN that might otherwise push the
peak LED ripple current too high.

Figure 4 shows the equivalent impedances presented to the
inductor current ripple when an output capacitor, CO, and its
equivalent series resistance (ESR) are placed in parallel with
the LED array. The entire inductor ripple current flows through
R

to provide the required 25 mV of ripple voltage for proper

SNS

operation of the CS comparator.

F

BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS

The buck converter is unique among non-isolated topologies
because of the direct connection of the inductor to the load
during the entire switching cycle. By definition an inductor will
control the rate of change of current that flows through it, and
this control over current ripple forms the basis for component
selection in both voltage regulators and current regulators. A
current regulator such as the LED driver for which the
LM3402/02HV was designed focuses on the control of the
current through the load, not the voltage across it. A constant
current regulator is free of load current transients, and has no
need of output capacitance to supply the load and maintain
output voltage. Referring to the Typical Application circuit on
the front page of this datasheet, the inductor and LED can
form a single series chain, sharing the same current. When
no output capacitor is used, the same equations that govern
inductor ripple current, ΔiL, also apply to the LED ripple cur­rent, ΔiF. For a controlled on-time converter such as
LM3402/02HV the ripple current is described by the following
expression:

A minimum ripple voltage of 25 mV is recommended at the
CS pin to provide good signal-to-noise ratio (SNR). The CS
pin ripple voltage, ΔV

, is described by the following:

SNS

ΔV

= ΔiF x R

SNS

SNS

BUCK CONVERTERS WITH OUTPUT CAPACITORS

A capacitor placed in parallel with the LED or array of LEDs
can be used to reduce the LED current ripple while keeping
the same average current through both the inductor and the
LED array. This technique is demonstrated in Design Exam­ple 1. With this topology the output inductance can be low-

20192115

FIGURE 4. LED and CO Ripple Current

To calculate the respective ripple currents the LED array is
represented as a dynamic resistance, rD. LED dynamic resis­tance is not always specified on the manufacturer’s
datasheet, but it can be calculated as the inverse slope of the
LED’s VF vs. IF curve. Note that dividing VF by IF will give an
incorrect value that is 5x to 10x too high. Total dynamic re­sistance for a string of n LEDs connected in series can be
calculated as the rD of one device multiplied by n. Inductor
ripple current is still calculated with the expression from Buck
Regulators without Output Capacitors. The following equa­tions can then be used to estimate ΔiF when using a parallel
capacitor:

The calculation for ZC assumes that the shape of the inductor
ripple current is approximately sinusoidal.

Small values of CO that do not significantly reduce ΔiF can
also be used to control EMI generated by the switching action
of the LM3402/02HV. EMI reduction becomes more important
as the length of the connections between the LED and the
rest of the circuit increase.

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LM3402/LM3402HV

INPUT CAPACITORS

Input capacitors at the VIN pin of the LM3402/02HV are se­lected using requirements for minimum capacitance and rms
ripple current. The input capacitors supply pulses of current
approximately equal to IF while the power MOSFET is on, and
are charged up by the input voltage while the power MOSFET
is off. Switching converters such as the LM3402/02HV have
a negative input impedance due to the decrease in input cur­rent as input voltage increases. This inverse proportionality of
input current to input voltage can cause oscillations (some­times called ‘power supply interaction’) if the magnitude of the
negative input impedance is greater the the input filter
impedance. Minimum capacitance can be selected by com­paring the input impedance to the converter’s negative resis­tance; however this requires accurate calculation of the input
voltage source inductance and resistance, quantities which
can be difficult to determine. An alternative method to select
the minimum input capacitance, C
imum voltage ripple which can be tolerated. This value,Δv

, is equal to the change in voltage across CIN during the

(MAX)

converter on-time, when CIN supplies the load current. C

can be selected with the following:

(MIN)

, is to select the max-

IN(MIN)

A good starting point for selection of CIN is to use an input
voltage ripple of 5% to 10% of VIN. A minimum input capaci­tance of 2x the C
LM3402/02HV circuits. To determine the rms current rating,

value is recommended for all

IN(MIN)

the following formula can be used:

Ceramic capacitors are the best choice for the input to the
LM3402/02HV due to their high ripple current rating, low ESR,
low cost, and small size compared to other types. When se­lecting a ceramic capacitor, special attention must be paid to
the operating conditions of the application. Ceramic capaci­tors can lose one-half or more of their capacitance at their
rated DC voltage bias and also lose capacitance with ex­tremes in temperature. A DC voltage rating equal to twice the
expected maximum input voltage is recommended. In addi­tion, the minimum quality dielectric which is suitable for
switching power supply inputs is X5R, while X7R or better is
preferred.

RECIRCULATING DIODE

The LM3402/02HV is a non-synchronous buck regulator that
requires a recirculating diode D1 (see the Typical Application
circuit) to carrying the inductor current during the MOSFET
off-time. The most efficient choice for D1 is a Schottky diode
due to low forward drop and near-zero reverse recovery time.
D1 must be rated to handle the maximum input voltage plus
any switching node ringing when the MOSFET is on. In prac­tice all switching converters have some ringing at the switch­ing node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average cur­rent, ID, calculated as:

ID = (1 – D) x I

F

This calculation should be done at the maximum expected
input voltage. The overall converter efficiency becomes more

IN

dependent on the selection of D1 at low duty cycles, where
the recirculating diode carries the load current for an increas-

IN

ing percentage of the time. This power dissipation can be
calculated by checking the typical diode forward voltage, VD,
from the I-V curve on the product datasheet and then multi­plying it by ID. Diode datasheets will also provide a typical
junction-to-ambient thermal resistance, θJA, which can be
used to estimate the operating die temperature of the Schot­tky. Multiplying the power dissipation (PD = ID x VD) by θ
gives the temperature rise. The diode case size can then be

JA

selected to maintain the Schottky diode temperature below
the operational maximum.

Design Example 1: LM3402

The first example circuit will guide the user through compo­nent selection for an architectural accent lighting application.
A regulated DC voltage input of 24V ±10% will power a single
1W white LED at a forward current of 350 mA ±5%. The typical
forward voltage of a 1W InGaN LED is 3.5V, hence the esti­mated average output voltage will be 3.7V. The objective of
this application is to place the complete current regulator and
LED in the compact space formerly occupied by an MR16
halogen light bulb. (The LED will be on a separate metal-core
PCB.) Switching frequency will be as fast as the 300 ns t
limit allows, with the emphasis on space savings over effi­ciency. Efficiency cannot be ignored, however, as the con­fined space with little air-flow requires a maximum tempera­ture rise of 40°C in each circuit component. A complete bill of
materials can be found in Table 1 at the end of this datasheet.

ON

15 www.national.com

LM3402/LM3402HV

20192119

FIGURE 5. Schematic for Design Example 1

RON and t

ON

To select RON the expression relating tON to input voltage from
the Controlled On-time Overview section can be re-written as:

Minimum on-time occurs at the maximum VIN, which is 24V x
110% = 26.4V. RON is therefore calculated as:

RON = (300 x 10-9 x 26.4) / 1.34 x 10

-10

= 59105 Ω

The closest 1% tolerance resistor is 59.0 kΩ. The switching
frequency of the circuit can then be found using the equation
relating RON to fSW:

fSW = 3.7 / (59000 x 1.34 x 10

-10

) = 468 kHz

USING AN OUTPUT CAPACITOR

The inductor will be the largest component used in this design.
Because the application does not require any PWM dimming,
an output capacitor can be used to greatly reduce the induc­tance needed without worry of slowing the potential PWM
dimming frequency. The total solution size will be reduced by
using an output capacitor and small inductor as opposed to
one large inductor.

OUTPUT INDUCTOR

Knowing that an output capacitor will be used, the inductor
can be selected for a larger current ripple. The desired max­imum value for ΔiL is ±30%, or 0.6 x 350 mA = 210 mA
Minimum inductance is selected at the maximum input volt-

P-P

age. Re-arranging the equation for current ripple selection
yields the following:

The closest standard inductor value is 33 µH. Off-the-shelf
inductors rated at 33 µH are available from many magnetics
manufacturers.

Inductor datasheets should contain three specifications which
are used to select the inductor. The first of these is the aver­age current rating, which for a buck regulator is equal to the
average load current, or IF. The average current rating is given
by a specified temperature rise in the inductor, normally 40°
C. For this example, the average current rating should be
greater than 350 mA to ensure that heat from the inductor
does not reduce the lifetime of the LED or cause the LM3402
to enter thermal shutdown.

The second specification is the tolerance of the inductance
itself, typically ±10% to ±30% of the rated inductance. In this
example an inductor with a tolerance of ±20% will be used.
With this tolerance the typical, minimum, and maximum in­ductor current ripples can be calculated:

Δi

= [(26.4 – 3.7) x 300 x 10-9] / 33 x 10

L(TYP)

= 206 mA

P-P

-6

Δi

= [(26.4 – 3.7) x 300 x 10-9] / 39.6 x 10

L(MIN)

= 172 mA

P-P

-6

Δi

= [(26.4 – 3.7) x 300 x 10-9] / 26.4 x 10

L(MAX)

= 258 mA

P-P

-6

The third specification for an inductor is the peak current rat­ing, normally given as the point at which the inductance drops

.

off by a given percentage due to saturation of the core. The
worst-case peak current occurs at maximum input voltage
and at minimum inductance, and can be determined with the
equation from the Design Considerations section:

L

= [(26.4 – 3.7) x 300 x 10-9] / (0.6 x 0.35) = 32.4 µH

MIN

www.national.com 16

I

= 0.35 + 0.258 / 2 = 479 mA

L(PEAK)

LM3402/LM3402HV

For this example the peak current rating of the inductor should
be greater than 479 mA. In the case of a short circuit across
the LED array, the LM3402 will continue to deliver rated cur­rent through the short but will reduce the output voltage to
equal the CS pin voltage of 200 mV. Worst-case peak current
in this condition is equal to:

Δi

L(LED-SHORT)

= [(26.4 – 0.2) x 300 x 10-9] / 26.4 x 10

= 298 mA

I

= 0.35 + 0.149 = 499 mA

L(PEAK)

P-P

-6

In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent in­ductor saturation during these short circuits the inductor’s
peak current rating must be above 735 mA. The device se­lected is an off-the-shelf inductor rated 33 µH ±20% with a
DCR of 96 mΩ and a peak current rating of 0.82A. The phys­ical dimensions of this inductor are 7.0 x 7.0 x 4.5 mm.

R

SNS

The current sensing resistor value can be determined by re­arranging the expression for average LED current from the
LED Current Accuracy section:

R

= 0.74Ω, t

SNS

= 220 ns

SNS

Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.75Ω resistor will give the best accuracy of the aver­age LED current. To determine the resistor size the power
dissipation can be calculated as:

P

= (IF)2 x R

SNS

P

= 0.352 x 0.75 = 92 mW

SNS

SNS

Standard 0805 size resistors are rated to 125 mW and will be
suitable for this application.

To select the proper output capacitor the equation from Buck
Regulators with Output Capacitors is re-arranged to yield the
following:

The target tolerance for LED ripple current is ±5% or 10%

= 35 mA

P

rD of 1.0Ω at 350 mA. The required capacitor impedance to
reduce the worst-case inductor ripple current of 258 mA
therefore:

, and the LED datasheet gives a typical value for

P-P

P-P

is

ZC = [0.035 / (0.258 - 0.035] x 1.0 = 0.157Ω

A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 468 kHz:

CO = 1/(2 x π x 0.157 x 4.68 x 105) = 2.18 µF

This calculation assumes that impedance due to the equiva­lent series resistance (ESR) and equivalent series inductance
(ESL) of CO is negligible. The closest 10% tolerance capacitor
value is 2.2 µF. The capacitor used should be rated to 10V or
more and have an X7R dielectric. Several manufacturers pro­duce ceramic capacitors with these specifications in the 0805
case size. A typical value for ESR of 1 mΩ can be read from
the curve of impedance vs. frequency in the product
datasheet.

INPUT CAPACITOR

Following the calculations from the Input Capacitor section,
Δv
pacitance is:

IN(MAX)

will be 1%

= 240 mV. The minimum required ca-

P-P

C

= (0.35 x 300 x 10-9) / 0.24 = 438 nF

IN(MIN)

In expectation that more capacitance will be needed to pre­vent power supply interaction a 1.0 µF ceramic capacitor
rated to 50V with X7R dielectric in a 1206 case size will be
used. From the Design Considerations section, input rms cur­rent is:

I

= 0.35 x Sqrt(0.154 x 0.846) = 126 mA

IN-RMS

Ripple current ratings for 1206 size ceramic capacitors are
typically higher than 1A, more than enough for this design.

RECIRCULATING DIODE

The first parameter for D1 which must be determined is the
reverse voltage rating. Schottky diodes are available at re­verse ratings of 30V and 40V, often in the same package, with
the same forward current rating. To account for ringing a 40V
Schottky will be used.

The next parameters to be determined are the forward current
rating and case size. In this example the low duty cycle (D =

3.7 / 24 = 15%) requires the recirculating diode D1 to carry
the load current much longer than the internal power MOS­FET of the LM3402. The estimated average diode current is:

ID = 0.35 x 0.85 = 298 mA

Schottky diodes are available at forward current ratings of

0.5A, however the current rating often assumes a 25°C am­bient temperature and does not take into account the appli­cation restrictions on temperature rise. A diode rated for
higher current may be needed to keep the temperature rise
below 40°C.To determine the proper case size, the dissipa-

P-

tion and temperature rise in D1 can be calculated as shown
in the Design Considerations section. VD for a small case size
such as SOD-123 in a 40V, 0.5A Schottky diode at 350 mA is
approximately 0.4V and the θJA is 206°C/W. Power dissipa­tion and temperature rise can be calculated as:

PD = 0.298 x 0.4 = 119 mW

T

= 0.119 x 206 = 24.5°C

RISE

According to these calculations the SOD-123 diode will meet
the requirements. Heating and dissipation are among the fac-

17 www.national.com

tors most difficult to predict in converter design. If possible, a
footprint should be used that is capable of accepting both
SOD-123 and a larger case size, such as SMA. A larger diode
with a higher forward current rating will generally have a lower
forward voltage, reducing dissipation, as well as having a
lower θJA, reducing temperature rise.

CB and C

LM3402/LM3402HV

The bootstrap capacitor CB should always be a 10 nF ceramic

F

capacitor with X7R dielectric. A 25V rating is appropriate for
all application circuits. The linear regulator filter capacitor C
should always be a 100 nF ceramic capacitor, also with X7R
dielectric and a 25V rating.

EFFICIENCY

To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. This term should not be confused with the op­tical efficacy of the circuit, which depends upon the LEDs
themselves.

Total output power, PO, is calculated as:

PO = IF x VO = 0.35 x 3.7 = 1.295W

Conduction loss, PC, in the internal MOSFET:

2

PC = (I

x R

F

) x D = (0.352 x 1.5) x 0.154 = 28 mW

DSON

Gate charging and VCC loss, PG, in the gate drive and linear
regulator:

PG = (600 x 10-6 + 468000 x 3 x 10-9) x 24 = 48 mW

PG = (I

+ fSW x QG) x V

IN-OP

IN

Switching loss, PS, in the internal MOSFET:

PS = 0.5 x 24 x 0.35 x (40 x 10-9) x 468000 = 78 mW

PS = 0.5 x VIN x IF x (tR + tF) x f

SW

AC rms current loss, P

, in the input capacitor:

CIN

2

P

= I

CIN

x ESR = (0.126)2 x 0.006 = 0.1 mW (negligible)

IN(rms)

DCR loss, PL, in the inductor

2

x DCR = 0.352 x 0.096 = 11.8 mW

PL = I

F

Recirculating diode loss, PD = 119 mW
Current Sense Resistor Loss, P

= 92 mW

SNS

Electrical efficiency, η = PO / (PO + Sum of all loss terms) =

1.295 / (1.295 + 0.377) = 77%

DIE TEMPERATURE

T

= (PC + PG + PS) x θ

T

LM3402

LM3402

= (0.028 + 0.05 + 0.078) x 200 = 31°C

JA

Design Example 2: LM3402HV

The second example application is an RGB backlight for a flat
screen monitor. A separate boost regulator provides a 60V

±5% DC input rail that feeds three LM3402HV current regu­lators to drive one series array each of red, green, and blue
1W LEDs. The target for average LED current is 350 mA ±5%
in each string. The monitor will adjust the color temperature
dynamically, requiring fast PWM dimming of each string with
external, parallel MOSFETs. 1W green and blue InGaN LEDs
have a typical forward voltage of 3.5V, however red LEDs use
AlInGaP technology with a typical forward voltage of 2.9V. In
order to match color properly the design requires 14 green
LEDs, twice as many as needed for the red and blue LEDs.
This example will follow the design for the green LED array,

F

providing the necessary information to repeat the exercise for
the blue and red LED arrays. The circuit schematic for Design
Example 2 is the same as the Typical Application on the front
page. The bill of materials (green array only) can be found in
Table 2 at the end of this datasheet.

OUTPUT VOLTAGE

Green Array: V

Blue Array: V

Red Array: V

= 14 x 3.5 + 0.2 = 49.2V

O(G)

= 7 x 3.5 + 0.2 = 24.7V

O(B)

= 7 x 2.9 + 0.2 = 20.5V

O(R)

RON and t

ON

A compromise in switching frequency is needed in this appli­cation to balance the requirements of magnetics size and
efficiency. The high duty cycle translates into large conduc­tion losses and high temperature rise in the IC. For best
response to a PWM dimming signal this circuit will not use an
output capacitor; hence a moderate switching frequency of
300 kHz will keep the inductance from becoming so large that
a custom-wound inductor is needed. This design will use only
surface mount components, and the selection of off-the-shelf
SMT inductors for switching regulators is poor at 1000 µH and
above. RON is selected from the equation for switching fre­quency as follows:

RON = 49.2 / (1.34 x 10

-10

x 3 x 105) = 1224 kΩ

The closest 1% tolerance resistor is 1.21 MΩ. The switching
frequency and on-time of the circuit can then be found using
the equations relating RON and tON to fSW:

fSW = 49.2 / (1210000 x 1.34 x 10

-10

) = 303 kHz

tON = (1.34 x 10

-10

x 1210000) / 60 = 2.7 µs

USING AN OUTPUT CAPACITOR

This application is dominated by the need for fast PWM dim­ming, requiring a circuit without any output capacitance.

OUTPUT INDUCTOR

In this example the ripple current through the LED array and
the inductor are equal. Inductance is selected to give the
smallest ripple current possible while still providing enough
Δv

signal for the CS comparator to operate correctly. De-

SNS

www.national.com 18

LM3402/LM3402HV

signing to a desired Δv
average inductor current will equal the desired average LED

of 25 mV and assuming that the

SNS

current of 350 mA yields the target current ripple in the in­ductor and LEDs:

ΔiF = ΔiL = Δv

SNS

/ R

SNS

, R

SNS

= V

SNS

/ I

F

ΔiF = 0.025 / 0.57 = 43.8 mA

With the target ripple current determined the inductance can
be chosen:

L

= [(60 – 49.2) x 2.7 x 10-6] / (0.044) = 663 µH

MIN

The closest standard inductor value is 680 µH. As with the
previous example, the average current rating should be
greater than 350 mA. Separation between the LM3402HV
drivers and the LED arrays mean that heat from the inductor
will not threaten the lifetime of the LEDs, but an overheated
inductor could still cause the LM3402HV to enter thermal
shutdown.

The inductance itself of the standard part chosen is ±20%.
With this tolerance the typical, minimum, and maximum in­ductor current ripples can be calculated:

Δi

= [(60 - 49.2) x 2.7 x 10-6] / 680 x 10

F(TYP)

= 43 mA

P-P

-6

Δi

= [(60 - 49.2) x 2.7 x 10-6] / 816 x 10

F(MIN)

= 36 mA

P-P

-6

Δi

= [(60 - 49.2) x 2.7 x 10-6] / 544 x 10

F(MAX)

= 54 mA

P-P

-6

The peak LED/inductor current is then estimated:

I

L(PEAK)

= IL + [Δi

L(MAX)

] / 2

I

= 0.35 + 0.027 = 377 mA

L(PEAK)

In the case of a short circuit across the LED array, the
LM3402HV will continue to deliver rated current through the
short but will reduce the output voltage to equal the CS pin
voltage of 200 mV. Worst-case peak current in this condition
would be equal to:

Δi

F(LED-SHORT)

= [(63 – 0.2) x 2.7 x 10-6] / 544 x 10

= 314 mA

I

= 0.35 + 0.156 = 506 mA

F(PEAK)

P-P

-6

In the case of a short at the switch node, the output, or from
the CS pin to ground the short circuit current limit will engage
at a typical peak current of 735 mA. In order to prevent in­ductor saturation during these fault conditions the inductor’s
peak current rating must be above 735 mA. A 680 µH off-the

shelf inductor rated to 1.2A (peak) and 0.72A (average) with
a DCR of 1.1Ω will be used for the green LED array.

R

SNS

A preliminary value for R
ΔiL. This value should be re-evaluated based on the calcula-

was determined in selecting

SNS

tions for ΔiF:

Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.56Ω device is the closest value, and a 0.125W, 0805
size device will handle the power dissipation of 69 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% toler­ance requirement. From the expression for LED current ac­curacy:

IF = 0.19 / 0.56 + 0.043 / 2 = 361 mA, 3% above 350 mA

INPUT CAPACITOR

Following the calculations from the Input Capacitor section,
Δv
pacitance is:

IN(MAX)

will be 1%

= 600 mV. The minimum required ca-

P-P

C

= (0.35 x 2.7 x 10-6) / 0.6 = 1.6 µF

IN(MIN)

In expectation that more capacitance will be needed to pre­vent power supply interaction a 2.2 µF ceramic capacitor
rated to 100V with X7R dielectric in an 1812 case size will be
used. From the Design Considerations section, input rms cur­rent is:

I

= 0.35 x Sqrt(0.82 x 0.18) = 134 mA

IN-RMS

Ripple current ratings for 1812 size ceramic capacitors are
typically higher than 2A, more than enough for this design.

RECIRCULATING DIODE

The input voltage of 60V ±5% requires Schottky diodes with
a reverse voltage rating greater than 60V. Some manufactur­ers provide Schottky diodes with ratings of 70, 80 or 90V;
however the next highest standard voltage rating is 100V.
Selecting a 100V rated diode provides a large safety margin
for the ringing of the switch node and also makes cross-ref­erencing of diodes from different vendors easier.

The next parameters to be determined are the forward current
rating and case size. In this example the high duty cycle (D =

49.2 / 60 = 82%) places less thermals stress on D1 and more
on the internal power MOSFET of the LM3402. The estimated
average diode current is:

ID = 0.361 x 0.18 = 65 mA

A Schottky with a forward current rating of 0.5A would be ad­equate, however at 100V the majority of diodes have a mini­mum forward current rating of 1A. To determine the proper
case size, the dissipation and temperature rise in D1 can be
calculated as shown in the Design Considerations section.
VD for a small case size such as SOD-123F in a 100V, 1A

19 www.national.com

Schottky diode at 350 mA is approximately 0.65V and the

θJA is 88°C/W. Power dissipation and temperature rise can be

calculated as:

PD = 0.065 x 0.65 = 42 mW

T

= 0.042 x 88 = 4°C

RISE

LM3402/LM3402HV

CB AND C

F

The bootstrap capacitor CB should always be a 10 nF ceramic
capacitor with X7R dielectric. A 25V rating is appropriate for
all application circuits. The linear regulator filter capacitor C
should always be a 100 nF ceramic capacitor, also with X7R
dielectric and a 25V rating.

EFFICIENCY

To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. Electrical efficiency, η, should not be confused
with the optical efficacy of the circuit, which depends upon the
LEDs themselves.

Total output power, PO, is calculated as:

PO = IF x VO = 0.361 x 49.2 = 17.76W

Conduction loss, PC, in the internal MOSFET:

2

PC = (I

F

x R

) x D = (0.3612 x 1.5) x 0.82 = 160 mW

DSON

Gate charging and VCC loss, PG, in the gate drive and linear
regulator:

PG = (600 x 10-6 + 3 x 105 x 3 x 10-9) x 60 = 90 mW

PG = (I

+ fSW x QG) x V

IN-OP

IN

Switching loss, PS, in the internal MOSFET:

PS = 0.5 x 60 x 0.361 x 40 x 10-9 x 3 x 105 = 130 mW

PS = 0.5 x VIN x IF x (tR + tF) x f

SW

AC rms current loss, P

, in the input capacitor:

CIN

2

P

= I

CIN

x ESR = (0.134)2 x 0.006 = 0.1 mW (negligible)

IN(rms)

DCR loss, PL, in the inductor

2

x DCR = 0.352 x 1.1 = 135 mW

PL = I

F

F

Recirculating diode loss, PD = 42 mW
Current Sense Resistor Loss, P

= 69 mW

SNS

Electrical efficiency, η = PO / (PO + Sum of all loss terms) =

17.76 / (17.76 + 0.62) = 96%
Temperature Rise in the LM3402HV IC is calculated as:

T

= (PC + PG + PS) x θJA = (0.16 + 0.084 + 0.13) x 200

LM3402

= 74.8°C

Layout Considerations

The performance of any switching converter depends as
much upon the layout of the PCB as the component selection.
The following guidelines will help the user design a circuit with
maximum rejection of outside EMI and minimum generation
of unwanted EMI.

COMPACT LAYOUT

Parasitic inductance can be reduced by keeping the power
path components close together and keeping the area of the
loops that high currents travel small. Short, thick traces or
copper pours (shapes) are best. In particular, the switch node
(where L1, D1, and the SW pin connect) should be just large
enough to connect all three components without excessive
heating from the current it carries. The LM3402/02HV oper­ates in two distinct cycles whose high current paths are shown
in Figure 6:

FIGURE 6. Buck Converter Current Loops

The dark grey, inner loop represents the high current path
during the MOSFET on-time. The light grey, outer loop rep­resents the high current path during the off-time.

GROUND PLANE AND SHAPE ROUTING

The diagram of Figure 6 is also useful for analyzing the flow
of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and off-

www.national.com 20

20192128

time are considered to be continuous current, while those that
carry current during the on-time or off-time only are pulsating
currents. Preference in routing should be given to the pulsat­ing current paths, as these are the portions of the circuit most
likely to emit EMI. The ground plane of a PCB is a conductor
and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the
ground net can be routed on the system ground plane with

LM3402/LM3402HV

less risk of injecting noise into other circuits. The path be­tween the input source and the input capacitor and the path
between the recirculating diode and the LEDs/current sense
resistor are examples of continuous current paths. In contrast,
the path between the recirculating diode and the input capac­itor carries a large pulsating current. This path should be
routed with a short, thick shape, preferably on the component
side of the PCB. Multiple vias in parallel should be used right
at the pad of the input capacitor to connect the component
side shapes to the ground plane. A second pulsating current
loop that is often ignored is the gate drive loop formed by the
SW and BOOT pins and capacitor CB. To minimize this loop
at the EMI it generates, keep CB close to the SW and BOOT
pins.

CURRENT SENSING

The CS pin is a high-impedance input, and the loop created
by R

, RZ (if used), the CS pin and ground should be made

SNS

as small as possible to maximize noise rejection. R
therefore be placed as close as possible to the CS and GND

SNS

should

pins of the IC.

REMOTE LED ARRAYS

In some applications the LED or LED array can be far away
(several inches or more) from the LM3402/02HV, or on a sep­arate PCB connected by a wiring harness. When an output
capacitor is used and the LED array is large or separated from
the rest of the converter, the output capacitor should be
placed close to the LEDs to reduce the effects of parasitic
inductance on the AC impedance of the capacitor. The current
sense resistor should remain on the same PCB, close to the
LM3402/02HV.

21 www.national.com

ID Part Number Type Size Parameters Qty Vendor

U1 LM3402 LED Driver MSOP-8 40V, 0.5A 1 NSC

L1 SLF7045T-330MR82 Inductor 7.0x7.0 x4.5mm

D1 CMHSH5-4 Schottky Diode SOD-123 40V, 0.5A 1 Central Semi

LM3402/LM3402HV

Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay

Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay

Cin C3216X7R1H105M Capacitor 1206 1µF 50V 1 TDK

Co C2012X7R1A225M Capacitor 0805 2.2 µF 10V 1 TDK

Rsns ERJ6BQFR75V Resistor 0805

Ron CRCW08055902F Resistor 0805

ID Part Number Type Size Parameters Qty Vendor

U1 LM3402HV LED Driver MSOP-8 75V, 0.5A 1 NSC

L1 DO5022P-684 Inductor 18.5x15.2 x7.1mm

D1 CMMSH1-100 Schottky Diode SOD-123F 100V, 1A 1 Central Semi

Cf VJ0805Y104KXXAT Capacitor 0805 100nF 10% 1 Vishay

Cb VJ0805Y103KXXAT Capacitor 0805 10nF 10% 1 Vishay

Cin C4532X7R2A225M Capacitor 1812 2.2µF 100V 1 TDK

Rsns ERJ6BQFR56V Resistor 0805

Ron CRCW08051214F Resistor 0805

TABLE 1. BOM for Design Example 1

33µH, 0.82A, 96mΩ

TABLE 2. BOM for Design Example 2

680µH, 1.2A, 1.1Ω

0.75Ω 1%

59.0 kΩ 1%

0.56Ω 1%

1.21MΩ 1%

1 TDK

1 Panasonic

1 Vishay

1 Coilcraft

1 Panasonic

1 Vishay

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Physical Dimensions inches (millimeters) unless otherwise noted

LM3402/LM3402HV

8-Lead MSOP Package

NS Package Number MUA08A

8-Lead PSOP Package

NS Package Number MRA08B

23 www.national.com

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LM3402/LM3402HV 0.5A Constant Current Buck Regulator for Driving High Power LEDs

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