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LM13700/LM13700A
Dual Operational Transconductance Amplifiers with
Linearizing Diodes and Buffers
General Description
The LM13700 series consists of two current controlled
transconductance amplifiers, each with differential inputs
and a push-pull output. The two amplifiers share common
supplies but otherwise operate independently.Linearizingdiodes are providedattheinputstoreducedistortionand allow
higher input levels. The result is a 10 dB signal-to-noise improvement referenced to 0.5 percent THD. High impedance
buffers are provided which are especially designed to
complement the dynamic range of the amplifiers.The output
buffers of the LM13700 differ from those of the LM13600 in
that their input bias currents (and hence their output DC levels) are independent of I
ABC
. This may result in performance
superior to that of the LM13600 in audio applications.
Features
n gmadjustable over 6 decades
n Excellent g
m
linearity
n Excellent matching between amplifiers
n Linearizing diodes
n High impedance buffers
n High output signal-to-noise ratio
Applications
n Current-controlled amplifiers
n Current-controlled impedances
n Current-controlled filters
n Current-controlled oscillators
n Multiplexers
n Timers
n Sample-and-hold circuits
Connection Diagram
Dual-In-Line and Small Outline Packages
DS007981-2
Top View
Order Number LM13700M, LM13700N or LM13700AN
See NS Package Number M16A or N16A
May 1998
LM13700/LM13700A Dual Operational Transconductance Amplifiers with Linearizing Diodes and
Buffers
© 1999 National Semiconductor Corporation DS007981 www.national.com
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Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage (Note 2)
LM13700 36 V
DC
or±18V
LM13700A 44 V
DC
or±22V
Power Dissipation (Note 3) T
A
=
25˚C
LM13700N, LM13700AN 570 mW
Differential Input Voltage
±
5V
Diode Bias Current (I
D
)2mA
Amplifier Bias Current (I
ABC
)2mA
Output Short Circuit Duration Continuous
Buffer Output Current (Note 4) 20 mA
Operating Temperature Range
LM13700N, LM13700AN 0˚C to +70˚C
DC Input Voltage +V
S
to −V
S
Storage Temperature Range −65˚C to +150˚C
Soldering Information
Dual-In-Line Package
Soldering (10 sec.) 260˚C
Small Outline Package
Vapor Phase (60 sec.) 215˚C
Infrared (15 sec.) 220˚C
See AN-450 “Surface Mounting Methods and Their Effect
on Product Reliability” for other methods of soldering
surface mount devices.
Electrical Characteristics (Note 5)
Parameter Conditions LM13700 LM13700A Units
Min Typ Max Min Typ Max
Input Offset Voltage (V
OS
) 0.4 4 0.4 1
Over Specified Temperature Range 2 mV
I
ABC
=
5 µA 0.3 4 0.3 1
V
OS
Including Diodes Diode Bias Current (ID)=500 µA 0.5 5 0.5 2 mV
Input Offset Change 5 µA ≤ I
ABC
≤ 500 µA 0.1 3 0.1 1 mV
Input Offset Current 0.1 0.6 0.1 0.6 µA
Input Bias Current Over Specified Temperature Range 0.4 5 0.4 5 µA
18 17
Forward 6700 9600 13000 7700 9600 12000 µmho
Transconductance (g
m
) Over Specified Temperature Range 5400 4000
g
m
Tracking 0.3 0.3 dB
Peak Output Current R
L
=
0, I
ABC
=
5µA 5 3 5 7
R
L
=
0, I
ABC
=
500 µA 350 500 650 350 500 650 µA
R
L
=
0, Over Specified Temp Range 300 300
Peak Output Voltage
Positive R
L
=
∞
,5µA≤I
ABC
≤ 500 µA +12 +14.2 +12 +14.2 V
Negative R
L
=
∞
,5µA≤I
ABC
≤ 500 µA −12 −14.4 −12 −14.4 V
Supply Current I
ABC
=
500 µA, Both Channels 2.6 2.6 mA
V
OS
Sensitivity
Positive ∆V
OS
/∆V
+
20 150 20 150 µV/V
Negative ∆V
OS
/∆V
−
20 150 20 150 µV/V
CMRR 80 110 80 110 dB
Common Mode Range
±12±
13.5
±12±
13.5 V
Crosstalk Referred to Input (Note 6) 100 100 dB
20 Hz
<f<
20 kHz
Differential Input Current I
ABC
=
0, Input
=
±
4V 0.02 100 0.02 10 nA
Leakage Current I
ABC
=
0 (Refer to Test Circuit) 0.2 100 0.2 5 nA
Input Resistance 10 26 10 26 kΩ
Open Loop Bandwidth 2 2 MHz
Slew Rate Unity Gain Compensated 50 50 V/µs
Buffer Input Current (Note 6) 0.5 2 0.5 2 µA
Peak Buffer Output Voltage (Note 6) 10 10 V
Note 1: “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits.
Note 2: For selections to a supply voltage above
±
22V, contact factory.
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Page 3
Electrical Characteristics (Note 5) (Continued)
Note 3: For operation at ambient temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance,
junction to ambient, as follows: LM13700N, 90˚C/W; LM13700M, 110˚C/W.
Note 4: Buffer output current should be limited so as to not exceed package dissipation.
Note 5: These specifications apply for V
S
=
±
15V,T
A
=
25˚C, amplifier bias current (I
ABC
)=500 µA, pins 2 and 15 open unless otherwise specified. The inputs to
the buffers are grounded and outputs are open.
Note 6: These specifications apply for V
S
=
±
15V, I
ABC
=
500 µA, R
OUT
=
5kΩconnected from the buffer output to −V
S
and the input of the buffer is connected
to the transconductance amplifier output.
Schematic Diagram
Typical Performance Characteristics
One Operational Transconductance Amplifier
DS007981-1
Input Offset Voltage
DS007981-38
Input Offset Current
DS007981-39
Input Bias Current
DS007981-40
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Typical Performance Characteristics (Continued)
Peak Output Current
DS007981-41
Peak Output Voltage and
Common Mode Range
DS007981-42
Leakage Current
DS007981-43
Input Leakage
DS007981-44
Transconductance
DS007981-45
Input Resistance
DS007981-46
Amplifier Bias Voltage vs
Amplifier Bias Current
DS007981-47
Input and Output Capacitance
DS007981-48
Output Resistance
DS007981-49
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Typical Performance Characteristics (Continued)
Circuit Description
The differential transistor pair Q4and Q5form a transconductance stage in that the ratio of their collector currents is
defined by the differential input voltage according to the
transfer function:
(1)
where V
IN
is the differential input voltage, kT/q is approxi-
mately 26 mV at 25˚C and I
5
and I4are the collector currents
of transistors Q
5
and Q4respectively. With the exception of
Distortion vs Differential
Input Voltage
DS007981-50
Voltage vs Amplifier
Bias Current
DS007981-51
Output Noise vs Frequency
DS007981-52
Unity Gain Follower
DS007981-5
Leakage Current Test Circuit
DS007981-6
Differential Input Current Test Circuit
DS007981-7
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Page 6
Circuit Description (Continued)
Q
3
and Q13, all transistors and diodes are identical in size.
Transistors Q
1
and Q2with Diode D1form a current mirror
which forces the sum of currents I
4
and I5to equal I
ABC
:
I
4+I5
=
I
ABC
(2)
where I
ABC
is the amplifier bias current applied to the gain
pin.
For small differential input voltages the ratio of I
4
and I5approaches unity and the Taylor series of the In function can be
approximated as:
(3)
(4)
Collector currents I
4
and I5are not very useful by themselves
and it is necessary to subtract one current from the other.
The remaining transistors and diodes formthree current mirrors that produce an output current equalto I
5
minus I4thus:
(5)
The term in brackets is then the transconductance of the amplifier and is proportional to I
ABC
.
Linearizing Diodes
For differential voltages greater than a few millivolts,
Equa-
tion (3)
becomes less valid and the transconductance be-
comes increasingly nonlinear.
Figure 1
demonstrates how
the internal diodes can linearize the transfer function of the
amplifier. For convenience assume the diodes are biased
with current sources and the input signal is in the form of current I
S
. Since the sum of I4and I5is I
ABC
and the difference
is I
OUT
, currents I4and I5can be written as follows:
Since the diodes and the input transistors have identical geometries and are subject to similar voltages and temperatures, the following is true:
(6)
Notice that in deriving
Equation (6)
no approximations have
been made and there are no temperature-dependent terms.
The limitations are that the signal current not exceed I
D
/2
and that the diodes be biased with currents. In practice, replacing the current sources with resistorswill generate insignificant errors.
Applications:
Voltage Controlled Amplifiers
Figure 2
shows how the linearizing diodes can be used in a
voltage-controlled amplifier. To understand the input biasing,
it is best to consider the 13 kΩ resistor as a current source
and use a Thevenin equivalent circuit as shown in
Figure 3
.
This circuit is similar to
Figure 1
and operates the same. The
potentiometer in
Figure 2
is adjusted to minimize the effects
of the control signal at the output.
For optimum signal-to-noise performance, I
ABC
should be as
large as possible asshown bythe OutputVoltage vs. Amplifier Bias Current graph. Larger amplitudes of input signal
also improve the S/N ratio. The linearizing diodes help here
by allowing larger input signals for the same output distortion
as shown by the Distortion vs. Differential Input Voltage
graph. S/N may be optimized by adjusting the magnitude of
the input signal via R
IN
(
Figure 2
) until the output distortion is
below some desired level. The output voltage swing can
then be set at any level by selecting R
L
.
Although the noise contribution of the linearizing diodes is
negligible relative to the contribution of the amplifier’s internal transistors, I
D
should be as large as possible. This mini-
mizes the dynamic junction resistance of the diodes (r
e
) and
DS007981-8
FIGURE 1. Linearizing Diodes
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Page 7
Applications:
Voltage Controlled Amplifiers
(Continued)
maximizes their linearizing action when balanced against
R
IN
. A value of 1 mA is recommended for IDunless the spe-
cific application demands otherwise.
Stereo Volume Control
The circuit of
Figure 4
uses the excellent matching of the two
LM13700 amplifiers to provide aStereo Volume Control with
a typical channel-to-channel gain tracking of 0.3 dB. R
P
is
provided to minimize the output offset voltage and may be
replaced with two 510Ω resistors in AC-coupled applications.
For the component values given, amplifier gain is derivedfor
Figure 2
as being:
If VCis derived from a second signal source then the circuit
becomes an amplitude modulator or two-quadrant multiplier
as shown in
Figure 5
, where:
The constant term in the above equation may be cancelled
by feeding I
SxIDRC
/2(V− + 1.4V) into IO. The circuit of
Fig-
ure 6
adds RMto provide this current, resulting in a
four-quadrant multiplier where R
C
is trimmed such that V
O
=
0V for V
IN2
=
0V. R
M
also serves as the load resistor for IO.
DS007981-9
FIGURE 2. Voltage Controlled Amplifier
DS007981-10
FIGURE 3. Equivalent VCA Input Circuit
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Stereo Volume Control (Continued)
DS007981-11
FIGURE 4. Stereo Volume Control
DS007981-12
FIGURE 5. Amplitude Modulator
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Stereo Volume Control (Continued)
Noting that the gain of the LM13700 amplifier of
Figure 3
may be controlled by varying the linearizing diode current I
D
as well as by varyingI
ABC
,
Figure 7
shows an AGC Amplifier
using this approach. As V
O
reaches a high enough amplitude
(3V
BE
) to turn on the Darlington transistors and the lineariz-
ing diodes, the increase in I
D
reduces the amplifier gain so
as to hold V
O
at that level.
Voltage Controlled Resistors
An Operational Transconductance Amplifier (OTA) may be
used to implement a Voltage Controlled Resistor as shown in
Figure 8
. A signal voltage applied at RXgenerates a VINto
the LM13700 which is then multiplied by theg
m
of the ampli-
fier to produce an output current, thus:
where gm≈ 19.2I
ABC
at 25˚C. Note that the attenuation of V
O
by R and RAis necessary to maintain VINwithin the linear
range of the LM13700 input.
Figure 9
shows a similar VCR where the linearizing diodes
are added, essentially improving the noise performance of
the resistor. A floating VCR is shown in
Figure 10
, where
each “end” of the “resistor” may be at any voltage within the
output voltage range of the LM13700.
DS007981-13
FIGURE 6. Four-Quadrant Multiplier
DS007981-14
FIGURE 7. AGC Amplifier
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Voltage Controlled Resistors (Continued)
Voltage Controlled Filters
OTA’s are extremely useful for implementing voltage controlled filters, with the LM13700 having the advantage that
the required buffers are included on the I.C. The VCLo-Pass
Filter of
Figure 11
performs as a unity-gain buffer amplifier at
frequencies below cut-off, with the cut-off frequency being
the point at which X
C/gm
equals the closed-loop gain of (R/
R
A
).At frequencies above cut-off the circuit provides a single
RC roll-off (6 dB per octave) of the input signal amplitude
with a −3 dB point defined by the given equation, where g
m
is again 19.2 x I
ABC
at room temperature.
Figure 12
shows a
VC High-Pass Filter which operates in much the same manner, providing a single RC roll-off below the defined cut-off
frequency.
Additional amplifiers may be usedto implement higher order
filters as demonstrated by the two-pole Butterworth Lo-Pass
Filter of
Figure 13
and the state variable filter of
Figure 14
.
Due to the excellent g
m
tracking of the two amplifiers, these
filters perform well over several decades of frequency.
DS007981-15
FIGURE 8. Voltage Controlled Resistor, Single-Ended
DS007981-16
FIGURE 9. Voltage Controlled Resistor with Linearizing Diodes
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Voltage Controlled Filters (Continued)
DS007981-17
FIGURE 10. Floating Voltage Controlled Resistor
DS007981-18
FIGURE 11. Voltage Controlled Low-Pass Filter
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Voltage Controlled Filters (Continued)
DS007981-19
FIGURE 12. Voltage Controlled Hi-Pass Filter
DS007981-20
FIGURE 13. Voltage Controlled 2-Pole Butterworth Lo-Pass Filter
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Voltage Controlled Filters (Continued)
Voltage Controlled Oscillators
The classic Triangular/Square Wave VCO of
Figure 15
is
one of a variety of Voltage Controlled Oscillators which may
be built utilizing the LM13700. With the component values
shown, this oscillator provides signals from 200 kHz to below
2HzasI
C
is varied from 1 mA to 10 nA. The output ampli-
tudes are set byI
AxRA
. Note that thepeak differentialinput
voltage must be less than 5V to prevent zenering the inputs.
A few modifications to this circuit produce the ramp/pulse
VCO of
Figure 16
. When VO2is high, IFis added to ICto increase amplifier A1’s bias current and thus to increase the
charging rate of capacitor C. When V
O2
is low, IFgoes to
zero and the capacitor discharge current is set by I
C
.
The VC Lo-Pass Filter of
Figure 11
may be used to produce
a high-quality sinusoidal VCO. The circuit of
Figure 16
employs two LM13700 packages, with three of the amplifiers
configured as lo-pass filters and the fourth as a limiter/
inverter. The circuit oscillates at the frequency at which the
loop phase-shift is 360˚ or 180˚ for the inverter and 60˚ per
filter stage. This VCO operates from 5 Hz to 50 kHz with less
than 1%THD.
DS007981-21
FIGURE 14. Voltage Controlled State Variable Filter
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Voltage Controlled Oscillators (Continued)
DS007981-22
FIGURE 15. Triangular/Square-Wave VCO
DS007981-23
FIGURE 16. Ramp/Pulse VCO
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Voltage Controlled Oscillators (Continued)
Additional Applications
Figure 19
presents an interesting one-shot which draws no
power supply current until it is triggered. A positive-going trigger pulse of at least 2V amplitude turns on the amplifier
through R
B
and pulls the non-inverting input high. The amplifier regenerates and latches its output high until capacitor C
charges to the voltage level on the non-inverting input. The
output then switches low, turning off the amplifier and discharging the capacitor. The capacitor discharge rate is
speeded up by shorting the diode biaspin to the invertinginput so that an additional discharge current flows through D
I
when the amplifier output switches low. A special feature of
this timer is that the other amplifier, when biased from V
O
,
can perform another function and draw zero stand-by power
as well.
DS007981-24
FIGURE 17. Sinusoidal VCO
DS007981-25
Figure 18
shows how to build a VCO using one amplifier when the other
amplifier is needed for another function.
FIGURE 18. Single Amplifier VCO
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Additional Applications (Continued)
The operation of the multiplexer of
Figure 20
is very straight-
forward. When A1 is turned on it holds V
O
equal to V
IN1
and
when A2 is supplied with bias current then itcontrols V
O.CC
and RCserve to stabilize the unity-gain configuration of amplifiers A1 and A2. The maximum clock rate is limited to
about 200 kHz by the LM13700 slew rate into 150 pF when
the (V
IN1–VIN2
) differential is at its maximum allowable value
of 5V.
The Phase-Locked Loop of
Figure 21
uses the four-quadrant
multiplier of
Figure 6
and the VCO of
Figure 18
to produce a
PLL with a
±
5%hold-in range and an input sensitivity of
about 300 mV.
DS007981-26
FIGURE 19. Zero Stand-By Power Timer
DS007981-27
FIGURE 20. Multiplexer
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Additional Applications (Continued)
The Schmitt Trigger of
Figure 22
uses the amplifier output
current into R to set the hysteresis of the comparator; thus
V
H
=
2xRxI
B
. VaryingIBwill produce a Schmitt Trigger with
variable hysteresis.
DS007981-28
FIGURE 21. Phase Lock Loop
DS007981-29
FIGURE 22. Schmitt Trigger
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Additional Applications (Continued)
Figure 23
shows a Tachometeror Frequency-to-Voltage converter. Whenever A1 is toggled by a positive-going input, an
amount of charge equal to(V
H–VL)Ct
is sourced into Cfand
R
t
. This once per cycle charge is then balanced by the cur-
rent of V
O/Rt
. The maximum FINis limited by the amount of
time required to charge C
t
from VLto VHwith a current of IB,
where V
L
and VHrepresent the maximum low and maximum
high output voltage swing of the LM13700. D1 is added to
provide a discharge path for C
t
when A1 switches low.
The Peak Detector of
Figure 24
usesA2 to turnon A1 when-
ever V
IN
becomes more positive than VO. A1 then charges
storage capacitor C to hold V
O
equal to VINPK. Pulling the
output of A2 low through D1 serves to turn off A1 so that V
O
remains constant.
The Ramp-and-Hold of
Figure 26
sources IBinto capacitor C
whenever the input toA1 is brought high, giving a ramp-rate
of about 1V/ms for the component values shown.
The true-RMS converter of
Figure 27
is essentially an automatic gain control amplifier which adjusts its gain such that
the AC power at the output of amplifier A1 is constant. The
output power of amplifierA1 is monitored by squaring amplifier A2 and the average compared to a reference voltage
with amplifier A3. The output of A3 provides bias current to
the diodes of A1 to attenuate the input signal. Because the
output power of A1 is held constant, the RMS value is constant and the attenuation is directly proportional to the RMS
value of the input voltage. The attenuation is also proportional to the diodebias current.Amplifier A4 adjusts the ratio
of currents through the diodes to be equal and therefore the
voltage at the output ofA4 is proportional to the RMS value
of the input voltage. The calibration potentiometer is set such
that V
O
reads directly in RMS volts.
DS007981-30
FIGURE 23. Tachometer
DS007981-31
FIGURE 24. Peak Detector and Hold Circuit
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Additional Applications (Continued)
DS007981-32
FIGURE 25. Sample-Hold Circuit
DS007981-33
FIGURE 26. Ramp and Hold
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Additional Applications (Continued)
The circuit of
Figure 28
is a voltage reference of variable
Temperature Coefficient. The 100 kΩ potentiometer adjusts
the output voltage which has a positive TC above 1.2V, zero
TC at about 1.2V, and negative TC below 1.2V. This is accomplished by balancing the TC of the A2 transfer function
against the complementary TC of D1.
The wide dynamic range of the LM13700 allows easy control
of the output pulse width in the Pulse Width Modulator of
Fig-
ure 29
.
For generating I
ABC
over a range of 4 to 6 decades of cur-
rent, the system of
Figure 30
provides a logarithmic current
out for a linear voltage in.
Since the closed-loop configurationensures thatthe inputto
A2 is held equal to 0V, the output current of A1 is equal to
I
3
=
−V
C/RC
.
The differential voltage between Q1 and Q2is attenuated by
the R1,R2 network so that A1 may be assumed to be operating within its linear range. From
Equation (5)
, the input volt-
age to A1 is:
The voltage on the base of Q1 is then
The ratio of the Q1 and Q2 collector currents is defined by:
Combining and solving for I
ABC
yields:
This logarithmic current can be used to bias the circuit of
Fig-
ure 4
to provide temperature independent stereo attenuation
characteristic.
DS007981-34
FIGURE 27. True RMS Converter
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Additional Applications (Continued)
DS007981-35
FIGURE 28. Delta VBE Reference
DS007981-36
FIGURE 29. Pulse Width Modulator
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Additional Applications (Continued)
DS007981-37
FIGURE 30. Logarithmic Current Source
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Physical Dimensions inches (millimeters) unless otherwise noted
S.O. Package (M)
Order Number LM13700M
NS Package Number M16A
Molded Dual-In-Line Package (N)
Order Number LM13700N or LM13700AN
NS Package Number N16A
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Page 24
Notes
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NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
National Semiconductor
Corporation
Americas
Tel: 1-800-272-9959
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Fax: 81-3-5639-7507
www.national.com
LM13700/LM13700A Dual Operational Transconductance Amplifiers with Linearizing Diodes and
Buffers
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
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