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LM13600
Dual Operational Transconductance Amplifiers with
Linearizing Diodes and Buffers
General Description
The LM13600 series consists of two current controlled
transconductance amplifiers each with differential inputs and
a push-pull output. The two amplifiers share common supplies but otherwise operate independently. Linearizing diodes are providedattheinputs to reduce distortion and allow
higher input levels. The result is a 10 dB signal-to-noise improvement referenced to 0.5 percent THD. Controlled impedance buffers which are especially designed to complement the dynamic range of the amplifiers are provided.
Features
n gmadjustable over 6 decades
n Excellent g
linearity
m
Connection Diagram
Dual-In-Line and Small Outline Packages
n Excellent matching between amplifiers
n Linearizing diodes
n Controlled impedance buffers
n High output signal-to-noise ratio
Applications
n Current-controlled amplifiers
n Current-controlled impedances
n Current-controlled filters
n Current-controlled oscillators
n Multiplexers
n Timers
n Sample and hold circuits
LM13600 Dual Operational Transconductance Amplifiers with Linearizing Diodes and Buffers
May 1998
DS007980-2
Order Number LM13600M, LM13600N or LM13600AN
See NS Package Number M16A or N16A
© 1999 National Semiconductor Corporation DS007980 www.national.com
Top View
Page 2
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage (Note 2)
LM13600 36 V
LM13600A 44 V
=
Power Dissipation (Note 3) T
25˚C 570 mW
A
Differential Input Voltage
Diode Bias Current (I
Amplifier Bias Current (I
)2mA
D
)2mA
ABC
Output Short Circuit Duration Continuous
DC
DC
or±18V
or±22V
±
Operating Temperature Range 0˚C to +70˚C
DC Input Voltage +V
Storage Temperature Range −65˚C to +150˚C
Soldering Information
Dual-In-Line Package
Soldering (10 seconds) 260˚C
Small Outline Package
Vapor Phase (60 seconds) 215˚C
5V
Infrared (15 seconds) 220˚C
See AN-450 “Surface Mounting Methods and Their Effect
on Product Reliability” for other methods of soldering
surface mount devices.
S
to −V
Buffer Output Current (Note 4) 20 mA
Electrical Characteristics (Note 5)
Parameter Conditions LM13600 LM13600A Units
Min Typ Max Min Typ Max
Input Offset Voltage (V
V
Including Diodes Diode Bias Current (ID)=500 µA 0.5 5 0.5 2 mV
OS
Input Offset Change 5 µA ≤ I
Input Offset Current 0.1 0.6 0.1 0.6 µA
Input Bias Current 0.4 5 0.4 5 µA
Forward
Transconductance (g
g
Tracking 0.3 0.3 dB
m
Peak Output Current R
Peak Output Voltage
Positive R
Negative R
Supply Current I
V
Sensitivity
OS
Positive ∆ V
Negative ∆ V
CMRR 80 110 80 110 dB
Common Mode Range
Crosstalk Referred to Input (Note 6) 100 100 dB
Differential Input Current I
Leakage Current I
Input Resistance 10 26 10 26 kΩ
Open Loop Bandwidth 2 2 MHz
Slew Rate Unity Gain Compensated 50 50 V/µs
Buffer Input Current (Note 6), Except I
Peak Buffer Output Voltage (Note 6) 10 10 V
) 0.4 4 0.4 1 mV
OS
Over Specified Temperature Range 2 mV
=
I
5 µA 0.3 4 0.3 1 mV
ABC
≤ 500 µA 0.1 3 0.1 1 mV
ABC
Over Specified Temperature Range 1 8 1 7 µA
) 6700 9600 13000 7700 9600 12000 µmho
m
Over Specified Temperature Range 5400 4000 µmho
=
L
=
R
L
=
R
L
Range
=
L
=
L
ABC
OS
OS
20 Hz
ABC
ABC
=
0, I
0, I
0, Over Specified Temp
∞
∞
=
5µA 5 3 5 7 µA
ABC
=
500 µA 350 500 650 350 500 650 µA
ABC
300 300 µA
,5µA≤I
,5µA≤I
≤ 500 µA +12 +14.2 +12 +14.2 V
ABC
≤ 500 µA −12 −14.4 −12 −14.4 V
ABC
500 µA, Both Channels 2.6 2.6 mA
/∆V+ 20 150 20 150 µV/V
/∆V− 20 150 20 150 µV/V
±12±
13.5
<f<
20 kHz
=
=
=
±
0, Input
4V 0.02 100 0.02 10 nA
0 (Refer to Test Circuit) 0.2 100 0.2 5 nA
=
0 µA 0.2 0.4 0.2 0.4 µA
ABC
±12±
13.5 V
S
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Page 3
Electrical Characteristics (Note 5) (Continued)
Note 1: “Absolute Maximum Ratings” indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits.
Note 2: For selections to a supply voltage above
Note 3: For operating at high temperatures, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of 175˚C/W
which applies for the device soldered in a printed circuit board, operating in still air.
Note 4: Buffer output current should be limited so as to not exceed package dissipation.
Note 5: These specifications apply for V
the buffers are grounded and outputs are open.
Note 6: These specifications apply for V
to the transconductance amplifier output.
=
±
S
=
±
S
±
22V, contact factory.
=
15V,T
25˚C, amplifier bias current (I
A
=
15V, I
ABC
500 µA, R
)=500 µA, pins 2 and 15 open unless otherwise specified. The inputs to
ABC
=
5kΩconnected from the buffer output to −V
OUT
and the input of the buffer is connected
S
Schematic Diagram
One Operational Transconductance Amplifier
Typical Performance Characteristics
Input Offset Voltage
DS007980-39
Input Offset Current
DS007980-40
DS007980-1
Input Bias Current
DS007980-41
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Page 4
Typical Performance Characteristics (Continued)
Peak Output Current
Input Leakage
Amplifier Bias Voltage vs
Amplifier Bias Current
DS007980-42
DS007980-45
Peak Output Voltage and
Common Mode Range
Transconductance
Input and Output Capacitance
DS007980-43
DS007980-46
Leakage Current
DS007980-44
Input Resistance
DS007980-47
Output Resistance
DS007980-48
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DS007980-49
DS007980-50
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Typical Performance Characteristics (Continued)
Distortion vs Differential
Input Voltage
DS007980-51
Voltage vs Amplifier Bias Current
DS007980-52
Unity Gain Follower
Output Noise vs Frequency
DS007980-53
Leakage Current Test Circuit
DS007980-5
Differential Input Current Test Circuit
DS007980-7
DS007980-6
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Page 6
Circuit Description
The differential transistor pair Q4and Q5form a transconductance stage in that the ratio of their collector currents is
defined by the differential input voltage according to the
transfer function:
(1)
where V
mately 26 mV at 25˚C and I
of transistors Q
Q
Transistors Q
which forces the sum of currents I
where I
pin.
For small differential input voltages the ratio of I
proaches unity and the Taylorseries of the In function can be
approximated as:
is the differential input voltage, kT/q is approxi-
IN
and Q4respectively. With the exception of
and Q13, all transistors and diodes are identical in size.
3
5
and Q2with Diode D1form a current mirror
1
is the amplifier bias current applied to the gain
ABC
and I4are the collector currents
5
and I5to equal I
4
=
I
I
4+I5
ABC
ABC
and I5ap-
4
;
(2)
(3)
(5)
The term in brackets is then the transconductance of the amplifier and is proportional to I
ABC
.
Linearizing Diodes
For differential voltages greater than a few millivolts,
tion (3)
becomes less valid and the transconductance be-
comes increasingly nonlinear.
Figure 1
demonstrates how
the internal diodes can linearize the transfer function of the
amplifier. For convenience assume the diodes are biased
with current sources and the input signal is in the form of current I
. Since the sum of I4and I5is I
S
is I
, currents I4and I5can be written as follows:
OUT
and the difference
ABC
Since the diodes and the input transistors have identical geometries and are subject to similar voltages and temperatures, the following is true:
Equa-
(4)
Collector currents I
and it is necessary to subtract one current from the other.
and I5are not very useful by themselves
4
The remaining transistors and diodes form three current mirrors that produce an output current equal to I
minus I4thus:
5
FIGURE 1. Linearizing Diodes
(6)
DS007980-8
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Page 7
Linearizing Diodes (Continued)
Notice that in deriving Equation 6 no approximations have
been made and there are no temperature-dependent terms.
The limitations are that the signal current not exceed I
and that the diodes be biased with currents. In practice, re-
D
placing the current sources with resistors will generate insignificant errors.
Controlled Impedance Buffers
The upper limit of transconductance is defined by the maximum value of I
the amplifier will function therefore determines the overall
dynamic range. At very low values of I
very low input bias current is desirable. An FET follower satisfies the low input current requirement, but is somewhat
non-linear for large voltage swing. The controlled impedance
buffer is a Darlington which modifies its input bias current to
suit the need. For low values of I
rent is minimal. At higher levels of I
up Q
with a current proportional to I
12
When I
ABC
buffer will shift. In audio applications where I
suddenly, this shift may produce an audible “pop”. For these
applications the LM13700 may produce superior results.
(2 mA). The lowest value of I
ABC
, a buffer which has
ABC
, the buffer’s input cur-
ABC
, transistor Q3biases
ABC
is changed, the DC level of the Darlington output
ABC
for which
ABC
for fast slew rate.
is changed
ABC
Applications-Voltage Controlled
Amplifiers
Figure 2
/2
shows how the linearizing diodes can be used in a
voltage-controlled amplifier. To understand the input biasing,
it is best to consider the 13 kΩ resistor as a current source
and use a Thevenin equivalent circuit as shown in
This circuit is similar to
potentiometer in
Figure 1
Figure 2
and operates the same. The
is adjusted to minimize the effects
Figure 3
of the control signal at the output.
For optimum signal-to-noise performance, I
large as possible as shown by the Output Voltage vs. Ampli-
should be as
ABC
fier Bias Current graph. Larger amplitudes of input signal
also improve the S/N ratio. The linearizing diodes help here
by allowing larger input signals for the same output distortion
as shown by the Distortion vs. Differential Input Voltage
graph. S/N may be optimized by adjusting the magnitude of
the input signal via R
below some desired level. The output voltage swing can
then be set at any level by selecting R
(
IN
Figure 2
) until the output distortion is
.
L
Although the noise contribution of the linearizing diodes is
negligible relative to the contribution of the amplifier’s internal transistors, I
mizes the dynamic junction resistance of the diodes (r
maximizes their linearizing action when balanced against
R
. A value of 1 mA is recommended for IDunless the spe-
IN
cific application demands otherwise.
should be as large as possible. This mini-
D
) and
e
.
FIGURE 2. Voltage Controlled Amplifier
DS007980-9
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Page 8
Applications-Voltage Controlled Amplifiers (Continued)
FIGURE 3. Equivalent VCA Input Circuit
Stereo Volume Control
The circuit of
LM13600 amplifiers to provide a Stereo VolumeControl with
a typical channel-to-channel gain tracking of 0.3 dB. R
provided to minimize the output offset voltage and may be
replaced with two 510Ω resistors in AC-coupled applications.
For the component values given, amplifier gain is derived for
Figure 2
Figure 4
as being:
uses the excellent matching of the two
P
If VCis derived from a second signal source then the circuit
becomes an amplitude modulator or two-quadrant multiplier
as shown in
Figure 5
is
DS007980-10
, where:
FIGURE 4. Stereo Volume Control
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DS007980-11
Page 9
Stereo Volume Control (Continued)
FIGURE 5. Amplitude Modulator
The constant term in the above equation may be cancelled
by feeding I
ure 6
adds RMto provide this current, resulting in a
four-quadrant multiplier where R
0V for V
IN2
/2 (V−+ 1.4V) into IO. The circuit of
SxIDRC
is trimmed such that V
=
also serves as the load resistor for IO.
0V. R
M
C
Noting that the gain of the LM13600 amplifier of
Fig-
O
Figure 3
may be controlled by varying the linearizing diode current I
as well as by varying I
using this approach. As V
(3 V
) to turn on the Darlington transistors and the lineariz-
BE
ing diodes, the increase in I
as to hold V
at that level.
O
,
Figure 7
ABC
O
shows an AGC Amplifier
reaches a high enough amplitude
reduces the amplifier gain so
D
Voltage Controlled Resistors
An Operational Transconductance Amplifier (OTA) may be
used to implement a VoltageControlled Resistor as shown in
DS007980-12
Figure 8
. A signal voltage applied at RXgenerates a VINto
the LM13600 which is then multiplied by the g
fier to produce an output current, thus:
=
D
where gm≈ 19.2 I
V
by R and RAis necessary to maintain VINwithin the linear
O
range of the LM13600 input.
Figure 9
shows a similar VCR where the linearizing diodes
at 25˚C. Note that the attenuation of
ABC
are added, essentially improving the noise performance of
the resistor. A floating VCR is shown in
each “end” of the “resistor” may be at any voltage within the
output voltage range of the LM13600.
of the ampli-
m
Figure 10
, where
FIGURE 6. Four-Quadrant Multiplier
DS007980-13
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Page 10
Voltage Controlled Resistors (Continued)
FIGURE 7. AGC Amplifier
DS007980-14
FIGURE 8. Voltage Controlled Resistor, Single-Ended
Voltage Controlled Filters
OTA’s are extremely useful for implementing voltage controlled filters, with the LM13600 having the advantage that
the required buffers are included on the I.C. The VC Lo-Pass
Filter of
Figure 11
frequencies below cut-off, with the cut-off frequency being
the point at which X
(R/R
). At frequencies above cut-off the circuit provides a
A
single RC roll-off (6 dB per octave) of the input signal amplitude with a −3 dB point defined by the given equation, where
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performs as a unity-gain buffer amplifier at
equals the closed-loop gain of
C/gm
DS007980-15
g
is again 19.2 x I
m
shows a VC High-Pass Filter which operates in much the
at room temperature.
ABC
Figure 12
same manner, providing a single RC roll-off below the defined cut-off frequency.
Additional amplifiers may be used to implement higher order
filters as demonstrated by the two-pole Butterworth Lo-Pass
Filter of
Figure 13
Due to the excellent g
varied bias of the buffer Darlingtons, these filters perform
and the state variable filter of
tracking of the two amplifiers and the
m
Figure 14
well over several decades of frequency.
.
Page 11
Voltage Controlled Filters (Continued)
FIGURE 9. Voltage Controlled Resistor with Linearizing Diodes
DS007980-16
FIGURE 10. Floating Voltage Controlled Resistor
FIGURE 11. Voltage Controlled Low-Pass Filter
DS007980-17
DS007980-18
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Voltage Controlled Filters (Continued)
FIGURE 12. Voltage Controlled Hi-Pass Filter
DS007980-19
FIGURE 13. Voltage Controlled 2-Pole Butterworth Lo-Pass Filter
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DS007980-20
Page 13
Voltage Controlled Filters (Continued)
FIGURE 14. Voltage Controlled State Variable Filter
Voltage Controlled Oscillators
The classic Triangular/Square Wave VCO of
one of a variety of Voltage Controlled Oscillators which may
be built utilizing the LM13600. With the component values
shown, this oscillator provides signals from 200 kHz to below
2HzasI
tudes are set by I
voltage must be less than 5V to prevent zenering the inputs.
is varied from 1 mA to 10 nA. The output ampli-
C
. Note that the peak differential input
AxRA
A few modifications to this circuit produce the ramp/pulse
VCO of
Figure 16
. When VO2is high, IFis added to ICto increase amplifier A1’s bias current and thus to increase the
charging rate of capacitor C. When V
zero and the capacitor discharge current is set by I
Figure 15
is low, IFgoes to
O2
.
C
The VC Lo-Pass Filter of
a high-quality sinusoidal VCO. The circuit of
is
ploys two LM13600 packages, with three of the amplifiers
Figure 11
may be used to produce
configured as lo-pass filters and the fourth as a limiter/
inverter. The circuit oscillates at the frequency at which the
loop phase-shift is 360˚ or 180˚ for the inverter and 60˚ per
filter stage. This VCO operates from 5 Hz to 50 kHz with less
than 1%THD.
DS007980-21
Figure 16
em-
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Voltage Controlled Oscillators (Continued)
FIGURE 15. Triangular/Square-Wave VCO
DS007980-22
FIGURE 16. Ramp/Pulse VCO
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DS007980-23
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Voltage Controlled Oscillators (Continued)
FIGURE 17. Sinusoidal VCO
DS007980-25
FIGURE 18. Single Amplifier VCO
Figure 18
when the other amplifier is needed for another function.
shows how to build a VCO using one amplifier
DS007980-24
when the amplifier output switches low. A special feature of
this timer is that the other amplifier, when biased from V
can perform another function and draw zero stand-by power
as well.
The operation of the multiplexer of
forward. When A1 is turned on it holds V
when A2 is supplied with bias current then it controls V
and RCserve to stabilize the unity-gain configuration of am-
Figure 20
O
is very straight-
equal to V
IN1
and
O.CC
plifiers A1 and A2. The maximum clock rate is limited to
about 200 kHz by the LM13600 slew rate into 150 pF when
the (V
of 5V.
The Phase-Locked Loop of
multiplier of
PLL with a
) differential is at its maximum allowable value
IN1-VIN2
Figure 6
Figure 21
and the VCO of
±
5%hold-in range and an input sensitivity of
uses the four-quadrant
Figure 18
to produce a
about 300 mV.
,
O
Additional Applications
Figure 19
power supply current until it is triggered.Apositive-going trigger pulse of at least 2V amplitude turns on the amplifier
through R
fier regenerates and latches its output high until capacitor C
charges to the voltage level on the non-inverting input. The
output then switches low, turning off the amplifier and discharging the capacitor. The capacitor discharge rate is increased by shorting the diode bias pin to the inverting input
so that an additional discharge current flows through D
presents an interesting one-shot which draws no
and pulls the non-inverting input high. The ampli-
B
DS007980-26
I
FIGURE 19. Zero Stand-By Power Timer
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Page 16
Additional Applications (Continued)
DS007980-27
FIGURE 20. Multiplexer
FIGURE 21. Phase Lock Loop
The Schmitt Trigger of
current into R to set the hysteresis of the comparator; thus
=
V
2xRxI
H
variable hysteresis.
Figure 23
B
shows a Tachometeror Frequency-to-Voltagecon-
Figure 22
. Varying IBwill produce a Schmitt Trigger with
uses the amplifier output
verter. Whenever A1 is toggled by a positive-going input, an
amount of charge equal to (V
R
. This once-per-cycle charge is then balanced by the cur-
t
rent of V
time required to charge C
where V
high output voltage swing of the LM13600. D1 is added to
. The maximum fINis limited by the amount of
O/Rt
and VHrepresent the maximum low and maximum
L
from VLto VHwith a current of IB,
t
provide a discharge path for C
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is sourced into Cfand
H−VL)Ct
when A1 switches low.
t
DS007980-28
The Peak Detector of
ever V
becomes more positive than VO. A1 then charges
IN
storage capacitor C to hold V
tion to observe when using this circuit: the Darlington transis-
Figure 24
usesA2 to turn on A1 when-
equal to VINPK. One precau-
O
tor used must be on the same side of the package as A2
since the A1 Darlington will be turned on and off with A1.
Pulling the output ofA2 low through D1 serves to turn off A1
so that V
remains constant.
O
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Additional Applications (Continued)
FIGURE 22. Schmitt Trigger
DS007980-29
FIGURE 23. Tachometer
FIGURE 24. Peak Detector and Hold Circuit
The Sample-Hold circuit of
Figure 25
also requires that the
Darlington buffer used be from the other (A2) half of the
package and that the corresponding amplifier be biased on
continuously. The Ramp-and-Hold of
DS007980-30
DS007980-31
Figure 26
sources I
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B
Page 18
Additional Applications (Continued)
into capacitor C whenever the input to A1 is brought high,
giving a ramp-rate of about 1 V/ms for the component values
shown.
The true-RMS converter of
matic gain control amplifier which adjusts its gain such that
the AC power at the output of amplifier A1 is constant. The
output power of amplifier A1 is monitored by squaring amplifier A2 and the average compared to a reference voltage
with amplifier A3. The output of A3 provides bias current to
the diodes of A1 to attenuate the input signal. Because the
output power of A1 is held constant, the RMS value is constant and the attentuation is directly proportional to the RMS
value of the input voltage. The attenuation is also proportional to the diode bias current. Amplifier A4 adjusts the ratio
of currents through the diodes to be equal and therefore the
voltage at the output of A4 is proportional to the RMS value
of the input voltage. The calibration potentiometer is set such
that V
reads directly in RMS volts.
O
Figure 27
is essentially an auto-
DS007980-32
FIGURE 25. Sample-Hold Circuit
FIGURE 26. Ramp and Hold
FIGURE 27. True RMS Converter
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DS007980-33
DS007980-34
Page 19
Additional Applications (Continued)
The circuit of
temperature coefficient. The 100 kΩ potentiometer adjusts
the output voltage which has a positive TC above 1.2V, zero
TC at about 1.2V and negative TC below 1.2V. This is accomplished by balancing the TC of the A2 transfer function
against the complementary TC of D1.
The log amplifier of
rents through buffer transistors Q3 and Q4. Zero temperature dependence for V
A2 transfer function is equal and opposite to the TC of the
logging transistors Q3 and Q4.
The wide dynamic range of the LM13600 allows easy control
of the output pulse width in the Pulse Width Modulator of
ure 30
For generating I
rent, the system of
out for a linear voltage in.
Since the closed-loop configuration ensures that the input to
A2 is held equal to 0V, the output current of A1 is equal
to I
3
The differential voltage between Q1 and Q2 is attenuated by
the R1, R2 network so that A1 may be assumed to be operating within its linear range. From
age to A1 is:
Figure 28
is a voltage reference of variable
Figure 29
responds to the ratio of cur-
is ensured because the TC of the
OUT
Fig-
.
over a range of 4 to 6 decades of cur-
ABC
Figure 31
=
−V
C/RC
.
provides a logarithmic current
Equation (5)
, the input volt-
The voltage on the base of Q1 is then
The ratio of the Q1 and Q2 collector currents is defined by:
Combining and solving for I
This logarithmic current can be used to bias the circuit of
ure 4
provide a temperature independent stereo attenuation
ABC
yields:
Fig-
characteristic.
FIGURE 28. Delta VBE Reference
DS007980-35
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Page 20
Additional Applications (Continued)
FIGURE 29. Log Amplifier
DS007980-36
FIGURE 30. Pulse Width Modulator
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DS007980-37
Page 21
Additional Applications (Continued)
FIGURE 31. Logarithmic Current Source
DS007980-38
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Page 22
22
Page 23
Physical Dimensions inches (millimeters) unless otherwise noted
S.O. Package (M)
Order Number LM13600M
NS Package Number M16A
Molded Dual-In-Line Package (N)
Order Number LM13600N or LM13600AN
NS Package Number N16A
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Page 24
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into
the body, or (b) support or sustain life, and whose failure to perform when properly used in accordance
2. A critical component is any component of a life support
device or system whose failure to perform can be reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
with instructions for use provided in the labeling, can
be reasonably expected to result in a significant injury
to the user.
National Semiconductor
Corporation
Americas
LM13600 Dual Operational Transconductance Amplifiers with Linearizing Diodes and Buffers
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
Tel: 1-800-272-9959
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Email: support@nsc.com
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