Datasheet IRU3037ACSPbF Specification

Data Sheet No. PD94173 revD

IRU3037/IRU3037A & (PbF)

8-PIN SYNCHRONOUS PWM CONTROLLER

FEATURES

Synchronous Controller in 8-Pin Package
Operating with single 5V or 12V supply voltage
Internal 200KHz Oscillator
(400KHz for IRU3037A)
Soft-Start Function
Fixed Frequency Voltage Mode
500mA Peak Output Drive Capability
Protects the output when control FET is shorted
SOIC 8-Lead also available LEAD-FREE

APPLICATIONS

DDR memory source sink Vtt application
Low cost on-board DC to DC such as 5V to 3.3V ,

2.5V or 1.8V
Graphic Card
Hard Disk Drive

TYPICAL APPLICA TION

12V

C3

1uF

C4
1uF

DESCRIPTION

The IRU3037 controller IC is designed to provide a low
cost synchronous Buck regulator for on-board DC to DC
converter applications. With the migration of today’s ASIC
products requiring low supply voltages such as 1.8V and
lower, together with currents in excess of 3A, traditional
linear regulators are simply too lossy to be used when
input supply is 5V or even in some cases with 3.3V
input supply . The IRU3037 together with dual N-channel
MOSFET s such as IRF7313, provide a low cost solution
for such applications. This device features an internal
200KHz oscillator (400KHz for "A" version), under-volt­age lockout for both Vcc and Vc supplies, an external
programmable soft-start function as well as output un­der-voltage detection that latches off the device when an
output short is detected.

5V

L1

C2
10TPB100M,
100uF, 55m

1uH

Ω

C1
47uF

Vcc Vc

HDrv

SS/SD

Comp

U1

IRU3037

Gnd

LDrv

Fb

R5

1.24K, 1%

C8

0.1uF

C9

2200pF

R4

24k

Figure 1 - Typical application of IRU3037 or IRU3037A.

P ACKAGE ORDER INFORMA TION

TA (°C) DEVICE LEADFREE DEVICE PACKAGE FREQUENCY

0 T o 70 IRU3037CF IRU3037CFPbF 8-Pin Plastic TSSOP (F) 200KHz
0 T o 70 IRU3037CS IRU3037CSPbF 8-Pin Plastic SOIC NB (S) 200KHz
0 T o 70 IRU3037ACF IRU3037ACFPbF 8-Pin Plastic TSSOP (F) 400KHz
0 T o 70 IRU3037ACS IRU3037ACSPbF 8-Pin Plastic SOIC NB (S) 400KHz

Q1
1/2 of IRF7313

D05022P-562, 5.6uH, 5.3A
Q2
1/2 of IRF7313

R3

249, 1%

L2

1.5V/5A

C7
2x 6TPC150M,
150uF, 40m

Ω

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1

IRU3037/IRU3037A & (PbF)

ABSOLUTE MAXIMUM RA TINGS

Vcc Supply Voltage .................................................. 25V

Vc Supply Voltage ...................................................... 30V (not rated for inductive load)

Storage Temperature Range ...................................... -65°C To 150°C

Operating Junction Temperature Range ..................... 0°C To 125°C

CAUTION: Stresses above those listed in "Absolute Maximum Ratings" may cause permanent damage to the device.

P ACKAGE INFORMA TION

8-PIN PLASTIC TSSOP (F) 8-PIN PLASTIC SOIC (S)

Fb

Vcc

LDrv

Gnd

1
2

3
4

θJA=124°C/W

8
7
6
5

SS/SD

Comp

Vc

HDrv

1

Fb

2

Vcc

3

LDrv

4

Gnd HDrv

θJA=160°C/W

8
7
6
5

SS/SD
Comp
Vc

(Also available LEAD-FREE)

ELECTRICAL SPECIFICA TIONS

Unless otherwise specified, these specifications apply over Vcc=5V , Vc=12V and TA=0 to 70°C. Typical values refer
to TA=25°C. Low duty cycle pulse testing is used which keeps junction and case temperatures equal to the ambient
temperature.

PARAMETER SYM TEST CONDITION MIN TYP MAX UNITS
Reference Voltage

Fb Voltage

Fb Voltage Line Regulation

UVLO

UVLO Threshold - Vcc
UVLO Hysteresis - Vcc
UVLO Threshold - Vc
UVLO Hysteresis - Vc
UVLO Threshold - Fb

UVLO Hysteresis - Fb

Supply Current

Vcc Dynamic Supply Current
Vc Dynamic Supply Current
Vcc Static Supply Current
Vc Static Supply Current

Soft-Start Section

Charge Current

VFB

LREG

UVLO Vcc

UVLO Vc

UVLO Fb

Dyn Icc

Dyn Ic

ICCQ

SSIB

IRU3037
IRU3037A
5<Vcc<12

Supply Ramping Up

Supply Ramping Up

Fb Ramping Down (IRU3037)
(IRU3037A)

Freq=200KHz, CL=1500pF
Freq=200KHz, CL=1500pF
SS=0V
SS=0V

ICQ

SS=0V

1.225

0.784

4.0

3.1

0.4

0.3

2
2
1

0.5

-10

1.250

0.800

0.2

4.2

0.25

3.3

0.2

0.6

0.4

0.1

5
7

3.3
1

-20

1.275

0.816

0.35

4.4

3.5

0.8

0.5

8

10

6

4.5

-30

V

%

V
V
V
V
V

V

mA
mA
mA
mA

µA

2

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IRU3037/IRU3037A & (PbF)

PARAMETER SYM TEST CONDITION MIN TYP MAX UNITS
Error Amp

Fb Voltage Input Bias Current
Fb Voltage Input Bias Current
Transconductance

Oscillator

Frequency

Ramp-Amplitude Voltage

V

Output Drivers

Rise Time
Fall Time
Dead Band Time
Max Duty Cycle
Min Duty Cycle

PIN DESCRIPTIONS

PIN# PIN SYMBOL PIN DESCRIPTION

1

Fb

This pin is connected directly to the output of the switching regulator via resistor divider to
provide feedback to the Error amplifier.

SS=3V, Fb=1V

IFB1

SS=0V, Fb=1V

IFB2

gm

IRU3037

Freq

IRU3037A

RAMP

CL=1500pF

Tr

C

L=1500pF

Tf

TDB

TON

Fb=1V , Freq=200KHz
Fb=1.5V

TOFF

410

180
345

1.225

50
85

0

-0.1

-64

600

200
400

1.25

50
50

150

90

0

830

220
440

1.275

100
100
250

95

µA
µA

µmho

KHz

V

ns
ns
ns

%
%

2

Vcc

This pin provides biasing for the internal blocks of the IC as well as power for the low side
driver. A minimum of 1µF, high frequency capacitor must be connected from this pin to

ground to provide peak drive current capability .
3
4

LDrv
Gnd

Output driver for the synchronous power MOSFET .

This pin serves as the ground pin and must be connected directly to the ground plane. A

high frequency capacitor (0.1 to 1µF) must be connected from V5 and V12 pins to this pin

for noise free operation.
5

HDrv

Output driver for the high side power MOSFET . Connect a diode, such as BA T54 or 1N4148,

from this pin to ground for the application when the inductor current goes negative (Source/

Sink), soft-start at no load and for the fast load transient from full load to no load.
6

Vc

This pin is connected to a voltage that must be at least 4V higher than the bus voltage of

the switcher (assuming 5V threshold MOSFET) and powers the high side output driver. A

minimum of 1µF , high frequency capacitor must be connected from this pin to ground to

provide peak drive current capability .
7

Comp

Compensation pin of the error amplifier. An external resistor and capacitor network is

typically connected from this pin to ground to provide loop compensation.
8

SS / SD

This pin provides soft-start for the switching regulator. An internal current source charges

an external capacitor that is connected from this pin to ground which ramps up the output

of the switching regulator, preventing it from overshooting as well as limiting the input

current. The converter can be shutdown by pulling this pin below 0.5V .

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3

IRU3037/IRU3037A & (PbF)

BLOCK DIAGRAM

3V

20uA

SS/SD

8

POR

1.25V
25K

Fb

Comp 7

25K

1

64uA Max

Error Amp

Vcc

4.0V

Vc

3.5V

0.5V

Ct

0.2V

0.2V

Osc i llator

Error Comp

FbLo Comp

POR

Bias

Generator

S

R

Reset Dom

Q

Figure 2 - Simplified block diagram of the IRU3037.

3V

1.25V

POR

6

Vc

5

HDrv

2

Vcc

3

LDrv

Gnd

4

THEORY OF OPERA TION

Introduction

The IRU3037 is a fixed frequency, voltage mode syn­chronous controller and consists of a precision refer­ence voltage, an error amplifier, an internal oscillator , a
PWM comparator, 0.5A peak gate driver , soft-start and
shutdown circuits (see Block Diagram).

The output voltage of the synchronous converter is set
and controlled by the output of the error amplifier; this is
the amplified error signal from the sensed output voltage
and the reference voltage.

This voltage is compared to a fixed frequency linear
sawtooth ramp and generates fixed frequency pulses of
variable duty-cycle, which drives the two N-channel ex­ternal MOSFET s.The timing of the IC is provided through
an internal oscillator circuit which uses on-chip capaci­tor to set the oscillation frequency to 200 KHz (400 KHz
for “A” version).

Soft-Start

The IRU3037 has a programmable soft-start to control
the output voltage rise and limit the current surge at the
start-up. To ensure correct start-up, the soft-start se­quence initiates when the Vc and Vcc rise above their
threshold (3.3V and 4.2V respectively) and generates

the Power On Reset (POR) signal. Soft-start function
operates by sourcing an internal current to charge an
external capacitor to about 3V . Initially , the soft-start func­tion clamps the E/A ’ s output of the PWM converter. As
the charging voltage of the external capacitor ramps up,
the PWM signals increase from zero to the point the
feedback loop takes control.

Short-Circuit Protection

The outputs are protected against the short-circuit. The
IRU3037 protects the circuit for shorted output by sens­ing the output voltage (through the external resistor di­vider). The IRU3037 shuts down the PWM signals, when
the output voltage drops below 0.6V (0.4V for IRU3037A).

The IRU3037 also protects the output from over-voltaging
when the control FET is shorted. This is done by turning
on the sync FET with the maximum duty cycle.

Under-V oltage Lockout

The under-voltage lockout circuit assures that the
MOSFET driver outputs remain in the off state whenever
the supply voltage drops below set parameters. Lockout
occurs if Vc or Vcc fall below 3.3V and 4.2V respec­tively . Normal operation resumes once Vc and Vcc rise
above the set values.

4

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APPLICA TION INFORMA TION

IRU3037/IRU3037A & (PbF)

Design Example:

The following example is a typical application for IRU3037,
the schematic is Figure 18 on page 14.

VIN = 5V
VOUT = 3.3V
IOUT = 4A
∆VOUT = 100mV
fS = 200KHz

Output Voltage Programming

Output voltage is programmed by reference voltage and
external voltage divider. The Fb pin is the inverting input
of the error amplifier, which is internally referenced to

1.25V (0.8V for IRU3037A). The divider is ratioed to pro­vide 1.25V at the Fb pin when the output is at its desired
value. The output voltage is defined by using the follow­ing equation:

R6

VOUT = VREF ×

When an external resistor divider is connected to the
output as shown in Figure 3.

1 +

( )

IRU3037

Figure 3 - Typical application of the IRU3037 for

programming the output voltage.

Equation (1) can be rewritten as:

VOUT

R6 = R5 ×

Choose R5 = 1KΩ
This will result to R6 = 1.65KΩ

If the high value feedback resistors are used, the input
bias current of the Fb pin could cause a slight increase
in output voltage. The output voltage set point can be
more accurate by using precision resistor.

( )

VREF

---(1)

R5

Fb

- 1

V

OUT

R

6

R

5

Soft-Start Programming

The soft-start timing can be programmed by selecting
the soft start capacitance value. The start up time of the
converter can be calculated by using:

tSTART = 75×Css (ms) ---(2)

Where:
CSS is the soft-start capacitor (µF)

For a start-up time of 7.5ms, the soft-start capacitor will
be 0.1µF. Choose a ceramic capacitor at 0.1µF.

Shutdown

The converter can be shutdown by pulling the soft-start
pin below 0.5V. The control MOSFET turns off and the
synchronous MOSFET turns on during shutdown.

Boost Supply Vc

T o drive the high-side switch it is necessary to supply a
gate voltage at least 4V greater than the bus voltage.
This is achieved by using a charge pump configuration
as shown in Figure 18. The capacitor is charged up to
approximately twice the bus voltage. A capacitor in the
range of 0.1µF to 1µF is generally adequate for most
applications. In application, when a separate voltage
source is available the boost circuit can be avoided as
shown in Figure 1.

Input Capacitor Selection

The input filter capacitor should be based on how much
ripple the supply can tolerate on the DC input line. The
larger capacitor, the less ripple expected but consider
should be taken for the higher surge current during the
power-up. The IRU3037 provides the soft-start function
which controls and limits the current surge. The value of
the input capacitor can be calculated by the following
formula:

IIN × ∆t

CIN = ---(3)

Where:
CIN is the input capacitance (µF)
IIN is the input current (A)

∆t is the turn on time of the high-side switch (µs)
∆V is the allowable peak to peak voltage ripple (V)

∆V

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5

IRU3037/IRU3037A & (PbF)

Assuming the following:

∆V = 1%(VIN), Efficiency(η) = 90%

1

∆t = D ×

O × IO

V

IIN =

η × VIN

∆t = 3.3µs

fS

IIN = 2.93A

By using equation (3), CIN = 193.3µF
For higher efficiency , low ESR capacitor is recommended.
Choose two 100µF capacitors.

The Sanyo TPB series PosCap capacitor 100µF, 10V
with 55mΩ ESR is a good choice.

Output Capacitor Selection

The criteria to select the output capacitor is normally
based on the value of the Effective Series Resistance
(ESR). In general, the output capacitor must have low
enough ESR to meet output ripple and load transient
requirements, yet have high enough ESR to satisfy sta­bility requirements. The ESR of the output capacitor is
calculated by the following relationship:

ESR ≤ ---(4)

∆VO

∆IO

Where:

∆VO = Output Voltage Ripple
∆IO = Output Current
∆VO=100mV and ∆IO=4A

Results to ESR=25mΩ

The Sanyo TPC series, PosCap capacitor is a good
choice. The 6TPC150M 150µF , 6.3V has an ESR 40mΩ.
Selecting two of these capacitors in parallel, results to
an ESR of ≅ 20mΩ which achieves our low ESR goal.

The capacitor value must be high enough to absorb the
inductor's ripple current. The larger the value of capaci­tor, the lower will be the output ripple voltage.

Inductor Selection

The inductor is selected based on output power, operat­ing frequency and efficiency requirements. Low inductor
value causes large ripple current, resulting in the smaller
size, but poor efficiency and high output noise. Gener­ally, the selection of inductor value can be reduced to
desired maximum ripple current in the inductor (∆i). The
optimum point is usually found between 20% and 50%
ripple of the output current.

For the buck converter, the inductor value for desired
operating ripple current can be determined using the fol­lowing relation:

IN - VOUT = L× ; ∆t = D× ; D =

V

L = (VIN - VOUT)× ---(5)

∆i
∆t

VOUT

VIN×∆i×fS

1
fS

VOUT

VIN

Where:
V

IN = Maximum Input Voltage

VOUT = Output Voltage
∆i = Inductor Ripple Current
fS = Switching Frequency
∆t = Turn On Time
D = Duty Cycle

If ∆i = 20%(IO), then the output inductor will be:

L = 7µH

The T oko D124C series provides a range of inductors in
different values, low profile suitable for large currents,
10µH, 4.2A is a good choice for this application. This
will result to a ripple approximately 14% of output cur­rent.

Power MOSFET Selection

The IRU3037 uses two N-Channel MOSFET s. The se­lections criteria to meet power transfer requirements is
based on maximum drain-source voltage (V

DSS), gate-

source drive voltage (VGS), maximum output current, On­resistance RDS(ON) and thermal management.

The MOSFET must have a maximum operating voltage
(VDSS) exceeding the maximum input voltage (VIN).

The gate drive requirement is almost the same for both
MOSFET s. Logic-level transistor can be used and cau­tion should be taken with devices at very low VGS to pre­vent undesired turn-on of the complementary MOSFET ,
which results a shoot-through current.

The total power dissipation for MOSFET s includes con­duction and switching losses. For the Buck converter
the average inductor current is equal to the DC load cur­rent. The conduction loss is defined as:

PCOND (Upper Switch) = ILOAD × RDS(ON) × D × ϑ
PCOND (Lower Switch) = ILOAD × RDS(ON) × (1 - D) × ϑ

2

2

ϑ = RDS(ON) T emperature Dependency

The RDS(ON) temperature dependency should be consid­ered for the worst case operation. This is typically given
in the MOSFET data sheet. Ensure that the conduction
losses and switching losses do not exceed the package
ratings or violate the overall thermal budget.

6

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IRU3037/IRU3037A & (PbF)

For this design, IRF7301 is a good choice. The device
provides low on-resistance in a compact SOIC 8-Pin
package.

The IRF7301 has the following data:

VDSS = 20V
ID = 5.2A
RDS(ON) = 0.05Ω

The total conduction losses will be:

PCON(TOTAL)=PCON(Upper Switch)+PCON(Lower Switch)

PCON(TOTAL) = ILOAD × RDS(ON) × ϑ

2

ϑ = 1.5 according to the IRF7301 data sheet for

150)C junction temperature

PCON(TOTAL) = 1.2W

The switching loss is more difficult to calculate, even
though the switching transition is well understood. The
reason is the effect of the parasitic components and
switching times during the switching procedures such
as turn-on / turnoff delays and rise and fall times. With a
linear approximation, the total switching loss can be ex­pressed as:

VDS(OFF)

PSW = ILOAD ---(6)

2

tr + tf

T

××

Where:
V

DS(OFF) = Drain to Source Voltage at of f time

tr = Rise Time
tf = Fall Time

T = Switching Period
ILOAD = Load Current

These values are taken under a certain condition test.
For more detail please refer to the IRF7301 data sheet.

By using equation (6), we can calculate the switching
losses.

SW = 0.186W

P

Feedback Compensation

The IRU3037 is a voltage mode controller; the control
loop is a single voltage feedback path including error
amplifier and error comparator. To achieve fast transient
response and accurate output regulation, a compensa­tion circuit is necessary . The goal of the compensation
network is to provide a closed loop transfer function with
the highest 0dB crossing frequency and adequate phase
margin (greater than 45)).

The output LC filter introduces a double pole, –40dB/
decade gain slope above its corner resonant frequency,
and a total phase lag of 180) (see Figure 5). The Reso­nant frequency of the LC filter expressed as follows:

1

FLC = ---(7)

2π× LO×CO

Figure 5 shows gain and phase of the LC filter. Since we
already have 180) phase shift just from the output filter,
the system risks being unstable.

Gain

0dB

-40dB/decade

Phase

)

0

The switching time waveform is shown in figure 4.

DS

V

90%

10%

GS

V

d

t

d

t

(ON)

(OFF)

r

t

f

t

Figure 4 - Switching time waveforms.

From IRF7301 data sheet we obtain:

tr = 42ns
tf = 51ns

The IRU3037’s error amplifier is a differential-input
transconductance amplifier. The output is available for
DC gain control or AC phase compensation.

The E/A can be compensated with or without the use of
local feedback. When operated without local feedback
the transconductance properties of the E/A become evi­dent and can be used to cancel one of the output filter
poles. This will be accomplished with a series RC circuit
from Comp pin to ground as shown in Figure 6.

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LC

F

Frequency

-180

LC

F

)

Figure 5 - Gain and phase of LC filter.

Frequency

7

IRU3037/IRU3037A & (PbF)

Note that this method requires that the output capacitor
should have enough ESR to satisfy stability requirements.
In general the output capacitor’s ESR generates a zero
typically at 5KHz to 50KHz which is essential for an
acceptable phase margin.

The ESR zero of the output capacitor expressed as fol­lows:

FESR = ---(8)

2π × ESR × Co

V

OUT

1

R

6

Fb

Comp

R

E/A

5

V

REF

Gain(dB)

Ve

C

9

R

4

H(s) dB

Frequency

F

Z

Figure 6 - Compensation network without local

feedback and its asymptotic gain plot.

The transfer function (Ve / V

H(s) = gm × × ---(9)

( )

R6 + R5

R5

OUT) is given by:

1 + sR4C9

sC9

The (s) indicates that the transfer function varies as a
function of frequency . This configuration introduces a gain
and zero, expressed by:

|H(s)| = gm× × R4 ---(10)

FZ = ---(11)

2π×R4×C9

R5

R6×R5

1

The gain is determined by the voltage divider and E/A's
transconductance gain.

Where:

IN = Maximum Input Voltage

V
VOSC = Oscillator Ramp Voltage
Fo = Crossover Frequency
FESR = Zero Frequency of the Output Capacitor
FLC = Resonant Frequency of the Output Filter
R5 and R6 = Resistor Dividers for Output Voltage
Programming
gm = Error Amplifier Transconductance

For:
VIN = 5V
VOSC = 1.25V
Fo = 30KHz
FESR = 26.52KHz
FLC = 2.9KHz
R5 = 1K
R6 = 1.65K
gm = 600µmho

This results to R

4=104.4KΩ. Choose R4=105KΩ

To cancel one of the LC filter poles, place the zero be­fore the LC filter resonant frequency pole:

FZ ≅ 75%FLC
FZ ≅ 0.75 × ---(13)

1

2π LO × CO
For:
Lo = 10µH
Co = 300µF
FZ = 2.17KHz
R4 = 86.6KΩ

Using equations (1 1) and (13) to calculate C9, we get:

C9 = 698pF
Choose C9 = 680pF

One more capacitor is sometimes added in parallel with
C9 and R4. This introduces one more pole which is mainly
used to supress the switching noise. The additional pole
is given by:

1

FP =

2π × R4 ×

C9 × CPOLE

C9 + CPOLE

First select the desired zero-crossover frequency (Fo):

Fo > FESR and FO ≤ (1/5 ~ 1/10)× fS

Use the following equation to calculate R4:

VOSC

R4 = ---(12)

VIN

Fo×FESR

2

FLC

×××

R5 + R6

R5

8

1

gm

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The pole sets to one half of switching frequency which
results in the capacitor CPOLE:

CPOLE =

for FP <<

1

π×R4×fS -

fS
2

1

C9

≅

π×R4×fS

1

IRU3037/IRU3037A & (PbF)

For a general solution for unconditionally stability for any
type of output capacitors, in a wide range of ESR values
we should implement local feedback with a compensa­tion network. The typically used compensation network
for voltage-mode controller is shown in Figure 7.

V

Z

IN

Gain(dB)

OUT

C

10

R

8

R

6

Fb

R

5

V

REF

R

7

E/A

C

12

C

11

Z

f

Ve

Comp

H(s) dB

F

F

1

Z

2

Z

F

2

P

Frequency

F

3

P

Figure 7 - Compensation network with local

feedback and its asymptotic gain plot.

In such configuration, the transfer function is given by:

Ve

VOUT

1 - gmZf

=

1 + gmZIN

The error amplifier gain is independent of the transcon­ductance under the following condition:

gmZf >> 1 and gmZIN >>1 ---(14)

By replacing Z
former function can be expressed as:

H(s)= ×

sR6(C12+C11)

As known, transconductance amplifier has high imped­ance (current source) output, therefore, consider should
be taken when loading the E/A output. It may exceed its
source/sink output current capability , so that the ampli­fier will not be able to swing its output voltage over the
necessary range.

The compensation network has three poles and two ze­ros and they are expressed as follows:

IN and Zf according to Figure 7, the trans-

1

(1+sR7C11)×[1+sC10(R6+R8)]

C12×C11

1+sR7

×(1+sR8C10)

[ ( )]

C12+C11

FP1 = 0
FP2 =

FP3 = ≅

FZ1 =

FZ2 = ≅

1

8×C10

2π×R

1

12×C11

2π×R7×

C

( )

C12+C11

1

2π×R

7×C11

1

2π×C10×(R6 + R8)

1

2π×R7×C12

1

2π×C10×R6

Cross Over Frequency:

FO = R7×C10× × ---(15)

VIN

VOSC12π×Lo×Co

Where:
VIN = Maximum Input Voltage
VOSC = Oscillator Ramp Voltage
Lo = Output Inductor
Co = T otal Output Capacitors

The stability requirement will be satisfied by placing the
poles and zeros of the compensation network according
to following design rules. The consideration has been
taken to satisfy condition (14) regarding transconduc­tance error amplifier.

1) Select the crossover frequency:

Fo < F

ESR and Fo ≤ (1/10 ~ 1/6)× fS

2) Select R7, so that R7 >>

2

gm

3) Place first zero before LC’s resonant frequency pole.

FZ1 ≅ 75% FLC
C11 =

1

2π × FZ1 × R7

4) Place third pole at the half of the switching frequency.

fS

FP3 =

2

C12 =

2π × R7 × FP3

1

C12 > 50pF
If not, change R7 selection.

5) Place R7 in (15) and calculate C10:

2π × Lo × Fo × Co

C10 ≤ ×

R7

VOSC

VIN

www.irf.com

9

IRU3037/IRU3037A & (PbF)

6) Place second pole at the ESR zero.
FP2 = FESR

R8 =

2π × C10 × FP2

1

1

Check if R8 >

gm

If R8 is too small, increase R7 and start from step 2.

7) Place second zero around the resonant frequency .
FZ2 = FLC

R6 = - R8

2π × C10 × FZ2

1

8) Use equation (1) to calculate R5.
R5 = × R6

VREF

VOUT - VREF

These design rules will give a crossover frequency ap­proximately one-tenth of the switching frequency. The
higher the band width, the potentially faster the load tran­sient speed. The gain margin will be large enough to
provide high DC-regulation accuracy (typically -5dB to ­12dB). The phase margin should be greater than 45) for
overall stability.

IC Quiescent Power Dissipation

Power dissipation for IC controller is a function of ap­plied voltage, gate driver loads and switching frequency .
The IC's maximum power dissipation occurs when the
IC operating with single 12V supply voltage (Vcc=12V
and Vc≅24V) at 400KHz switching frequency and maxi­mum gate loads.

Figures 9 and 10 show voltage vs. current, when the
gate drivers loaded with 470pF , 1 150pF and 1540pF ca­pacitors. The IC's power dissipation results to an exces­sive temperature rise. This should be considered when
using IRU3037A for such application.

Layout Consideration

The layout is very important when designing high fre­quency switching converters. Layout will affect noise
pickup and can cause a good design to perform with
less than expected results.

Start to place the power components, make all the con­nection in the top layer with wide, copper filled areas.
The inductor, output capacitor and the MOSFET should
be close to each other as possible. This helps to reduce
the EMI radiated by the power traces due to the high
switching currents through them. Place input capacitor
directly to the drain of the high-side MOSFET , to reduce
the ESR replace the single input capacitor with two par­allel units. The feedback part of the system should be
kept away from the inductor and other noise sources,
and be placed close to the IC. In multilayer PCB use
one layer as power ground plane and have a control cir­cuit ground (analog ground), to which all signals are ref­erenced. The goal is to localize the high current path to
a separate loop that does not interfere with the more
sensitive analog control function. These two grounds
must be connected together on the PC board layout at a
single point.

Figure 8 shows a suggested layout for the critical com­ponents, based on the schematic on page 14.

PGnd PGnd

C1

Vin

L1

D

D

D

2

3

1

Analog Gnd

C
5

C2A, B C7A, B

1

2

U1

6

C4

IRU3037

PGnd

7

C9

R4

8

C8

Q1

PGnd

5678

4

3

45

3

2

C3
R5

R6

1

Single Point
Analog Gnd
Connect to
Power Ground plane

Analog Gnd

Vout

L2

Figure 8 - Suggested layout.

(T opside shown only)

10

www.irf.com

IRU3037/IRU3037A & (PbF)

TYPICAL PERFORMANCE CHARACTERISTICS

IR U 3037A

V cc vs. Icc

@ 470PF, 1150PF and 1540P F G ate Load

14
12
10

8
6

Icc ( m A)

4
2
0

02468101214

Vcc ( V)

Figure 9 - Vcc vs. Icc

CLOAD=1540pF

CLOAD=470pF

TA = 25)C

LOAD=1150pF

C

IRU3037A

TA = 25)C

Vc v s . Ic

@470PF , 1150PF and 1540PF G ate Load

30
25
20
15

Ic (ma)

10

5
0

0 2 4 6 8 101214161820222426

Vc (V)

Figure 10 - Vc vs. Ic

CLOAD=1540pF

CLOAD=1150pF

CLOAD=470pF

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11

IRU3037/IRU3037A & (PbF)

e

TYPICAL PERFORMANCE CHARACTERISTICS

IRU3037

Output Voltage

1.3

1.28

Max

1.26

Volts

1.24

Min

1.22

1.2

-40°C 0°C +50°C +100°C +150°C
Ou tput Voltage Spec Max. Spec Min.

Figure 1 1 - Output Voltage Figure 12 - Output Frequency

IRU3037

Output Frequency

240

230

220

210

200

Kilo Hertz

190

180

170

160

-40°C 0°C +50°C +100°C +150°C

Os c illation Frequency Spec Max. Spec Min.

Max

Min

IRU3037

Maximum Duty Cycle

92.0%

90.0%

88.0%

86.0%

Percent Duty Cycl

84.0%

82.0%

80.0%

-40°C -25°C 0°C +25°C +50°C +75°C +100°C +125°C +150°C

Max Duty Cy c l e

Figure 13 - Maximum Duty Cycle

12

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IRU3037/IRU3037A & (PbF)

TYPICAL PERFORMANCE CHARACTERISTICS

IRU3037A

Output V oltage

820

Max

810

800

790

milli Volts

780

770

760

-40°C -25°C 0°C +25°C +50°C +75°C +100°C +150°C
Output Voltage Spec Max. Spec Min.

Min

IRU3037A

Output Frequency

460

440

420

400

380

Kilo Hertz

360

340

320

300

-40°C -25°C 0°C + 25°C +50°C +75°C + 100°C +150°C

Os c illation Frequenc y Spec Max. Spec Min.

Max

Min

Figure 15 - Output FrequencyFigure 14 - Output Voltage

IRU 3037 / IRU3037A

Transconductance ( GM )

1000

900

800

700

600

500

micro Mho's

400

300

200

100

0

-40°C -25°C 0°C +25°C +50°C +75°C +100°C

Positive load GM Negative load GM

Figure 16 - Transconductance

www.irf.com

13

IRU3037/IRU3037A & (PbF)
TYPICAL APPLICATION

Single Supply 5V Input

D1
1N4148

D3

C8

0.1uF

C9

680pF

105K

R4

C3

1uF

1N4148

Vcc Vc

SS/SD

IRU3037

Comp

HDrv

U1

LDrv

Gnd

C4

1uF

Fb

D2
1N4148

C5

0.1uF

R5
1K, 1%

Q1
1/2 of IRF7301

D05022P-103, 10uH, 4.3A
Q2
1/2 of IRF7301

R6

1.65K, 1%

C2
2x 10TPB100ML,
100uF, 55m

Ω

L2

L1

1uH

5V

C1
47uF
Tantalum

3.3V

@ 4A
C7
2x 6TPC150M,
150uF, 40m

Ω

Figure 17 - Typical application of IRU3037 in an on-board DC-DC converter

using a single 5V supply .

14

www.irf.com

TYPICAL APPLICA TION

Dual Supply , 5V Bus and 12V Bias Input

5V

12V

C7

0.1uF

C8

2200pF

14K

R2

C4

1uF

Vcc

SS/SD

IRU3037

Comp

Vc

U1

Gnd

C1
1uF

HDrv

LDrv

Fb

R3
1K

1%

IRU3037/IRU3037A & (PbF)

C2
10TPB100M, 100uF,
55m

Q1
1/2 of IRF7752

CTX5-2P, 3.5uH @ 2.5A

Q2
1/2 of IRF7752

R1

1K, 1%

L1

1uH

, 1.5A rms

Ω

L2

L2

CTX5-2P, 3.5uH @ 2.5A
CTX10-5P, 5.7uH @ 2.5A

C1
47uF

IRU1206-18

C6
6TPB150M, 150uF, 55m

C6

6TPB150M, 150uF, 55mΩ (Qty 2 )
6TPB150M, 150uF, 55m

1.8V/1A

C3
47uF

2.5V/2A

Ω

(Qty 1 )

Ω

C9
10TPB100M, 100uF,

, 1.5A rms

55m

Ω

C13

0.1uF

C14

2200pF

14K

R5

C10

1uF

Vcc

SS/SD

IRU3037

Comp

Vc

U2

Gnd

HDrv

LDrv

C11
1uF

Q3
1/2 of IRF7752

Q4
1/2 of IRF7752

Fb

R6
1K, 1%

R4

1.65K, 1%

L3

CTX5-1P, 3.4uH @ 2A

C12
6TPB150M, 150uF, 55m

3.3V/1.8A

Ω

Figure 18 - Typical application of IRU3037 or IRU3037A in an on-board DC-DC converter providing the Core,

GTL+, and Clock supplies for the Pentium II microprocessor.

www.irf.com

15

IRU3037/IRU3037A & (PbF)

TYPICAL APPLICA TION

1.8V to 7.5V / 0.5A Boost Converter

Vpwr (1.5V Min)

Vc/Vcc

R2
10K

R3
20K

R1
20K

Q1
2N2222

Q2
2N2222

C10

100pF

C4

0.01uF
R4

25K

C5
1uF

C5

0.1uF

SS/SD

Fb

R5 R6

1K, 1%

Comp

U1

IRU3037

Vcc

LDrv

C8
1uF

5K, 1%

Vc

L1
1uH (CoilTronics UP2B-1R0)

C1
2x 68uF

Q3
IRF7 402

HDrv

Gnd

D1

1N5817

V

OUT

(7.5V / 0.5A)

C9
2x 47uF

Gnd

Figure 19 - Typical application of IRU3037 as a boost converter .

16

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DEMO-BOARD APPLICA TION

5V or 12V to 3.3V @ 10A

L1

V

IN

5V or 12V

Gnd

C1
33uF
16V

C8

1uF

C5

0.1uF

C4

2200pF

20K

R3

1uH

D4

LL4148

Vcc Vc

SS/SD

U1

IRU3037

Comp

Gnd

HDrv

LDrv

Fb

C2A

47uF

16V

C3
1uF

IRU3037/IRU3037A & (PbF)

C7
470pF

R4

4.7

D1
LL4148

C6

0.1uF

L2

V

C13
1uF

OUT

3.3V
@ 10A

Gnd

3.3uH

C9C

C9B

150uF

Ω

150uF

6.3V

6.3V

C2B
47uF
16V

C19
1uF

Q2

IRF7460

R6

1.65K

R5
1K

LL4148

Q1
IRF7457

D2

Figure 20 - Demo-board application of IRU3037.

Application Parts List
Ref Desig Description Value Qty Part# Manuf

Q1
Q2
U1
D1, D2, D4
L1
L2
C1
C2A, C2B
C9B, C9C
C5, C6
C3
C4
C7
C8, C13, C19
R3
R4
R5
R6

MOSFET
MOSFET
Controller
Diode
Inductor
Inductor
Capacitor, Tantalum
Capacitor, Poscap
Capacitor, Poscap
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Capacitor, Ceramic
Resistor
Resistor
Resistor
Resistor

20V , 7mΩ, 15A
20V , 10mΩ, 12A
Synchronous PWM
Fast Switching
1µH, 10A

3.3µH, 12A
33µF, 16V
47µF, 16V , 70mΩ
150µF, 6.3V , 40mΩ

0.1µF, Y5V, 25V
1µF , X7R, 25V
2200pF , X7R, 50V
470pF , X7R
1µF, Y5V, 16V
20K, 5%

4.7Ω, 5%
1K, 1%

1.65K, 1%

1

IRF7457

1

IRF7460

1

IRU3037

3

LL4148

1

D03316P-102HC

1

D05022P-332HC

1

ECS-T1CD336R

2

16TPB47M

2

6TPC150M

2

ECJ-2VF1E104Z

1

ECJ-3YB1E105K

1

ECJ-2VB1H222K

1

ECJ-2VB2D471K

3

ECJ-2VF1C105Z
1
1
1
1

IR
IR
IR

Coilcraft
Coilcraft
Panasonic
Sanyo
Sanyo
Panasonic
Panasonic
Panasonic
Panasonic
Panasonic

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17

IRU3037/IRU3037A & (PbF)

DEMO-BOARD W A VEFORMS

VIN=5.0V, V

100

90

OUT

=3.3V

IRU3037

VIN

80

Efficiency (%)

70

01234567891011

Output Current (A)

Figure 21 - Efficiency for IRU3037 Evaluation Board.

Vss

IRU3037

VOUT

IOUT = 5V

VOUT

Figure 22 - Start-up time @ IOUT=5A.

IRU3037

Figure 23 - Shutdown the output by

pulling down the soft-start.

IRU3037

2A

0A

Figure 24 - 3.3V output voltage ripple @ IOUT=5A.

IRU3037

4A

0A

Figure 25 - Transient response @ IOUT = 0 to 2A. Figure 26 - Transient response @ IOUT = 0 to 4A.

18

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IRU3037/IRU3037A & (PbF)

(F) TSSOP Package

8-Pin

A

L

Q

R1

C

1.0 DIA

B

R

E

F

PIN NUMBER 1

M

D

G

J

H

K

---

0)

8-PIN

NOM

0.65 BSC

4.40

6.40 BSC

---

1.00

1.00

3.00

---

0.90

---

12) REF

12) REF

---

1.00 REF

0.60

0.20

---

---

MAX

4.50

0.30

3.10

1.10

0.95

0.15

8)

0.75

---

---

SYMBOL

DESIG

A
B
C
D
E
F
G
H

J
K
L
M
N
O
P
Q
R

R1

MIN

4.30

0.19

2.90

0.85

0.05

0.50

0.09

0.09

P

O

DETAIL A

N

DETAIL A

NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS.

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19

IRU3037/IRU3037A & (PbF)

(S) SOIC Package

8-Pin Surface Mount, Narrow Body

A

B C

E

H

PIN NO. 1

DETAIL-A

L

D

0.38±0.015 x 45

T

F

G

8-PIN

SYMBOL

A

B

C

D

E

F

G

H

I

J
K
L
T

K

MIN

4.80

1.27 BSC

0.53 REF

0.36

3.81

1.52

0.10

7) BSC

0.19

5.80

0)

0.41

1.37

)

J

MAX

4.98

0.46

3.99

1.72

0.25

0.25

6.20

8)

1.27

1.57

DETAIL-A

I

20

NOTE: ALL MEASUREMENTS ARE IN MILLIMETERS.

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IRU3037/IRU3037A & (PbF)

P ACKAGE SHIPMENT METHOD

PKG

DESIG

F

S

P ACKAGE

DESCRIPTION

TSSOP Plastic
SOIC, Narrow Body

1 11 1 11

Feed Direction

Figure A

LEADFREE PART MARKING INFORMATION

Part number

PIN

COUNT

8
8

IRxxxxxx

P ARTS

PER TUBE

100

95

P ARTS

PER REEL

2500
2500

Feed Direction

Figure B

T & R

Orientation

Fig A
Fig B

Date code

Pin 1
Identifier

?

MARKING CODE

P

Lead Free Released
Non-Lead Free

Released

IR WORLD HEADQUARTERS: 233 Kansas St., El Segundo, CA 90245, USA T el: [phone] T AC Fax: [phone]

YWW?

?XXXX

(Prod mode - 4 digit SPN code)

Assembly site code

Visit us at www.irf.com for sales contact information

Data and specifications subject to change without notice. 10/24/2005

IR logo

Lot Code

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21