Datasheet DA807A-155BRRL7, DA807A-155BRRL, DA807A-155BR Datasheet (Analog Devices)

REV. B

Information furnished by Analog Devices is believed to be accurate and
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use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

a

AD807

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2000

Fiber Optic Receiver with Quantizer and

Clock Recovery and Data Retiming

FEATURES
Meets CCITT G.958 Requirements

for STM-1 Regenerator—Type A
Meets Bellcore TR-NWT-000253 Requirements for OC-3
Output Jitter: 2.0 Degrees RMS
155 Mbps Clock Recovery and Data Retiming
Accepts NRZ Data, No Preamble Required
Phase-Locked Loop Type Clock Recovery—

No Crystal Required
Quantizer Sensitivity: 2 mV
Level Detect Range: 2.0 mV to 30 mV
Single Supply Operation: +5 V or –5.2 V
Low Power: 170 mW
10 KH ECL/PECL Compatible Output
Package: 16-Lead Narrow 150 mil SOIC

reliance on external components such as a crystal or a SAW
filter, to aid frequency acquisition.

The AD807 acquires frequency and phase lock on input data
using two control loops that work without requiring external
control. The frequency acquisition control loop initially acquires
the frequency of the input data, acquiring frequency lock on
random or scrambled data without the need for a preamble. At
frequency lock, the frequency error is zero and the frequency
detector has no further effect. The phase acquisition control
loop then works to ensure that the output phase tracks the input
phase. A patented phase detector has virtually eliminated pattern
jitter throughout the AD807.

The device VCO uses a ring oscillator architecture and patented
low noise design techniques. Jitter is 2.0 degrees rms. This low
jitter results from using a fully differential signal architecture,
Power Supply Rejection Ratio circuitry and a dielectrically
isolated process that provides immunity from extraneous signals
on the IC. The device can withstand hundreds of millivolts of
power supply noise without an effect on jitter performance.

The user sets the jitter peaking and acquisition time of the PLL
by choosing a damping factor capacitor whose value determines
loop damping. CCITT G.958 Type A jitter transfer require­ments can easily be met with a damping factor of 5 or greater.

Device design guarantees that the clock output frequency will
drift by less than 20% in the absence of input data transitions.
Shorting the damping factor capacitor, C

D

, brings the clock

output frequency to the VCO center frequency.

The AD807 consumes 170 mW and operates from a single
power supply at either +5 V or –5.2 V.

FUNCTIONAL BLOCK DIAGRAM

COMPENSATING

ZERO

LOOP

FILTER

SIGNAL

LEVEL

DETECTOR

VCO

⌽

DET

+

–

+

–

F

DET

⌺

RETIMING

DEVICE

PHASE-LOCKED LOOP

AD807

LEVEL

DETECT

COMPARATOR/

BUFFER

QUANTIZER

PIN

NIN

THRADJ

SDOUT

CLKOUTP

CLKOUTN

DATAOUTP

DATAOUTN

CF1 CF2

PRODUCT DESCRIPTION

The AD807 provides the receiver functions of data quantization,
signal level detect, clock recovery and data retiming for 155 Mbps
NRZ data. The device, together with a PIN diode/preamplifier
combination, can be used for a highly integrated, low cost, low
power SONET OC-3 or SDH STM-1 fiber optic receiver.

The receiver front end signal level detect circuit indicates when
the input signal level has fallen below a user adjustable thresh­old. The threshold is set with a single external resistor. The
signal level detect circuit 3 dB optical hysteresis prevents chatter
at the signal level detect output.

The PLL has a factory-trimmed VCO center frequency and a
frequency acquisition control loop that combine to guarantee
frequency acquisition without false lock. This eliminates a

REV. B

–2–

AD807–SPECIFICATIONS

Parameter Condition Min Typ Max Unit

QUANTIZER–DC CHARACTERISTICS

Input Voltage Range @ P

IN

or N

IN

2.5 V

CC

V

Input Sensitivity, V

SENSE

PIN–NIN, Figure 1, BER = ≤ 1 × 10

–10

2mV

Input Overdrive, V

OD

Figure 1, BER = ≤ 1 × 10

–10

0.001 2.5 V

Input Offset Voltage 50 500 µV
Input Current 510µA
Input RMS Noise BER = ≤ 1 × 10

–10

50 µV

Input Peak-to-Peak Noise BER = ≤ 1 × 10

–10

650 µV

QUANTIZER–AC CHARACTERISTICS

Upper –3 dB Bandwidth 180 MHz
Input Resistance 1MΩ
Input Capacitance 2pF
Pulsewidth Distortion 100 ps

LEVEL DETECT

Level Detect Range R

THRESH

= INFINITE 0.8 2 4.0 mV

R

THRESH

= 49.9 kΩ 4 5 7.4 mV

R

THRESH

= 3.4 kΩ 14 20 25 mV

Response Time DC-Coupled 0.1 1.5 µs
Hysteresis (Electrical) R

THRESH

= INFINITE 2.3 4.0 10.0 dB

R

THRESH

= 49.9 kΩ 3.0 5.0 9.0 dB

R

THRESH

= 3.4 kΩ 3.0 7.0 10.0 dB

SDOUT Output Logic High Load = +4 mA 3.6 V
SDOUT Output Logic Low Load = –1.2 mA 0.4 V

PHASE-LOCKED LOOP NOMINAL

CENTER FREQUENCY 155.52 MHz

CAPTURE RANGE 155 156 MHz

TRACKING RANGE 155 156 MHz

STATIC PHASE ERROR 27–1 PRN Sequence 4 20 Degrees

SETUP TIME (tSU) Figure 2 3.0 3.2 3.5 ns

HOLD TIME (tH) Figure 2 3.0 3.1 3.3 ns

PHASE DRIFT 240 Bits, No Transitions 40 Degrees

JITTER 2

7

–1 PRN Sequence 2.0 Degrees RMS

223–1 PRN Sequence 2.0 2.7 Degrees RMS

JITTER TOLERANCE f = 10 Hz 3000 Unit Intervals

f = 6.5 kHz 4.5 7.6 Unit Intervals
f = 65 kHz 0.45 1.0 Unit Intervals
f = 1.3 MHz 0.45 0.67 Unit Intervals

JITTER TRANSFER

Peaking (Figure 11) C

D

= 0.15 µF 0.08 dB

CD = 0.33 µF 0.04 dB
Bandwidth 65 92 130 kHz
Acquisition Time

C

D

= 0.1 µF2

23

–1 PRN Sequence, TA = 25°C4 × 1052 × 106Bit Periods

CD = 0.33 µFV

CC

= 5 V, VEE = GND 2 × 10

6

Bit Periods

POWER SUPPLY VOLTAGE V

MIN

to V

MAX

4.5 5.5 Volts

POWER SUPPLY CURRENT VCC = 5.0 V, VEE = GND, TA = 25°C 25 34.5 39.5 mA

PECL OUTPUT VOLTAGE LEVELS

Output Logic High, V

OH

VCC = 5.0 V, VEE = GND, TA = 25°C –1.2 –1.0 –0.7 Volts
Output Logic Low, V

OL

Referenced to V

CC

–2.0 –1.8 –1.7 Volts

SYMMETRY (Duty Cycle) ρ = 1/2, T

A

= 25°C,

Recovered Clock Output, Pin 5 VCC = 5 V, VEE = GND 50.1 54.1 %

OUTPUT RISE / FALL TIMES

Rise Time (t

R

) 20%–80% 1.1 1.5 ns

Fall Time (tF) 80%–20% 1.1 1.5 ns

Specifications subject to change without notice.

(TA = T

MIN

to T

MAX

, VCC = V

MIN

to V

MAX

, CD = 0.1 ␮F, unless otherwise noted.)

REV. B

AD807

–3–

ABSOLUTE MAXIMUM RATINGS*

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12 V

Input Voltage (Pin 12 or Pin 13) . . . . . . . . . . . . . V

CC

+ 0.6 V

Maximum Junction Temperature . . . . . . . . . . . . . . . . . 165°C

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

Lead Temperature Range (Soldering 10 sec) . . . . . . . . . 300°C

ESD Rating (Human Body Model) . . . . . . . . . . . . . . . . . 500 V

*Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
Thermal Characteristics:
16-Lead Narrow Body SOIC Package: θJA = 110°C/W.

1

0

OUTPUT

NOISE

OFFSET

OVERDRIVE

SENSITIVITY

INPUT (V)

Figure 1. Input Sensitivity, Input Overdrive

SETUP

t

SU

HOLD

t

H

DATAOUTP

(PIN 2)

CLKOUTP

(PIN 5)

Figure 2. Setup and Hold Time

PIN FUNCTION DESCRIPTIONS

Pin
No. Mnemonic Description

1 DATAOUTN Differential Retimed Data Output

2 DATAOUTP Differential Retimed Data Output

3V

CC2

Digital VCC for ECL Outputs

4 CLKOUTN Differential Recovered Clock Output

5 CLKOUTP Differential Recovered Clock Output

6V

CC1

Digital VCC for Internal Logic

7 CF1 Loop Damping Capacitor

8 CF2 Loop Damping Capacitor

9AV

EE

Analog V

EE

10 THRADJ Level Detect Threshold Adjust

11 AV

CC1

Analog VCC for PLL

12 NIN Quantizer Differential Input

13 PIN Quantizer Differential Input

14 AV

CC2

Analog VCC for Quantizer

15 SDOUT Signal Detect Output

16 V

EE

Digital VEE for Internal Logic

PIN CONFIGURATION

TOP VIEW

(Not to Scale)

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CLKOUTP

V

CC1

CF1

CF2

V

EE

SDOUT

AV

CC2

PIN

NIN

AV

CC1

THRADJ

AV

EE

AD807

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD807A-155BR –40°C to +85°C 16-Lead Narrowbody SOIC R-16A
AD807A-155BRRL7 –40°C to +85°C 750 Pieces, 7" Reel R-16A
AD807A-155BRRL –40°C to +85°C 2500 Pieces, 13" Reel R-16A

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD807 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

WARNING!

ESD SENSITIVE DEVICE

REV. B

AD807

–4–

DEFINITION OF TERMS
Maximum, Minimum and Typical Specifications

Specifications for every parameter are derived from statistical
analyses of data taken on multiple devices from multiple wafer
lots. Typical specifications are the mean of the distribution of
the data for that parameter. If a parameter has a maximum (or a
minimum), that value is calculated by adding to (or subtracting
from) the mean six times the standard deviation of the distribution.
This procedure is intended to tolerate production variations: if the
mean shifts by 1.5 standard deviations, the remaining 4.5 standard
deviations still provide a failure rate of only 3.4 parts per million.
For all tested parameters, the test limits are guardbanded to
account for tester variation to thus guarantee that no device is
shipped outside of data sheet specifications.

Input Sensitivity and Input Overdrive

Sensitivity and Overdrive specifications for the Quantizer involve
offset voltage, gain and noise. The relationship between the logic
output of the quantizer and the analog voltage input is shown in
Figure 1.

For sufficiently large positive input voltage the output is always
Logic 1 and similarly, for negative inputs, the output is always
Logic 0. However, the transitions between output Logic Levels 1
and 0 are not at precisely defined input voltage levels, but occur
over a range of input voltages. Within this Zone of Confusion,
the output may be either 1 or 0, or it may even fail to attain
a valid logic state. The width of this zone is determined by the
input voltage noise of the quantizer (650 µV at the 1 × 10

–10

confidence level). The center of the Zone of Confusion is the
quantizer input offset voltage (±500 µV maximum). Input Over­drive is the magnitude of signal required to guarantee correct
logic level with 1 × 10

–10

confidence level.

With a single-ended PIN-TIA (Figure 3), ac coupling is used and
the inputs to the Quantizer are dc biased at some common-mode
potential. Observing the Quantizer input with an oscilloscope
probe at the point indicated shows a binary signal with average
value equal to the common-mode potential and instantaneous
values both above and below the average value. It is convenient
to measure the peak-to-peak amplitude of this signal and call the
minimum required value the Quantizer Sensitivity. Referring to
Figure 1, since both positive and negative offsets need to be
accommodated, the Sensitivity is twice the Overdrive. The AD807
Quantizer has 2 mV Sensitivity.

With a differential TIA (Figure 3), Sensitivity seems to improve
from observing the Quantizer input with an oscilloscope probe.
This is an illusion caused by the use of a single-ended probe. A
1 mV peak-to-peak signal appears to drive the AD807 Quantizer.
However, the single-ended probe measures only half the signal.
The true Quantizer input signal is twice this value since the
other Quantizer input is a complementary signal to the sig­nal being observed.

Response Time

Response time is the delay between removal of the input signal
and indication of Loss of Signal (LOS) at SDOUT. The response
time of the AD807 (1.5 µs maximum) is much faster than the
SONET/SDH requirement (3 µs

≤

response time ≤ 100 µs). In

practice, the time constant of the ac coupling at the Quantizer
input determines the LOS response time.

Nominal Center Frequency

This is the frequency at which the VCO will oscillate with the
loop damping capacitor, C

D

, shorted.

Tracking Range

This is the range of input data rates over which the AD807 will
remain in lock.

Capture Range

This is the range of input data rates over which the AD807 will
acquire lock.

Static Phase Error

This is the steady-state phase difference, in degrees, between the
recovered clock sampling edge and the optimum sampling instant,
which is assumed to be halfway between the rising and falling
edges of a data bit. Gate delays between the signals that define
static phase error, and IC input and output signals prohibit
direct measurement of static phase error.

Data Transition Density, ρ

This is a measure of the number of data transitions, from “0” to
“1” and from “1” to “0,” over many clock periods. ρ is the ratio
(0 ≤ ρ ≤ 1) of data transitions to bit periods.

Jitter

This is the dynamic displacement of digital signal edges from
their long term average positions, measured in degrees rms or
Unit Intervals (UI). Jitter on the input data can cause dynamic
phase errors on the recovered clock sampling edge. Jitter on the
recovered clock causes jitter on the retimed data.

Output Jitter

This is the jitter on the retimed data, in degrees rms, due to a
specific pattern or some pseudorandom input data sequence
(PRN Sequence).

Jitter Tolerance

Jitter Tolerance is a measure of the AD807’s ability to track a
jittery input data signal. Jitter on the input data is best thought
of as phase modulation, and is usually specified in unit intervals.

The PLL must provide a clock signal that tracks the phase
modulation in order to accurately retime jittered data. In order
for the VCO output to have a phase modulation that tracks the
input jitter, some modulation signal must be generated at the
output of the phase detector. The modulation output from the
phase detector can only be produced by a phase error between
its data input and its clock input. Hence, the PLL can never
perfectly track jittered data. However, the magnitude of the
phase error depends on the gain around the loop. At low fre­quencies, the integrator of the AD807 PLL provides very high
gain, and thus very large jitter can be tracked with small phase
errors between input data and recovered clock. At frequencies
closer to the loop bandwidth, the gain of the integrator is much
smaller, and thus less input jitter can be tolerated. The AD807
output will have a bit error rate less than 1 × 10

–10

when in lock
and retiming input data that has the CCITT G.958 specified
jitter applied to it.

Jitter Transfer (Refer to Figure 11)

The AD807 exhibits a low-pass filter response to jitter applied to
its input data.

REV. B

AD807

–5–

Bandwidth

This describes the frequency at which the AD807 attenuates
sinusoidal input jitter by 3 dB.

Peaking

This describes the maximum jitter gain of the AD807 in dB.

Damping Factor, ζ

Damping factor, ζ describes the compensation of the second
order PLL. A larger value of ζ corresponds to more damping
and less peaking in the jitter transfer function.

Acquisition Time

This is the transient time, measured in bit periods, required for
the AD807 to lock onto input data from its free-running state.

Symmetry—Recovered Clock Duty Cycle

Symmetry is calculated as (100 × on time)/period, where on
time equals the time that the clock signal is greater than the
midpoint between its “0” level and its “1” level.

Bit Error Rate vs. Signal-to-Noise Ratio

AD807 Bit Error Rate vs. Signal-to-Noise Ratio performance is
shown in TPC 6. Wideband amplitude noise is summed with
the input data signal as shown in Figure 4. Performance is
shown for input data levels of 5 mV and 10 mV.

V

CM

2mV p-p

SCOPE
PROBE

AD807 QUANTIZER

EPITAXX ERM504

V

CM

BINARY

OUTPUT

a. Single-Ended Input Application

V

CM

1mV p-p

SCOPE
PROBE

AD807 QUANTIZER

AD8015

DIFFERENTIAL

OUTPUT TIA

V

CM

BINARY

OUTPUT

+OUT

–OUT

b. Differential Input Application
Figure 3. (a–b) Single-Ended and Differential Input
Applications

⌺

+

+

⌺

50⍀

50⍀

0.47␮F

0.47␮F

75⍀ 1.0␮F

100⍀

GND

5V

+

–

POWER

COMBINER

POWER

COMBINER

PIN

NIN

DIFFERENTIAL

SIGNAL

SOURCE

POWER

SPLITTER

NOISE

SOURCE

FILTER100MHz

D.U.T.

AD807

Figure 4. Bit Error Rate vs. Signal-to-Noise Ratio
Test: Block Diagram

VBE 0.8V

AV

CC2

DIFFERENTIAL

INPUT

CURRENT SOURCES

HEADROOM

0.7V

0.5mA 1mA 0.5mA

AV

EE

400⍀ 400⍀

a. Quantizer Differential Input Stage

5.9k⍀

1.2V +V

BE

AV

EE

THRADJ

94.6k⍀

b. Threshold Adjust

150⍀

V

EE

SDOUT

150⍀

V

CC1

I

OH

I

OL

c. Signal Detect Output (SDOUT)

450⍀ 450⍀

V

CC2

DIFFERENTIAL
INPUT

2.5mA

V

EE

d. PLL Differential Output Stage—DATAOUT(N),
CLKOUT(N)

Figure 5. (a–d) Simplified Schematics

REV. B

AD807

–6–

–Typical Performance Characteristics

SIGNAL DETECT LEVEL – mV

200.0E+3

0.0E+0

0.0 30.05.0

R

THRESH

– ⍀

10.0 15.0 20.0 25.0

160.0E+3

140.0E+3

100.0E+3

80.0E+3

120.0E+3

180.0E+3

60.0E+3

40.0E+3

20.0E+3

35.0

TPC 1. Signal Detect Level vs. R

THRESH

TEMPERATURE – ⴗC

35.0E–3

0.0E+0
–40 80–20

SIGNAL DETECT LEVEL – Volts

0204060

25.0E–3

15.0E–3

20.0E–3

30.0E–3

10.0E–3

5.0E–3

100

R

THRESH

= OPEN

R

THRESH

= 49.9k⍀

R

THRESH

= 0⍀

TPC 2. Signal Detect Level vs. Temperature

TEMPERATURE – ⴗC

9.00

3.00
–40 80–20

ELECTRICAL HYSTERESIS – dB

0204060

7.00

5.00

6.00

8.00

4.00

100

R

THRESH

= OPEN

R

THRESH

= 49.9k⍀

R

THRESH

= 0⍀

TPC 3. Signal Detect Hysteresis vs. Temperature

SUPPLY VOLTAGE – Volts

35.000E–3

0.000E+0

4.4 5.64.6

SIGNAL DETECT LEVEL – Volts

4.8 5.0 5.2 5.4

25.000E–3

20.000E–3

10.000E–3

5.000E–3

15.000E–3

30.000E–3

R

THRESH

= OPEN

R

THRESH

= 49.9k⍀

R

THRESH

= 0⍀

TPC 4. Signal Detect Level vs. Supply Voltage

POWER SUPPLY – V

8.00

0.00

4.4 5.64.6

ELECTRICAL HYSTERESIS – dB

4.8 5.0 5.2 5.4

6.00

5.00

2.00

1.00

4.00

7.00

3.00

R

THRESH

= OPEN

R

THRESH

= 49.9k⍀

R

THRESH

= 0⍀

TPC 5. Signal Detect Hysteresis vs. Power Supply

1
2

1

2 2

S
N

)

(

erfc

10 12 14 16 18 20 22 24

S/N – dB

1E–1

5E–2
3E–2

2E–2

1E–2

1E–3

1E–4

1E–5
1E–6

1E–8

1E–10
1E–12

BIT ERROR RATE

1278

NSN

1276

1279

1277

TPC 6. Bit Error Rate vs. Signal-to-Noise Ratio

REV. B

AD807

–7–

RMS JITTER – Degrees

30

0

1.4 2.31.5

PERCENTAGE – %

1.6 1.7 1.8 2.2

20

5

15

25

10

1.9 2.0 2.1

TEST CONDITIONS
WORST-CASE:
–40ⴗC, 4.5V

TPC 7. Output Jitter Histogram

FREQUENCY – Hz

1E+3

100E–3

10E+0 10E+6

JITTER TOLERANCE – UI

10E+0

100E+0 1E+3 10E+3 100E+3

100E+0

1E+0

1E+6

AD807

SONET MASK

TPC 8. Jitter Tolerance

NOISE – V p-p @ 311MHz

3.0

0

0 0.60.1

JITTER – ns p-p

0.2 0.3 0.4 0.5

2.0

1.0

1.00.7 0.8 0.9

PSR – NO FILTER

CMR

PSR – WITH FILTER

TPC 9. Output Jitter vs. Supply Noise and
Output Jitter vs. Common Mode Noise

THEORY OF OPERATION
Quantizer

The quantizer (comparator) has three gain stages, providing a
net gain of 350. The quantizer takes full advantage of the Extra
Fast Complementary Bipolar (XFCB) process. The input stage
uses a folded cascode architecture to virtually eliminate pulse
width distortion, and to handle input signals with common­mode voltage as high as the positive supply. The input offset
voltage is factory trimmed and guaranteed to be less than 500 µV.

XFCB’s dielectric isolation allows the different blocks within
this mixed-signal IC to be isolated from each other, hence the
2 mV Sensitivity is achieved. Traditionally, high speed compara­tors are plagued by crosstalk between outputs and inputs, often
resulting in oscillations when the input signal approaches 10 mV.
The AD807 quantizer toggles at ±650 µV (1.3 mV sensitivity) at
the input without making bit errors. When the input signal is
lowered below ± 650 µV, circuit performance is dominated by
input noise, and not crosstalk.

0.1␮F

0.1␮F

0.1␮F

0.1␮F

0.1␮F

FERRITE BEAD

OPTIONAL FILTER

0.1␮F

50⍀

309⍀

50⍀50⍀

3.65k⍀

+5V

10␮F

CHOKE

“BIAS TEE”

311MHz
NOISE
INPUT

0.1␮F

0.1␮F
500⍀

500⍀

13

12

14

11

6

3

PIN

NIN

AV

CC2

AV

CC1

V

CC1

V

CC2

AD807

0.1␮F

QUANTIZER
INPUT

Figure 6. Power Supply Noise Sensitivity Test Circuit

0.1␮F

0.1␮F

0.1␮F

0.1␮F

0.1␮F

50⍀

309⍀

50⍀50⍀

3.65k⍀

+5V

10␮F

CHOKE

“BIAS TEE”

311MHz
NOISE
INPUT

0.1␮F

0.1␮F
500⍀

500⍀

13

12

14

11

6

3

PIN

NIN

AV

CC2

AV

CC1

V

CC1

V

CC2

AD807

0.1␮F

QUANTIZER
INPUT

Figure 7. Common-Mode Rejection Test Circuit

Signal Detect

The input to the signal detect circuit is taken from the first stage
of the quantizer. The input signal is first processed through a gain
stage. The output from the gain stage is fed to both a positive
and a negative peak detector. The threshold value is subtracted
from the positive peak signal and added to the negative peak signal.
The positive and negative peak signals are then compared. If the
positive peak, POS, is more positive than the negative peak,
NEG, the signal amplitude is greater than the threshold, and the
output, SDOUT, will indicate the presence of signal by remain­ing low. When POS becomes more negative than NEG, the
signal amplitude has fallen below the threshold, and SDOUT
will indicate a loss of signal (LOS) by going high. The circuit
provides hysteresis by adjusting the threshold level higher by a
factor of two when the low signal level is detected. This means
that the input data amplitude needs to reach twice the set LOS
threshold before SDOUT will signal that the data is again valid.
This corresponds to a 3 dB optical hysteresis.

REV. B

AD807

–8–

AD807
COMPARATOR
STAGES AND
CLOCK RECOVERY
PLL

PIN

NIN

THRESHOLD

BIAS

⌺

+

+

IHYS

ITHR

SDOUT

POSITIVE

PEAK

DETECTOR

NEGATIVE

PEAK

DETECTOR

LEVEL-

SHIFT

UP

LEVEL-

SHIFT
DOWN

Figure 8. Signal Level Detect Circuit Block Diagram

Phase-Locked Loop

The phase-locked loop recovers clock and retimes data from
NRZ data. The architecture uses a frequency detector to aid
initial frequency acquisition; refer to Figure 9 for a block diagram.
Note the frequency detector is always in the circuit. When the
PLL is locked, the frequency error is zero and the frequency
detector has no further effect. Since the frequency detector is
always in the circuit, no control functions are needed to initiate
acquisition or change mode after acquisition.

⌺

⌽

DET

F

DET

DATA

INPUT

S + 1

RETIMING

DEVICE

1
S

VCO

RECOVERED CLOCK
OUTPUT

RETIMED DATA
OUTPUT

Figure 9. PLL Block Diagram

The frequency detector delivers pulses of current to the charge
pump to either raise or lower the frequency of the VCO. During
the frequency acquisition process the frequency detector output
is a series of pulses of width equal to the period of the VCO.
These pulses occur on the cycle slips between the data frequency
and the VCO frequency. With a maximum density data pattern
(1010 . . . ), every cycle slip will produce a pulse at the frequency
detector output. However, with random data, not every cycle
slip produces a pulse. The density of pulses at the frequency
detector output increases with the density of data transitions. The
probability that a cycle slip will produce a pulse increases as the
frequency error approaches zero. After the frequency error has
been reduced to zero, the frequency detector output will have
no further pulses. At this point the PLL begins the process of phase
acquisition, with a settling time of roughly 2000 bit periods.

Jitter caused by variations of density of data transitions (pattern
jitter) is virtually eliminated by use of a new phase detector
(patented). Briefly, the measurement of zero phase error does
not cause the VCO phase to increase to above the average run
rate set by the data frequency. The jitter created by a 2

7

–1
pseudorandom code is 1/2 degree, and this is small compared to
random jitter.

The jitter bandwidth for the PLL is 0.06% of the center fre­quency. This figure is chosen so that sinusoidal input jitter at
92 kHz will be attenuated by 3 dB.

The damping ratio of the PLL is user programmable with a
single external capacitor. At 155 MHz, a damping ratio of 5
is obtained with a 0.15 µF capacitor. More generally, the damp­ing ratio scales as (f

DATA

× CD)

1/2

.

A lower damping ratio allows a faster frequency acquisition;
generally the acquisition time scales directly with the capacitor
value. However, at damping ratios approaching one, the acquisi­tion time no longer scales directly with capacitor value. The
acquisition time has two components: frequency acquisition and
phase acquisition. The frequency acquisition always scales with
capacitance, but the phase acquisition is set by the loop band­width of the PLL and is independent of the damping ratio.
Thus, the 0.06% fractional loop bandwidth sets a minimum
acquisition time of 2000 bit periods. Note the acquisition time
for a damping factor of one is 15,000 bit periods. This comprises
13,000 bit periods for frequency acquisition and 2,000 bit peri­ods for phase acquisition. Compare this to the 400,000 bit
periods acquisition time specified for a damping ratio of 5; this
consists entirely of frequency acquisition, and the 2,000 bit
periods of phase acquisition is negligible.

While a lower damping ratio affords faster acquisition, it also
allows more peaking in the jitter transfer response (jitter peaking).
For example, with a damping ratio of 10, the jitter peaking is

0.02 dB, but with a damping ratio of 1, the peaking is 2 dB.

Center Frequency Clamp (Figure 10)

An N-channel FET circuit can be used to bring the AD807 VCO
center frequency to within ±10% of 155 MHz when SDOUT
indicates a Loss of Signal (LOS). This effectively reduces the
frequency acquisition time by reducing the frequency error
between the VCO frequency and the input data frequency at
clamp release. The N-FET can have “on” resistance as high as
1 kΩ and still attain effective clamping. However, the chosen
N-FET should have greater than 10 MΩ “off” resistance and
less than 100 nA leakage current (source and drain) so as not to
alter normal PLL performance.

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

AD807

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CLKOUTP

V

CC1

CF1

CF2

V

EE

SDOUT

AV

CC2

PIN

NIN

AV

CC1

THRADJ

AV

EE

N_FET

C

D

Figure 10. Center Frequency Clamp Schematic

FREQUENCY – Hz

10 20k

0.02dB/DIV

100 1k 10k

CD PEAK

0.1

0.15

0.22

0.33

0.12

0.08

0.06

0.04

Figure 11. Jitter Transfer vs. C

D

REV. B

AD807

–9–

AD807

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CF1

CF2

V

EE

SDOUT

PIN

NIN

AV

CC1

THRADJ

AV

EE

50⍀ STRIP LINE
EQUAL LENGTH

C7–C10 ARE 0.1␮F BYPASS
CAPACITORS

RIGHT ANGLE SMA CONNECTOR
OUTER SHELL TO GND PLANE

ALL RESISTORS ARE 1% 1/8 WATT
SURFACE MOUNT

TPx

O

TEST POINTS ARE VECTORBOARD

K24A/M PINS

NOTES:

TP7 TP8

J5

SDOUT

C9

R13

301⍀

R14

49.9⍀

R15

49.9⍀

C13

0.1␮F

C14

0.1␮F

R16

3.65k⍀

C12

0.1␮F

J6

J7

PIN

NIN

VECTOR PINS SPACED FOR RN55C
TYPE RESISTOR; COMPONENT
SHOWN FOR REFERENCE ONLY

V

CC1

CLKOUTP

AV

CC2

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

NOTE:
INTERCONNECTION
RUN UNDER DUT

C10

R

THRESH

TP5

TP6

TP1

TP2

CD

VECTOR PINS SPACED THROUGH-HOLE
CAPACITOR ON VECTOR CUPS; COMPONENT
SHOWN FOR REFERENCE ONLY

R5 100⍀

R6 100⍀

50⍀ STRIP LINE
EQUAL LENGTH

R10
154⍀

R9

154⍀

C7

R7 100⍀

R8 100⍀

C8

R12
154⍀

R11

154⍀

C11

10␮F

TP3

TP4

5V GND

R2

100⍀

R1

100⍀

C3 0.1␮F

C4 0.1␮F
C5 0.1␮F

C6 0.1␮F

R4

100⍀

R3

100⍀

C2

0.1␮F

DATAOUTN

DATAOUTP

CLKOUTN

CLKOUTP

C1 0.1␮F

J1

J2
J3

J4

Figure 12. Evaluation Board Schematic

Figure 13. Evaluation Board Pictorials

CIRCUIT SIDE

08-002901-02

REV A

SILKSCREEN TOP

08-002901-03

REV A

INT2

08-002901-08

REV A

COMPONENT SIDE

08-002901-01

REV A

INT1

08-002901-07

REV A

SOLDERMASK TOP

08-002901-04

REV A

REV. B

AD807

–10–

AD807

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CF1

CF2

V

EE

SDOUT

PIN

NIN

AV

CC1

THRADJ

AV

EE

SDOUT

TP7

R14
50⍀

R15
50⍀

V

CC1

CLKOUTP

AV

CC2

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

TP5

TP1

TP2

CD

R5 100⍀

R6 100⍀

R10
154⍀

R9

154⍀

C7

R7 100⍀

R8 100⍀

C8

R12
154⍀

R11

154⍀

TP4

R2

100⍀

R1

100⍀

C2 0.1␮F

C3 0.1␮F
C4 0.1␮F

C5 0.1␮F

R4

100⍀

R3

100⍀

C2

0.1␮F

DATAOUTN

DATAOUTP

CLKOUTN

CLKOUTP

NOTES:

1. ALL CAPACITORS ARE CHIP,
15pF ARE MICA.

2. 150nH ARE SMT

3. C7, C8, C10, C11 ARE 0.1␮F
BYPASS CAPACITORS

C1 0.1␮F

J1

J2
J3

J4

TP6
R13

THRADJ

C10

C11

C15

0.1␮F

C14

0.1␮F

R16
301⍀

C12

2.2␮F

R17

3.65k⍀

C13

0.1␮F

NC

I

IN

NC

V

BYP

+V

S

+OUT

–V

S

–OUT

AD8015

1

2

3

4

8

7

6

5

ABB HAFO 1A227

FC HOUSING

0.8A/W, 0.7pF

2.5GHz

0.01␮F0.1␮F

NC = NO CONNECT

0.1␮F

10␮F

15pF

150nH

150nH

15pF

TP3

5V

C9

10␮F

50⍀
LINE

50⍀
LINE

Figure 14. Low Cost 155 Mbps Fiber Optic Receiver Schematic

Table I. AD807—AD8015 Fiber Optic Receiver Circuit:

Output Bit Error Rate and Output Jitter vs. Input Power

Average Optical
Input Power Output Bit Output Jitter
(dBm) Error Rate (ps rms)

–6.4 Loses Lock
–6.5 7.5 × 10

–3

–6.6 9.4 × 10

–4

–6.7 0 × 10

–14

–7.0 to –35.5 0 × 10

–14

<40

–36.0 3 × 10

–12

<40

–36.5 4.8 × 10

–10

–37.0 2.8 × 10

–8

–38.0 1.3 × 10

–5

–39.0 1.0 × 10

–3

–39.2 1.9 × 10

–3

–39.3 Loses Lock

APPLICATIONS
Low Cost 155 Mbps Fiber Optic Receiver

The AD807 and AD8015 can be used together for a complete
155 Mbps Fiber Optic Receiver (Quantizer and Clock Recovery,
and Transimpedance Amplifier) as shown in Figure 14.

The PIN diode front end is connected to a single mode 1300 nm
laser source. The PIN diode has 3.3 V reverse bias, 0.8 A/W
responsively, 0.7 pF capacitance, and 2.5 GHz bandwidth.

The AD8015 outputs (P

OUT

and N

OUT

) drive a differential,

constant impedance (50 Ω) low-pass filter with a 3 dB cutoff
of 100 MHz. The outputs of the low-pass filter are ac coupled
to the AD807 inputs (PIN and NIN). The AD807 PLL damp­ing factor is set at 7 using a 0.22 µF capacitor.

503mV

100mV/

DIV

–497mV

48.12ns 1ns/DIV 58.12ns

Figure 15. Receiver Output (Data) Eye Diagram,
–7.0 dBm Optical Input

503mV

100mV/

DIV

–497mV

49.12ns 1ns/DIV 59.12ns

Figure 16. Receiver Output (Data) Eye Diagram,
–36.0 dBm Optical Input

REV. B

AD807

–11–

AD807

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CF1

CF2

V

EE

SDOUT

PIN

NIN

AV

CC1

THRADJ

AV

EE

SDOUT

V

CC1

CLKOUTP

AV

CC2

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

CD

R5 100⍀

R6 100⍀

R10
154⍀

R9

154⍀

C7 0.1␮F

R7 100⍀

R8 100⍀

C8

0.1␮F

R12
150⍀

R11

150⍀

R2

100⍀

R1

100⍀

C2 0.1␮F

C3 0.1␮F
C4 0.1␮F

C5 0.1␮F

R4

100⍀

R3

100⍀

C6

0.1␮F

C1 0.1␮F

J1

J2

J3

J4

R13
THRADJ

C11 0.1␮F

C10 0.1␮F

R14
47⍀

R15
47⍀

R16

330⍀

C9

10␮F

5V

C13

0.1␮F

C14

0.1␮F

30pF 30pF

NOISE FILTER

120nH

R17

3.9k⍀

C12

0.1␮F

EPITAXX
ERM504

1

2

PIN TIA

1␮F

NOTE:
PIN TIA PIN 4 (CASE)
IS CONNECTED TO
GROUND

Figure 17. AD807 Application with Epitaxx PIN—Transimpedance Amplifier Module

The entire circuit was enclosed in a shielded box. Table I sum­marizes results of tests performed using a 2

23

-1 PRN Sequence,

and varying the average power at the PIN diode.

The circuit acquires and maintains lock with an average input
power as low as –39.25 dBm.

Table II. AD807—Epitaxx ERM504 PIN TIA 155 Mbps
Fiber Optic Receiver Circuit: Output Bit Error Rate and
Output Jitter vs. Average Input Power

Average Optical
Input Power Output Bit Output Jitter
(dBm) Error Rate (ps rms)

0 0.0 × 10

–10

29

–3 0.0 × 10

–10

35

–10 0.0 × 10

–10

40

–20 0.0 × 10

–10

37

–30 0.0 × 10

–10

33

–32 0.0 × 10

–10

35

–34 0.0 × 10

–10

36

–35 0.0 × 10

–10

39

–35.5 0.0 × 10

–10

40

–36 0.0 × 10

–10

41

–37.0 0.0 × 10

–10

42

–37.6 0.5 × 10

–10

43

–38.0 4 × 10

–6

50

SONET (OC-3)/SDH (STM-1) Fiber Optic Receiver Circuit

A light wave receiver circuit for SONET/SDH application at
155 Mbps is shown in Figure 17, with test results given in Table
II. The circuit operates from a single 5 V supply, and uses two
major components: an Epitaxx ERM504 PIN-TIA module with
AGC, and the AD807 IC.

A 120 MHz, third order, low-pass Butterworth filter at the
output of the PIN-TIA module provides adequate bandwidth
(70% of the bit rate), and attenuates high frequency (out of
band) noise.

250mV

50mV/

DIV

–250mV

38.12ns 1ns/DIV 48.12ns

Figure 18. Receiver Output (Data) Eye Diagram,
0 dBm Optical Input

250mV

50mV/

DIV

–250mV

38.12ns 1ns/DIV 48.12ns

Figure 19. Receiver Output (Data) Eye Diagram,
–38 dBm Optical Input

REV. B

–12–

C00862–0–12/00 (rev. B)

PRINTED IN U.S.A.

AD807

USING THE AD807
Ground Planes

Use of one ground plane for connections to both analog and
digital grounds is recommended.

Power Supply Connections

Use of a 10 µF capacitor between VCC and ground is recom­mended. Care should be taken to isolate the 5 V power trace to
V

CC2

(Pin 3). The V

CC2

pin is used inside the device to provide

the CLKOUT and DATAOUT signals.

Use of 0.1 µF capacitors between IC power supply and ground
is recommended. Power supply decoupling should take place as
close to the IC as possible. Refer to the schematic, Figure 12,
for recommended connections.

Transmission Lines

Use of 50 Ω transmission lines are recommended for PIN, NIN,
CLKOUT, and DATAOUT signals.

Terminations

Termination resistors should be used for PIN, NIN, CLKOUT,
and DATAOUT signals. Metal, thick film, 1% tolerance resistors
are recommended. Termination resistors for the PIN, NIN signals
should be placed as close as possible to the PIN, NIN pins.

Connections from 5 V to load resistors for PIN, NIN, CLKOUT,
and DATAOUT signals should be individual, not daisy chained.
This will avoid crosstalk on these signals.

Loop Damping Capacitor, C

D

A ceramic capacitor may be used for the loop damping capaci­tor. Using a 0.15 µF,

+20% capacitor for a damping factor of

five provides < 0.1 dB jitter peaking.

AD807 Output Squelch Circuit

A simple P-channel FET circuit can be used in series with the
Output Signal ECL Supply (V

CC2

, Pin 3) to squelch clock and
data outputs when SDOUT indicates a loss of signal (Figure 20).
The V

CC2

supply pin draws roughly 61 mA (14 mA for each of 4

ECL loads, plus 5 mA for all 4 ECL output stages). This means
that selection of a FET with ON RESISTANCE of 0.5 Ω will
affect the common mode of the ECL outputs by only 31 mV.

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

AD807

DATAOUTN

DATAOUTP

V

CC2

CLKOUTN

CLKOUTP

V

CC1

CF1

CF2

V

EE

SDOUT

AV

CC2

PIN

NIN

AV

CC1

THRADJ

AV

EE

P_FET

5V

BYPASS

CAP

TO V

CC1

, AVCC, AV

CC2

Figure 20. Squelch Circuit Schematic

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

16-Lead Small Outline IC Package

(R-16A)

16 9

81

0.1574 (4.00)

0.1497 (3.80)

0.3937 (10.00)

0.3859 (9.80)

0.050 (1.27)
BSC

PIN 1

0.2440 (6.20)

0.2284 (5.80)

SEATING
PLANE

0.0098 (0.25)

0.0040 (0.10)

0.0192 (0.49)

0.0138 (0.35)

0.0688 (1.75)

0.0532 (1.35)

8ⴗ
0ⴗ

0.0196 (0.50)

0.0099 (0.25)

ⴛ 45ⴗ

0.0500 (1.27)

0.0160 (0.41)

0.0099 (0.25)

0.0075 (0.19)