Datasheet CS5111YDWFR24, CS5111YDWF24 Datasheet (Cherry Semiconductor)

1

Features

■ Linear Regulator

5V ± 2% @ 100mA

■ Switching Regulator

1.4A Peak Internal
Switch

120kHz Maximum
Switching Frequency

5V to 26V Operating
Supply Range

■ Smart Functions

Watchdog

■ Protection

Overvoltage

Overtemperature

Current Limit

■ 54V Peak Transient

Capability

ENABLE

RESET

Package Option

24 Lead SO Wide

(Internally Fused Leads)

CS5111

1.4A Switching Regulator with 5V, 100mA Linear
Regulator with Watchdog, RESET and ENABLE

V

REG

V

LIN

I

BIAS

Gnd
Gnd

Gnd
Gnd

RESET
C

Delay

WDI
C

OSC

V

IN

NC
NC

V

SW

Gnd
Gnd

Gnd
Gnd

V

FB1

V

FB2

SELECT

COMP

ENABLE

CS5111

Description

Over

Temperature

V

IN

Linear
Error Amplifier

1.25V

V

REG

1.4A

V

SW

COMP

V

FB1

V

FB2

SELECT

V

LIN

I

BIAS

C

DELAY

Over Voltage

RESET &

Watchdog Timer

Current

Limit

WDI

C

OSC

Base
Drive

RESET

Gnd

Bandgap

Reference

Oscillator

Multiplexer

+

-

COMP

Logic

+

-

+

-

+

­Switcher Shutdown

Switcher
Error Amplifier

Current Sense Amplifier

ENABLE

Block Diagram

The CS5111 is a dual output power sup­ply integrated circuit. It contains a 5V
±2%, 100mA linear regulator, a watchdog
timer, a linear output voltage monitor to
provide a Power On Reset (POR) and a

1.4A current mode PWM switching reg­ulator.

The 5V linear regulator is comprised of
an error amplifier, reference, and super­visory functions. It has low internal sup­ply current consumption and provides

1.2V (typical) dropout voltage at maxi­mum load current.

The watchdog timer circuitry monitors
an input signal (WDI) from the micro­processor. It responds to the falling
edge of this watchdog signal. If a correct
watchdog signal is not received within
the externally programmable time, a
reset signal is issued.

The externally programmable active
reset circuit operates correctly for an out­put voltage (V

LIN

) as low as 1V. During

power up, or if the output voltage shifts

below the regulation limit, tog­gles low and remains low for the duration
of the delay after proper output voltage
regulation is restored. Additionally a reset
pulse is issued if the correct watchdog is
not received within the programmed
time. Reset pulses continue until the cor­rect watchdog signal is received. The
reset pulse width and frequency, as well
as the Power On Reset delay, are set by
one external RC network.

The current mode PWM switching regu­lator is comprised of an error amplifier
with selectable feedback inputs, a cur­rent sense amplifier, an adjustable oscil­lator, and a 1.4A output power switch
with anti-saturation control. The switch­ing regulator can be configured in a
variety of topologies.

The CS5111 is load dump capable and
has protection circuitry which includes
overvoltage shutdown, current limit on
the linear and switcher outputs, and an
overtemperature limiter.

RESET

Rev. 12/28/98

Cherry Semiconductor Corporation

2000 South County Trail, East Greenwich, RI 02818

Tel: [phone] Fax: [phone]

Email: [e-mail]

Web Site: www.cherry-semi.com

A Company

®

1

2

CS5111

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Absolute Maximum Ratings

Logic Inputs/Outputs ( , SELECT, WDI, ) ................................................................................-0.3V to V

LIN

V

LIN

................................................................................................................................................................................-0.3V to 10V

VIN, V

REG:

DC Input Voltage .................................................................................................................................................-0.3V to 26V

Peak Transient Voltage (40V Load Dump @ 14V VIN)....................................................................................-0.3V to 54V

VSWPeak Transient Voltage .....................................................................................................................................................54V

C

OSC

, C

Delay

, COMP,V

FB1

, V

FB2

..................................................................................................................................-0.3V to V

LIN

Power Dissipation.............................................................................................................................................Internally Limited

V

LIN

Output Current ........................................................................................................................................Internally Limited

VSWOutput Current .........................................................................................................................................Internally Limited

Output Sink Current ..................................................................................................................................................5mA

ESD Susceptibility (Human Body Model)..............................................................................................................................2kV

ESD Susceptibility (Machine Model).....................................................................................................................................200V

Storage Temperature...................................................................................................................................................-65 to 150°C

Lead Temperature Soldering: Reflow (SMD styles only) ..........................................60 sec. max above 183°C, 230°C peak

RESET

RESETENABLE

Electrical Characteristics: 5V ≤ V

IN

≤ 26V and -40°C ≤ TJ ≤ 150°C, C

OUT

= 100µF (ESR≤8Ω), C

Delay

= 0.1µF, R

BIAS

= 64.9kΩ,

C

OSC

= 390 pF, C

COMP

= 0.1µF unless otherwise specified.

■ General

I

IN

Off Current 6.6V ≤ VIN≤ 26V, I

SW

= 0A 2.0 mA

IINOn Current 6.6V ≤ VIN≤ 26V, I

SW

= 1.4A 30 70 mA

I

REG

Current I

LIN

= 100mA, 6.6V ≤ V

REG

≤ 26V 6 mA

Thermal Limit Guaranteed by design 160 210 °C

■ 5V Regulator Section

V

LIN

Output Voltage 6.6V ≤ V

REG

≤ 26V, 1mA ≤ I

LIN

≤ 100mA 4.9 5.0 5.1 V

Dropout Voltage (V

REG

- V

LIN

) @ I

LIN

= 100mA 1.2 1.5 V
Overvoltage Shutdown 30 34 38 V
Line Regulation 6.6V ≤ V

REG

≤ 26V, I

LIN

= 5mA 5 25 mV

Load Regulation V

REG

= 19V, 1mA ≤ I

LIN

≤ 100mA 5 25 mV

Current Limit 6.6V ≤ V

REG

≤ 26V 120 mA

DC Ripple Rejection 14V ≤ V

REG

≤ 24V 60 75 dB

■ Section

Low Threshold (V

RTL

)V

LIN

Decreasing 4.05 4.25 4.45 V

High Threshold (V

RTH

)V

LIN

Increasing 4.20 4.45 4.70 V

Hysteresis V

RTH

- V

RTL

140 190 240 mV

Active High V

LIN

> V

RTH

, I

RESET

= -25µA V

LIN

- 0.5 V

Active Low V

LIN

= 1V, 10kΩ pullup from to V

LIN

0.4 V

V

LIN

= 4V, I

RESET

= 1mA 0.7 V
Delay Invalid WDI 6.25 8.78 11.0 ms
Power On Delay V

LIN

crossing V

RTH

6.25 ms

RESET

RESET

3

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

CS5111

Electrical Characteristics: 5V ≤ VIN≤ 26V and -40°C ≤ TJ≤ 150°C, C

OUT

=100µF(ESR ≤ 8Ω), C

Delay

= 0.1µF, R

BIAS

= 64.9k,

C

OSC

= 390 pF, C

COMP

= 0.1µF unless otherwise specified.

■ Watchdog Input (WDI)

VIH Peak WDI needed to activate 2.0 V
VIL 0.8 V
Hysteresis Note 1 25 50 mV
Pull-Up Resistor WDI=0V 20 50 100 kΩ
Low Threshold 6.25 8.78 11.0 ms
Floating Input Voltage 3.5 V
WDI Pulse Width 5µs

■ Switcher Section

Minimum Operating 5.0 V

Input Voltage
Switching Frequency Refer to Figure 1d. 80 95 110 kHz
Switch Saturation Voltage I

SW

= 1.4A 0.7 1.1 1.6 V
Output Current Limit 1.4 2.5 A
Max Switching Frequency V

SW

= 7.5V with 50Ω load, 120 kHz

Refer to Figure 1d.

V

FB1

Regulation Voltage 1.206 1.25 1.294 V

V

FB2

Regulation Voltage 1.206 1.25 1.294 V

V

FB1

, V

FB2

Input Current V

FB1

= V

FB2

= 5V 1 µA

Oscillator Charge Current C

OSC

= 0V 35 40 45 µA

Oscillator Discharge Current C

OSC

= 4V 270 320 370 µA

C

Delay

Charge Current C

Delay

= 0V 35 40 45 µA

Switcher Max Duty Cycle VSW= 5V with 50Ω load, 72 85 95 %

V

FB1

= V

FB2

= 1V

Current Sense Amp Gain I

SW

= 2.3A 7
Error Amp DC Gain 67 dB
Error Amp Transconductance 2700 µA/V

■ Input

VIL 0.8 1.24 V
VIH 1.30 2.0 V
Hysteresis 60 mV
Input Impedance 10 20 40 kΩ

■ Select Input

VIL (Selects V

FB1

) 4.9 ≤ V

LIN

≤ 5.1 0.8 1.25 V

VIH (Selects V

FB2

) 4.9 ≤ V

LIN

≤ 5.1 1.25 2.0 V
SELECT Pull-Up SELECT = 0V 10 24 50 kΩ
Floating Input Voltage 3.5 4.5 V

Note 1: Guaranteed by Design, not 100% tested in production.

ENABLE

RESET

4

Typical Performance Characteristics

0A 20mA 40mA 60mA 80mA 100mA

3.5mA

4.0mA

4.5mA

I

LIN

I

REG

- I

LIN

-30mA

-40mA
0A 0.5A

I

SW

-20mA

-10mA

0A

1.0A 1.5A 2.0A

I

IN

I

SW

0A 0.5A 1.0A 1.5A 2.0A

0.4V

0.0V

0.6V

0.8V

1.2V

1.0V

V

SW

0.2V

1.4V

Package Lead Description

PACKAGE LEAD # LEAD SYMBOL FUNCTION

CS5111

Figure 1a. 5V Regulator Bias Current vs. Load Current. Figure 1b. Supply Current vs. Switch Current.

Figure 1c. Switch Saturation Voltage.

Figure 1d. Oscillator Frequency (kHz) vs. C

OSC

(pF), assuming R

BIAS

=

64.9kΩ.

24 Lead SO Wide

1V

IN

Supply Voltage.

2, 3 NC No connection.

4V

SW

Collector of NPN power switch for switching regulator section.

5,6,7,8,17,18,19,20 Gnd Connected to the heat removing leads.

9V

FB1

Feedback input voltage 1 (referenced to 1.25V)

10 V

FB2

Feedback input voltage 2 (referenced to 1.25V)

11 SELECT Logic level input that selects either V

FB1

or V

FB2

. An open selects

V

FB2

. Connect to Gnd to select V

FB1

.

12 COMP Output of the transconductance error amplifier.

13 C

OSC

A capacitor connected to Gnd sets the switching frequency.
Refer to Figure 1d.

14 WDI Watchdog input. Active on falling edge.

15 C

Delay

A capacitor connected to Gnd sets the Power On Reset and
Watchdog time.

16 output. Active low if V

LIN

is below the regulation limit.

If watchdog timeout is reached, a reset pulse train is issued.

21 I

BIAS

A resistor connected to Gnd sets internal bias currents as well as
the C

OSC

and C

Delay

charge currents.

22 V

LIN

Regulated 5V output from the linear regulator section.

23 V

REG

Input voltage to the linear regulator and the internal supply cir­cuitry.

24 Logic level input to shut down the switching regulator.

ENABLE

RESETRESET

180
160

140
120

100

80
60

40

Frequency (kHz)

20

0

500 1500 2500

0

1000

C

OSC

2000 3000

(pF)

The 5V linear regulator consists of an error amplifier,
bandgap voltage reference, and a composite pass transistor.

The 5V linear regulator circuitry is shown in Figure 2.
When an unregulated voltage greater than 6.6V is applied
to the V

REG

input, a 5V regulated DC voltage will be pre-

sent at V

LIN

. For proper operation of the 5V linear regula-

tor, the I

BIAS

lead must have a 64.9kΩ pull down resistor to
ground. A 100µF or larger capacitor with an ESR <8Ω
must be connected between V

LIN

and ground. To operate
the 5V linear regulator as an independent regulator (i.e.
separate from the switching supply), the input voltage
must be tied to the V

REG

lead.

As the voltage at the V

REG

input is increased, Q1is turned

on. Q

1

provides base drive for Q2which in turn provides

base current for Q

3

. As Q3is turned on, the output voltage,

V

LIN

, begins to rise as Q3’s output current charges the out-

put capacitor, C

OUT

. Once V

LIN

rises to a certain level, the
error amplifier becomes biased and provides the appropri­ate amount of base current to Q

1

. The error amplifier mon­itors the scaled output voltage via an internal voltage
divider, R

2

through R5, and compares it to the bandgap
voltage reference. The error amplifier output or error sig­nal is an output current equal to the error amplifier’s input
differential voltage times the transconductance of the
amplifier. Therefore, the error amplifier varies the base
current to Q1, which provides bias to Q2and Q3, based on
the difference between the reference voltage and the
scaled V

LIN

output voltage.

The watchdog timer circuitry monitors an input signal
(WDI) from the microprocessor. It responds to the falling
edge of this watchdog signal which it expects to see within
an externally programmable time (see Figure 3).

The watchdog time is given by:

t

WDI

= 1.353 × C

DelayRBIAS

Using C

Delay

= 0.1µF and R

BIAS

= 64.9kΩ gives a time rang­ing from 6.25ms to 11ms assuming ideal components. Based
on this, the software must be written so that the watchdog
arrives at least every 6.25ms. In practice, the tolerance of
C

Delay

and R

BIAS

must be taken into account when calculat-

ing the minimum watchdog time (t

WDI

).

Figure 3. Timing diagram for normal regulator operation.

Figure 4. Timing diagram when WDI fails to appear within the preset
time interval, t

WDI

.

V

LIN

WDI

RESET

V

REG

t

POR

A

B

A: Watchdog waiting for
low-going transition on
WDI

50% Duty

Cycle

B: RESET stays low for
t

WDI

time.

Control Functions

5V Linear Regulator

Circuit Description

CS5111

V

Figure 2. Block diagram of 5V linear regulator portion of the CS5111.

5

REG

Over Voltage

1.25V

I

BIAS

R

BIAS

64.9kΩ

C

delay

Bandgap

Reference

Watchdog Timer

WDI

+

-

RESET &

R
Linear
Error
Amplifier

Current

Limit

Over

Temperature

1

Q

2

Q

Q

1

3

R

2

V

LIN

C

= 100µF

OUT

ESR < 8Ω

R

3

R

4

R

5

RESET

V

RESET

REG

WDI

V

LIN

t

POR

Normal Operation

6

Circuit Description: continued

CS5111

If a correct watchdog signal is not received within the
specified time a reset pulse train is issued until the correct
watchdog signal is received. The nominal reset signal in
this case is a 5 volt square wave with a 50% duty cycle as
shown in Figure 4.

The signal frequency is given by:

f

RESET

=

The Power On Reset (POR) and low voltage use
the same circuitry and issue a reset when the linear output
voltage is below the regulation limit. After V

LIN

rises
above the minimum specified value, remains low
for a fixed period t

POR

as shown in Figure 5.

The POR delay (t

POR

) is given by:

t

POR

= 1.353 × C

Delay RBIAS

Figure 5a. The power on reset time interval (t

POR

) begins when V

LIN

rises above 4.45V (typical).

Figure 5b. signal is issued whenever V

LIN

falls below 4.25V

(typical).

The current mode PWM switching voltage regulator con­tains an error amplifier with selectable feedback inputs, a
current sense amplifier, an adjustable oscillator and a 1.4A
output power switch with antisaturation control. The
switching regulator and external components, connected
in a boost configuration, are shown in Figure 6.

The switching regulator begins operation when V

REG

and

VINare raised above 5 volts. V

REG

is required since the
switching supply’s control circuitry is powered through
V

LIN

. VINsupplies the base drive to the switcher output

transistor.
The output transistor turns on when the oscillator starts to

charge the capacitor on C

OSC

. The output current will
develop a voltage drop across the internal sense resistor
(RS). This voltage drop produces a proportional voltage at
the output of the current sense amplifier, which is com­pared to the output of the error amplifier. The error ampli­fier generates an output voltage which is proportional to
the difference between the scaled down output boost volt­age (V

FB1

or V

FB2

)

and the internal bandgap voltage refer­ence. Once the current sense amplifier output exceeds the
error amplifier’s output voltage, the output transistor is
turned off.

The energy stored in the inductor during the output tran­sistor on time is transferred to the load when the output
transistor is turned off. The output transistor is turned
back on at the next rising edge of the oscillator. On a cycle
by cycle basis, the current mode controller in a discontinu­ous mode of operation charges the inductor to the appro­priate amount of energy, based on the energy demand of
the load. Figure 7 shows the typical current and voltage
waveforms for a boost supply operating in the discontinu­ous mode.

NOTES:

1. Refer to Figure 1d to determine oscillator frequency.

2. The switching regulator can be disabled by providing a

logic high at the input.

3. The boost output voltage can be controlled dynamically

by the feedback select input. If select is open, V

FB2

is

selected. If select is low, then V

FB1

is selected.

If the input voltage at V

REG

is increased above the over­voltage threshold, the drive to the linear and switcher out­put transistors is shut off. Therefore, V

LIN

is disabled and

VSWcan not be pulled low.
The current out of V

LIN

is sensed in order to limit exces­sive power dissipation in the linear output transistor over
the output range of 0V to regulation. Also, the current into
VSWis sensed in order to provide the current limit func­tion in the switcher output transistor.

If the die temperature is increased above 160°C, either due
to excessive ambient temperature or excessive power dis­sipation, the drive to the linear output transistor is
reduced proportionally with increasing die temperature.
Therefore, V

LIN

will decrease with increasing die tempera­ture above 160°C. Since the switcher control circuitry is
powered through V

LIN

, the switcher performance, includ-

ing current limit, will be affected by the decrease in V

LIN

.

Protection Circuitry

ENABLE

Current Mode PWM Switching Circuitry

RESET

RESET

RESET

1

2(t

WDI

)

RESET

V

LIN

4.45V

4.25V

RESET

V

R

V

R

PEAK

V

4.25V

LO

LIN

5V

t

POR

RESET

5V

t

POR

7

Application Notes

CS5111

Circuit Description: continued

This section outlines a procedure for designing a boost switch­ing power supply operating in the discontinuous mode.

Step 1

Determine the output power required by the load.

P

OUT

= I

OUTVOUT

(1)

Step 2

Choose C

OSC

based on the target oscillator frequency with an

external resistor value, R

BIAS

= 64.9kΩ. (See Figure 1d).

Figure 7: Voltage and current waveforms for boost topology in CS5111.

Step 3

Next select the output voltage feedback sense resistor
divider as follows (Figure 8).

For V

FB1

active, choose a value for R1and then solve for

REQwhere:

R

EQ

=

.

(3a)

For V

FB2

active, find:

V

FB1

= V

OUT

, (3b)

and then calculate R2where:

R2= =

. (3c)

Then find R

3

, where:

R

3

= REQ- R2. (3d)

V

FB1

-

V

FB2

V

FB1/REQ

V

R2

I

R2

)

R

EQ

R1+ R

EQ

(

R

1

Design Procedure for Boost Topology

Figure 6: Block diagram of the 1.4A current mode control switching regulator portion of the CS5111 in a boost configuration.

Figure 8. Feedback sense
resistor divider connected
between V

OUT

and ground.

-1

V

OUT

V

FB1

V

LIN

I

BIAS

R

BIAS

64.9kΩ
C

OSC

Oscillator

V

REG

Over Voltage

COMP

COMP

Current Sense Amplifier

ENABLE

Switcher
Error
Amplifier

-

+

Logic

+

­Switcher Shutdown

Multiplexer

Base Drive

+

-

1.25V

V

V

FB1

FB2

V

IN

V

SW

1.4A

R

S

Bandgap

Reference

SELECT

Gnd

V

C

OUT

R

R

R

OUT

1

2

3

V

V

OUT

V

SAT

0

I

Peak

0

I

Peak

SW

V

IN

I

SW

I

D

t

t

R

EQ

V

OUT

V

R2

{

R

1

V

FB1

R

2

V

FB2

R

3

0

t

8

Application Notes: continued

CS5111

Step 4

Determine the maximum on time at the minimum oscilla­tor frequency and VIN. For discontinuous operation, all of
the stored energy in the inductor is transferred to the load
prior to the next cycle. Since the current through the
inductor cannot change instantaneously and the induc­tance is constant, a volt-second balance exists between the
on time and off time. The voltage across the inductor dur­ing the on cycle is VINand the voltage across the inductor
during the off cycle is V

OUT

- VIN. Therefore:

VINton= (V

OUT-VIN)toff

(4a)

where the maximum on time is:

t

on(max)

≈ . (4b)

Step 5

Calculate the maximum inductance allowed for discontin­uous operation:

L

(max)

= (5)

where η = efficiency.

Usually η = 0.75 is a good starting point. The IC’s power
dissipation should be calculated after the peak current has
been determined in Step 6. If the efficiency is less than
originally assumed, decrease the efficiency and recalculate
the maximum inductance and peak current.

Step 6

Determine the peak inductor current at the minimum
inductance, minimum V

IN

and maximum on time to make

sure the inductor current doesn’t exceed 1.4A.

I

pk

=

(6)

Step 7

Determine the minimum output capacitance and maxi­mum ESR based on the allowable output voltage ripple.

C

OUT(min)

=

(7a)

ESR

(min)

=

(7b)

In practice, it is normally necessary to use a larger capaci­tance value to obtain a low ESR. By placing capacitors in
parallel, the equivalent ESR can be reduced.

Step 8

Compensate the feedback loop to guarantee stability
under all operating conditions. To do this, we calculate the
modulator gain and the feedback resistor network attenu­ation and set the gain of the error amplifier so that the

overall loop gain is 0dB at the crossover frequency, f

CO

. In
addition, the gain slope should be -20dB/decade at the
crossover frequency.

The low frequency gain of the modulator (i.e. error ampli­fier output to output voltage) is:

=

√

,

(8a)

where

I

pk(max)

=

=

=2.3A.

The V

OUT/VEA

transfer function has a pole at:

fp= 1/(πR

LoadCOUT

) , (8b)

and a zero due to the output capacitor’s ESR at:

f

z

= 1/(2πESR C

OUT

). (8c)

Since the error amplifier reference voltage is 1.25V, the
output voltage must be divided down or attenuated
before being applied to the input of the error amplifier.
The feedback resistor divider attenuation is:

.

The error amplifier in the CS5111 is an operational transcon­ductance amplifier (OTA), with a gain given by:

G

OTA

= gmZ

OUT

(8d)

where:

gm = . (8e)

For the CS5111, gm = 2700µA/V typical.

One possible error amplifier compensation scheme is
shown in Figure 9. This gives the error amplifier a gain
plot as shown in Figure 10.

For the error amplifier gain shown in Figure 10, a low fre­quency pole is generated by the error amplifier output
impedance and C

1

. This is shown by the line AB with a ­20dB/decade slope in Figure 12. The slope changes to zero
at point B due to the zero at:

fz= 1/(2πR4C1). (8f)

Figure 9. RC network used to compensate the error amplifier (OTA).

∆I

OUT

∆V

IN

1.25V
V

OUT

(2.4V)/(7)

150mΩ

V

EA(max)/GCSA

R

S

R

Load

Lf

2

I

pk(max)

V

EA(max)

∆V

OUT

∆V

EA

∆V

ripple

I

pk

I

pk

8f∆V

ripple

V

IN(min)ton(max)

L

(min)

f

SW(min)VIN2(min)ton2(max)

2 P

OUT

/η

]

1

f

SW(min)

[

]

1 -

V

IN(min)

V

OUT(max)

[

V

OUT

R

V

1

FB1

R

2

V

FB2

R

3

1.25V

M
U
X

+

–

Error
Amplifier

C

C

2

1

R

4

SELECT

Figure 10. Bode plot of error amplifier (OTA) gain and modulator gain
added to the feedback resistor divider attenuation.

A pole at point C:

fp= 1/(πR4C2), (8g)

offsets the zero set by the ESR of the output capacitors.

An alternative scheme uses a single capacitor as shown in
Figure 11, to roll the gain off at a relatively low frequency.

Figure 11. A typical application diagram with external components con­figured in a boost topology.

Step 9

Finally the watchdog timer period and Power on Reset
time is determined by:

t

Delay

= 1.353 × C

DelayRBIAS

. (9)

V

IN

NC
NC
V

SW

Gnd
Gnd

Gnd
Gnd

V

FB1

V

FB2

SELECT
COMP

V

REG

V

LIN

I

BIAS

Gnd
Gnd
Gnd
Gnd

C

Delay

WDI

C

OSC

R

BIAS

= 64.9kΩ

100µF
ESR<8Ω

0.1µF

C

COMP

0.33µF

L=33µH

V

IN

C

OUT

88µF

(2)

100kΩ

946Ω

7.5kΩ

R

1

R

2

R

3

C

delay

390pF

C

OSC

CS-5111

RESET

ENABLE

(1)

V

OUT

= 18V, Select > 2V

V

OUT

= 16V, Select < 0.8V

MICROPROCESSOR

5V

Pole due to error amplifier
output impedance and C

1

G

0

fz = 1/2πR4C

1

+G

B

A

C

fP = 1/π R

LoadCOUT

error amplifier gain

f

CO

fz = 1/2π ESR C

OUT

fP = 1/πR4C

2

-20dB/dec

-G

Gain (dB)

modulator gain + feedback resistor divider attenuation

9

CS5111

Application Notes: continued

Worst Case Switcher Worst Case Switcher

Linear Power Power Available Power Available

V

REG

V

IN

I

LIN

Dissipation (ΘJA= 55°C/W) (ΘJA= 35°C/W)

(V) (V) (mA) (W) (W) (W)

20 14 25 0.44 0.74 1.42
20 14 50 0.83 0.35 1.03
20 14 75 1.22 * 0.64
20 14 100 1.60 * 0.26
25 14 25 0.60 0.58 1.26
25 14 50 1.11 0.07 0.75
25 14 75 1.62 * 0.24
25 14 100 2.14 * *

Linear Regulator Output Current vs. Input Voltage

Figure 12: The shaded area shows the safe operating area of the CS5111 as a function of I

LIN

, V

REG

, and ΘJA. Refer to the table below for typical

loads and voltages.

* Subjecting the CS5111 to these conditions will exceed the maximum total power that the part can handle, thereby forcing it into thermal limit.

100

75

Θ

= 55°C/W

(mA)

50

LIN

I

25

0

05

JA

VIN = 14V
Max Total Power = 1.18W

10 20 25

15 30

V

(V)

REG

100

75

(mA)

50

LIN

I

25

0

05

Θ

VIN = 14V
Max Total Power = 1.86W

= 35°C/W

JA

10 20 25

15 30

V

(V)

REG

Part Number Description

CS5111YDWF24 24 Lead SO Wide

(internally fused leads)

CS5111YDWFR24 24 Lead SO Wide

(internally fused leads) (tape & reel)

10

Rev. 12/28/98

Thermal Data 24 Lead SO Wide

R

ΘJC

typ 9 ˚C/W

R

ΘJA

typ 55 ˚C/W

Package Specification

PACKAGE THERMAL DATA

Ordering Information

D

Lead Count Metric English

Max Min Max Min

24 Lead SO Wide 15.60 15.20 .614 .598

(internally fused leads)

PACKAGE DIMENSIONS IN mm (INCHES)

CS5111

© 1999 Cherry Semiconductor Corporation

Cherry Semiconductor Corporation reserves the right to
make changes to the specifications without notice. Please
contact Cherry Semiconductor Corporation for the latest
available information.

1.27 (.050) BSC

7.60 (.299)

7.40 (.291)

10.65 (.419)

10.00 (.394)

D

0.32 (.013)

0.23 (.009)

1.27 (.050)

0.40 (.016)

REF: JEDEC MS-013

2.49 (.098)

2.24 (.088)

0.51 (.020)

0.33 (.013)

2.65 (.104)

2.35 (.093)

0.30 (.012)

0.10 (.004)

Surface Mount Wide Body (DW); 300 mil wide