Datasheet ADP3164 Datasheet (Analog Devices)

5-Bit Programmable 4-Phase

a

FEATURES
ADOPT™ Optimal Positioning Technology for Superior

Load Transient Response and Fewest Output

Capacitors
Complies with VRM 9.1 with Lowest System Cost
4-Phase Operation at up to 500 kHz per Phase
Quad Logic-Level PWM Outputs for Interface to

External High-Power Drivers
Active Current Balancing between All Output Phases
Accurate Multiple VRM Module Current Sharing
5-Bit Digitally Programmable 1.1 V to 1.85 V Output
Total Output Accuracy ⴞ0.8% Over Temperature
Current-Mode Operation
Short Circuit Protection
Enhanced Power Good Output Detects Open Outputs

in Multi-VRM Power Systems
Overvoltage Protection Crowbar Protects

Microprocessors with No Additional

External Components

APPLICATIONS
Desktop PC Power Supplies for:

Intel Pentium

VRM Modules

®

4 Processors

Synchronous Buck Controller

ADP3164

FUNCTIONAL BLOCK DIAGRAM

VCC

REF

GND

CT

SHARE

COMP

UVLO

& BIAS

3.0V

REFERENCE

OSCILLATOR

SOFT

START

ADP3164

VID4

SET

RESET

CROWBAR

CMP

VID

DAC

VID3 VID2 VID1

CMP

4-PHASE

DRIVER

LOGIC

POWER

VID0

DAC + 20%

GOOD

DAC – 20%

g

m

PWM1

PWM2

PWM3

PWM4

PGND

PWRGD

CS–

CS+

FB

GENERAL DESCRIPTION

The ADP3164 is a highly efficient 4-phase synchronous buck
switching regulator controller optimized for converting a 12 V
main supply into the core supply voltage required by high per­formance Intel processors. The ADP3164 uses an internal 5-bit
DAC to read a voltage identification (VID) code directly from
the processor, which is used to set the output voltage between

1.1 V and 1.85 V. The ADP3164 uses a current mode PWM
architecture to drive the logic-level outputs at a programmable
switching frequency that can be optimized for VRM size and
efficiency. The four output phases share the dc output current
to reduce overall output voltage ripple. An active current bal­ancing function ensures that all phases carry equal portions of
the total load current, even under large transient loads, to mini­mize the size of the inductors.

ADOPT is a trademark of Analog Devices, Inc.
Pentium is a registered trademark of Intel Corporation.

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices.

The ADP3164 also uses a unique supplemental regulation tech­nique called active voltage positioning (ADOPT) to enhance
load transient performance. Active voltage positioning results in
a dc/dc converter that meets the stringent output voltage specifi­cations for high-performance processors, with the minimum
number of output capacitors and smallest footprint. Unlike
voltage-mode and standard current-mode architectures, active
voltage positioning adjusts the output voltage as a function of the
load current so that it is always optimally positioned for a system
transient. The ADP3164 also provides accurate and reliable short
circuit protection, adjustable current limiting, and an enhanced
Power Good output that can detect open outputs in any phase for
single or multi-VRM systems.

The ADP3164 is specified over the commercial temperature
range of 0°C to 70°C and is available in a 20-lead TSSOP package.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] © Analog Devices, Inc., 2001

1

ADP3164–SPECIFICATIONS

(VCC = 12 V, I

Parameter Symbol Conditions Min Typ Max Unit

FEEDBACK INPUT

Accuracy V

FB

1.1 V Output 1.091 1.1 1.109 V

1.6 V Output 1.587 1.6 1.613 V

1.85 V Output 1.835 1.85 1.865 V

Line Regulation ∆V
Input Bias Current I

FB

Crowbar Trip Point V

FB

CROWBAR

VCC = 10 V to 14 V 0.01 %

% of Nominal Output 115 120 125 %
Crowbar Reset Point % of Nominal Output 40 50 60 %
Crowbar Response Time t

CROWBAR

Overvoltage to PWM Going Low 400 ns

REFERENCE

Output Voltage V
Output Current I

REF

REF

VID INPUTS

Input Low Voltage V
Input High Voltage V
Input Current I
Pull-Up Resistance R

IL(VID)

IH(VID)

VID

VID

VID(X) = 0 V 70 90 µA

Internal Pull-Up Voltage 2.7 3.0 3.3 V

OSCILLATOR

Maximum Frequency
Frequency Variation f

CT Charge Current I

2

f

CT(MAX)

CT

CT

TA = 25°C, CT = 150 pF 475 575 675 kHz

= 25°C, CT = 68 pF 850 1000 1250 kHz

T

A

T

= 25°C, CT = 47 pF 1100 1300 1500 kHz

A

TA = 25°C, VFB in Regulation 260 300 340 µA

TA = 25°C, VFB = 0 V 40 65 80 µA

ERROR AMPLIFIER

Output Resistance R
Transconductance g
Output Current I
Maximum Output Voltage V
Output Disable Threshold V
–3 dB Bandwidth BW

O(ERR)

m(ERR)

O(ERR)

COMP(MAX)

COMP(OFF)

ERR

FB = 0 V 575 µA

FB Forced to V

COMP = Open 500 kHz

CURRENT SENSE

Threshold Voltage V

CS(TH)

CS+ = VCC, 143 158 173 mV

FB Forced to V

FB ≤ 750 mV 80 92 108 mV

0.8 V ≤ SHARE ≤ 1 V 0 5 mV

Input Bias Current I
Response Time t

CS+

CS

, I

CS–

CS+ = CS– = VCC 1 5 µA

CS+ – (CS–) ≥ 173 mV 50 ns

to PWM Going Low

CURRENT SHARING

Output Source Current 2 mA
Output Sink Current 300 400 µA
Maximum Output Voltage V

SHARE(MAX)

FB Forced to V

POWER GOOD COMPARATOR

Undervoltage Threshold V
Overvoltage Threshold V
Output Voltage Low V

PWRGD(UV)

PWRGD(OV)

OL(PWRGD)IPWRGD(SINK)

Percent of Nominal Output 75 80 85 %

Percent of Nominal Output 115 120 125 %

Response Time 250 ns

PWM OUTPUTS

Output Voltage Low V
Output Voltage High V
Duty Cycle Limit Per Phase

2

OL(PWM)

OH(PWM)

DC 25 %

I

PWM(SINK)

I

PWM(SOURCE)

= 150 ␮A, TA = 0ⴗC to 70ⴗC, unless otherwise noted.)

REF

550 nA

2.952 3.00 3.048 V
300 µA

0.8 V

2.0 V

33 43 kΩ

4000 kHz

1MΩ

2.0 2.2 2.45 mmho

– 3% 3.0 V

OUT

800 875 mV

– 3%

OUT

– 3% 3.0 V

OUT

= 1 mA 375 525 mV

= 400 µA 100 500 mV

= 400 µA 4.0 5.0 V

–2–

REV. 0

ADP3164

WARNING!

ESD SENSITIVE DEVICE

Parameter Symbol Conditions Min Typ Max Unit

SUPPLY

DC Supply Current

Normal Mode I
No CPU Mode I
UVLO Mode I

UVLO Threshold Voltage V

CC

CC(NO CPU)

CC(UVLO)

UVLO

VID4 – VID0 = Open 3.5 5.5 mA
VCC ≤ V

, VCC Rising 350 500 µA

UVLO

5.9 6.4 6.9 V

UVLO Hysteresis 0.5 0.8 1.0 V

NOTES

1

All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC).

2

Guaranteed by design, not tested in production.

Specifications subject to change without notice.

3.75 5.5 mA

ABSOLUTE MAXIMUM RATINGS*

VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +15 V

CS+, CS– . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VCC +0.3 V

All Other Inputs and Outputs . . . . . . . . . . . . –0.3 V to +10 V

Operating Ambient Temperature Range . . . . . . . 0°C to 70°C

Operating Junction Temperature . . . . . . . . . . . . . . . . . . 125°C

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 143°C/W

θ

JA

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C

Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . 215°C

Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C

*This is a stress rating only; operation beyond these limits can cause the device to

be permanently damaged. Unless otherwise specified, all voltages are referenced
to PGND.

ORDERING GUIDE

Model Temperature Range Package Description Package Option

ADP3164JRU 0°C to 70°C Thin Shrink Small Outline RU-20 (TSSOP-20)

PIN CONFIGURATION

RU-20

VID4

1

2

VID3

3

VID2

4

VID1

FB

CT

5

6

7

8

9

10

ADP3164

TOP VIEW

(Not to Scale)

VID0

SHARE

COMP

GND

20

19

18

17

16

15

14

13

12

11

VCC

REF

PWM1

PWM2

PWM3

PWM4

PGND

CS–

CS+

PWRGD

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the ADP3164 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

REV. 0

–3–

ADP3164

PIN FUNCTION DESCRIPTIONS

Pin Mnemonic Function

1–5 VID4 – Voltage Identification DAC Inputs. These pins are pulled up to an internal 3 V reference, providing a

VID0 Logic 1 if left open. The DAC output programs the FB regulation voltage from 1.1 V to 1.85 V. Leaving all five

DAC inputs open results in the ADP3164 going into a “No CPU” mode, shutting off its PWM outputs.

6 SHARE Current Sharing Output. This pin is connected to the SHARE pins of other ADP3164s in multiple VRM systems

to ensure proper current sharing between the converters. The voltage at this output programs the output current

control level between CS+ and CS–.
7 COMP Error Amplifier Output and Compensation Point.
8 GND Ground. FB, REF, and the VID DAC of the ADP3164 are referenced to this ground. This is a low current ground

that can also be used as a return for the FB pin in remote voltage sensing applications.
9 FB Feedback Input. Error amplifier input for remote sensing of the output voltage.
10 CT External capacitor CT connection to ground sets the frequency of the device.
11 PWRGD Open drain output that signals when the output voltage is outside of the proper operating range or when a phase

is not supplying current even if the output voltage is in specification.
12 CS+ Current Sense Positive Node. Positive input for the current comparator. The output current is sensed as a voltage

at this pin with respect to CS–.
13 CS– Current Sense Negative Node. Negative input for the current comparator.
14 PGND Power Ground. All internal biasing and logic output signals of the ADP3164 are referenced to this ground.
15 PWM4 Logic-Level Output for the Phase 4 Driver.
16 PWM3 Logic-Level Output for the Phase 3 Driver.
17 PWM2 Logic-Level Output for the Phase 2 Driver.
18 PWM1 Logic-Level Output for the Phase 1 Driver.
19 REF 3.0 V Reference Output.
20 VCC Supply Voltage for the ADP3164.

ADP3164

VCC

REF

PWM1

PWM2

PWM3

PWM4

PGND

CS–

CS+

PWRGD

1.2V

20

19

20k⍀

18

17

16

15

14

13

12

11

+

V

FB

1␮F 100nF

12V

100⍀

100nF

5-BIT CODE

1

2

3

4

5

6

7

8

9

10

VID4

VID3

VID2

VID1

VID0

SHARE

COMP

GND

FB

CT

AD820

Figure 1. Closed-Loop Output Voltage Accuracy Test Circuit

–4–

REV. 0

Typical Performance Characteristics–ADP3164

10

1

FREQUENCY – MHz

0.1
0 10050

150 250200 300

CT CAPACITANCE – pF

TPC 1. Oscillator Frequency vs. Timing Capacitor (CT)

4.5

4.4

4.3

25

20

15

10

NUMBER OF PARTS – %

5

0

–0.5

OUTPUT ACCURACY – % of Nominal

0 0.5

TPC 3. Output Accuracy Distribution

TA = 25ⴗC
V

= 1.6V

OUT

4.2

SUPPLY CURRENT – mA

4.1

4.0
0 1000500

OSCILLATOR FREQUENCY – kHz

1500 25002000 3000

TPC 2. Supply Current vs. Oscillator Frequency

REV. 0

–5–

ADP3164

THEORY OF OPERATION

The ADP3164 combines a current-mode, fixed frequency PWM
controller with multiphase logic outputs for use in a 4-phase syn­chronous buck power converter. Multiphase operation is important
for switching the high currents required by high performance
microprocessors. Handling the high current in a single-phase
converter would place unreasonable requirements on the power
components such as inductor wire size and MOSFET ON­resistance and thermal dissipation. The ADP3164’s high side
current sensing topology ensures that the load currents are bal­anced in each phase, such that no single phase has to carry
more than it’s share of the power. An additional benefit of high
side current sensing over output current sensing is that the
average current through the sense resistor is reduced by the duty
cycle of the converter allowing the use of a lower power, lower
cost resistor. The outputs of the ADP3164 are logic drivers only
and are not intended to directly drive external power MOSFETs.
Instead, the ADP3164 should be paired with drivers such as
the ADP3413.

The frequency of the ADP3164 is set by an external capacitor
connected to the CT pin. The error amplifier and current sense
comparator control the duty cycle of the PWM outputs to main­tain regulation. The maximum duty cycle per phase is inherently
limited to 25%. While one phase is on, all other phases remain
off. In no case can more than one output be high at any time.

Output Voltage Sensing

The output voltage is sensed at the FB pin allowing for remote
sensing. To maintain the accuracy of the remote sensing, the
GND pin should also be connected close to the load. A voltage
error amplifier (g
voltage and a programmable reference voltage. The reference
voltage is programmed between 1.1 V and 1.85 V by an internal
5-bit DAC, which reads the code at the voltage identification
(VID) pins. (Refer to Table I for the output voltage versus VID
pin code information.)

Active Voltage Positioning

The ADP3164 uses Analog Devices Optimal Positioning Tech­nology (ADOPT), a unique supplemental regulation technique

Table I. Output Voltage vs. VID Code

VID4 VID3 VID2 VID1 VID0 V

OUT(NOM)

11111No CPU

111101.100 V

111011.125 V

111001.150 V

110111.175 V

110101.200 V

110011.225 V

110001.250 V

101111.275 V

101101.300 V

101011.325 V

101001.350 V

100111.375 V

100101.400 V

100011.425 V

100001.450 V

011111.475 V

011101.500 V

011011.525 V

011001.550 V

010111.575 V

010101.600 V

010011.625 V

010001.650 V

001111.675 V

001101.700 V

001011.725 V

001001.750 V

000111.775 V

000101.800 V

000011.825 V

000001.850 V

that uses active voltage positioning and provides optimal com­pensation for load transients. When implemented, ADOPT adjusts
the output voltage as a function of the load current, so that it is
always optimally positioned for a load transient. Standard (passive)
voltage positioning has poor dynamic performance, rendering it
ineffective under the stringent repetitive transient conditions
required by high performance processors. ADOPT, however,
provides a bandwidth for transient response that is limited only
by parasitic output inductance. This yields optimal load tran­sient response with the minimum number of output capacitors.

Reference Output

A 3.0 V reference is available on the ADP3164. This reference
is normally used to accurately set the voltage positioning using a
resistor divider to the COMP pin. In addition, the reference can
be used for other functions such as generating a regulated voltage
with an external amplifier. The reference is bypassed with a 1 nF
capacitor to ground. It is not intended to drive larger capacitive
loads, and it should not be used to provide more than 300 µA of
output current.

Cycle-by-Cycle Operation

During normal operation (when the output voltage is regulated),
the voltage-error amplifier and the current comparator are the
main control elements. The free running oscillator ramps between
0 V and 3 V. When the voltage on the CT pin reaches 3 V, the
oscillator sets the driver logic, which sets PWM1 high. During
the ON time of Phase 1, the driver IC turns on the Phase 1 high
side MOSFET. The CS+ and CS– pins monitor the current
through the sense resistor that feeds all of the high side MOSFETs.
When the voltage between the two pins exceeds the threshold
level, the driver logic is reset and the PWM1 output goes low.
This signals the driver IC to turn off the Phase 1 high side
MOSFET and turn on the Phase 1 low side MOSFET. On the
next cycle of the oscillator, the driver logic toggles and sets
PWM2 high. The current is then steered through the second
phase. This cycle continues for each of the PWM outputs.

) amplifies the difference between the output

m

–6–

REV. 0

ADP3164

On each of the following cycles of the oscillator, the outputs
cycle between each of the active PWM outputs. In each case,
the current comparator resets the PWM output low when the
VT1 is reached. The current of each phase is sensed with the
same resistor and the same comparator, so the current is
inherently balanced. As the load current increases, the output
voltage starts to decrease. This causes an increase in the output
of the voltage error amplifier (g

), which in turn leads to an

m

increase in the current comparator threshold VT1, thus tracking
the load current.

Active Current Sharing

The ADP3164 ensures current balance in all the active phases
by sensing the current through a single sense resistor. During
one phase’s ON time, the current through the respective high
side MOSFET and inductor is measured through the sense
resistor. When the comparator threshold is reached, the high
side MOSFET turns off. On the next cycle the ADP3164
switches to the next phase. The current is measured with the
same sense resistor and the same internal comparator, ensuring
accurate matching. This scheme is immune to imbalances in the
MOSFET’s R

and inductor parasitic resistance.

DS(ON)

If for some reason one of the phases has a short circuit failure,
the other phases will still be limited to their maximum output
current (one over the total number phases times the total short
circuit current limit). If this is not sufficient to supply the load,
the output voltage will droop and cause the PWRGD output to
signal that the output voltage has fallen out of its specified
range. If one of the phases has an open circuit failure, the
ADP3164 will detect the open phase and signal the problem via
the PWRGD pin (see Power Good Monitoring section).

Current Sharing in Multi-VRM Applications

The ADP3164 includes a SHARE pin to allow multiple VRMs
to accurately share load current. In multiple VRM applications,
the SHARE pins should be connected together. This pin is a
low impedance buffered output of the COMP pin voltage. The
output of the buffer is internally connected to set the threshold
of the current sense comparator. The buffer has a 400 µA sink
current, and a 2 mA sourcing capability. The strong pull-up
allows one VRM to control the current threshold set point for
all ADP3164s connected together. The ADP3164’s high accu­racy current set threshold ensures good current balance between
VRMs. Also, the low impedance of the buffer minimizes noise
pickup on this trace which is routed to multiple VRMs. This
circuit operates in addition to the active current sharing between
phases of each VRM described above.

Short Circuit Protection

The ADP3164 has multiple levels of short circuit protection to
ensure fail-safe operation. The sense resistor and the maximum
current sense threshold voltage given in the specifications set the
peak current limit.

When the load current exceeds the current limit, the excess
current discharges the output capacitor. When the output volt­age is below the foldback threshold, V

, the maximum

FB(LOW)

deliverable output current is cut by reducing the current sense
threshold from the current limit threshold, V
back threshold, V

CS(FOLD)

. Along with the resulting current

, to the fold-

CS(CL)

foldback, the oscillator frequency is reduced by a factor of five
when the output is 0 V. This further reduces the average current
in short circuit.

Power Good Monitoring

The power good comparator monitors the output voltage of the
supply via the FB pin. The PWRGD pin is an open drain output
whose high level (when connected to a pull-up resistor) indicates
that the output voltage is within the specified range of the nomi­nal output voltage requested by the VID DAC. PWRGD will go
low if the output is outside this range.

Short circuits in a VRM power path are relatively easy to detect
in applications where multiple VRMs are connected to a com­mon power plane. VRM power train open failures are not as
easily spotted, since the other VRMs may be able to supply
enough total current to keep the output voltage within the
power good voltage specification even when one VRM is not
functioning. The ADP3164 addresses this problem by monitor­ing both the output voltage and the switch current to determine
the state of the PWRGD output.

The output voltage portion of the power good monitor domi­nates; as long as the output voltage is outside the specified
window, PWRGD will remain low. If the output voltage is
within specification, a second circuit checks to make sure that
current is being delivered to the output by each phase. If no
current is detected in a phase for three consecutive cycles, it is
assumed that an open circuit exists somewhere in the power
path, and PWRGD will be pulled low.

Output Crowbar

The ADP3164 includes a crowbar comparator that senses when
the output voltage rises higher than the specified trip threshold,
V

CROWBAR

. This comparator overrides the control loop and sets
both PWM outputs low. The driver ICs turn off the high side
MOSFETs and turn on the low side MOSFETs, thus pulling
the output down as the reversed current builds up in the induc­tors. If the output overvoltage is due to a short of the high side
MOSFET, this action will current-limit the input supply or blow
its fuse, protecting the microprocessor from destruction. The
crowbar comparator releases when the output drops below the
specified reset threshold, and the controller returns to normal
operation if the cause of the overvoltage failure does not persist.

Output Disable

The ADP3164 includes an output disable function that turns off
the control loop to bring the output voltage to 0 V. Because an
extra pin is not available, the disable feature is accomplished by
pulling the COMP pin to ground. When the COMP pin drops
below 0.8 V, the oscillator stops and all PWM signals are driven
low. This function does not place the part in low current shut­down and the reference voltage is still available. The COMP
pin should be pulled down with an open drain type of output
capable of sinking at least 2 mA.

REV. 0

–7–

ADP3164

V

12V

IN

V

RTN

IN

R

A

26.7k⍀

C

OC

1.2nF

1.5k⍀

R

B

10.5k⍀

R

Z

C11

100pF

L1

1␮H

OS-CON SP SERIES

C1

VID4

1

2

VID3

VID2

3

4

VID1

VID0

5

SHARE

6

COMP

7

8

GND

FB

9

CT

10

C10
100pF

R3

1k⍀

270␮F/16V x 3

+

+

C2

U1

ADP3164

PWM1

PWM2

PWM3

PWM4

PGND

PWRGD

4.7␮F

VCC

REF

CS–

CS+

+

C3

R4

10⍀

C5

20

19

18

17

16

15

14

13

12

11

R5
20⍀

C7

10nF

C8

1nF

C6

4.7␮F

R6
2k⍀

Z1

ZMM5263BCT

10␮Fⴛ2

MLCC

Q1
FZ649TA

R7

5m⍀

MBR052LTI

C13

4.7␮F

MBR052LTI

C16

4.7␮F

MBR052LTI

C19

4.7␮F

MBR052LTI

C22

4.7␮F

D1

1

BST

2

IN

3

NC

4

VCC

D2

1

BST

2

IN

3

NC

4

VCC

D3

1

BST

2

IN

3

NC

4

VCC

D5

1

BST

2

IN

3

DLY

4

VCC

U2

ADP3414

DRVH

PGND

DRVL

U3

ADP3414

DRVH

PGND

DRVL

U4

ADP3414

DRVH

PGND

DRVL

U5

ADP3414

DRVH

PGND

DRVL

C12

100nF

8

7

SW

6

5

Q6

FDB8030L

C15

100nF

8

7

SW

6

5

Q7

FDB8030L

C18

100nF

8

7

SW

6

5

Q8

FDB8030L

C21

100nF

8

7

SW

6

5

Q9

FDB8030L

Q2
FDB7030L

600nH

C14
15nF

R8
2⍀

Q3
FDB7030L

600nH

C17
15nF

R9
2⍀

Q4
FDB7030L

600nH

C20
15nF

R10
2⍀

Q5
FDB7030L

600nH

C23
15nF

R11
2⍀

L2

OS-CON SP SERIES

L3

L4

L5

820␮F/4V x 13

12m⍀ESR (EACH)

+ +

C24

C37

V

CC(CORE)

1.1V – 1.85V

80A
V

CC(CORE)RTN

NC = NO CONNECT

Figure 2. 80 A Intel VRM 9.1-Compliant CPU Supply Circuit

–8–

REV. 0

ADP3164

APPLICATION INFORMATION

The design parameters for a typical VRM 9.1-compliant CPU
application are as follows:

Input voltage (V
VID setting voltage (V
Nominal output voltage at no load (V
Nominal output voltage at 80 A load (V

) = 12 V

IN

) = 1.475 V

VID

) = 1.4605 V

ONL

) = 1.3845 V

OFL

Static output voltage drop based on a 0.95 mΩ load line

) from no load to full load (V∆) = V

(R

OUT

ONL

– V

OFL

=

1.4605 V – 1.3845 V = 76 mV
Maximum Output Current (I

) = 81 A

O

Number of Phases (n) = 4

CT Selection—Choosing the Clock Frequency

The ADP3164 uses a fixed-frequency control architecture. The
frequency is set by an external timing capacitor, CT. The clock
frequency determines the switching frequency, which relates
directly to switching losses and the sizes of the inductors and
input and output capacitors. A clock frequency of 800 kHz sets
the switching frequency of each phase, f

, to 200 kHz, which

SW

represents a practical trade-off between the switching losses and
the sizes of the output filter components. To achieve an 800 kHz
oscillator frequency, the required timing capacitor value is 100 pF.
For good frequency stability and initial accuracy, it is recom­mended to use a capacitor with low temperature coefficient
and tight tolerance, e.g., an MLC capacitor with NPO dielec­tric and with 5% or less tolerance.

Inductance Selection

The choice of inductance determines the ripple current in the
inductor. Less inductance leads to more ripple current, which
increases the output ripple voltage and the conduction losses in
the MOSFETs, but allows using smaller-size inductors and, for
a specified peak-to-peak transient deviation, output capacitors
with less total capacitance. Conversely, a higher inductance
means lower ripple current and reduced conduction losses, but
requires larger-size inductors and more output capacitance for
the same peak-to-peak transient deviation. In a 4-phase con­verter, a practical value for the peak-to-peak inductor ripple
current is under 50% of the dc current in the same inductor. A
choice of 50% for this particular design example yields a total
peak-to-peak output ripple current of 8% of the total dc output
current. The following equation shows the relationship between
the inductance, oscillator frequency, peak-to-peak ripple current
in an inductor and input and output voltages.

VV V

(– )

IN OUT OUT

L

=

Vf I

××

IN SW L RIPPLE

×

()

(1)

For 10 A peak-to-peak ripple current, which is 50% of the
20 A full-load dc current in an inductor, Equation 1 yields an
inductance of:

L

800

V

××

12

VV V

(–. ).12 1 475 1 475

kHz

4

×

A

10

=

646

nH=

A 600 nH inductor can be used, which gives a calculated ripple
current of 10.8 A at no load. The inductor should not saturate
at the peak current of 26 A, and should be able to handle the

sum of the power dissipation caused by the average current of
20 A in the winding and the core loss.

The output ripple current is smaller than the inductor ripple
current due to the four phases partially canceling. This can be
calculated as follows:

nV V nV

I

×× ×

4 1 475 12 4 1 475

I

=

O∆∆

Designing an Inductor

×× ×

=

O

VV V

.(–.)

V nH kHz

××

12 600 800

(– )

OUT IN OUT

VLf

××

IN OSC

=

.

625

(2)

A

Once the inductance is known, the next step is either to design
an inductor or find a standard inductor that comes as close as
possible to meeting the overall design goals. The first decision in
designing the inductor is to choose the core material. There are
several possibilities for providing low core loss at high frequencies.
Two examples are the powder cores (e.g., Kool-Mµ

®

from
Magnetics, Inc.) and the gapped soft ferrite cores (e.g., 3F3 or
3F4 from Philips). Low frequency powdered iron cores should
be avoided due to their high core loss, especially when the induc­tor value is relatively low and the ripple current is high.

Two main core types can be used in this application. Open
magnetic loop types, such as beads, beads on leads, and rods
and slugs, provide lower cost but do not have a focused mag­netic field in the core. The radiated EMI from the distributed
magnetic field may create problems with noise interference in
the circuitry surrounding the inductor. Closed-loop types, such
as pot cores, PQ, U, and E cores, or toroids, cost more, but
have much better EMI/RFI performance. A good compromise
between price and performance are cores with a toroidal shape.

There are many useful references for quickly designing a power
inductor. Table II gives some examples.

Table II. Magnetics Design References

Magnetic Designer Software
Intusoft (http://www.intusoft.com)

Designing Magnetic Components for High-Frequency DC-DC
Converters

McLyman, Kg Magnetics
ISBN 1-883107-00-08

Selecting a Standard Inductor

The companies listed in Table III can provide design consulta­tion and deliver power inductors optimized for high power
applications upon request.

Table III. Power Inductor Manufacturers

Coilcraft
[phone]
http://www.coilcraft.com

Coiltronics
[phone]
http://www.coiltronics.com

Sumida Electric Company
[phone]
http://www.sumida.com

REV. 0

–9–

ADP3164

R

SENSE

The value of R

is based on the maximum required output

SENSE

current. The current comparator of the ADP3164 has a mini­mum current limit threshold of 143 mV. Note that the 143 mV
value cannot be used for the maximum specified nominal cur­rent, as headroom is needed for ripple current and tolerances.

The current comparator threshold sets the peak of the inductor
current yielding a maximum output current, IO, which equals
twice the peak inductor current value less half of the peak-to­peak inductor ripple current. From this, the maximum value of

is calculated as:

R

SENSE

R

SENSE

≤

V

I

O

n

CSCL MIN

()

I

L RIPPLE

()

+

2

=

mV

143

AA

80410 8

.

+

2

m

=Ω

.

56

(3)

In this case, 5 mΩ was chosen as the closest standard value.
Once R
where current limit is reached, I

has been chosen, the output current at the point

SENSE

, can be calculated using

OUT(CL)

the maximum current sense threshold of 173 mV:

V

In

OUT CL

()

I

OUT CL

()

CSCL MAX

=× −

=× −

R

SENSE

mV

173

4

m

5

() ( )

Ω

nI

×

.

×

4108

2

L RIPPLE

2

A

=

116 8

A

.

(4)

At output voltages below 750 mV, the current sense threshold is
reduced to 108 mV, and the ripple current is negligible. There­fore, at dead short the output current is reduced to:

V

CS SC

In

OUT SC

()

R

()

SENSE

108

5

m

mV

Ω

.=× =× =4

86 4

A

(5)

To safely carry the current under maximum load conditions, the
sense resistor must have a power rating of at least:

PI R

=×

R SENSE RMS SENSE

SENSE

2

()

(6)

where:

2

I

I

SENSE RMS

2

=×

()

V

O OUT

n

V

×η

IN

(7)

In this formula, n is the number of phases, and η is the con­verter efficiency, in this case assumed to be 85%. Combining
Equations 6 and 7 yields:

2

AV

P

=×

R

SENSE

80

4

1 475

.

×

085 12

.

mW

×=

512

V

.Ω

Output Resistance

This design requires that the regulator output voltage measured
at the CPU drop when the output current increases. The speci­fied voltage drop corresponds to a dc output resistance of:

VV

−

R

OUT

ONL OFL

=

I

=

∆

O

VV

80

−

A

1 4605 1 3845

..

m

=Ω

095

.

(8)

The required dc output resistance can be achieved by terminating
the g

amplifier with a resistor. The value of the total termina-

m

tion resistance that will yield the correct dc output resistance:

nR

×

R

where n
the g

I SENSE

=

T

ng R

××

m OUT

is the division ratio from the output voltage signal of

I

amplifier to the PWM comparator CMP1, gm is the

m

=

transconductance of the g

12 5 5

××Ω

422 095

..

amplifier itself, and n is the number

m

m

×Ω

.

mmho m

k

=Ω

748

.

(9)

of phases.

Output Offset

Intel’s VRM 9.1 specification requires that at no load the nominal
output voltage of the regulator be offset to a lower value than the
nominal voltage corresponding to the VID code to make sure that
circuit tolerances never cause the output voltage to exceed the
VID value. The offset is introduced by realizing the total termina­tion resistance of the g

amplifier with a divider connected between

m

the REF pin and ground. The resistive divider introduces an
offset to the output of the g
through the gain of the g

amplifier that, when reflected back

m

stage, accurately positions the output

m

voltage near its allowed maximum at light load. Furthermore, the
output of the g

amplifier sets the current sense threshold voltage.

m

At no load, the current sense threshold is increased by the peak of
the ripple current in the inductor and reduced by the delay between
sensing when the current threshold has been reached and when
the high side MOSFET actually turns off. These two factors are
combined with the inherent voltage (V
g

amplifier that commands a current sense threshold of 0 mV:

m

IRn

VV

=+

GNL GNL

nt R n

×× ×

D SENSE I

VV

1

=+

GNL

ns m V

4 60 5 12 5 1 074

××Ω×=

L RIPPLE SENSE I

0

10 8 5 12 5212 1 475

...

The divider resistors (R

××

()

2

Am V V

×Ω×

..

for the upper and RB for the lower)

A

), at the output of the

GNL0

VV

−

IN OUT

−

L

−

−

600

nH

×

(10)

×

can now be calculated, assuming that the internal resistance of

amplifier (R

the g

m

R

=

B

VV

−

REF GNL

R

T

R

=

B

VV

−

.

3 1 074

k

Ω

.

748

Choosing the nearest 1% resistor value gives R
Finally, R

R

A

is calculated:

A

=

1

11111

−−

RR R k M k

T OGM B

) is 1 MΩ:

OGM

V

REF

gV V

−× −

()

m ONL VID

V

3

mmho V V

−×−

.(. .)

22 14605 1475

=

−

Ω

74811

..

1

−

Ω

10 5

k

=Ω

.

10 37

= 10.5 kΩ.

B

k

=Ω

26 7

.

Ω

(12)

(11)

Choosing the nearest 1% resistor value gives RA = 26.7 kΩ.

–10–

REV. 0

C

Selection

OUT

The required equivalent series resistance (ESR) and capacitance
drive the selection of the type and quantity of the output capaci­tors. The ESR must be less than or equal to the specified output
resistance (R

), in this case 0.95 mΩ. The capacitance must

OUT

be large enough that the voltage across the capacitors, which is
the sum of the resistive and capacitive voltage deviations, does
not deviate beyond the initial resistive step while the inductor
current ramps up or down to the value corresponding to the
new load current.

One can, for example, use thirteen SP-Type OS-CON capaci­tors from Sanyo, with 820 µF capacitance, a 4 V voltage rating,
and 12 mΩ ESR. The ten capacitors have a maximum total ESR
of 0.92 mΩ when connected in parallel.

As long as the capacitance of the output capacitor bank is above
a critical value and the regulating loop is compensated with
Analog Devices’ proprietary compensation technique (ADOPT),
the actual capacitance value has no influence on the peak-to­peak deviation of the output voltage to a full step change in the
load current. The critical capacitance can be calculated as follows:

I

C

OUT CRIT

..

0 95 1 475

=

()

A

80

mV

×

Ω

O

RVLn

OUT OUT

×=

×

nH

600

×=

4

.

856

mF

(13)

The critical capacitance limit for this circuit is 8.56 mF, while
the actual capacitance of the thirteen OS-CON capacitors is
13 × 820 µF = 10.66 mF. In this case, the capacitance is safely
above the critical value.

Multilayer ceramic capacitors are also required for high-frequency
decoupling of the processor. The exact number of these MLC
capacitors is a function of the board layout space and parasitics.
Typical designs use twenty to thirty 10 µF MLC capacitors
located as close to the processor power pins as is practical.

Feedback Loop Compensation Design for ADOPT

Optimized compensation of the ADP3164 allows the best pos­sible containment of the peak-to-peak output voltage deviation.
Any practical switching power converter is inherently limited by
the inductor in its output current slew rate to a value much less
than the slew rate of the load. Therefore, any sudden change of
load current will initially flow through the output capacitors,
and assuming that the capacitance of the output capacitor is
larger than the critical value defined by Equation 13, this will
produce a peak output voltage deviation equal to the ESR of the
output capacitor times the load current change.

The optimal implementation of voltage positioning, ADOPT,
will create an output impedance of the power converter that is
entirely resistive over the widest possible frequency range, includ­ing dc, and equal to the maximum acceptable ESR of the output
capacitor array. With the resistive output impedance, the output
voltage will droop in proportion with the load current at any load
current slew rate; this ensures the optimal positioning and allows
the minimization of the output capacitor bank.

ADP3164

With an ideal current-mode-controlled converter, where the
average inductor current would respond without delay to the
command signal, the resistive output impedance could be
achieved by having a single-pole roll-off of the voltage gain of
the voltage-error amplifier. The pole frequency must coincide
with the ESR zero of the output capacitor bank. The ADP3164
uses constant frequency current-mode control, which is known
to have a nonideal, frequency-dependent command signal to
inductor current transfer function. The frequency dependence
manifests in the form of a pair of complex conjugate poles at
one-half of the switching frequency. A purely resistive output
impedance could be achieved by canceling the complex conjugate
poles with zeros at the same complex frequencies and adding a
third pole equal to the ESR zero of the output capacitor. Such a
compensating network would be quite complicated. Fortunately, in
practice it is sufficient to cancel the pair of complex conjugate
poles with a single real zero placed at one-half of the switching
frequency. Although the end result is not a perfectly resistive
output impedance, the remaining frequency dependence causes
only a small percentage of deviation from the ideal resistive
response. The single-pole and single-zero compensation can easily
be implemented by terminating the g
parallel combination of a resistor (R
The value of the terminating resistor R
mined; the capacitance and resistance of the series RC network
are calculated as follows:

C

=

OC

C

=

OC

CR

×

OUT OUT

R

mF m

10 7 0 92

..

−

T OSC T

×

kkHzk

Ω

748

..

n

fR

××

π

Ω

−
π

The nearest standard value of COC is 1 nF. The resistance of the
zero-setting resistor in series with the compensating capacitor is:

R

=

Z

n

f C kHz nF

××=××

ππ

OSC OC

The nearest standard 5% resistor value is 1.5 kΩ. Note that this
resistor is only required when C
25% or less). In this example, C

should be included.

R

Z

Power MOSFETs

In this example, eight N-channel power MOSFETs must be used;
four as the main (control) switches, and the remaining four as
the synchronous rectifier switches. The main selection parameters
for the power MOSFETs are V
minimum gate drive voltage (the supply voltage to the ADP3414)
dictates whether standard threshold or logic-level threshold
MOSFETs must be used. Since V
old MOSFETs (V

< 2.5 V) are strongly recommended.

GS(TH)

error amplifier with the

m

) and a series RC network.

T

was previously deter-

T

4

××Ω

800 7 48

4

800 1

approaches C

OUT

is approaching C

OUT

, QG and R

GS(TH)

<8 V, logic-level thresh-

GATE

=Ω

159.

CRIT

DS(ON)

=

11

.

k

(within

CRIT

. The

nF

(14)

(15)

, so

REV. 0

–11–

ADP3164

The maximum output current IO determines the R

DS(ON)

require­ment for the power MOSFETs. When the ADP3164 is operating
in continuous mode, the simplifying assumption can be made
that in each phase one of the two MOSFETs is always conducting
the average inductor current. For VIN =12 V and V

= 1.475 V,

OUT

the duty ratio of the high-side MOSFET is:

V

D

OUT

== =

HSF

V

IN

1 475

12

V

V

12 3..%

(16)

The duty ratio of the low-side (synchronous rectifier) MOSFET is:

DD

LSF MAX HSF MAX() ()

.%=− =1877

(17)

The maximum rms current of the high-side MOSFET during
normal operation is:



I

I

HSF MAX

()

AA

80

××+

4

O

=× ×+

.

0 123 1

D






HSF

10 8

×

380

n

I

L

1






2



.



2

A



2

I

×

.

702




2



O



A

RIPPLE

()

3

=

=

(18)

The maximum rms current of the low-side MOSFET during
normal operation is:

D

II

LSF MAX HFS M AX

.

The R

DS(ON)

=×=

() ()

.

0 877

AA

×=702

.

0 123

for each MOSFET can be derived from the allowable

18 75

LSF

D

HSF

(19)

.

dissipation. If 10% of the maximum output power is allowed for
MOSFET dissipation, the total dissipation in the eight MOSFETs
of the 4-phase converter will be:

PVI

FET TOTAL MIN O

()

PVAW

FET TOTAL

()

.

=× ×

01

.. .

=× × =

0 1 1 3845 80 11 08

(20)

Allocating half of the total dissipation for the four high-side
MOSFETs and half for the four low-side MOSFETs, and
assuming that the resistive and switching losses of the high-side
MOSFETs are equal, the required maximum MOSFET resis­tances will be:

P

FET TOTAL

R

DS ON HSF

()

R

DS ON HSF

()

=

4

=

44702

××

nI

11 08

.

××

()

HSF MAX

W

.

2

()

=Ω

14

2

A

m

(21)

and:

P

FET TOTAL

R

DS ON LSF

()

R

DS ON LSF

()

=

2

=

241875

××

nI

11 08

.

××

()

LSF MAX

2

()

W

=Ω

2

A

.

394

.

m

(22)

Note that there is a trade-off between converter efficiency and
cost. Larger MOSFETs reduce the conduction losses and allow
higher efficiency, but increase the system cost. A Fairchild
FDB7030L (R
the high-side and a Fairchild FDB8030L (R

= 7 mΩ nominal, 10 mΩ worst-case) for

DS(ON)

DS(ON)

= 3.1 mΩ

nominal, 5.6 mΩ worst-case) for the low-side are good choices.
The high-side MOSFET dissipation is:

PR I

=×+

HSF DS ON HSF HSF MAX

VI Qf

IN L PK G SW

Pm A

HSF

12 150 200 1 95

+× × =

() ( )

×××

()

2

I

×

G

10 7 02

=Ω× +

.

VnCkHz W

2

VQ f

+× ×

IN RR SW

12 26 35 200

V A nC kHz

2

×× ×

21

A

×

.

(23)

Where the first term is the conduction loss of the MOSFET, the
second term represents the turn-off loss of the MOSFET and
the third term represents the turn-on loss due to the stored
charge in the body diode of the low-side MOSFET. In the sec­ond term, Q
turn-off and I
for the FDB7030L the value of Q

is the gate charge to be removed from the gate for

G

is the gate turn-off current. From the data sheet,

G

is about 35 nC and the peak

G

gate drive current provided by the ADP3414 is about 1 A. In
the third term, Q

, is the charge stored in the body diode of

RR

the low-side MOSFET at the valley of the inductor current. The
data sheet of the FDB8030L does not give that information, so
an estimated value of 150 nC is used. This estimate is based on
information found on data sheets of similar devices. In both
terms, f
or 200 kHz. I

is the actual switching frequency of the MOSFETs,

SW

is the peak current in the inductor, or 26 A.

L(PK)

The worst-case low-side MOSFET dissipation is:

PR I

=×

LSF DS ON LSF LSF MAX

Pm AW

LSF

() ( )

...

5 6 18 75 1 97

=Ω× =

2

2

(24)

Note that there are no switching losses in the low-side MOSFET.

CIN Selection and Input Current di/dt Reduction

In continuous inductor-current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty
ratio equal to V

OUT/VIN

and an amplitude of one-half of the
maximum output current. To prevent large voltage transients, a
low ESR input capacitor sized for the maximum rms current
must be used. The maximum rms capacitor current is given by:

I

C RMS

()

I

C RMS

()

O

=×× ×

=××−× =

nD nD

n

A

80

4

–( )

HSF HSF

.(.)

4 0 123 4 0 123 10

2

2

(25)

A

I

Note that the capacitor manufacturer’s ripple current ratings are
often based on only 2000 hours of life. This makes it advisable
to further derate the capacitor, or to choose a capacitor rated at
a higher temperature than required. Several capacitors may be
placed in parallel to meet size or height requirements in the
design. In this example, the input capacitor bank is formed by
three 270 µF, 16 V OS-CON capacitors with a ripple current
rating of 4.4 A each.

–12–

REV. 0

ADP3164

The ripple voltage across the three paralleled capacitors is:

V

C RIPPLE

()

80

4

=× +



18

Am

×






I

ESR

OCCHSF




n

Ω

+

3

3 270 200

××

n

××

nC f

CINSW

0 123

.

µ

F kHz

D



=






=

135

mV




(26)

Multilayer ceramic input capacitors are also required. These
capacitors should be placed between the Input side of the cur­rent sense resistor and the sources of the low-side synchronous
MOSFETs. These capacitors decouple the high-frequency lead­ing edge current spike that supplies the reverse recovery charge
of the low-side MOSFET’s body diode. The exact number
required is a function of the board layout. Typical designs will
use two 10 µF MLC capacitors. To reduce the input-current di/
dt to below the recommended maximum of 0.1 A/µs, an addi-
tional small inductor (L > 1 µH @ 15 A) should be inserted
between the converter and the supply bus. That inductor also
acts as a filter between the converter and the primary power source.

LAYOUT AND COMPONENT PLACEMENT GUIDELINES

The following guidelines are recommended for optimal perfor­mance of a switching regulator in a PC system.

General Recommendations

1. For good results, at least a four-layer PCB is recommended.
This should allow the needed versatility for control circuitry
interconnections with optimal placement, a signal ground
plane, power planes for both power ground and the input
power (e.g., 12 V), and wide interconnection traces in the
rest of the power delivery current paths. Keep in mind that
each square unit of 1 ounce copper trace has a resistance of
~0.53 mΩ at room temperature.

2. Whenever high currents must be routed between PCB
layers, vias should be used liberally to create several parallel
current paths so that the resistance and inductance intro­duced by these current paths is minimized and the via
current rating is not exceeded.

3. If critical signal lines (including the voltage and current
sense lines of the ADP3164) must cross through power
circuitry, it is best if a signal ground plane can be inter­posed between those signal lines and the traces of the
power circuitry. This serves as a shield to minimize noise
injection into the signals at the expense of making signal
ground a bit noisier.

4. The power ground plane should not extend under signal
components, including the ADP3164 itself. If necessary,
follow the preceding guideline to use the signal ground
plane as a shield between the power ground plane and the
signal circuitry.

5. The GND pin of the ADP3164 should be connected first to
the timing capacitor (on the CT pin), and then into the
signal ground plane. In cases where no signal ground plane
can be used, short interconnections to other signal ground
circuitry in the power converter should be used.

6. The output capacitors of the power converter should be
connected to the signal ground plane even though power

current flows in the ground of these capacitors. For this
reason, it is advised to avoid critical ground connections
(e.g., the signal circuitry of the power converter) in the signal
ground plane between the input and output capacitors. It is
also advised to keep the planar interconnection path short
(i.e., have input and output capacitors close together).

7. The output capacitors should also be connected as closely
as possible to the load (or connector) that receives the power
(e.g., a microprocessor core). If the load is distributed, the
capacitors should also be distributed, and generally in pro­portion to where the load tends to be more dynamic.

8. Absolutely avoid crossing any signal lines over the switching
power path loop, described below.

Power Circuitry

9. The switching power path should be routed on the PCB to
encompass the smallest possible area in order to minimize
radiated switching noise energy (i.e., EMI). Failure to take
proper precautions often results in EMI problems for the
entire PC system as well as noise-related operational problems
in the power converter control circuitry. The switching power
path is the loop formed by the current path through the
input capacitors, the power MOSFETs, and the power
Schottky diode, if used (see next), including all intercon­necting PCB traces and planes. The use of short and wide
interconnection traces is especially critical in this path for
two reasons: it minimizes the inductance in the switching
loop, which can cause high-energy ringing, and it accommo­dates the high current demand with minimal voltage loss.

10. MLC input capacitors should be placed between V

IN

and
Power Ground as close as possible to the sources of the
low-side MOSFETs.

11. To dampen ringing, an RC Snubber circuit should be placed
from the SW node of each phase to ground.

12. An optional power Schottky diode (3 A–5 A dc rating)
from each lower MOSFET’s source (anode) to drain (cath­ode) will help to minimize switching power dissipation in
the upper MOSFETs. In the absence of an effective Schot­tky diode, this dissipation occurs through the following
sequence of switching events. The lower MOSFET turns
off in advance of the upper MOSFET turning on (necessary
to prevent cross-conduction). The circulating current in
the power converter, no longer finding a path for current
through the channel of the lower MOSFET, draws cur­rent through the inherent body diode of the MOSFET. The
upper MOSFET turns on, and the reverse recovery charac­teristic of the lower MOSFET’s body diode prevents the
drain voltage from being pulled high quickly. The upper
MOSFET then conducts very large current while it momen­tarily has a high voltage forced across it, which translates
into added power dissipation in the upper MOSFET. The
Schottky diode minimizes this problem by carrying a major­ity of the circulating current when the lower MOSFET is
turned off, and by virtue of its essentially nonexistent
reverse recovery time. The Schottky diode has to be con­nected with very short copper traces to the MOSFET to
be effective.

REV. 0

–13–

ADP3164

13. Whenever a power dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias,
both directly on the mounting pad and immediately sur­rounding it, is recommended. Two important reasons for
this are: improved current rating through the vias, and
improved thermal performance from vias extended to the
opposite side of the PCB where a plane can more readily
transfer the heat to the air.

14. The output power path, though not as critical as the switch­ing power path, should also be routed to encompass a small
area. The output power path is formed by the current path
through the inductor, the current sensing resistor, the out­put capacitors, and back to the input capacitors.

15. For best EMI containment, the power ground plane should
extend fully under all the power components except the out­put capacitors. These components are: the input capacitors,
the power MOSFETs and Schottky diodes, the inductors,

the current sense resistor, and any snubbing element that
might be added to dampen ringing. Avoid extending the
power ground under any other circuitry or signal lines,
including the voltage and current sense lines.

Signal Circuitry

16. The output voltage is sensed and regulated between the FB
pin and the GND pin (which connects to the signal ground
plane). The output current is sensed (as a voltage) by the
CS+ and CS– pins. In order to avoid differential mode
noise pickup in the sensed signal, the loop area should be
small. Thus the FB trace should be routed atop the signal
ground plane, and the CS+ and CS– pins (the CS+ pin
should be over the signal ground plane as well).

17. The CS+ and CS– traces should be Kelvin-connected to
the current sense resistor, so that the additional voltage
drop due to current flow on the PCB at the current sense
resistor connections, does not affect the sensed voltage.

–14–

REV. 0

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

20-Lead TSSOP

(RU-20)

0.260 (6.60)

0.252 (6.40)

ADP3164

PIN 1

0.006 (0.15)

0.002 (0.05)

SEATING

PLANE

20

0.0256 (0.65)
BSC

11

0.177 (4.50)

0.169 (4.30)

101

0.0433 (1.10)
MAX

0.0118 (0.30)

0.0075 (0.19)

0.256 (6.50)

0.246 (6.25)

0.0079 (0.20)

0.0035 (0.090)

8ⴗ
0ⴗ

0.028 (0.70)

0.020 (0.50)

REV. 0

–15–

C02484–1–10/01(0)

–16–

PRINTED IN U.S.A.