Datasheet ADP3161 Datasheet (Analog Devices)

4-Bit Programmable 2-Phase

a

FEATURES
ADOPT™ Optimal Positioning Technology for Superior

Load Transient Response and Fewest Output

Capacitors
Active Current Balancing Between Both Output Phases
VRM 8.4-Compatible Digitally Programmable 1.3 V to

2.05 V Output

Dual Logic-Level PWM Outputs for Interface to External

High-Power Drivers
Total Output Accuracy ⴞ0.8% Over Temperature
Current-Mode Operation
Short Circuit Protection
Power-Good Output
Overvoltage Protection Crowbar Protects

Microprocessors with No Additional

External Components

APPLICATIONS
Desktop PC Power Supplies for:

Intel Pentium

VRM Modules

®

III Processors

Synchronous Buck Controller

ADP3161

FUNCTIONAL BLOCK DIAGRAM

VCC

CROWBAR

CMP

CMP1

SET

RESET

CMP3

CMP2

CMP

2-PHASE

DRIVER

LOGIC

DAC+24%

DAC–18%

g

m

REF

GND

COMP

CT

UVLO

& BIAS

3.0V

REFERENCE

OSCILLATOR

ADP3161

PWM1

PWM2

PWRGD

CS–

CS+

FB

GENERAL DESCRIPTION

The ADP3161 is a highly efficient dual output synchronous buck
switching regulator controller optimized for converting a 5 V or
12 V main supply into the core supply voltage required by high­performance processors such as Pentium

®

III. The ADP3161
uses an internal 4-bit DAC to read a voltage identification (VID)
code directly from the processor, which is used to set the output
voltage between 1.3 V and 2.05 V. The ADP3161 uses a current
mode PWM architecture to drive two logic-level outputs at a
programmable switching frequency that can be optimized for
VRM size and efficiency. The output signals are 180 degrees out of
phase, allowing for the construction of two complementary buck
switching stages. These two stages share the dc output current
to reduce overall output voltage ripple. An active current bal­ancing function ensures that both phases carry equal portions
of the total load current, even under large transient loads, to
minimize the size of the inductors.

ADOPT is a trademark of Analog Devices Inc.
Pentium is a registered trademark of Intel Corporation

VID

DAC

VID3 VID2 VID1 VID0

The ADP3161 also uses a unique supplemental regulation tech­nique called active voltage positioning to enhance load transient
performance. Active voltage positioning results in a dc/dc con­verter that meets the stringent output voltage specifications for
high-performance processors, with the minimum number of
output capacitors and smallest footprint. Unlike voltage-mode and
standard current-mode architectures, active voltage positioning
adjusts the output voltage as a function of the load current so
that it is always optimally positioned for a system transient. The
ADP3161 also provides accurate and reliable short circuit protec­tion and adjustable current limiting.

The ADP3161 is specified over the commercial temperature
range of 0°C to 70°C and is available in a 16-lead narrow body
SOIC package.

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2001

1

ADP3161–SPECIFICATIONS

(VCC = 12 V, I

Parameter Symbol Conditions Min Typ Max Unit

FEEDBACK INPUT

Accuracy V

FB

1.3 V Output TPC 1 1.290 1.3 1.310 V

1.6 V Output TPC 1 1.587 1.6 1.613 V

2.05 V Output TPC 1 2.034 2.05 2.066 V

Line Regulation ∆V
Input Bias Current I

FB

Crowbar Trip Point V

FB

CROWBAR

VCC = 10 V to 14 V 0.05 %

Percent of Nominal Output 114 124 134 %
Crowbar Reset Point Percent of Nominal Output 50 60 70 %
Crowbar Response Time t
FB Low Comparator Threshold V

CROWBAR

FB(LOW)

Overvoltage to PWM Going Low 300 ns

REFERENCE

Output Voltage V
Output Current I

REF

REF

VID INPUTS

Input Low Voltage V
Input High Voltage V
Input Current I
Pull-Up Resistance R

IL(VID)

IH(VID)

VID

VID

VID(X) = 0 V 180 250 µA

Internal Pull-Up Voltage 4.5 5.0 5.5 V

OSCILLATOR

Maximum Frequency
Frequency Variation ∆f
CT Charge Current I

2

f

CT(MAX)

CT

CT

TA = 25°C, CT = 91 pF 430 500 570 kHz

TA = 25°C, VFB in Regulation 130 150 170 µA

TA = 25°C, VFB = 0 V 263646 µA

ERROR AMPLIFIER

Output Resistance R
Transconductance g
Output Current I
Maximum Output Voltage V
Output Disable Threshold V
–3 dB Bandwidth BW

O(ERR)

m(ERR)

O(ERR)

COMP(MAX)

COMP(OFF)

ERR

VFB = 0 V 1 mA

FB Forced to V

COMP = Open 500 kHz

CURRENT SENSE

Threshold Voltage V

CS(TH)

CS+ = VCC, 69 79 89 mV

FB Forced to V

0.8 V ≤ COMP ≤ 1 V 0 15 mV

FB ≤ 375 mV 37 47 58 mV

1 V ≤ V

COMP

CS+ = CS– = VCC 0.5 5 µA

CS+ – (CS–) ≥ 89 mV 50 ns

∆V

COMP

/∆V

CS

Input Bias Current I
Response Time t

V

CS(FOLD)

n

i

, I

CS+

CS

CS–

to PWM Going Low

POWER GOOD COMPARATOR

Undervoltage Threshold V
Overvoltage Threshold V
Output Voltage Low V

PWRGD(UV)

PWRGD(OV)

OL(PWRGD)IPWRGD(SINK)

Percent of Nominal Output 76 82 88 %

Percent of Nominal Output 114 124 134 %

Response Time FB Going High 2 µs

FB Going Low 200 ns

PWM OUTPUTS

Output Voltage Low V
Output Voltage High V
Output Current I
Duty Cycle Limit

2

OL(PWM)

OH(PWM)

PWM

DC Per Phase, Relative to f

I

PWM(SINK)

I

PWM(SOURCE)

= 150 ␮A, TA = 0ⴗC to 70ⴗC, unless otherwise noted.)

REF

550 nA

375 425 500 mV

2.952 3.0 3.048 V
300 µA

0.6 V

2.2 V

20 28 kΩ

2000 kHz

200 kΩ

2.0 2.2 2.45 mmho

– 3% 3.0 V

OUT

560 720 800 mV

– 3%

OUT

≤ 3 V 25 V/V

= 100 µA 30 200 mV

= 400 µA 100 500 mV

= 400 µA 4.5 5.0 5.5 V

0.4 1 mA

CT

50 %

–2– REV. 0

ADP3161

WARNING!

ESD SENSITIVE DEVICE

Parameter Symbol Conditions Min Typ Max Unit

SUPPLY

DC Supply Current

Normal Mode I
UVLO Mode I

UVLO Threshold Voltage V

CC

CC(UVLO)

UVLO

VCC ≤ V

, VCC Rising 220 400 µA

UVLO

UVLO Hysteresis 0.1 0.4 0.6 V

NOTES

1

All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods.

2

Guaranteed by design, not tested in production.

Specifications subject to change without notice.

3.8 5.5 mA

5.9 6.4 6.9 V

ABSOLUTE MAXIMUM RATINGS*

VCC . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to +15 V

CS+, CS– . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VCC + 0.3 V

All Other Inputs and Outputs . . . . . . . . . . . . –0.3 V to +10 V

Operating Ambient Temperature Range . . . . . . . 0°C to 70°C

Operating Junction Temperature . . . . . . . . . . . . . . . . . . 125°C

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

θ

JA

Two-Layer Board . . . . . . . . . . . . . . . . . . . . . . . . . . 125°C/W

Four-Layer Board . . . . . . . . . . . . . . . . . . . . . . . . . . 81°C/W

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C

Vapor Phase (60 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . 215°C

Infrared (15 sec) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 220°C

*This is a stress rating only; operation beyond these limits can cause the device to

be permanently damaged. Unless otherwise specified, all voltages are referenced
to GND.

ORDERING GUIDE

Temperature Package Package

Model Range Description Option

ADP3161JR 0°C to 70°C Narrow Body SOIC R-16A (SO-16)

PIN CONFIGURATION

R-16A

1

NC

VID3

2

3

VID2

4

VID1

VID0

5

COMP

6

7

FB

8

CT

NC = NO CONNECT

ADP3161

TOP VIEW

(Not to Scale)

16

15

14

13

12

11

10

9

VCC

REF

CS–

PWM1

PWM2

CS+

PWRGD

GND

PIN FUNCTION DESCRIPTIONS

Pin
No. Name Function

1 NC No Connect.
2–5 VID3– Voltage Identification DAC Inputs.

VID0 These pins are pulled up to an internal refer-

ence, providing a Logic 1 if left open. The
DAC output programs the FB regulation
voltage from 1.3 V to 2.05 V.

6 COMP Error Amplifier Output and Compensation

Point. The voltage at this output programs
the output current control level between
CS+ and CS–.

7 FB Feedback Input. Error amplifier input for

remote sensing of the output voltage.

8 CT External capacitor CT connection to ground

sets the frequency of the device.

9 GND Ground. All internal signals of the ADP3161

are referenced to this ground.

10 PWRGD Open drain output that signals when the output

voltage is in the proper operating range.

11 CS+ Current Sense Positive Node. Positive input

for the current comparator. The output
current is sensed as a voltage at this pin with

respect to CS–.
12 PWM2 Logic-level output for the phase 2 driver.
13 PWM1 Logic-level output for the phase 1 driver.
14 CS– Current Sense Negative Node. Negative

input for the current comparator.
15 REF 3.0 V Reference Output.
16 VCC Supply Voltage for the ADP3161.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP3161 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

–3–REV. 0

ADP3161

4-BIT CODE

100

100nF

–Typical Performance Characteristics

ADP3161

16

1

2

3

4

5

6

7

8

NC

VID3

VID2

VID1

VID0

COMP

FB

CT

VCC

REF

CS–

PWM1

PWM2

CS+

PWRGD

GND

15

14

13

12

11

10

9

+

1 F

100nF

NC = NO CONNECT

12V

4.10

4.05

4.00

3.95

V

FB

AD820

1.2V

TPC 1. Closed-Loop Output Voltage Accuracy Test Circuit

10000

1000

OSCILLATOR FREQUENCY – kHz

100

0

100

200 300 400 500

CT CAPACITOR – pF

TPC 2. Oscillator Frequency vs. Timing Capacitor

SUPPLY CURRENT – mA

3.90

3.85
250 500 750 1250 1500 1750

0 2000

OSCILLATOR FREQUENCY – kHz

1000

TPC 3. Supply Current vs. Oscillator Frequency

30

25

20

15

10

NUMBER OF PARTS – %

5

0
–0.5 0.5

OUTPUT ACCURACY – % OF NOMINAL

0

TA = 25⬚C

V

OUT

= 1.6V

TPC 4. Output Accuracy Distribution

5V

OR

12V

VID INPUTS

12V

ADP3161

2-PHASE

SYNCHRONOUS

BUCK

CONTROLLER

5V

I

L1

PWM1

PWM2

ADP3412

SYNCHRONOUS

DRIVER

5V

ADP3412

SYNCHRONOUS

DRIVER

5V OR 12V

OUT

+

I

L2

TPC 5. 2-Phase CPU Supply System Level Block Diagram

–4– REV. 0

I

OUT

I

L2

PWM2

PWM1

I

L1

ADP3161

Table I. Output Voltage vs. VID Code

VID3 VID2 VID1 VID0 V

OUT(NOM)

1 1 1 1 1.30 V
1 1 1 0 1.35 V
1 1 0 1 1.40 V
1 1 0 0 1.45 V
1 0 1 1 1.50 V
1 0 1 0 1.55 V
1 0 0 1 1.60 V
1 0 0 0 1.65 V
0 1 1 1 1.70 V
0 1 1 0 1.75 V
0 1 0 1 1.80 V
0 1 0 0 1.85 V
0 0 1 1 1.90 V
0 0 1 0 1.95 V
0 0 0 1 2.00 V
0 0 0 0 2.05 V

THEORY OF OPERATION

The ADP3161 combines a current-mode, fixed frequency PWM
controller with antiphase logic outputs in a controller for a two­phase synchronous buck power converter. Two-phase operation
is important for switching the high currents required by high
performance microprocessors. Handling the high current in a
single-phase converter would place difficult requirements on the
power components such as inductor wire size, MOSFET ON­resistance, and thermal dissipation. The ADP3161’s high-side
current sensing topology ensures that the load currents are
balanced in each phase, such that neither phase has to carry
more than half of the power. An additional benefit of high­side current sensing over output current sensing is that the
average current through the sense resistor is reduced by the duty
cycle of the converter, allowing the use of a lower power, lower
cost resistor. The outputs of the ADP3161 are logic drivers only
and are not intended to directly drive external power MOS­FETs. Instead, the ADP3161 should be paired with drivers such
as the ADP3412, ADP3413, or ADP3414. A system level block
diagram of a 2-phase power supply for high current CPUs is
shown in TPC 5.

The frequency of the ADP3161 is set by an external capacitor
connected to the CT pin. Each output phase of the ADP3161
operates at half of the frequency set by the CT pin. The error
amplifier and current sense comparator control the duty cycle of
the PWM outputs to maintain regulation. The maximum duty
cycle per phase is inherently limited to 50% because the PWM
outputs toggle in two-phase operation. While one phase is on,
the other phase is off. In no case can both outputs be high at the
same time.

Output Voltage Sensing

The output voltage is sensed at the FB pin allowing for remote
sensing. To maintain the accuracy of the remote sensing, the
GND pin should also be connected close to the load. A voltage
error amplifier (g

) amplifies the difference between the output

m

voltage and a programmable reference voltage. The reference
voltage is programmed between 1.3 V and 2.05 V by an inter­nal 5-bit DAC, which reads the code at the voltage identification
(VID) pins. (Refer to Table I for the output voltage versus VID pin
code information.)

Active Voltage Positioning

The ADP3161 uses Analog Devices Optimal Positioning Technol­ogy (ADOPT), a unique supplemental regulation technique that
uses active voltage positioning and provides optimal compensa­tion for load transients. When implemented, ADOPT adjusts the
output voltage as a function of the load current, so that it is always
optimally positioned for a load transient. Standard (passive) volt­age positioning has poor dynamic performance, rendering it
ineffective under the stringent repetitive transient conditions
required by high performance processors. ADOPT, however,
provides optimal bandwidth for transient response that yields
optimal load transient response with the minimum number of
output capacitors.

Reference Output

A 3.0 V reference is available on the ADP3161. This reference
is normally used to set the voltage positioning accurately using a
resistor divider to the COMP pin. In addition, the reference can be
used for other functions such as generating a regulated voltage
with an external amplifier. The reference is bypassed with a 1 nF
capacitor to ground. It is not intended to supply current to large
capacitive loads, and it should not be used to provide more than
1 mA of output current.

Cycle-by-Cycle Operation

During normal operation (when the output voltage is regulated),
the voltage-error amplifier and the current comparator are the
main control elements. The voltage at the CT pin of the oscilla­tor ramps between 0 V and 3 V. When that voltage reaches 3 V,
the oscillator sets the driver logic, which sets PWM1 high. Dur­ing the ON time of Phase 1, the driver IC turns on the high-side
MOSFET. The CS+ and CS– pins monitor the current through
the sense resistor that feeds both high-side MOSFETs. When
the voltage between the two pins exceeds the threshold level
set by the voltage error amplifier (g

), the driver logic is reset

m

and the PWM output goes low. This signals the driver IC to turn
off the high-side MOSFET and turn on the low-side MOSFET.
On the next cycle of the oscillator, the driver logic toggles and sets
PWM2 high. On each following cycle of the oscillator, the outputs
toggle between PWM1 and PWM2. In each case, the current
comparator resets the PWM output low when the current compara­tor threshold is reached. As the load current increases, the output
voltage starts to decrease. This causes an increase in the output
of the g

amplifier, which in turn leads to an increase in the

m

current comparator threshold, thus programming more current to
be delivered to the output so that voltage regulation is maintained.

Active Current Sharing

The ADP3161 ensures current balance in the two phases by
actively sensing the current through a single sense resistor. During
one phase’s ON time, the current through the respective high-side
MOSFET and inductor is measured through the sense resistor
(R4 in TPC 6). When the comparator (CMP1 in the Functional
Block Diagram) threshold programmed by the g

amplifier is

m

reached, the high-side MOSFET turns off. In the next cycle the
ADP3161 switches to the second phase. The current is measured
with the same sense resistor and the same internal comparator,
ensuring accurate matching. This scheme is immune to imbalances
in the MOSFETs’ R

and inductors’ parasitic resistances.

DS(ON)

If for some reason one of the phases fails, the other phase will still
be limited to its maximum output current (one-half of the short
circuit current limit). If this is not sufficient to supply the load,

–5–REV. 0

ADP3161

the output voltage will droop and cause the PWRGD output to
signal that the output voltage has fallen out of its specified range.

Short Circuit Protection

The ADP3161 has multiple levels of short circuit protection to
ensure fail-safe operation. The sense resistor and the maximum
current sense threshold voltage given in the specifications set the
peak current limit.

When the load current exceeds the current limit, the excess
current discharges the output capacitor. When the output voltage is
below the foldback threshold V

, the maximum deliverable

FB(LOW)

output current is cut by reducing the current sense threshold
from the current limit threshold, V
threshold, V

CS(FOLD)

. Along with the resulting current foldback,

, to the foldback

CS(CL)

the oscillator frequency is reduced by a factor of five when
the output is 0 V. This further reduces the average current in
short circuit.

Power-Good Monitoring

The Power-Good comparator monitors the output voltage of the
supply via the FB pin. The PWRGD pin is an open drain output
whose high level (when connected to a pull-up resistor) indicates
that the output voltage is within the specified range of the nominal
output voltage requested by the VID DAC. PWRGD will go low if
the output is outside this range.

Output Crowbar

The ADP3161 includes a crowbar comparator that senses when
the output voltage rises higher than the specified trip thresh­old, V

CROWBAR

. This comparator overrides the control loop and
sets both PWM outputs low. The driver ICs turn off the high side
MOSFETs and turn on the low-side MOSFETs, thus pulling the
output down as the reversed current builds up in the inductors. If
the output overvoltage is due to a short of the high side MOSFET,

this action will current limit the input supply or blow its fuse,
protecting the microprocessor from destruction. The crowbar
comparator releases when the output drops below the specified
reset threshold, and the controller returns to normal operation if
the cause of the overvoltage failure does not persist.

Output Disable

The ADP3161 includes an output disable function that turns off
the control loop to bring the output voltage to 0 V. Because
an extra pin is not available, the disable feature is accomplished
by pulling the COMP pin to ground. When the COMP pin drops
below 0.56 V, the oscillator stops and both PWM signals are
driven low. This function does not place the part in a low quiescent
current shutdown state, and the reference voltage is still available.
The COMP pin should be pulled down with an open collector
or open drain type of output capable of sinking at least 2 mA.

APPLICATION INFORMATION
A VRM 8.4–Compliant Design Example

The design parameters for a typical high-performance Intel
Pentium III CPU application designed to meet Intel’s VRM 8.4
FMB specification (see Figure 1) are as follows:

Input Voltage (V
Nominal Output Voltage (V
Static Output Voltage Tolerance (V

(–80 mV) = 120 mV

Average Output Voltage (V

= 1.780 V

Maximum Output Current (I

) = 5 V

IN

OUT

) = V

AVG

) = 26 A

O

) = 1.8 V

) = (V+) – (V–) = 40 mV –

∆

()(–)VV++

+

OUT

2

Output Current di/dt < 20 A/µs.

17.8k

R7

Z1

U3

R5

2.4k

DRVH

SW

PGND

DRVL

20

C10
1

F

8

7

6

5

Q5
2N3904

C5

F

1

D1

MBR052LTI

C7

15pF

Q3
IRL3803

Q4
IRL3803

BST

1

2

IN

3

DLY

4

VCC

L2

1

R4

4m

C9

1

U2

ADP3412

DRVH

SW

PGND

DRVL

1000 F 9

RUBYCON ZA SERIES

24m ESR (EACH)

H

• • •

C14 TO C22

F

C20

Q1
IRL3803

Q2
IRL3803

V

1.3V – 2.05V
26A

V

1 H

CC(CORE)

CC(CORE)

L1

RTN

8

7

6

5

V

IN

5V

V

RTN

IN

12V V

CC

12V V

RTN

CC

R

A

15.1k

C

OC

2.7nF

R

R

B

740pF

Z

560

C2

+

C11
1000
16V

ADP3161

NC

1

2

VID3

3

VID2

4

VID1

VID0

5

6

COMP

7

FB

8

CT

C1
150pF

R1
1k

++

C12
1000 F

F

16V

U1

VCC

16

REF

15

14

CS–

13

PWM1

12

PWM2

CS+

11

10

PWRGD

9

GND

NC = NO CONNECT

C4

4.7

F

C25 1nF

10

C13
1000
16V

R6

C6

1 F

F

C23

15nF

R8
330

C24

330pF

D2

MBR052LTI

C8

15pF

C26

4.7

F

ZMM5236BCT

ADP3412

BST

1

2

IN

3

DLY

4

VCC

Figure 1. 26 A Pentium III CPU Supply Circuit

–6– REV. 0

ADP3161

CT Selection—Choosing the Clock Frequency

The ADP3161 uses a fixed-frequency control architecture. The
frequency is set by an external timing capacitor, C

for a given clock frequency can be selected using the graph in

C

T

. The value of

T

TPC 2.

The clock frequency determines the switching frequency, which
relates directly to switching losses and the sizes of the inductors
and input and output capacitors. A clock frequency of 400 kHz
sets the switching frequency of each phase, f

, to 200 kHz,

SW

which represents a practical trade-off between the switching
losses and the sizes of the output filter components. From
TPC 2, for 400 kHz the required timing capacitor value is 150 pF.
For good frequency stability and initial accuracy, it is recom­mended to use a capacitor with low temperature coefficient and
tight tolerance, e.g., an MLC capacitor with NPO dielectric and
with 5% or less tolerance.

Inductance Selection

The choice of inductance determines the ripple current in the
inductor. Less inductance leads to more ripple current, which
increases the output ripple voltage and the conduction losses in
the MOSFETs, but allows using smaller-size inductors and, for
a specified peak-to-peak transient deviation, output capacitors
with less total capacitance. Conversely, a higher inductance means
lower ripple current and reduced conduction losses, but requires
larger-size inductors and more output capacitance for the same
peak-to-peak transient deviation. In a two-phase converter a practi­cal value for the peak-to-peak inductor ripple current is under
50% of the dc current in the same inductor. A choice of 46%
for this particular design example yields a total peak-to-peak output
ripple current of 23% of the total dc output current. The follow­ing equation shows the relationship between the inductance,
oscillator frequency, peak-to-peak ripple current in an inductor
and input and output voltages.

VV V

(– )

IN AVG AVG

L

=

Vf I

××

IN SW L RIPPLE

×

()

(1)

For 6 A peak-to-peak ripple current, which corresponds to
just under 50% of the 13 A full-load dc current in an induc­tor, Equation 1 yields an inductance of

VV V

(–. ).

5 1 780 1 780

L

V kHz A

××

5 400 2 6

×

/

nH=

=

955

A 1 µH inductor can be used, which gives a calculated ripple
current of 5.7 A at no load. The inductor should not saturate at
the peak current of 18.7 A and should be able to handle the sum
of the power dissipation caused by the average current of 15 A
in the winding and the core loss.

The output ripple current is smaller than the inductor ripple
current due to the two phases partially canceling. This can be
calculated as follows:

VV V

××

22

I

=

O

∆

×××

2 1 780 5 2 1 780

Designing an Inductor

VV V

.(–.)

V H kHz

×µ ×

5 1 400

(– )

AVG IN AVG

VLf

××

IN OSC

=

=

26

.

A

(2)

Once the inductance is known, the next step is either to design an
inductor or find a standard inductor that comes as close as

possible to meeting the overall design goals. The first decision
in designing the inductor is to choose the core material. There
are several possibilities for providing low core loss at high frequen­cies. Two examples are the powder cores (e.g., Kool-Mµ

®

from
Magnetics) and the gapped soft ferrite cores (e.g., 3F3 or 3F4
from Philips). Low frequency powdered iron cores should be
avoided due to their high core loss, especially when the inductor
value is relatively low and the ripple current is high.

Two main core types can be used in this application. Open mag­netic loop types, such as beads, beads on leads, and rods and
slugs, provide lower cost but do not have a focused magnetic
field in the core. The radiated EMI from the distributed magnetic
field may create problems with noise interference in the circuitry
surrounding the inductor. Closed-loop types, such as pot cores,
PQ, U, and E cores, or toroids, cost more, but have much better
EMI/RFI performance. A good compromise between price and
performance are cores with a toroidal shape.

There are many useful references for quickly designing a power
inductor. Table II gives some examples.

Table II. Magnetics Design References

Magnetic Designer Software
Intusoft (http://www.intusoft.com)

Designing Magnetic Components for High-Frequency DC-DC

Converters

McLyman, Kg Magnetics
ISBN 1-883107-00-08

Selecting a Standard Inductor

The companies listed in Table III can provide design consul­tation and deliver power inductors optimized for high power
applications upon request.

Table III. Power Inductor Manufacturers

Coilcraft
[phone]
http://www.coilcraft.com

Coiltronics
[phone]
http://www.coiltronics.com

Sumida Electric Company
[phone]
http://www.sumida.com

C

Selection—Determining the ESR

OUT

The required equivalent series resistance (ESR) and capacitance
drive the selection of the type and quantity of the output capaci­tors. The ESR must be small enough to contain the voltage
deviation caused by a maximum allowable CPU transient cur­rent within the specified voltage limits, giving consideration also
to the output ripple and the regulation tolerance. The capaci­tance must be large enough that the voltage across the capacitor,
which is the sum of the resistive and capacitive voltage deviations,
does not deviate beyond the initial resistive deviation while the
inductor current ramps up or down to the value corresponding
to the new load current. The maximum allowed ESR also repre­sents the maximum allowed output resistance, R

OUT

.

–7–REV. 0

ADP3161

The cumulative errors in the output voltage regulations cut into
the available regulation window, V

. When considering dynamic

WIN

load regulation this relates directly to the ESR. When consider­ing dc load regulation, this relates directly to the programmed
output resistance of the power converter.

Some error sources, such as initial voltage accuracy and ripple
voltage, can be directly deducted from the available regulation
window, while other error sources scale proportionally to the
amount of voltage positioning used, which, for an optimal design,
should utilize the maximum that the regulation window will allow.
The error determination is a closed-loop calculation, but it can
be closely approximated. To maintain a conservative design while
avoiding an impractical design, various error sources should be
considered and summed statistically.

The output ripple voltage can be factored into the calculation by
summing the output ripple current with the maximum output
current to determine an effective maximum dynamic current

The critical capacitance for the four OS-CON capacitors with
an equivalent ESR of 2.75 mΩ is 2.6 mF, while the equivalent
capacitance of those capacitors is 4.8 mF. The critical capacitance
for the nine ZA series Rubycon capacitors is 2.8 mF while the
equivalent capacitance is 9 mF. With both choices, the capaci­tance is safely above the critical value.

R

SENSE

The value of R

is based on the maximum required output

SENSE

current. The current comparator of the ADP3161 has a mini­mum current limit threshold of 69 mV. Note that the 69 mV
value cannot be used for the maximum specified nominal cur­rent, as headroom is needed for ripple current and tolerances.

The current comparator threshold sets the peak of the inductor
current yielding a maximum output current, I
twice the peak inductor current value less half of the peak-to­peak inductor ripple current. From this the maximum value of

is calculated as:

R

SENSE

change. The remaining errors are summed separately according
to the formula:

VVV k

=××

(–( ))

WIN VID VID

∆

2

(3)



I






where k

O

1

–.

II

+

OO

∆

= 0.7% is the initial programmed voltage tolerance

VID

k

RCS

2

+

from the graph of TPC 4, k
current sense resistor, k

CSF

the current sense filter components, k
the two termination resistors added at the COMP pin, and k

2





k

CSF









2

= 2% is the tolerance of the

RCS

22

kk mV

++

RT EA




83 5

=




= 20% is the summed tolerance of

= 2% is the tolerance of

RT

EA

=

8% accounts for the IC current loop gain tolerance including

tolerance.

the g

m

The remaining window is then divided by the maximum output
current plus the ripple to determine the maximum allowed ESR

R

SENSE

In this case, 4 mΩ was chosen as the closest standard value.

Once R
where current limit is reached, I
the maximum current sense threshold of 89 mV:

I

OUT CL

At output voltages below 425 mV, the current sense threshold is
reduced to 58 mV, and the ripple current is negligible. There­fore, at dead short the output current is reduced to:

V

CS CL MIN

()( )

≤

SENSE

=×

2

I

I

L RIPPLE

()

O

+

22

has been chosen, the output current at the point

V

CS CL MAX

2

R

mV

–. .

m

Ω

()( )

SENSE

5 7 38 8

=×

89

4

and output resistance:

58

4

RR

E MAX OUT MAX

() ()

V

WIN

IImVAA

OO

26 2 6

∆

83 5

.

.

+

=

29

.==+=

m

Ω

The output filter capacitor bank must have an ESR of less than

2.9 mΩ. One can, for example, use four SP-Type OS-CON

I

(4)

OUT SC()

To safely carry the current under maximum load conditions, the
sense resistor must have a power rating of at least:

capacitors from Sanyo, with 1.2 mF capacitance, a 2.5 V voltage
rating, and 11 mΩ ESR. The four capacitors have a maximum
total ESR of 2.75 mΩ when connected in parallel. Another
possibility is the ZA series from Rubycon. The trade-off is
size versus cost. Nine 1 mF capacitors would give an ESR of

2.67 mΩ. These capacitors take up more space than four
OS-CON capacitors, but are significantly less expensive.

C

—Checking the Capacitance

OUT

As long as the capacitance of the output capacitor is above a
critical value and the regulating loop is compensated with ADOPT,
the actual value has no influence on the peak-to-peak deviation
of the output voltage to a full step change in the load current.
The critical capacitance can be calculated as follows:

C

OUT CRIT

=

..

275 1812

()

A

26

m

Ω×

=

I

O

RV

×

E OUT

H

µ

×

=

×

.

26

L

2

mF

(5)

PI R

=×

R SENSE RMS SENSE

SENSE

()

where:

I

2

I

SENSE RMS

()

O OUT

=×

n

In this formula, n is the number of phases, and η is the con­verter efficiency, in this case assumed to be 85%. Combining
Equations 9 and 10 yields:

P

R

26

=××=

SENSE

Power MOSFETs

In the standard two-phase application, two pairs of N-channel
power MOSFETs must be used with the ADP3161 and ADP3412,
one pair as the main (control) switches, and the other pair as
the synchronous rectifier switches. The main selection parameters

mV

69

=

AA

13 2 85

OUT(CL)

–

AA

=

mV

mV

2

AV

2

2

V

2

.=× =2

29 0

V

×η

IN

18

085 5

.

.

+

, can be calculated using

I

L RIPPLE()

()

A

.

V

, which equals

O

m

.

44

=Ω

mW

573

(6)

(7)

(8)

(9)

(10)

–8– REV. 0

ADP3161

for the power MOSFETs are V
mum gate drive voltage (the supply voltage to the ADP3412)
dictates whether standard threshold or logic-level threshold
MOSFETs must be used. Since V
MOSFETs (V

The maximum output current I

< 2.5 V) are strongly recommended.

GS(TH)

O

ment for the power MOSFETs. When the ADP3161 is operating
in continuous mode, the simplifying assumption can be made
that in each phase one of the two MOSFETs is always conduct­ing the average inductor current. For V

1.8 V, the duty ratio of the high-side MOSFET is:

V

D

HSF

OUT

==36%

V

IN

The duty ratio of the low-side (synchronous rectifier) MOSFET is:

DD

==164–%

LSF HSF

The maximum rms current of the high-side MOSFET during
normal operation is:

and R

GS(TH)

< 8 V, logic-level threshold

GATE

determines the R

DS(ON)

DS(ON)

= 5 V and V

IN

. The mini-

require-

=

OUT

(11)

(12)

the peak gate drive current provided by the ADP3412 is about
1 A. In the third term, Q
of the low-side MOSFET at the valley of the inductor current.
The data sheet of the IRL3803 gives 450 nC for the stored charge
at 71 A. That value corresponds to a stored charge of 80 nC
at the valley of the inductor current. In both terms f
actual switching frequency of the MOSFETs, or 200 kHz. I
is the peak current in the inductor, or 15.85 A.)

Substituting the above data in Equation 18, and using the worst­case value for the MOSFET resistance, yields a conduction loss
of 0.56 W, a turn-off loss of 1.1 W, and a turn-on loss of 0.08 W.
Thus the worst-case total loss in a high-side MOSFET is 1.74 W.

The worst-case low-side MOSFET dissipation is:

PR I m AW

LSF DS ON LS

(Note that there are no switching losses in the low-side MOSFET.)

CIN Selection and Input Current di/dt Reduction

In continuous inductor-current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty

I

HSF MAX

()



I

O

D

HSF

2

I

L RIPPLE

()

1




×

3




2



I

O

A

.=×+

=

79

(13)

2

The maximum rms current of the low-side MOSFET during

ratio equal to V
maximum output current. To prevent large voltage transients, a
low ESR input capacitor sized for the maximum rms current
must be used. The maximum rms capacitor current is given by:

normal operation is:

D

II

LSF MAX HSF MAX

() ()

The R

for each MOSFET can be derived from the allowable

DS(ON)

LSF

D

HSF

A

.==10 5

(14)

dissipation. If 10% of the maximum output power is allowed for
MOSFET dissipation, the total dissipation in the four MOSFETs
of the two-phase converter will be:

I

Note that the capacitor manufacturer’s ripple current ratings are
often based on only 2000 hours of life. This makes it advisable to
further derate the capacitor, or to choose a capacitor rated at a
higher temperature than required. Several capacitors may be

PVIVAW

MOSFET TOTAL OUT O()

....=× ×=× × =01 01 18 26 47

Allocating half of the total dissipation for the pair of high-side
MOSFETs and half for the pair of low-side MOSFETs, and
assuming that the resistive and switching losses of the high-side

(15)

placed in parallel to meet size or height requirements in the
design. In this example, the input capacitor bank is formed by
three 1000 µF, 16 V Rubycon capacitors.

The ripple voltage across the three paralleled capacitors is:

MOSFET are equal, the required maximum MOSFET resis­tances will be:

P

MOSFET TOTAL

R

DS ON HS MAX

()( )

R

DS ON LS MAX

()( )

P

4

()

22

I

×

8

HSF MAX

MOSFET TOTAL

()

22

I

×

LSF MAX

=

()

=

()

An IRL3803 MOSFET from International Rectifier (R
6 mΩ nominal, 9 mΩ worst case) is a good choice for both the
high-side and low-side. The high-side MOSFET dissipation is:

PR I

=×

()

HSF DS ON HS HFS MAX

() ( )

2

VQ f

+××

()

IN RR SW

+

where the second term represents the turn-off loss of the MOSFET
and the third term represents the turn-on loss due to the stored
charge in the body diode of the low-side MOSFET. (In the
second term, Q
for turn-off and I

is the gate charge to be removed from the gate

G

is the gate turn-off current. From the data

G

sheet, for the IRL3803 the value of Q

W

.

47

A

(. )

×

879

W

.

47

A

(. )

×

4105

VI Qf

×××

IN L PK G

()

×

2

is about 140 nC and

G

m

.=

=Ω

94

m

.=

=Ω

10 6

DS(ON)

SW

I

G

(16)

(17)

=

(18)

To reduce the input-current di/dt to below the recommended
maximum of 0.1 A/µs, an additional small inductor (L > 1 µH
@ 15 A) should be inserted between the converter and the sup­ply bus. That inductor also acts as a filter between the converter
and the primary power source.

Feedback Loop Compensation Design for ADOPT

Optimized compensation of the ADP3161 allows the best pos­sible containment of the peak-to-peak output voltage deviation.
Any practical switching power converter is inherently limited by
the inductor in its output current slew rate to a value much less
than the slew rate of the load. Therefore, any sudden change of
load current will initially flow through the output capacitors,
and assuming that the capacitance of the output capacitor is
larger than the critical value defined by Equation 5, this will
produce a peak output voltage deviation equal to the ESR of the
output capacitor times the load current change.

is the charge stored in the body diode

RR

is the

SW

L(PK)

=× =Ω×=

()

=×−× =

C RMS

A

26

2

V

C RIPPLE

()



26

24

Am

×



2



22

LSF MAX

()

OUT/VIN

I

××=

2036 2036 58

3

and an amplitude of one-half of the

O

2

DD

22

.–( .) .



I

OCCHSF

=× +




n

Ω

+

3 1000 200

××

91051

()

HSF HSF()

ESR

n

036

.

µ

FkHz

(. )

2

2

A

D

××

nC f

CINSW



=

112






=




mV

(19)

(20)

(21)

–9–REV. 0

ADP3161

The optimal implementation of voltage positioning, ADOPT,
will create an output impedance of the power converter that is
entirely resistive over the widest possible frequency range, includ­ing dc, and equal to the maximum acceptable ESR of the output
capacitor array. With the resistive output impedance, the output
voltage will droop in proportion with the load current at any load
current slew rate; this ensures the optimal positioning and allows
the minimization of the output capacitor.

With an ideal current-mode-controlled converter, where the
average inductor current would respond without delay to the
command signal, the resistive output impedance could be achieved
by having a single-pole roll-off of the voltage gain of the voltage­error amplifier. The pole frequency must coincide with the ESR
zero of the output capacitor. The ADP3161 uses constant frequency
current-mode control, which is known to have a nonideal, fre­quency dependent command signal to inductor current transfer
function. The frequency dependence manifests in the form of a
pair of complex conjugate poles at one-half of the switching fre­quency. A purely resistive output impedance could be achieved
by canceling the complex conjugate poles with zeros at the same
complex frequencies and adding a third pole equal to the ESR
zero of the output capacitor. Such a compensating network would
be quite complicated. Fortunately, in practice it is sufficient to
cancel the pair of complex conjugate poles with a single real zero
placed at one-half of the switching frequency. Although the end
result is not a perfectly resistive output impedance, the remaining
frequency dependence causes only a few percentage of deviation
from the ideal resistive response. The single-pole and single­zero compensation can be easily implemented by terminating

error amplifier with the parallel combination of a resistor

the g

m

and a series RC network.

The first step in the design of the feedback loop compensation is
to determine the targeted output resistance, R

E(MAX)

of the power
converter using Equation 4. The compensation can then be
tailored to create that output impedance for the power converter,
and the quantity of output capacitors can be chosen to create a
net ESR that is less than or equal to R

E(MAX)

.

The next step is to determine the total termination resistance of
the g

amplifier that will yield the correct output resistance:

m

nR

×

R

T

where n
g

m

ductance of the g

I SENSE

=

gR

××

m E MAX

()

is the division ratio from the output voltage signal of the

I

=

mmho m

..

2

22 29 2

amplifier to the PWM comparator (CMP1), gm is the transcon-

amplifier itself, and the factor of 2 is the result

m

m

×Ω

25 4

×Ω×

k

=Ω

.

784

(22)

of the two-phase configuration.

Once R
from the REF pin to output of the g

is known, the two resistors that make up the divider

T

amplifier (COMP pin)

m

must be calculated. The resistive divider introduces an offset to
the output of the g
the gain of the g

amplifier that, when reflected back through

m

stage, accurately positions the output voltage

m

near its allowed maximum at light load. Furthermore, the output
of the g

amplifier sets the current sense threshold voltage.

m

At no load, the current sense threshold is increased by the peak
of the ripple current in the inductor and reduced by the delay
between sensing when the current threshold has been reached
and when the high-side MOSFET actually turns off. These
two factors are combined with the inherent voltage at the output

amplifier that commands a current sense threshold of

of g

m

0 mV (V

VV

=+

GNL GNL

The output voltage at no load (V

):

GNL0

IRn

L RIPPLE CS I

0

VV

−

5178

−

1

××

()

2

VV

=+

1

GNL

.

×× × Ω× =

2 60 4 25 1 25

H

µ

VV

−

IN AVG

−

L

Am

×Ω×

.

57 4 25

2

ns m V

) can be calculated by start-

ONL

tR n

×× ×

2

()

DCSI

(23)

.

ing with the VID setting, adding in the positive offset (V+),
subtracting half the ripple voltage, and then subtracting the
dominant error terms:

2



V

WIN




V

VID

(24)

V

for

A

) is 200 kΩ:

(25)

mV

V

Ω×

2



2




2



=

1 824

.




RI

∆

EO

VVV

=+−×−× + ×

ONL VID

–. (. ) .

+

2

VVmV

18 40

.

=+ −

ONL



V

18 0007 002

×+×

2




2

Vkk

VID VID RT

mA

29 26

..

83 5

.

18

.

With these two terms calculated, the divider resistors (R
the upper, and R
that the internal resistance of the g

R

B

R

=

B

VV

−

3125

.

784

for the lower) can be calculated. Assuming

B

=

–

VV

REF GNL

R

T

.

−×

k

Ω

=Ω

V

−−

3

mmho V V

.(.–.)

22 1824 18

.

17 6

amplifier (R

m

REF

()

gV V

m ONL VID

OGM

V

k

Choosing the nearest 1% resistor value gives RB = 17.8 kΩ.
Finally, R

R

Again, choosing the nearest 1% resistor value gives R

is calculated:

A

=

A

1

11111

−−

RR R k k k

T OGM B

=

−

Ω

7841200

..

−

Ω

17 8

=Ω

15 1

1

Ω

= 15.0 kΩ.

A

k

.

(26)

The compensating capacitor can be calculated from the equation

CR

mF m

9267

C

=

OC

OC

×Ω

784

..

RfR

T OSC T

.

−

k kHz k

Ω

××Ω

400 7 84

π

×

OUT E

C

=

−
××

π

2

2

(27)

nF

=

286

.

Choosing the nearest standard value yields 2.7 nF.

The resistance of the zero-setting resistor in series with the
compensating capacitor is

R

=

Z

22

C f nF kHz

××

ππ.

OC OSC

=

××

2 7 400

k

=Ω

590

(28)

The nearest 2.7 standard 5% resistor value is 560 Ω.

–10– REV. 0

ADP3161

LAYOUT AND COMPONENT PLACEMENT GUIDELINES

The following guidelines are recommended for optimal perfor­mance of a switching regulator in a PC system.

General Recommendations

1. For good results, at least a four-layer PCB is recommended.
This should allow the needed versatility for control circuitry
interconnections with optimal placement, a signal ground
plane, power planes for both power ground and the input
power (e.g., 5 V), and wide interconnection traces in the
rest of the power delivery current paths. Keep in mind that
each square unit of 1 ounce copper trace has a resistance of
~ 0.53 mΩ at room temperature.

2. Whenever high currents must be routed between PCB layers,
vias should be used liberally to create several parallel current
paths so that the resistance and inductance introduced by
these current paths is minimized and the via current rating is
not exceeded.

3. If critical signal lines (including the voltage and current
sense lines of the ADP3161) must cross through power
circuitry, it is best if a signal ground plane can be inter­posed between those signal lines and the traces of the power
circuitry. This serves as a shield to minimize noise injection
into the signals at the expense of making signal ground a
bit noisier.

4. The power ground plane should not extend under signal com­ponents, including the ADP3161 itself. If necessary, follow
the preceding guideline to use the signal ground plane as a
shield between the power ground plane and the signal circuitry.

5. The GND pin of the ADP3161 should be connected first to
the timing capacitor (on the CT pin), and then into the
signal ground plane. In cases where no signal ground plane
can be used, short interconnections to other signal ground
circuitry in the power converter should be used.

6. The output capacitors of the power converter should be con­nected to the signal ground plane even though power current
flows in the ground of these capacitors. For this reason, it is
advised to avoid critical ground connections (e.g., the signal
circuitry of the power converter) in the signal ground plane
between the input and output capacitors. It is also advised to
keep the planar interconnection path short (i.e., have input
and output capacitors close together).

7. The output capacitors should also be connected as closely as
possible to the load (or connector) that receives the power
(e.g., a microprocessor core). If the load is distributed, the
capacitors should also be distributed, and generally in pro­portion to where the load tends to be more dynamic.

8. Absolutely avoid crossing any signal lines over the switching
power path loop, described below.

Power Circuitry

9. The switching power path should be routed on the PCB to
encompass the smallest possible area in order to minimize
radiated switching noise energy (i.e., EMI). Failure to take
proper precautions often results in EMI problems for the
entire PC system as well as noise-related operational problems

in the power converter control circuitry. The switching power
path is the loop formed by the current path through the input
capacitors, the power MOSFETs, and the power Schottky
diode, if used (see next), including all interconnecting PCB
traces and planes. The use of short and wide interconnection
traces is especially critical in this path for two reasons: it
minimizes the inductance in the switching loop, which can
cause high-energy ringing, and it accommodates the high
current demand with minimal voltage loss.

10. An optional power Schottky diode (3 A–5 A dc rating) from
each lower MOSFET’s source (anode) to drain (cathode) will
help to minimize switching power dissipation in the upper
MOSFETs. In the absence of an effective Schottky diode,
this dissipation occurs through the following sequence of
switching events. The lower MOSFET turns off in advance
of the upper MOSFET turning on (necessary to prevent
cross-conduction). The circulating current in the power
converter, no longer finding a path for current through the
channel of the lower MOSFET, draws current through the
inherent body diode of the MOSFET. The upper MOSFET
turns on, and the reverse recovery characteristic of the lower
MOSFET’s body diode prevents the drain voltage from being
pulled high quickly. The upper MOSFET then conducts
very large current while it momentarily has a high voltage
forced across it, which translates into added power dissipa­tion in the upper MOSFET. The Schottky diode minimizes
this problem by carrying a majority of the circulating current
when the lower MOSFET is turned off, and by virtue of its
essentially nonexistent reverse recovery time. The Schottky
diode has to be connected with very short copper traces to
the MOSFET to be effective.

11. A small ferrite bead inductor placed in series with the drain
of the lower MOSFET can also help to reduce this previously
described source of switching power loss.

12. Whenever a power dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias, both
directly on the mounting pad and immediately surrounding
it, is recommended. Two important reasons for this are:
improved current rating through the vias, and improved
thermal performance from vias extended to the opposite side
of the PCB where a plane can more readily transfer the heat
to the air.

13. The output power path, though not as critical as the switch­ing power path, should also be routed to encompass a small
area. The output power path is formed by the current path
through the inductor, the current sensing resistor, the out­put capacitors, and back to the input capacitors.

14. For best EMI containment, the power ground plane should
extend fully under all the power components except the output
capacitors. These components are: the input capacitors, the
power MOSFETs and Schottky diodes, the inductors, the
current sense resistor, and any snubbing element that might
be added to dampen ringing. Avoid extending the power
ground under any other circuitry or signal lines, including
the voltage and current sense lines.

–11–REV. 0

ADP3161

Signal Circuitry

15. The output voltage is sensed and regulated between the FB
pin and the GND pin (which connects to the signal ground
plane). The output current is sensed (as a voltage) by the
CS+ and CS– pins. In order to avoid differential mode noise
pickup in the sensed signal, the loop area should be small.
Thus the FB trace should be routed atop the signal ground
plane, and the CS+ and CS– pins should be routed as a

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

16-Lead SOIC

(R-16A/SO-16)

0.3937 (10.00)

0.3859 (9.80)

0.1574 (4.00)

0.1497 (3.80)

PIN 1

16

1

0.050 (1.27)
BSC

9

0.2440 (6.20)

0.2284 (5.80)

8

0.0688 (1.75)

0.0532 (1.35)

closely coupled pair (the CS+ pin should be over the signal
ground plane as well).

16. The CS+ and CS– traces should be Kelvin-connected to the
current sense resistor, so that the additional voltage drop due
to current flow on the PCB at the current sense resistor
connections, does not affect the sensed voltage.

0.0196 (0.50)

0.0099 (0.25)

ⴛ 45ⴗ

C01033–2.5–1/01 (rev. 0)

0.0098 (0.25)

0.0040 (0.10)

0.0192 (0.49)

0.0138 (0.35)

SEATING
PLANE

0.0099 (0.25)

0.0075 (0.19)

8ⴗ
0ⴗ

0.0500 (1.27)

0.0160 (0.41)

PRINTED IN U.S.A.

–12–

REV. 0