Datasheet ADP3156 Datasheet (Analog Devices)

Dual Power Supply Controller

V

CC

+12V

1mF

22mF

V

IN

+5V

+

C

IN

L

+

C

O

V

O

R

SENSE

1nF

C

COMP

35kV

20kV

V

O2

1000mF

R1

200pF

V

CC

SD

DRIVE1

SENSE+
SENSE–

DRIVE2

PGNDAGND

C

T

CMP

ADP3156

FB

V

LDO

R2

a

FEATURES
Active Voltage Positioning with Gain and Offset

Adjustment

Optimal Compensation for Superior Load Transient

Response
Fixed 1.5 V, 1.8 V and 2.5 V Output Versions
Dual N-Channel Synchronous Driver
On-Board Linear Regulator Controller
Total Output Accuracy ⴞ1% Over Temperature
High Efficiency, Current-Mode Operation
Short Circuit Protection
Overvoltage Protection Crowbar Protects Loads with

No Additional External Components
Power Good Output
SO-16 Package

APPLICATIONS
Desktop Computer Supplies
ACPI-Compliant Power Systems
General Purpose DC-DC Converters

SD

for Desktop Systems

ADP3156

FUNCTIONAL BLOCK DIAGRAM

V

REF

CMPI

V

IN

SENSE–

AGND

+15%

DELAY

V

REF

V

T1

PWRGD

+5% V

g

m

DRIVE1 DRIVE2 PGND

V

CC

NONOVERLAP

DRIVE

CROWBAR

IN

OFF

S

Q

R

V

T2

C

T

CMPT

OFF-TIME
CONTROL

SENSE+

–5%

REF

V

REF

REFERENCE

1.20V

ADP3156

SENSE–

2R

R

V

LDO

FB

GENERAL DESCRIPTION

The ADP3156 is a highly efficient synchronous buck switching
regulator controller optimized for converting the 3.3 V or 5 V
main supply into lower supply voltages required on the mother­boards of Pentium

®

III and other high performance processor
systems. The ADP3156 uses a current mode, constant off-time
architecture to drive two external N-channel MOSFETs at a
programmable switching frequency that can be optimized for
size and efficiency. It also uses a unique supplemental regulation
technique called active voltage positioning to enhance load
transient performance. Active voltage positioning results in a
DC/DC converter that provides the best possible transient re­sponse using the minimum number of output capacitors and
smallest footprint. Unlike voltage-mode and standard current­mode architectures, active voltage positioning adjusts the output
voltage as a function of the load current so that it is always
optimally positioned for a system transient.

The ADP3156 provides accurate and reliable short circuit
protection and adjustable current limiting. It also includes an
integrated overvoltage crowbar function to protect the micro­processor from destruction in case the core supply exceeds the
nominal programmed voltage by more than 15%.

Pentium is a registered trademark of Intel Corporation.
All other trademarks are the property of their respective holders.

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

CMP

The ADP3156 contains a linear regulator controller that is
designed to drive an external N-channel MOSFET. This linear
regulator is used to generate the auxiliary voltages (AGP, GTL,
etc.) required in most motherboard designs, and has been de­signed to provide a high bandwidth load-transient response. A
pair of external feedback resistors sets the linear regulator out­put voltage.

Figure 1. Typical Application

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 1999

ADP3156–SPECIFICATIONS

(0ⴗC ⱕ TA ⱕ +70ⴗC, VCC = 12 V, VIN = 5 V, unless otherwise noted)

1

Parameter Symbol Conditions Min Typ Max Units

OUTPUT ACCURACY

ADP3156-1.5 V V

O

1.480 1.500 1.520 V
ADP3156-1.8 V 1.777 1.800 1.823 V
ADP3156-2.5 V 2.475 2.500 2.525 V

I

OUTPUT VOLTAGE LINE ∆V

O

= 10 A (Figure 2)

LOAD

REGULATION VIN = 4.75 V to 5.25 V 0.05 %

INPUT DC SUPPLY CURRENT

Normal Mode I
Shutdown T

2

Q

VSD = 0.6 V 4.1 5.5 mA

= +25°C, VSD = 2.0 V 140 250 µA

A

CURRENT SENSE THRESHOLD

VOLTAGE V

C

PIN DISCHARGE CURRENT I

T

OFF-TIME t

DRIVER OUTPUT TRANSITION t

SENSE(TH)VSENSE–

T

OFF

, t

R

F

TIME T

POSITIVE POWER GOOD TRIP POINT3V

NEGATIVE POWER GOOD TRIP POINT3V

POWER GOOD RESPONSE TIME t

CROWBAR TRIP POINT V

PWRGD

PWRGD

PWRGD

CROWBAR

Forced to V

T

= +25°C

A

in Regulation 65 µA

V

OUT

V

= 0 V 2 10 µA

OUT

C

= 150 pF 1.8 2.45 3.2 µs

T

– 3% 125 145 165 mV

OUT

CL = 7000 pF (DRIVE1, 2)

= +25°C 120 200 ns

A

% Above Output Voltage 5 8 %

% Below Output Voltage –8 –5 %

500 µs

% Above Output Voltage 9 15 24 %

ERROR AMPLIFIER

OUTPUT IMPEDANCE RO

ERR

275 kΩ

ERROR AMPLIFIER

TRANSCONDUCTANCE g

m(ERR)

2.2 mmho

ERROR AMPLIFIER MINIMUM

OUTPUT VOLTAGE V

CMPMIN

V

SENSE–

Forced to V

+ 3% 0.8 V

OUT

ERROR AMPLIFIER MAXIMUM

OUTPUT VOLTAGE V

CMPMAX

ERROR AMPLIFIER BANDWIDTH –3 dB BW

ERR

V

SENSE–

Forced to V

– 3% 2.4 V

OUT

CMP = Open 500 kHz

LINEAR REGULATOR FEEDBACK

CURRENT I

FB

LINEAR REGULATOR Figure 2, V

OUTPUT VOLTAGE V

O2

R

= 5 kΩ, R2 = 20 kΩ, 1.47 1.5 1.53 V

PROG

I

= 1 A

O2

LDOIN

= 1.8 V

0.35 1 µA

SHUTDOWN (SD) PIN

Low Threshold SD
High Threshold SD
Input Current SD

NOTES

1

All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods. Specifications subject to change without
notice.

2

Dynamic supply current is higher due to the gate charge being delivered to the external MOSFETs.

3

The trip point is for the output voltage coming into regulation.

Specifications subject to change without notice.

L

H

IC

Part Active 0.6 V
Part in Shutdown 2.0 V

10 µA

–2–

REV. 0

ADP3156

TOP VIEW

(Not to Scale)

16

15

14

13

12

11

10

9

1

2

3

4

5

6

7

8

NC = NO CONNECT

NC
NC

AGND

SD
FB

V

LDO

SENSE–
SENSE+

PGND
NC
DRIVE1
DRIVE2
V

CC

PWRGD
CMP
C

T

ADP3156

WARNING!

ESD SENSITIVE DEVICE

PIN FUNCTION DESCRIPTIONS

Pin Mnemonic Function

1, 2, 15 NC No Connect.

3 AGND Analog Ground. All internal signals of the ADP3156 are referenced to this ground.

4 SD Shutdown. A logic high will place the ADP3156 in shutdown and disable both outputs. This pin

is internally pulled down.

5 FB Feedback connection for the linear controller. Connect this pin to the resistor divider network to

set the output voltage of the linear regulator.

6V

LDO

7 SENSE– Connects to the internal resistor divider that senses the output voltage. This pin is also the refer-

8 SENSE+ (+) input for the current comparator. The output current is sensed as a voltage at this pin with

9C

T

10 CMP Error Amplifier output and compensation point. The voltage at this output programs the

11 PWRGD Power Good. An open drain signal indicates the output voltage is within a ±5% regulation band.

12 V

CC

13 DRIVE2 Gate Drive for the (bottom) Synchronous Rectifier N-channel MOSFET. The voltage at DRIVE2

14 DRIVE1 Gate Drive for the buck switch N-channel MOSFET. The voltage at DRIVE1 swings from

16 PGND Power Ground. The drivers turn off the buck and synchronous MOSFETs by discharging their

Gate Drive for the linear regulator N-channel MOSFET.

ence input for the current comparator.

respect to SENSE–.

External Capacitor CT connection to ground sets the off-time of the device.

output current control level between the SENSE pins.

Supply Voltage to ADP3156.

swings from ground to V

ground to V

CC

.

CC

.

gate capacitances to this pin. PGND should have a low impedance path to the source of the syn­chronous MOSFET.

ABSOLUTE MAXIMUM RATINGS*

PIN CONFIGURATION

Input Supply Voltage (VCC) . . . . . . . . . . . . . . .–0.3 V to +16 V

Shutdown Input Voltage . . . . . . . . . . . . . . . . –0.3 V to +16 V

Operating Ambient Temperature Range . . . . . . 0°C to +70°C

Junction Temperature Range . . . . . . . . . . . . . . 0°C to +150°C

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 90°C/W

θ

JA

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . +300°C

*This is a stress rating only; operation beyond these limits can cause the device to

be permanently damaged.

ORDERING GUIDE

Buck Converter Package

Model Output Voltage Option

ADP3156JR-1.5 1.5 V R-16A/SO-16
ADP3156JR-1.8 1.8 V R-16A/SO-16
ADP3156JR-2.5 2.5 V R-16A/SO-16

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily

–3–

accumulate on the human body and test equipment and can discharge without detection.
Although the ADP3156 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

REV. 0

16-Lead SOIC

ADP3156

SYSTEM

V

O2

+1.5V

4A

1000mF

RTN

mP

100kV

IRL3103

22V

ESR = 34mV

2700mF 3 2

22mF

1mF

(10V)

ADP3156-1.8

PGND

NC
DRIVE1
DRIVE2

V

PWRGD

CMP

16

15

14

13

12

CC

11

10

9

C

T

C

T

200pF

R1
110kV

R2
16kV

C
2nF

IRL3103

IRL3103

COMP

L1

3mH

10BQ015

R

SENSE

12.9mV

220V

1nF

220V

R4
5kV

R3
20kV

2kV
47pF

1

2

3

4

5

6

7

8

NC
NC
AGND
SD
FB
V

LDO

SENSE–
SENSE+

Figure 2. ADP3156 Typical VRM8.4 AGP and GTL Chipset DC/DC Converter Circuit

1mF

L2

1.7mH

ESR = 60mV

470mF 3 4

12V

5V
5V RTN

12V RTN

V

O

1.8V
7A

RTN

SD

DRIVE1 DRIVE2 PGND

V

CC

AGND

PWRGD

SENSE+

SENSE–

DELAY

NONOVERLAP

DRIVE

CROWBAR

IN

OFF

S

Q

R

V

T2

C

T

CMPT

OFF-TIME
CONTROL

V

REF

CMPI

V

IN

SENSE–

+15%

2R

V

REF

V

+5% V

T1

g

m

–5%

REF

V

REF

REFERENCE

R

V

LDO

FB

ADP3156

CMP

Figure 3. Functional Block Diagram

–4–

REV. 0

Typical Performance Characteristics–ADP3156

100

95

90

85

EFFICIENCY – %

80

75

0.5 1.5

V

= 2.5V

OUT

V

OUT

V

= 1.5V

OUT

SEE FIGURE 2

2.5 3.5 4.5 6.55.5

OUTPUT CURRENT – Amps

= 1.8V

Figure 4. Efficiency vs. Output
Current

I

OUT

PRIMARY

N-DRIVE

1

2

DRIVER OUTPUT

SECONDARY

N-DRIVE

DRIVER OUTPUT

SEE FIGURE 2

= 10A

450
400
350

300
250
200
150

FREQUENCY – kHz

100

50

0

50 100 800

200 300 400 500 600 700
TIMING CAPACITOR – pF

Figure 5. Frequency vs. Timing
Capacitor

SEE FIGURE 2

VCC = +12V

= +5V

V

IN

I

= 10A

OUT

45

40
35
30
25
20
15

SUPPLY CURRENT – mA

10

5
0

45 397

Q

GATE(TOTAL)

58 83 134

OPERATING FREQUENCY – kHz

= 100nC

Figure 6. Supply Current vs.
Operating Frequency

OUTPUT VOLTAGE
50mV/DIV

OUTPUT CURRENT
7A TO 1A

DRIVE 1 AND 2 = 5V/DIV

500ns/DIV

Figure 7. Gate Switching Waveforms

OUTPUT VOLTAGE
50mV/DIV

OUTPUT CURRENT
1A TO 7A

10ms/DIV

Figure 10. Transient Response,
1 A–7 A of Figure 2 Circuit

100ns/DIV

Figure 8. Driver Transition
Waveforms

VCC VOLTAGE

5V/DIV

3

REGULATOR

OUTPUT VOLTAGE

1V/DIV

4

10ms/DIV

Figure 11. Power-On Start-Up
Waveforms

10ms/DIV

Figure 9. Transient Response,
7 A–1 A of Figure 2 Circuit

REV. 0

–5–

ADP3156

12V

SENSE+
SENSE–

PGNDAGND

OP27

0.1mF

V

DRIVE1
DRIVE2

CC

1mF

0.1mF

V

OUT

1kV

4700pF

1.2V

SD

CMP

C

T

100kV

ADP3156

Figure 12. Closed-Loop Test Circuit for Accuracy

THEORY OF OPERATION

The ADP3156 uses a current-mode, constant-off-time control
technique to switch a pair of external N-channel MOSFETs in a
synchronous buck topology. Constant off-time operation offers
several performance advantages, including that no slope com­pensation is required for stable operation. A unique feature of
the constant-off-time control technique is that since the off-time
is fixed, the converter’s switching frequency is a function of the
ratio of input voltage to output voltage. The fixed off-time is
programmed by the value of an external capacitor connected to

pin. The on-time varies in such a way that a regulated

the C

T

output voltage is maintained as described below in the cycle-by­cycle operation. Under fixed operating conditions the on-time
does not vary, and it varies only slightly as a function of load.
This means that switching frequency is fairly constant in stan­dard VRM applications. In order to maintain a ripple current in
the inductor, which is independent of the output voltage (which
also helps control losses and simplify the inductor design), the
off-time is made proportional to the value of the output voltage.
Normally, the output voltage is constant and therefore the off­time is constant as well.

Active Voltage Positioning

The output voltage is sensed at the SENSE– pin. SENSE– is
connected to an internal voltage divider. The output of the
divider is then compared to the internal reference. A unique
supplemental regulation technique called active voltage posi­tioning with optimal compensation adjusts the output voltage as
a function of the load current so that it is always optimally posi­tioned for a load transient. Standard (passive) voltage position­ing, sometimes recommended for use with other architectures,
has poor dynamic performance which renders it ineffective
under the stringent repetitive transient conditions specified in
Intel VRM documents. Consequently, such techniques do not
allow the minimum possible number of output capacitors to be
used. Optimally compensated active voltage positioning, as used
in the ADP3156, provides a bandwidth for transient response
that is limited only by parasitic output inductance. This yields
an optimal load transient response with the minimum number
of output capacitors.

Cycle-by-Cycle Operation

During normal operation (when the output voltage is regulated),
the voltage-error amplifier and the current comparator (CMPI)
are the main control elements. (See the block diagram of Figure

3). During the on-time of the high side MOSFET, CMPI

monitors the voltage between the SENSE+ and SENSE– pins.
When the voltage level between the two pins reaches the thresh­old level V

, the high side drive output is switched to ground,

T1

which turns off the high side MOSFET. The timing capacitor

is then discharged at a rate determined by the off-time con-

C

T

troller. While the timing capacitor is discharging, the low side
drive output goes high, turning on the low side MOSFET. When
the voltage level on the timing capacitor has discharged to the
threshold voltage level V

, comparator CMPT resets the SR

T2

flip-flop. The output of the flip-flop forces the low side drive
output to go low and the high side drive output to go high. As a
result, the low side switch is turned off and the high side switch
is turned on. The sequence is then repeated. As the load current
increases, the output voltage starts to decrease. This causes an
increase in the output of the voltage-error amplifier, which, in
turn, leads to an increase in the current comparator threshold

, thus tracking the load current. To prevent cross conduc-

V

T1

tion of the external MOSFETs, feedback is incorporated to
sense the state of the driver output pins. Before the low side
drive output can go high, the high side drive output must be
low. Likewise, the high side drive output is unable to go high
while the low side drive output is high.

Power Good

The ADP3156 has an internal monitor that senses the output
voltage and drives the PWRGD pin of the device. This pin is an
open drain output whose high level (when connected to a pull­up resistor) indicates that the output voltage has been within a

±5% regulation band of the targeted value for more than 500 µs.

The PWRGD pin will go low if the output is outside the regula-

tion band for more than 500 µs.

Output Crowbar

An added feature of using an N-channel MOSFET as the syn­chronous switch is the ability to crowbar the output with the
same MOSFET. If the output voltage is 15% greater than the
targeted value, the ADP3156 will turn on the lower MOSFET,
which will current-limit the source power supply or blow its
fuse, pull down the output voltage, and thus save the micropro­cessor from destruction. The crowbar function releases at ap­proximately 50% of the nominal output voltage. For example, if
the output is programmed to 2.0 V, but is pulled up to 2.3 V or
above, the crowbar will turn on the lower MOSFET. If in this
case the output is pulled down to less than 1.0 V, the crowbar
will release, allowing the output voltage to recover to 2.0 V if
the fault condition has been removed.

Shutdown

The ADP3156 has a shutdown (SD) pin that is pulled down by
an internal resistor. In this condition the device functions nor­mally. This pin should be pulled high to disable the output
drives.

APPLICATION INFORMATION

A number of power conversion requirements must be consid­ered when designing an ACPI compliant system. In normal
operating mode, 12 V, 5 V and 3.3 V are available from the
main supply. These voltages need to be converted into the
appropriate supply voltages for the Northbridge core, the
Southbridge core and RAMBUS memory, as well as supplies for
GTL and I/O drivers, CMOS memory and clock and graphics
(AGP) circuits.

–6–

REV. 0

ADP3156

During the standby operating state, the 12 V, 5 V and 3.3 V
power supply outputs are disabled, and only a low power 5 V
rail (5VSB) is available. The circuits that must remain active in
standby must be able to run from 5VSB. To accomplish this,
power routing is required to allow switching between normal
and standby supplies. Lack of a 12 V rail in standby makes control
of linear outputs difficult, and with up to 8 A demand from the

1.5 V and 1.8 V rails, an all-linear solution is inefficient.

Figure 13 shows a typical ACP-compliant Pentium III / chipset
power management system using the ADP3155 and ADP3156.
The ADP3155 provides VID switched output and two linear
regulators for standby operation. A charge-pump-doubled 5VSB is
ORed into the supply rail to supply the linear regulators during
standby operation. The VID output collapses when the main
5 V rail collapses, but the N-channel MOSFET linear regu­lators can continue to supply current from the ~9 V supply.
The ADP3156 provides 1.8 V via its main switching regulator,
and allows efficient linear regulation of 1.5 V rail by using the

1.8 V output as its source.

The design parameters for an ACPI-compliant Pentium III
peripheral system depend on what peripherals are used
(e.g., AGP) and what their specifications are. The following is
an example where the higher of two low system voltages (1.8 V
and 1.5 V) is created directly with the main buck converter, and
also used to supply power for the lower output voltage using the
ADP3156’s linear regulator controller.

Input voltage (power source): V

Auxiliary voltage: V

= 12 V

CC

Output voltages and tolerances: V

= 5 V

IN

= 1.8 V ± 5%, V2 = 1.5 V ±

1

5%

Maximum output currents: I

1MAX

= 3 A, I

2MAX

= 4 A

Slew rate of load current change: di

1

/dt = di

/dt >10 A/µs

2

The absence of an inductor on the 1.5 V linear regulated output
allows the output current to respond quickly and the linear
regulator MOSFET’s resistance to be modulated quickly. This,
and some small bypassing capacitors, essentially insulates the

1.5 V output from transient activity on the 1.8 V output. How­ever, this same fast response characteristic means that any 1.5 V
transient activity will be passed straight through the linear regu­lator to the 1.8 V output. This means that the 1.8 V output filter
capacitor selection must consider both 1.8 V and 1.5 V load
transients.

In this design example, worst case consideration requires that
the 1.8 V output be designed for transient current loading of
I

1MAX

+ I

= 7 A. Also, because a practical switching regula-

2MAX

tor design will have a current slew rate of <1 A/µs due to the

inductor, nearly the entire 7 A transient current must be ab­sorbed by the output capacitors.

CT Selection for Operating Frequency

The ADP3156 uses a constant-off-time architecture with t

OFF

determined by an external timing capacitor CT. Each time the
high side N-channel MOSFET switch turns on, the voltage
across C
C

T

is reset to approximately 3.3 V. During the off-time,

T

is discharged by a constant current of 65 µA. Once C

reaches

T

2.3 V, a new on-time cycle is initiated. The value of the off-time
is calculated using the continuous-mode operating frequency.
Assuming a nominal operating frequency of f

= 200 kHz at

NOM

an output voltage of 1.8 V, the corresponding off-time is:

t

OFF



=






V

Vf

1

O




IN NOM

=1

32–.µ

s

The timing capacitor can be calculated from the equation:

tA

×

65

OFF

C

=

T

µ

V

1

=

208

pF

ATX

(OR NLX)

POWER
SUPPLY

ATX_SHUTDOWN

REV. 0

12V

5V

3.3V
5V_ALWAYS
ATX_PGOOD

GND

1.5V OR 3.3V

VDDQ FOR AGP

TYPEDET# FOR

AGP SELECT

12V
5V

3.3V
5V_PM
ATX_POWERGOOD
ATX_SHUTDOWN

1.8V FOR

SB CORE,

MEM, ETC

1.5V VTT

FOR GTL

POWER MANAGEMENT

STATE COMMAND

ATX_POWER GOOD

DUAL

OUTPUT

SUPPLY

SWITCHER

OUT

LINEAR

OUT

VDDQ

POWER ROUTING

SELECT

CTRLS

CTRLS

3.3V_IN

1.5V_IN

5V_PM

IN

IN

12V

V

CC

MAIN_

CTRLS

PMSC

5V_PM

ATXPG

ADP3156

LIN_

CTRLS

5V

3.3V

POWER

MANAGEMENT

FUNCTIONS

ADP3155

LIN#2_
CTRLS

5V

5V_PM

5V_PM

LIN#1_
CTRLS

12V

V

CC

V

CC

VID_4:0

MAIN_

CTRLS

IN

CTRLS

IN

CTRLS

IN

CTRLS

Figure 13. ACPI-Compliant Pentium III System Block Diagram

–7–

TRIPLE

OUTPUT

VID

SUPPLY

SWITCHER

LINEAR#1

LINEAR#2

OUT

OUT

OUT

CPU

@ VID

V

CORE

3.3V_PM
FOR POWER
MANAGEMENT

2.5V_PM
FOR CMOS,
CLOCK, MEMORY

ADP3156

The converter operates at the nominal operating frequency only
at the V

specified above, and at light load. At higher load

OUT

conditions, the operating frequency decreases due to the para­sitic voltage drops across the power devices. The actual mini­mum frequency at V

= 1.8 V is calculated from Equation 1,

OUT

and is a function of the finite resistances of various components
in the power converter.

C

Selection—Determining the ESR

OUT

The required ESR and capacitance drive the selection of the
type and quantity of the output capacitors. The ESR must be
small enough that both the resistive voltage deviation due to a
step change in the load current and the output ripple voltage
stay below the values defined in the specification of the supplied
circuitry. The capacitance must be large enough that the output
is held up while the inductor current ramps up or down to the
value corresponding to the new load current.

Taking into account the ±1% setpoint accuracy of the ADP3156,

and assuming a 1% (or 15␣ mV) peak-to-peak ripple, the allowed
static voltage deviation of the output voltage when the load
changes between the minimum and maximum values is:

(2 × 5% × 1.5 V) – (2 × 1% × 1.5 V) – (1% × 1.5 V) = 105 mV
This sets the maximum ESR at 105 mV/7 A = 15 mΩ. Four

parallel capacitors of 470 µF with a maximum ESR of 60 mΩ
will achieve the 15 mΩ maximum net ESR. Whether or not the

capacitance is sufficient must be determined after the inductor
is selected.

Inductor Selection

The minimum inductor value can be calculated from ESR, off­time, dc output voltage and allowed peak-to-peak ripple voltage
using the following equation:

Vt R

L

MIN

O OFF E MAX

1

V

()

RIPPLE p p

15 32 7

−

××

..

sm

µΩ

mV

15

H

224==

=µ

.

The minimum inductance gives a peak-to-peak ripple current of

2.14 A, or 35% of the maximum dc output current I

OMAX

. The

inductor peak current in normal operation is:

I

LPEAK

= I

OMAX

+ I

/2 = 8.07 A

RPP

The inductor valley current is:

I

LVALLEY

= I

LPEAK

– I

RPP

/2 = 5.93 A

The inductor for this application should have an inductance of

not less than 2.24 µH at full load current and should not satu-

rate at the worst-case overload or short circuit current at the
maximum specified ambient temperature. For this example, it is
assumed the inductance might drop as much as 33% due to

load current, so its initial value might be as high as 3.36 µH.

Tips for Selecting the Inductor Core

Ferrite designs have very low core loss, so the design should
focus on copper loss and on preventing saturation. Molypermalloy,
or MPP, is a low loss core material for toroids, and it yields the
smallest size inductor, but MPP cores are more expensive than

ferrite cores or the Kool Mµ® cores from Magnetics, Inc. The

lowest cost core is made of powdered iron, for example the #52
material from Micrometals, Inc., but yields a larger size inductor.

C

Selection—Determining the Capacitance

OUT

The minimum capacitance of the output capacitor is determined
from the requirement that the output be held up while the in­ductor current ramps up (or down) to the new value. The mini­mum capacitance should produce an initial dv/dt which is equal
(but opposite in sign) to the dv/dt obtained by multiplying the
di/dt in the inductor and the ESR of the capacitor.

II

–

C

MIN

C

MIN

OMAX OMIN

=

Rdidt

(/)

E

AA

−

=

70

mV H

15 18 336

./.

Ωµ

()

F

871

µ

=

In the above equation the value of di/dt is calculated as the
smaller voltage across the inductor (i.e., the smaller of

– V

OUT

and V

V

IN

) divided by the maximum inductance

OUT

(3.36 µH) of the inductor. The four parallel-connected
470 µF capacitors have a total capacitance of 1880 µF, so the

minimum capacitance requirement is met with ample margin.

R

SENSE

The value of R

is based on the required output current.

SENSE

The current comparator of the ADP3156 has a threshold range
that extends from 0 mV to 125 mV (minimum). Note that the
full 125 mV range cannot be used for the maximum specified
nominal current, as headroom is needed for current ripple and
transients.

The current comparator threshold sets the peak of the inductor
current yielding a maximum output current I

, which equals

OMAX

the peak value less half of the peak-to-peak ripple current. Solv­ing for R

and allowing a 20% margin for overhead, and

SENSE

using the minimum current sense threshold of 125 mV yields:

R

= (125 mV)/[1.2(I

SENSE

Once R
I

SC(PK)

I

SC(PK)

has been chosen, the peak short-circuit current

SENSE

can be predicted from the following equation:

= (145 mV)/R

= (145 mV)/(12.9 mΩ)= 11.2 A

SENSE

OMAX

+ I

RPP

/2)] = 12.9 m

Ω

The actual short-circuit current is less than the above-calculated

I

value because the off-time rapidly increases when the

SC(PK)

output voltage drops below 1 V. The relationship between the
off-time and the output voltage is:

CV

×

1

≈

360

T

V

O

k

Ω

A

+

2

µ

t

OFF

With a short circuit across the output, the off-time will be about

104 µs. During that time the inductor current gradually decays.

The amount of decay depends on the L/R time constant in the

output circuit. With an inductance of 2.24 µH and total resis-
tance of 40 mΩ (the inductor’s series resistance plus the sense
resistor), the time constant will be 56 µs. This yields a valley

f

=×

MIN

VIRI R R RV

1

t

VIRI R R RR

OFF

––( )–

IN IN IN OMAX DS ON HSF SENSE L O

––( – )

IN IN IN OMAX DS ON HSF SENSE L DS ON LSF

()

() ()

++

++

–8–

(1)

REV. 0

ADP3156

current of 1.7 A and an average short-circuit current of about

6.5 A—meaning that there is actually a small degree of short­circuit current foldback. To safely carry the maximum current,
the sense resistor must have a power rating of at least (11.2 A –

1.07 A)

2

× 12.9 mW = 1.3 W.

Current Transformer Option

An alternative to using a low value and high power current sense
resistor is to reduce the sensed current by using a low cost cur­rent transformer and a diode. The current can then be sensed
with a small size, low cost SMT resistor. Using a transformer
with one primary and 50 secondary turns reduces the worst-case
resistor dissipation to a few mW. Another advantage of using
this option is the separation of the current and voltage sensing,
which makes the voltage sensing more accurate.

Power MOSFETs

Two external N-channel power MOSFETs must be selected for
use with the ADP3156, one for the main switch, and an identi­cal one for the synchronous switch. The main selection param­eters for the power MOSFETs are the threshold voltage V
and the on resistance R

. The minimum input voltage

DS(ON)

GS(TH)

dictates whether standard threshold or logic-level threshold
MOSFETs must be used. For V
MOSFETs (V

< 4 V) may be used. If VIN is expected to

GS(TH)

drop below 8 V, logic-level threshold MOSFETs (V

> 8 V, standard threshold

IN

GS(TH)

<

2.5 V) are strongly recommended. Only logic-level MOSFETs
with V

ratings higher than the absolute maximum of V

GS

CC

should be used.

The maximum output current I

determines the R

OMAX

DS(ON)

requirement for the two power MOSFETs. When the ADP3156
is operating in continuous mode, the simplifying assumption can
be made that one of the two MOSFETs is always conducting
the average load current. For V

IN

= 5 V and V

= 1.8 V, the

OUT

maximum duty ratio of the high side FET is:

D

MAXHF

= (1 – f

MIN

× t

) = (1 – 150 kHz × 3.2

OFF

µ

s) = 52%

The duty ratio of the low side (synchronous rectifier) FET un­der the maximum load condition is:

D

MAXLF

= 1 – D

MAXHF

= 48%

The maximum rms current of the high side FET is:

I

RMSHS

= [D

MAXHF

(I

LVALLEY

2 + I

LPEAK

2 + I

LVALLEYILPEAK

)/3]

0.5

= 7.32 A rms

The maximum rms current of the low side FET is:

I

RMSLS

= [D

MAXLF

(I

LVALLEY

2 + I

LPEAK

2 + I

LVALLEYILPEAK

)/3]

0.5

= 7.03 A rms

The R

for each FET can be derived from the allowable

DS(ON)

dissipation. Allowing 8% of the maximum output power for
FET dissipation, the total dissipation will be:

P

FETALL

= 0.08 VOI

OMAX

= 1.0 W

Allocating half of the total dissipation for the high side FET and
half for the low side FET, the required minimum FET resis­tances will be:

R

DS(ON)HSF(MIN)

R

DS(ON)LSF(MIN)

= 1 W × 52%/(7.32A)2 = 9.7 mΩ

= 1 W × 48%/(7.03A)2 = 9.7 mΩ

Note that there is a trade-off between converter efficiency and
cost. Larger FETs reduce the conduction losses and allow
higher efficiency, but increase the system cost. If efficiency is
not a major concern, the International Rectifier IRL3103 is an

economical choice for both the high side and low side positions.
Those devices have an R

of 14 mΩ at V

DS(ON)

= 10 V and at

GS

+25°C. The low side FET is turned on with at least 10 V. The

high side FET, however, is turned on with only 12 V – 5 V =
7 V. Checking the typical output characteristics of the device in
the data sheet, shows that for an output current of 10 A, and at

of 7 V, the VDS is 0.15 V. This gives an R

a V

GS

slightly above the one specified at a V

GS

of 10 V, so the resis-

DS(ON)

only

tance increase due to the reduced gate drive can be neglected.
The specified R

temperature of +140°C must be modified by an R

at the expected highest FET junction

DS(ON)

DS(ON)

multi-

plier, using the graph in the data sheet. In this case:

R

DS(ON)MULT

Using this multiplier, the expected R

= 1.7

DS(ON)

at +140°C is

1.7 × 14 = 24 mΩ.

The high side FET dissipation is:

P

DFETHS

= I

RMSHS

2

R

DS(ON)

+ 0.5 VINI

LPEAKQGfMIN/IG

~ 2.54 W

where the second term represents the turn-off loss of the FET.
(In the second term, Q
the gate for turn-off and I
sheet, Q

is about 50 nC–70 nC and the gate drive current

G

is the gate charge to be removed from

G

is the gate current. From the data

G

provided by the ADP3156 is about 1 A.)

The low side FET dissipation is:

P

DFETLS

= I

rmsls

2

R

DS(ON)

= 0.49 W

(Note that there are no switching losses in the low side FET.)
To maintain an acceptable MOSFET junction temperature,
proper heat sinking should be used. The heat sink and airflow
are chosen based on how low the impedance must be reduced in
order to keep the MOSFET’s junction temperature at an ac­ceptably low level, according to the formula:

θ

= [(T

HA

where θ

is the thermal resistance from the heat sink to ambi-

HA

ent air (and depends on airflow), T

JMAXOP

– TA)¸P

] – θJC – θ

DFET

is the user-deter-

JMAXOP

CH

mined maximum acceptable operating temperature of the
MOSFET, and the last two factors are the thermal resistance
from junction-to-case of the device, and case-to-heat sink. Typi-

cally, the junction-to-case thermal resistance is 2°C/W, and the
case-to-heat sink resistance is 0.5°C/W.

CIN Selection and Input Current di/dt Reduction

In continuous-inductor-current mode, the source current of the
high side MOSFET is a square wave with a duty ratio of V

O/VlN

.
To keep the input ripple voltage at a low value, one or more
capacitors with low equivalent series resistance (ESR) and ad­equate ripple-current rating must be connected across the input
terminals. The maximum rms current of the input bypass ca­pacitors is:

I

CINRMS

= I

OMAX

[D

MAX

× (1–D

MAX

0.5

)]

= 3.5 A

For an FA-type capacitor with 2700 mF capacitance and 10 V

voltage rating, the ESR is 34 mΩ and the allowed ripple current
at 100 kHz (and similar frequencies) is 1.94 A. At +105°C, two

such capacitors may be connected in parallel to handle the cal­culated ripple current.

To further reduce the effect of the ripple voltage on the system
supply voltage bus and to reduce the input-current di/dt to

below the recommended maximum of 0.1 A/µs, an additional

small inductor should be inserted between the converter and the
supply bus (see Figure 2).

REV. 0

–9–

ADP3156

Feedback Loop Compensation Design for Active Voltage
Positioning

Optimized compensation of the ADP3156 allows the best pos­sible containment of the peak-to-peak output voltage deviation.
Any practical switching power converter is inherently limited by
the inductor in its output current slew rate to a value much less
than the slew rate of the load. Therefore, any sudden change of
load current will initially flow through the output capacitors,
and this will produce an output voltage deviation equal to the
ESR of the output capacitor array times the load current change.
To correctly implement active voltage positioning, the low fre­quency output impedance (i.e., the output resistance) of the
converter should be made equal to the maximum ESR of the
output capacitor array. This can be achieved by having a single
pole roll-off of the voltage gain of the g

error amplifier, where

m

the pole frequency coincides with the ESR zero of the output
capacitor. A gain with single pole roll-off requires that the g

m

amplifier output pin be terminated by the parallel combination
of a resistor and capacitor. The required resistor value can be
calculated from the equation:

kRt

×275

Ω

R

=

C

kRt

275

Ω –

TOTAL

TOTAL

where:

kR I

××16 4.–Ω

Rt

TOTAL

=

CS OMAX

VV

HI LO

where the quantities 16.4 kΩ and 275 kΩ are characteristics of

the ADP3156, the value of the current sense resistor, R
already been determined as above, and where V

and VLO are

HI

CS

, has

the respective upper and lower limits allowed for regulation.
Although a single termination resistor equal to R

would yield

C

the proper voltage positioning gain, the dc biasing of that resis­tor would determine how the regulation band is centered (i.e.,
note that sometimes the specified regulation band is asymmetri­cal with respect to the nominal VID voltage.) With the ADP3156,
the offset is already considered as part of the design procedure—
no special provision is required. To accomplish the dc biasing, it
is simplest to use two resistors to terminate the g

output, with

m

the lower resistor tied to ground and the upper resistor to the
12 V supply of the IC. The values of these resistors can be cal­culated using:

V

RR

=×

UPPER C

DIV

V

OS

and:

V

RR

=×

LOWER C

where V
ommended 12 V), and V

is the resistor divider supply voltage (e.g., the rec-

DIV

is the offset voltage required on the

OS

amplifier to produce the desired offset at the output. V

OS

VV

–

DIV OS

OS

is

calculated using Equation 2, where V

is the offset from

OUT(OS)

the nominal VID-programmed value to the center of the speci­fied regulation window for the output voltage. (Note this may be
either positive or negative.) For clarification, that offset is given
by:

VVVVID

OUT OS HI LO()

1

()–=+

2

Finally, the compensating capacitance is determined from the
equality of the pole frequency of the error amplifier gain and the
zero frequency of the impedance of the output capacitor:

C ESR

×

C

COMP

O

=

Rt

TOTAL

Trade-Offs Between DC Load Regulation and AC Load
Regulation

Casual observation of the circuit operation—e.g., with a volt­meter—would make it appear that the dc load regulation ap­pears to be rather poor compared to a conventional regulator.
This would be especially noticeable under very light or very
heavy loads where the voltage is “positioned” near one of the
extremes of the regulation window rather than near the nominal
center value. It must be noted and understood that this low gain
characteristic (i.e., loose dc load regulation) is inherently re­quired to allow improved transient containment (i.e., to achieve
tighter ac load regulation). That is, the dc load regulation is
intentionally sacrificed (but kept within specification) in order
to minimize the number of capacitors required to contain the
load transients produced by the CPU.

Linear Regulator

The ADP3156 linear regulator provides a low cost, convenient
and versatile solution for generating a lower supply rail in addi­tion to the main output. The maximum output load current is
determined by the size and thermal impedance of the external
N-channel power MOSFET that is controlled by the ADP3156.
The output voltage, V

in Figure 14, is sensed at the FB pin of

O2

the ADP3156 and compared to an internal 1.2 V reference in a
negative feedback loop which keeps the output voltage in regula­tion. If the load is being reduced or increased, the FET drive
will also be reduced or increased by the ADP3156 to provide a

well regulated ±1% accurate output voltage. The output voltage

is programmed by adjusting the value of the external resistor

, shown in Figure 14.

R

PROG

VIN = +1.8V

ADP3156

V

R

PROG

5kV

LDO

FB

20kV

VO2 = +1.5V

I

= 4.0A

O2

IRL3103

1000mF/10V

Figure 14. Linear Regulator Configuration

V

R

=×+

OS

Rt

TOTAL



C

08

VV






()

OUT OS




136





Rt

TOTAL TOTAL

–.



.

k

ΩΩ



17



V




Rt

275



+



k



–10–

6.

RI

CS OMAX







(2)

REV. 0

ADP3156

Efficiency of the Linear Regulator

The efficiency and corresponding power dissipation of the linear
regulator are not determined by the ADP3156. Rather, these are
a function of input and output voltage and load current. Effi­ciency is approximated by the formula:

η

= 100% × (V

OUT

⫼ VIN)

The corresponding power dissipation in the MOSFET, together
with any resistance added in series from input to output is given
by:

P

LDO

= (V

IN(LDO)

– V

OUT(LDO)

) × I

OUT(LDO)

Minimum power dissipation and maximum efficiency are ac­complished by choosing the lowest available input voltage that
exceeds the desired output voltage. However, if the chosen
input source is itself generated by a linear regulator, its power
dissipation will be increased in proportion to the additional
current it must now provide. For most PC systems, the lowest
available input source for the linear regulators, which is not
itself generated by a linear regulator, is 3.3 V from the main
power supply. However, in this case, the main output of the
ADP3156 creates a lower voltage that may be used as the source
supply for the linear regulator. Assuming that a 1.8 V main
output is used to provide power for a 1.5 V linear regulator

output, the efficiency will nominally be 1.5 V ÷ 1.8 V = 83%.

If the 1.5 V output must supply a 4 A maximum load (a total of
6 W), the steady state dissipation in the MOSFET may be as
high as:

P

LDO(MAX)

= (VIN – V

OUT

) × I

OUT(MAX)

= (1.8 V – 1.5 V) × 4 A

= 1.2 W

The minimum acceptable on resistance of the MOSFET that
would deliver the 4 A load with only a 0.3 V difference between
input and output is:

R

DS(ON, MAX)

= (V

OUT

– V

IN

) ÷ I

OUT(MAX)

= (1.8 V – 1.5 V) ÷ 4 A

= 75 mΩ

There are many MOSFETs to choose from that can support the
maximum power dissipation without need for a heat sink and
without exceeding the calculated maximum on-resistance. For
simplicity it may be desirable to use the same MOSFET as is
used for the main power converter.

The output voltage may be programmed by the R

PROG

resistor

as follows:

R

PROG



V

=



12

.



O

2

V



×Ω= −

120

–






kkk






15

.

12

.

×=Ω

120 5




Ω

The output filter capacitor maximum allowed ESR is:

where V

ESR~V

TR2/IOMAX

is the maximum allowed transient deviation on the

TR2

= 0.036/0.5 = 0.072

Ω

output. This requirement is met using a 1000 µF/10 V LXV

series capacitor from United Chemicon. For applications requir­ing higher output current, a heat sink and/or a larger MOSFET
should be used to reduce the MOSFET’s junction-to-ambient
thermal impedance.

LAYOUT AND COMPONENT PLACEMENT GUIDELINES

The following guidelines are recommended for optimal perfor­mance of a switching regulator in a PC system:

General Recommendations

1. For best results, a four-layer (minimum) PCB is recom-

mended. This should allow the needed versatility for control
circuitry interconnections with optimal placement, a signal
ground plane, power planes for both power ground and the
input power (e.g., 5 V), and wide interconnection traces in
the rest of the power delivery current paths. Each square

unit of 1 ounce copper trace has a resistance of ~0.53 mΩ at

room temperature.

2. Whenever high currents must be routed between PCB layers

vias should be used liberally to create several parallel current
paths so that the resistance and inductance introduced by
these current paths is minimized and the via current rating is
not exceeded.

3. The power and ground planes should overlap each other as

little as possible. It is generally the easiest (although not
necessary) to have the power and signal ground planes on
the same PCB layer. The planes should be connected near­est to the first input capacitor where the input ground cur­rent flows from the converter back to the power source (e.g.,
5 V).

4. If critical signal lines (including the voltage and current

sense lines of the ADP3156) must cross through power
circuitry, it is best if a signal ground plane can be interposed
between those signal lines and the traces of the power cir­cuitry. This serves as a shield to minimize noise injection
into the signals at the expense of making signal ground a bit
noisier.

5. The PGND pin of the ADP3156 should connect first to a

ceramic bypass capacitor (on the V

pin) and then into the

CC

power ground plane using the shortest possible trace. How­ever, the power ground plane should not extend under other
signal components, including the ADP3156 itself. If neces­sary, follow the preceding guideline to use the signal plane
as a shield between the power ground plane and the signal
circuitry.

6. The AGND pin of the ADP3156 should connect first to the

timing capacitor (on the C

pin), and then into the signal

T

ground plane. In cases where no signal ground plane can be
used, short interconnections to other signal ground circuitry
in the power converter should be used—the compensation
capacitor being the next most critical.

7. The output capacitors of the power converter should be

connected to the signal ground plan even though power
current flows in the ground of these capacitors. For this
reason, it is advised to avoid critical ground connections (e.g.,
the signal circuitry of the power converter) in the signal ground
plane in between the input and output capacitors. It is also
advised to keep the planar interconnection path short (i.e.,
have input and output capacitors close together).

8. The output capacitors should also be connected as closely as

possible to the load (or connector) which receives the power
(e.g., a microprocessor core). If the load is distributed, the
capacitors also should be distributed, and generally in pro­portion to where the load tends to be more dynamic.

REV. 0

–11–

ADP3156

9. Absolutely avoid crossing any signal lines over the switching
power path loop, described below.

Power Circuitry

10. The switching power path should be routed on the PCB to
encompass the smallest possible area in order to minimize
radiated switching noise energy (i.e., EMI). Failure to take
proper precaution often results in EMI problems for the
entire PC system as well as noise-related operational prob­lems in the power converter control circuitry. The switching
power path is the loop formed by the current path through
the input capacitors, the two FETs, and the power Schottky
diode if used, including all interconnecting PCB traces and
planes. The use of short and wide interconnection traces is
especially critical in this path for two reasons: it minimizes
the inductance in the switching loop, which can cause high­energy ringing, and it accommodates the high current de­mand with minimal voltage loss.

11. A power Schottky diode (1~2 Adc rating) placed from the
lower FET’s source (anode) to drain (cathode) will help to
minimize switching power dissipation in the upper FET. In
the absence of an effective Schottky diode, this dissipation
occurs through the following sequence of switching events.
The lower FET turns off in advance of the upper FET turn­ing on (necessary to prevent cross-conduction). The circu­lating current in the power converter, no longer finding a
path for current through the channel of the lower FET,
draws current through the inherent body-drain diode of the
FET. The upper FET turns on, and the reverse recovery
characteristic of the lower FET’s body-drain diode prevents
the drain voltage from being pulled high quickly. The upper
FET then conducts very large current while it momentarily
has a high voltage forced across it, which translates into
added power dissipation in the upper FET. The Schottky
diode minimizes this problem by carrying a majority of the
circulating current when the lower FET is turned off, and by
virtue of its essentially nonexistent reverse recovery time.

12. A small ferrite bead inductor placed in series with the drain
of the lower FET can also help to reduce this previously
described source of switching power loss.

13. Whenever a power dissipating component (e.g., a power
MOSFET) is soldered to a PCB, the liberal use of vias both
directly on the mounting pad and immediately surrounding
it is recommended. Two important reasons for this are:
improved current rating through the vias (if it is a current
path), and improved thermal performance—especially if the
vias extended to the opposite side of the PCB where a plane
can more readily transfer the heat to the air.

14. The output power path, though not as critical as the switch­ing power path, should also be routed to encompass a small
area. The output power path is formed by the current path
through the inductor, the current sensing resistor, the out­put capacitors, and back to the input capacitors.

15. For best EMI containment, the power ground plane should
extend fully under all the power components except the
output capacitors. These are: the input capacitors, the power
MOSFETs and Schottky diode, the inductor, the current
sense resistor, and any snubbing elements that might be
added to dampen ringing. Avoid extending the power ground
under any other circuitry or signal lines, including the volt­age and current sense lines.

Signal Circuitry

16. The output voltage is sensed and regulated between the
AGND pin (which connects to the signal ground plane) and
the SENSE– pin. The output current is sensed (as a voltage)
and regulated between the SENSE– pin and the SENSE+
pin. In order to avoid differential mode noise pickup in
those sensed signals, their loop areas should be small. Thus
the SENSE– trace should be routed atop the signal ground
plane, and the SENSE+ and SENSE– traces should be routed
as a closely coupled pair (SENSE+ should be over the signal
ground plane as well).

17. The SENSE+ and SENSE– traces should be Kelvin con­nected to the current sense resistor so that the additional
voltage drop due to current flow on the PCB at the current
sense resistor connections does not affect the sensed voltage.
It is desirable to both have the ADP3156 close to the output
capacitor bank and not in the output power path so that any
voltage drop between the output capacitors and the AGND
pin is minimized, and voltage regulation is not compromised.

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

16-Lead Standard Small Outline Package (SOIC)

(R-16A)

0.3937 (10.00)

0.3859 (9.80)

0.1574 (4.00)

0.1497 (3.80)

PIN 1

0.0098 (0.25)

0.0040 (0.10)

16 9

0.050 (1.27)
BSC

0.0192 (0.49)

0.0138 (0.35)

0.2440 (6.20)

0.2284 (5.80)

81

0.0688 (1.75)

0.0532 (1.35)

SEATING
PLANE

0.0099 (0.25)

0.0075 (0.19)

0.0196 (0.50)

0.0099 (0.25)

88
08

0.0500 (1.27)

0.0160 (0.41)

3 458

C3598–2–4/99

PRINTED IN U.S.A.

–12–

REV. 0