Datasheet ADP2105, ADP2106, ADP2107 Datasheet (ANALOG DEVICES)

查询ADP2105供应商查询ADP2105供应商

1 Amp/1.5 Amp/2 Amp Synchronous,

FEATURES

Extremely high 97% efficiency
Ultralow quiescent current: 20 μA

1.2 MHz switching frequency

0.1 μA shutdown supply current
Maximum load current:

ADP2105: 1 A
ADP2106: 1.5 A

ADP2107: 2 A
Input voltage: 2.7 V to 5.5 V
Output voltage: 0.8 V to V
Maximum duty cycle: 100%
Smoothly transitions into low dropout (LDO) mode
Internal synchronous rectifier
Small 16-lead 4 mm × 4 mm LFCSP_VQ package
Optimized for small ceramic output capacitors
Enable/shutdown logic input
Undervoltage lockout
Soft start

APPLICATIONS

Mobile handsets
PDAs and palmtop computers
Telecommunication/networking equipment
Set top boxes
Audio/video consumer electronics

TYPICAL PERFORMANCE CHARACTERISTICS

100

VIN=3.3V

95

90

VIN=5V

85

EFFICIENCY (%)

80

75

0 2000

200 400 600 800 1000 1200 1400 1600 1800

Figure 1. Efficiency vs. Load Current for the ADP2107 with V

IN

VIN=3.6V

LOAD CURRENT (mA)

V

OUT

=2.5V

= 2.5 V

OUT

Step-Down DC-to-DC Converters

ADP2105/ADP2106/ADP2107

GENERAL DESCRIPTION

The ADP2105/ADP2106/ADP2107 are low quiescent current,
synchronous, step-down dc-to-dc converters in a compact 4 mm ×
4 mm LFCSP_VQ package. At medium to high load currents,
these devices use a current-mode, constant-frequency pulse­width modulation (PWM) control scheme for excellent stability
and transient response. To ensure the longest battery life in
portable applications, the ADP2105/ADP2106/ADP2107 use a
pulse frequency modulation (PFM) control scheme under light
load conditions that reduces switching frequency to save power.

The ADP2105/ADP2106/ADP2107 run from input voltages of

2.7 V to 5.5 V, allowing single Li+/Li− polymer cell, multiple
alkaline/NiMH cells, PCMCIA, and other standard power sources.
The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
adjustable from 0.8 V to the input voltage, whereas the
ADP2105/ADP2106/ADP2107-xx are available in preset output
voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V. Each of these
variations is available in three maximum current levels, 1 A
(ADP2105), 1.5 A (ADP2106), and 2 A (ADP2107). The power
switch and synchronous rectifier are integrated for minimal
external part count and high efficiency. During logic-controlled
shutdown, the input is disconnected from the output, and it
draws less than 0.1 µA from the input source. Other key features
include undervoltage lockout to prevent deep battery discharge
and programmable soft start to limit inrush current at startup.

TYPICAL OPERATING CIRCUIT

0.1F

FB

16 15 14 13

FB PWIN1

ON

OFF

06079-001

1

2

3

4

120pF

GND

EN

GND

ADP2107-ADJ

GND

GND

SS

COMP

5 6 7 8

70k

AGND

1nF

Figure 2. Circuit Configuration of ADP2107 with V

VININPUT VOLTAGE = 2.7V TO 5.5V

10

10F

IN

LX2

PGND

LX1

PWIN2

NC

OUTPUT VOLTAGE = 2.5V

12

2H

11

85k

10

FB

V

IN

40k

9

10F

NC = NO CONNECT

10F

OUT

4.7F

LOAD

0A TO 2A

= 2.5 V

06079-002

Rev. A

Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her
rights of third parties that may result from its use. Specifications subject to change without notice. No
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One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] ©2006–2007 Analog Devices, Inc. All rights reserved.

ADP2105/ADP2106/ADP2107

TABLE OF CONTENTS

Features.............................................................................................. 1

Applications....................................................................................... 1

General Description ......................................................................... 1

Typical Performance Characteristics ............................................. 1

Typical Operating Circuit................................................................ 1

Revision History ............................................................................... 2

Specifications..................................................................................... 3

Absolute Maximum Ratings............................................................ 5

Thermal Resistance ...................................................................... 5

Boundary Condition .................................................................... 5

ESD Caution.................................................................................. 5

Pin Configuration and Function Descriptions............................. 6

Typical Performance Characteristics ............................................. 7

Theory of Operation ...................................................................... 13

Control Scheme .......................................................................... 13

PWM Mode Operation.............................................................. 13

PFM Mode Operation................................................................ 13

Pulse-Skipping Threshold......................................................... 13

100% Duty Cycle Operation (LDO Mode)............................. 13

Slope Compensation .................................................................. 14

Features........................................................................................ 14

Applications Information .............................................................. 16

External Component Selection ................................................ 16

Setting the Output Voltage........................................................ 16

Inductor Selection...................................................................... 17

Output Capacitor Selection....................................................... 18

Input Capacitor Selection.......................................................... 18

Input Filter................................................................................... 19

Soft Start ...................................................................................... 19

Loop Compensation .................................................................. 19

Bode Plots.................................................................................... 20

Load Transient Response .......................................................... 21

Efficiency Considerations ......................................................... 22

Thermal Considerations............................................................ 22

Design Example.......................................................................... 23

External Component Recommendations.................................... 24

Circuit Board Layout Recommendations ................................... 26

Evaluation Board............................................................................ 27

Evaluation Board Schematic (ADP2107-1.8V)...................... 27

Recommended PCB Board Layout

(Evaluation Board Layout)........................................................ 27

Application Circuits ....................................................................... 29

Outline Dimensions....................................................................... 31

Ordering Guide .......................................................................... 31

REVISION HISTORY

3/07—Rev. 0 to Rev. A

Updated Format..................................................................Universal

Changes to Output Characteristics and

LX (Switch Node) Characteristics Sections .................................. 3

Changes to Typical Performance Characteristics Section........... 7

Changes to Load Transient Response Section ............................ 21

7/06—Revision 0: Initial Version

Rev. A | Page 2 of 32

ADP2105/ADP2106/ADP2107

SPECIFICATIONS

VIN = 3.6 V @ TA = 25°C, unless otherwise noted.1 Bold values indicate −40°C ≤ TJ ≤ +125°C.

Table 1.

Parameter Conditions Min Typ Max Unit

INPUT CHARACTERISTICS

Input Voltage Range
Undervoltage Lockout Threshold VIN rising
V
Undervoltage Lockout Hysteresis

2

IN

falling

2.7 5.5

2.2

2.0

2.4

2.2
200

2.6

2.5

OUTPUT CHARACTERISTICS

Output Regulation Voltage ADP210x-3.3, load = 10 mA 3.267 3.3 3.333 V

ADP210x-3.3, VIN = 3.6 V to 5.5 V, no load to full load

3.201

3.3

3.399

ADP210x-1.8, load = 10 mA 1.782 1.8 1.818 V
ADP210x-1.8, VIN = 2.7 V to 5.5 V, no load to full load

1.746

1.8

1.854

ADP210x-1.5, load = 10 mA 1.485 1.5 1.515 V

ADP210x-1.5, VIN = 2.7 V to 5.5 V, no load to full load

1.455

1.5

1.545

ADP210x-1.2, load = 10 mA 1.188 1.2 1.212 V
ADP210x-1.2, VIN = 2.7 V to 5.5 V, no load to full load
Load Regulation ADP2105
ADP2106
ADP2107
Line Regulation

3

ADP2105, measured in servo loop

1.164

1.2

0.4

0.5

0.6

1.236

0.1 0.33 %/V

ADP2106 and ADP2107, measured in servo loop 0.1 0.3 %/V

Output Voltage Range ADP210x-ADJ 0.8 VIN V

FEEDBACK CHARACTERISTICS

OUT_SENSE Bias Current

ADP210x-1.2 3
ADP210x-1.5 4
ADP210x-1.8 5

ADP210x-3.3 10
FB Regulation Voltage ADP210x-ADJ
FB Bias Current ADP210x-ADJ

0.784

−0.1

INPUT CURRENT CHARACTERISTICS

IN Operating Current ADP210x-ADJ, VFB = 0.9 V 20

ADP210x-xx, output voltage 10% above regulation voltage 20

IN Shutdown Current

5

VEN = 0 V 0.1 1 µA

LX (SWITCH NODE) CHARACTERISTICS

LX On Resistance

4

P-channel switch, ADP2105 190
P-channel switch, ADP2106 and ADP2107 100
N-channel synchronous rectifier, ADP2105 160
N-channel synchronous rectifier, ADP2106 and ADP2107 90
LX Leakage Current
LX Peak Current Limit
P-channel switch, ADP2106
P-channel switch, ADP2105
LX Minimum On-Time

ENABLE CHARACTERISTICS

EN Input High Voltage VIN = 2.7 V to 5.5 V
EN Input Low Voltage VIN = 2.7 V to 5.5 V
EN Input Leakage Current VIN = 5.5 V, VEN = 0 V, 5.5 V

OSCILLATOR FREQUENCY VIN = 2.7 V to 5.5 V

4, 5

4

P-channel switch, ADP2107

2.6

2.0

1.3

VIN = 5.5 V, VLX = 0 V, 5.5 V 0.1 1 µA

4

In PWM mode of operation, VIN = 5.5 V

2

0.4

−1 −0.1 +1
1

0.8

2.9

2.25

1.5

1.2

6
8
10
20

0.816
+0.1

30
30

270
165
230
140

3.3

2.6

1.8
100

1.4

SOFT START PERIOD CSS = 1 nF 750 1000 1200 µs

V
V
V
mV

V

V

V

V
%/A
%/A
%/A

µA
µA
µA
µA
V
µA

µA
µA

mΩ
mΩ
mΩ
mΩ

A
A
A
ns

V
V
µA
MHz

Rev. A | Page 3 of 32

ADP2105/ADP2106/ADP2107

Parameter Conditions Min Typ Max Unit

THERMAL CHARACTERISTICS

Thermal Shutdown Threshold 140
Thermal Shutdown Hysteresis 40

COMPENSATOR TRANSCONDUCTANCE (Gm) 50 µA/V

2

CURRENT SENSE AMPLIFIER GAIN (GCS)

ADP2105 1.875 A/V
ADP2106 2.8125 A/V
ADP2107 3.625 A/V

1

All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C.

2

Guaranteed by design.

3

The ADP2015/ADP2106/ADP2107 line regulation was measured in a servo loop on the ATE that adjusts the feedback voltage to achieve a specific comp voltage.

4

All LX (switch node) characteristics are guaranteed only when the LX1 and LX2 pins are tied together.

5

These specifications are guaranteed from −40°C to +85°C.

°C
°C

Rev. A | Page 4 of 32

ADP2105/ADP2106/ADP2107

ABSOLUTE MAXIMUM RATINGS

Table 2.

Parameter Rating

IN, EN, SS, COMP, OUT_SENSE/FB to

AGND

LX1, LX2 to PGND −0.3 V to (VIN + 0.3 V)
PWIN1, PWIN2 to PGND −0.3 V to +6 V
PGND to AGND −0.3 V to +0.3 V
GND to AGND −0.3 V to +0.3 V
PWIN1, PWIN2 to IN −0.3 V to +0.3 V
Operating Junction Temperature Range −40°C to +125°C
Storage Temperature Range −65°C to +150°C
Soldering Conditions JEDEC J-STD-020

−0.3 V to +6 V

THERMAL RESISTANCE

θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.

Table 3. Thermal Resistance

Package Type θ

16-Lead LFCSP_VQ/QFN 40 °C/W
Maximum Power Dissipation 1 W

JA

Unit

BOUNDARY CONDITION

Natural convection, 4-layer board, exposed pad soldered to the PCB.

Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.

ESD CAUTION

Rev. A | Page 5 of 32

ADP2105/ADP2106/ADP2107

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

15 GND

16 OUT_SENSE/FB

14 IN

13 PWIN1

PIN 1

EN

GND

GND

GND

1

2

3

4

INDICATOR

ADP2105/
ADP2106/
ADP2107

TOP VIEW

(Not to Scale)

6

5

7

SS

AGND

COMP

NC = NO CONNECT

Figure 3. Pin Configuration

Table 4. Pin Function Descriptions

Mnemonic
Pin No. ADP210x-xx ADP210x-ADJ Description

1 EN EN

Enable Input. Drive EN high to turn on the ADP2105/ADP2106/ADP2107. Drive EN low to turn
it off and reduce the input current to 0.1 μA.

2, 3, 4,
15

GND GND

Test Pins. These pins are used by Analog Devices, Inc. for internal testing and are not ground
return pins. Tie these pins to the AGND plane as close to the ADP2105/ADP2106/ADP2107 as
possible.

5 COMP COMP

Feedback Loop Compensation Node. COMP is the output of the internal transconductance
error amplifier. Place a series RC network from COMP to AGND to compensate the converter.
See the Loop Compensation section.

6 SS SS

Soft Start Input. Place a capacitor from SS to AGND to set the soft start period. A 1 nF capacitor
sets a 1 ms soft start period.

7 AGND AGND

Analog Ground. Connect the ground of the compensation components, soft start capacitor,
and the voltage divider on the FB pin to the AGND pin as close as possible to the ADP2105/

ADP2106/ADP2107. Also connect AGND to the exposed pad of ADP2105/ADP2106/ADP2107.
8 NC NC No Connect. Not internally connected. Can be connected to other pins or left unconnected.
9, 13

PWIN2,
PWIN1

PWIN2, PWIN1

Power Source Inputs. The source of the PFET high-side switch. Bypass each PWIN pin to the nearest

PGND plane with a 4.7 μF or greater capacitor as close as possible to the ADP2105/ADP2106/

ADP2107. See the Input Capacitor Selection section.
10, 12 LX1, LX2 LX1, LX2

Switch Outputs. The drain of the P-channel power switch and N-channel synchronous rectifier.

Tie the two LX pins together and connect the output LC filter between LX and the output

voltage.
11 PGND PGND

Power Ground. Connect the ground return of all input and output capacitors to PGND pin,

using a power ground plane as close as possible to the ADP2105/ADP2106/ADP2107. Also

connect PGND to the exposed pad of the ADP2105/ADP2106/ADP2107.
14 IN IN

ADP2105/ADP2106/ADP2107 Power Input. The power source for the ADP2105/ADP2106/

ADP2107 internal circuitry. Connect IN and PWIN1 with a 10 Ω resistor as close as possible to

the ADP2105/ADP2106/ADP2107. Bypass IN to AGND with a 0.1 μF or greater capacitor. See

the Input Filter section.
16 OUT_SENSE FB

Output Voltage Sense or Feedback Input. For fixed output versions, connect OUT_SENSE to the

output voltage. For adjustable versions, FB is the input to the error amplifier. Drive FB through

a resistive voltage divider to set the output voltage. The FB regulation voltage is 0.8 V.

8
NC

12 LX2

11 PGND

10 LX1

9PWIN2

6079-003

Rev. A | Page 6 of 32

ADP2105/ADP2106/ADP2107

TYPICAL PERFORMANCE CHARACTERISTICS

100

100

95

90

85

80

75

EFFICIENCY (%)

70

65

60

11

VIN = 2.7V

VIN = 5.5V

VIN = 3.6V

VIN = 4.2V

INDUCTOR: SD14, 2.5µH
DCR: 60m
T

= 25°C

A

10 100

LOAD CURRENT (mA)

06079-084

000

Figure 4. Efficiency—ADP2105 (1.2 V Output)

100

95

90

85

80

75

EFFICIENCY (%)

70

65

60

1 1000

VIN = 4.2V

10 100

LOAD CURRENT (mA)

VIN = 3.6V

VIN = 5.5V

INDUCTOR: CDRH5D18, 4. 1H
DCR: 43m
T

= 25°C

A

06079-085

Figure 5. Efficiency—ADP2105 (3.3 V Output)

100

95

90

VIN=2.7V

85

80

75

70

EFFICIENCY (%)

65

60

55

50

1 10000

VIN=3.6V

VIN=4.2V

VIN=5.5V

INDUCTOR: D62LCB, 2µH
DCR: 28m
T

= 25°C

A

10 100 1000

LOAD CURRENT (mA)

06079-062

Figure 6. Efficiency—ADP2106 (1.8 V Output)

95

90

85

80

EFFICIE NCY (%)

75

70

65

1 1000

Figure 7. Efficiency—ADP2105 (1.8 V Output)

100

95

90

VIN=3.6V

85

80

75

70

EFFICIENCY (%)

65

60

55

50

1 10000

Figure 8. Efficiency—ADP2106 (1.2 V Output)

100

95

90

85

80

VIN=4.2V

75

70

EFFICIENCY (%)

65

60

55

50

VIN=3.6V

1 10000

Figure 9. Efficiency—ADP2106 (3.3 V Output)

VIN = 2.7V

VIN = 3.6V

VIN = 4.2V

VIN = 5.5V

INDUCTOR: SD3814, 3.3µH
DCR: 93m
T

= 25°C

A

10 100

LOAD CURRENT (mA)

VIN=2.7V

VIN=4.2V

VIN=5.5V

INDUCTOR: D62LCB, 2µH
DCR: 28m
T

=25°C

A

10 100 1000

LOAD CURRENT (mA)

VIN=5.5V

INDUCTOR: D62LCB, 3.3µ H
DCR: 47m
T

= 25°C

A

10 100 1000

LOAD CURRENT (mA)

06079-086

06079-008

06079-053

Rev. A | Page 7 of 32

ADP2105/ADP2106/ADP2107

100

95

90

VIN=3.6V

85

80

75

70

EFFICIENCY (%)

65

60

55

50

1 10000

VIN=2.7V

VIN=4.2V

VIN=5.5V

INDUCTOR: SD12, 1.2µH
DCR: 37m
T

=25°C

A

10 100 1000

LOAD CURRENT (mA)

Figure 10. Efficiency—ADP2107 (1.2 V)

100

95

90

85

80

VIN=4.2V

75

70

EFFICIENCY (%)

65

60

55

50

1 10000

VIN=3.6V

10 100 1000

LOAD CURRENT (mA)

VIN=5.5V

INDUCTOR: CDRH5D28, 2.5µH
DCR: 13m
T

= 25°C

A

Figure 11. Efficiency—ADP2107 (3.3 V)

1.85

1.83

06079-010

06079-054

100

95

90

85

80

75

70

EFFICIENCY (%)

65

60

55

50

1 10000

VIN=2.7V

VIN=4.2V

10 100 1000

VIN=3.6V

VIN=5.5V

INDUCTOR: D62LCB, 1.5µ H
DCR: 21m
T

=25°C

A

LOAD CURRENT (mA)

Figure 13. Efficiency—ADP2107 (1.8 V)

1.23

2.7V, –40°C 2.7V, +25°C 2.7V, +125°C

3.6V, –40°C 3.6V, +25°C 3.6V, +125°C

5.5V, –40°C 5.5V, +25°C

1.22

1.21

1.20

1.19

OUTPUT VOLTAGE (V)

1.18

1.17

0.01 10000

0.1 1 10 100 1000

LOAD CURRENT (mA)

5.5V, +125°C

Figure 14. Output Voltage Accuracy—ADP2107 (1.2 V)

3.38

3.6V, –40°C 3.6V, +25°C 3.6V, +125°C

5.5V, –40°C 5.5V, +25°C

3.36

3.34

5.5V, + 125°C

06079-063

06079-082

1.81

1.79

OUTPUT VOLTAGE (V)

1.77

2.7V, –40°C 2.7V, +25°C 2.7V, +125°C

3.6V, –40°C 3.6V, +25°C 3.6V, +125°C

5.5V, –40°C 5.5V, +25°C

1.75

0.1 10000

1 10 100 1000

LOAD CURRENT (mA)

5.5V, +125°C

Figure 12. Output Voltage Accuracy—ADP2107 (1.8 V)

06079-064

Rev. A | Page 8 of 32

3.32

3.30

3.28

OUTPUT VOLTAGE (V)

3.26

3.24

3.22

0.01 10000

0.1 1 10 100 1000

LOAD CURRENT (mA)

Figure 15. Output Voltage Accuracy—ADP2107 (3.3 V)

06079-081

ADP2105/ADP2106/ADP2107

10000

1000

INPUT CURRENT (µ A)

0.802

0.801

0.800

0.799

0.798

0.797

FEEDBACK VOLT AGE (V)

0.796

+25°C

100

10

1

0.8

–40°C

+125°C

1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0 4.4 4.8 5.2

INPUT VOLTAGE (V)

Figure 16. Quiescent Current vs. Input Voltage

06079-016

190

180

170

160

150

140

130

NMOS SYNCHRONOUS RECTIFI ER

120

SWITCH ON RESISTANCE (m)

110

100

2.7 3.0 3.3 3.6 3.9 4.2 4.5 5.1 5.44.8

PMOS POWER SWITCH

INPUT VOLTAGE (V)

Figure 19. Switch On Resistance vs. Input Voltage—ADP2105

120

100

80

60

40

SWITCH ON RESISTANCE (m)

20

PMOS POWER SWITCH

NMOS SYNCHRONOUS RECTIFI ER

06079-093

0.795
–40 125

–20 0 20 40 60 80 100 120

TEMPERATURE (°C)

Figure 17. Feedback Voltage vs. Temperature

1.75

1.70

1.65

1.60

1.55

1.50

1.45

1.40

PEAK CURRENT LIMIT (A)

1.35

1.30

1.25

2.7 5.7

3.0 3.3 3.6 3.9 4.2 4. 5 4.8 5.1 5.4

INPUT VOLTAGE (V)

ADP2105 (1A)

Figure 18. Peak Current Limit of ADP2105

TA= 25°C

06079-017

0

2.7 5.4

3.03.33.63.94.24.54.85.1

INPUT VOLTAGE (V)

TA = 25°C

06079-018

Figure 20. Switch On Resistance vs. Input Voltage—ADP2106 and ADP2107

1260

1250

1240

1230

1220

1210

SWITCHING FREQUENCY ( kHz)

1200

06079-073

1190

2.7 5.4

3.03.33.63.94.24.54.85.1

+125°C

–40°C

INPUT VOLTAGE (V)

+25°C

06079-021

Figure 21. Switching Frequency vs. Input Voltage

Rev. A | Page 9 of 32

ADP2105/ADP2106/ADP2107

2.35

2.30

2.25

2.20

2.15

2.10

2.05

2.00

PEAK CURRENT LIMIT (A)

1.95

1.90

1.85

2.7 5.7

3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4

INPUT VOLTAGE (V)

ADP2106 (1.5A)

Figure 22. Peak Current Limit of ADP2106

3.00

2.95

2.90

2.85

2.80

2.75

2.70

2.65

PEAK CURRENT LIMIT (A)

2.60

2.55

2.50

2.7 5.7

3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4

INPUT VOLTAGE (V)

ADP2107 (2A)

Figure 23. Peak Current Limit of ADP2107

150

135

120

105

90

75

60

V

45

30

15

PULSE-SKIP PING THRESHO LD CURRENT (mA)

0

2.7 5.7

V

= 1.2V

OUT

V

= 2.5V

= 1.8V

OUT

3.0 3.3 3.6 3.9 4.2 4. 5 4.8 5. 1 5.4

INPUT VOLTAGE (V)

OUT

TA= 25°C

TA=25°C

TA = 25°C

3

1

4

06079-072

135

120

105

90

75

60

45

30

15

06079-071

PULSE-SKIP PING THRESHO LD CURRENT (mA)

Figure 26. Pulse-Skipping Threshold vs. Input Voltage for ADP2105

195

180

165

150

135

120

105

90

75

60

45

30

15

06079-067

PULSE-SKIP PING THRESHO LD CURRENT (mA)

LX NODE (SWITCH NODE)

CH1 1V

INDUCTOR CURRENT

OUTPUT V OLTAGE

M 10µs A CH1 1.78V

45.8%CH4 1ACH3 5V

T

: 260mV
@: 3.26V

Figure 25. Short Circuit Response at Output

V

= 1.2V

OUT

V

= 1.8V

OUT

0

2.7 5.7

3.0 3.3 3.6 3.9 4.2 4. 5 4.8 5. 1 5.4

INPUT VOLTAGE (V)

0

2.7 5.7

3.0 3.3 3.6 3.9 4.2 4.5 4.8 5.1 5.4

INPUT VOLTAGE (V)

V

= 2.5V

OUT

TA = 25°C

V

= 1.2V

OUT

V

= 1.8V

OUT

V

= 2.5V

OUT

TA = 25°C

06079-074

06079-066

06079-068

Figure 24. Pulse-Skipping Threshold vs. Input Voltage for ADP2106

Figure 27. Pulse-Skipping Threshold vs. Input Voltage for ADP2107

Rev. A | Page 10 of 32

ADP2105/ADP2106/ADP2107

250

230

210

190

170

150

130

110

90

SWITCH ON RESISTANCE (m)

70

50

–40 –20 0 20 40 60 80 100 120

Figure 28. Switch On Resistance vs. Temperature—ADP2105

140

120

100

80

60

40

SWITCH ON RESISTANCE (m)

20

0

–40

–20 0 20 40 60 80 100 120

Figure 29. Switch On Resistance vs. Temperature—ADP2106 and ADP2107

PMOS POWER SWITCH

NMOS SYNCHRONOUS RECTIFIER

JUNCTION TEM PERATURE (°C)

PMOS POWER SWITCH

NMOS SYNCHRONOUS RECTI FIER

JUNCTION T EMPERATURE (°C)

3

LX NODE (SWITCH NODE)

1

4

06079-093

CH1 50mV

OUTPUT V OLTAGE (AC-COUPLED)

INDUCTOR CURRENT

M 400ns A CH3 3.88V

17.4%CH4 200mACH3 2V

T

06079-033

Figure 31. DCM Mode of Operation at Light Load (100 mA)

LX NODE (SWITCH NODE)

3

1

OUTPUT V OLTAGE (AC-COUPLED)

06079-083

4

CH1 20mV

INDUCTOR CURRENT

M 2µs A CH3 1.84V

13.4%CH4 1ACH3 2V

T

06079-034

Figure 32. Minimum Off Time Control at Dropout

LX NODE

3

1

OUTPUT VOLTAGE (AC-COUPLED)

4

INDUCTOR CURRENT

CH1 50mV

(SWITCH NODE)

M 2µs A CH3 3.88V

6%CH4 200mACH3 2V

T

Figure 30. PFM Mode of Operation at Very Light Load (10 mA)

06079-030

Rev. A | Page 11 of 32

LX NODE (SWITCH NODE)

3

1

OUTPUT VOLTAGE (AC-COUPLED)

INDUCTOR CURRENT

4

CH1 20mV

M 1µs A CH3 3.88V

17.4%CH4 1ACH3 2V

T

Figure 33. PWM Mode of Operation at Medium/Heavy Load (1.5 A)

06079-031

ADP2105/ADP2106/ADP2107

LX NO DE (S WIT CH NOD E)

3

ENABLE VOL TAGE

3

CHANNEL 3
FREQUENCY
= 336.6kHz

INDUCTOR CURRENT

OUTPUT V OLTAG E

1

4

CH1 1V

M 4µs A CH3 1.8V

45%CH4 1ACH3 5V

T

: 2.86A
@: 2.86A

06079-032

Figure 34. Current Limit Behavior of ADP2107 (Frequency Foldback)

OUTPUT V OLTAGE

1

INDUCTOR CURRENT

4

CH1 1V

Figure 35. Startup and Shutdown Waveform (C

M 400µs A CH1 1.84V

20.2%CH4 500mACH3 5V

T

= 1 nF → SS Time = 1 ms)

SS

06079-035

Rev. A | Page 12 of 32

ADP2105/ADP2106/ADP2107

THEORY OF OPERATION

The ADP2105/ADP2106/ADP2107 are step-down, dc-to-dc
converters that use a fixed frequency, peak current-mode
architecture with an integrated high-side switch and low-side
synchronous rectifier. The high 1.2 MHz switching frequency
and tiny 16-lead, 4 mm × 4 mm LFCSP_VQ package allow for
a small step-down dc-to-dc converter solution. The integrated
high-side switch (P-channel MOSFET) and synchronous rectifier
(N-channel MOSFET) yield high efficiency at medium to heavy
loads. Light load efficiency is improved by smoothly transitioning
to variable frequency PFM mode.

The ADP2105/ADP2106/ADP2107-ADJ operate with an input
voltage from 2.7 V to 5.5 V and regulate an output voltage down
to 0.8 V. The ADP2105/ADP2106/ADP2107 are also available with
preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V.

CONTROL SCHEME

The ADP2105/ADP2106/ADP2107 operate with a fixed
frequency, peak current-mode PWM control architecture at
medium to high loads for high efficiency, but shift to a variable
frequency PFM control scheme at light loads for lower quies­cent current. When operating in fixed frequency PWM mode,
the duty cycle of the integrated switches is adjusted to regulate
the output voltage, but when operating in PFM mode at light loads,
the switching frequency is adjusted to regulate the output voltage.

The ADP2105/ADP2106/ADP2107 operate in the PWM mode
only when the load current is greater than the pulse-skipping
threshold current. At load currents below this value, the converter
smoothly transitions to the PFM mode of operation.

PWM MODE OPERATION

In PWM mode, the ADP2105/ADP2106/ADP2107 operate at
a fixed frequency of 1.2 MHz set by an internal oscillator. At the
start of each oscillator cycle, the P-channel MOSFET switch is
turned on, putting a positive voltage across the inductor. Current
in the inductor increases until the current sense signal crosses
the peak inductor current level that turns off the P-channel
MOSFET switch and turns on the N-channel MOSFET synchro­nous rectifier. This puts a negative voltage across the inductor,
causing the inductor current to decrease. The synchronous
rectifier stays on for the rest of the cycle, unless the inductor
current reaches zero, which causes the zero-crossing comparator
to turn off the N-channel MOSFET, as well. The peak inductor
current is set by the voltage on the COMP pin. The COMP pin
is the output of a transconductance error amplifier that compares
the feedback voltage with an internal 0.8 V reference.

PFM MODE OPERATION

The ADP2105/ADP2106/ADP2107 smoothly transition to the
variable frequency PFM mode of operation when the load current
decreases below the pulse-skipping threshold current, switching
only as necessary to maintain the output voltage within regulation.
When the output voltage dips below regulation, the ADP2105/
ADP2106/ADP2107 enter PWM mode for a few oscillator cycles
to increase the output voltage back to regulation. During the wait
time between bursts, both power switches are off, and the output
capacitor supplies all the load current. Because the output voltage
dips and recovers occasionally, the output voltage ripple in this
mode is larger than the ripple in the PWM mode of operation.

PULSE-SKIPPING THRESHOLD

The output current at which the ADP2105/ADP2106/ADP2107
transition from variable frequency PFM control to fixed frequency
PWM control is called the pulse-skipping threshold. The pulse­skipping threshold has been optimized for excellent efficiency
over all load currents. The variation of pulse-skipping threshold
with input voltage and output voltage is shown in

Figure 24,

Figure 26, and Figure 27.

100% DUTY CYCLE OPERATION (LDO MODE)

As the input voltage drops, approaching the output voltage,
the ADP2105/ADP2106/ADP2107 smoothly transition to 100%
duty cycle, maintaining the P-channel MOSFET switch on continu­ously. This allows the ADP2105/ADP2106/ADP2107 to regulate
the output voltage until the drop in input voltage forces the
P-channel MOSFET switch to enter dropout, as shown in the
following equation:

= I

V

IN(MIN)

OUT

× (R

The ADP2105/ADP2106/ADP2107 achieve 100% duty cycle
operation by stretching the P-channel MOSFET switch on time
if the inductor current does not reach the peak inductor current
level by the end of the clock cycle. Once this happens, the oscil­lator remains off until the inductor current reaches the peak
inductor current level, at which time the switch is turned off and
the synchronous rectifier is turned on for a fixed off time. At
the end of the fixed off time, another cycle is initiated. As the
ADP2105/ADP2106/ADP2107 approach dropout, the switching
frequency decreases gradually to smoothly transition to 100%
duty cycle operation.

DS(ON) − P

+ DCR

IND

) + V

OUT(NOM)

Rev. A | Page 13 of 32

ADP2105/ADP2106/ADP2107

SLOPE COMPENSATION

Slope compensation stabilizes the internal current control loop
of the ADP2105/ADP2106/ADP2107 when operating beyond
50% duty cycle to prevent subharmonic oscillations. It is imple­mented by summing a fixed scaled voltage ramp to the current
sense signal during the on time of the P-channel MOSFET switch.

The slope compensation ramp value determines the minimum
inductor that can be used to prevent subharmonic oscillations
at a given output voltage. The slope compensation ramp values
for ADP2105/ADP2106/ADP2107 follow. For more information,
see the

Inductor Selection section.

For the ADP2105:

Slope Compensation Ramp Value = 0.72 A/µs

For the ADP2106:

Slope Compensation Ramp Value = 1.07 A/µs

For the ADP2107:

Slope Compensation Ramp Value = 1.38 A/µs

FEATURES

Enable/Shutdown

Drive EN high to turn on the ADP2105/ADP2106/ADP2107.
Drive EN low to turn off the ADP2105/ADP2106/ADP2107,
reducing input current below 0.1 μA. To force the ADP2105/
ADP2106/ADP2107 to automatically start when input power
is applied, connect EN to IN. When shut down, the ADP2105/
ADP2106/ADP2107 discharge the soft start capacitor, causing
a new soft start cycle every time they are re-enabled.

Synchronous Rectification

In addition to the P-channel MOSFET switch, the ADP2105/
ADP2106/ADP2107 include an integrated N-channel MOSFET
synchronous rectifier. The synchronous rectifier improves
efficiency, especially at low output voltage, and reduces cost and
board space by eliminating the need for an external rectifier.

Current Limit

The ADP2105/ADP2106/ADP2107 have protection circuitry to
limit the direction and amount of current flowing through the
power switch and synchronous rectifier. The positive current
limit on the power switch limits the amount of current that can
flow from the input to the output, and the negative current limit
on the synchronous rectifier prevents the inductor current from
reversing direction and flowing out of the load.

Short Circuit Protection

The ADP2105/ADP2106/ADP2107 include frequency foldback
to prevent output current runaway on a hard short. When the
voltage at the feedback pin falls below 0.3 V, indicating the possi­bility of a hard short at the output, the switching frequency is
reduced to 1/4 of the internal oscillator frequency. The reduction
in the switching frequency gives more time for the inductor to
discharge, preventing a runaway of output current.

Undervoltage Lockout (UVLO)

To protect against deep battery discharge, undervoltage lockout
circuitry is integrated on the ADP2105/ADP2106/ADP2107.
If the input voltage drops below the 2.2 V UVLO threshold, the
ADP2105/ADP2106/ADP2107 shutdown, and both the power
switch and synchronous rectifier turn off. Once the voltage rises
again above the UVLO threshold, the soft start period is initiated,
and the part is enabled.

Thermal Protection

In the event that the ADP2105/ADP2106/ADP2107 junction
temperatures rise above 140°C, the thermal shutdown circuit turns
off the converter. Extreme junction temperatures can be the
result of high current operation, poor circuit board design, and/or
high ambient temperature. A 40°C hysteresis is included so that
when thermal shutdown occurs, the ADP2105/ADP2106/
ADP2107 do not return to operation until the on-chip tempera­ture drops below 100°C. When coming out of thermal
shutdown, soft start is initiated.

Soft Start

The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect the soft start
capacitor (C

) from SS to AGND. When the ADP2105/ADP2106/

SS

ADP2107 are disabled, or if the input voltage is below the under­voltage lockout threshold, C
ADP2105/ADP2106/ADP2107 are enabled, C

is internally discharged. When the

SS

is charged through

SS

an internal 0.8 µA current source, causing the voltage at SS to rise
linearly. The output voltage rises linearly with the voltage at SS.

Rev. A | Page 14 of 32

ADP2105/ADP2106/ADP2107

COMP

FB

OUT_SENSE

AGND

GND

GND

GND

GND

5

SS

NC

EN

1

START

1

16

1

16

7

FOR PRESET

VOLTAGES

OPTIONS ONLY

2

3

4

8

15

1

FB FOR ADP210x- ADJ (ADJUSTABLE VERSION) AND O UT_SENSE F OR ADP210x-xx (FIXED VERSI ON).

SOFT

6

REFERENCE

SLOPE

COMPENSATION

OSCILLATOR

0.8V

GM ERROR

AMP

PWM/

PFM

CONTROL

CURRENT SENSE

AMPLIFI ER

CURRENT

LIMIT

DRIVER

AND

ANTI-

SHOOT

THROUGH

ZERO CROSS
COMPARATOR

THERMAL

SHUTDOWN

Figure 36. Block Diagram of the ADP2105/ADP2106/ADP2107

14

9

13

10

12

11

IN

PWIN2

PWIN1

LX1

LX2

PGND

06079-037

Rev. A | Page 15 of 32

ADP2105/ADP2106/ADP2107

V

APPLICATIONS INFORMATION

EXTERNAL COMPONENT SELECTION

The external component selection for the ADP2105/ADP2106/
ADP2107 application circuits shown in

Figure 37 and Figure 38
depend on input voltage, output voltage, and load current
requirements. Additionally, trade-offs between performance
parameters like efficiency and transient response can be made
by varying the choice of external components.

SETTING THE OUTPUT VOLTAGE

The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
externally set by a resistive voltage divider from the output
voltage to FB. The ratio of the resistive voltage divider sets the
output voltage, and the absolute value of those resistors sets the
divider string current. For lower divider string currents, the
small 10 nA (0.1 A maximum) FB bias current should be taken

0.1F

V

OUT

16 15 14 13

OUT_SENSE

ON

1

OFF

EN

2

GND

3

GND

4

GND

COMP

5 6 7 8

R

COMP

C

COMP

ADP2105/
ADP2106/
ADP2107

SS

AGND

C

SS

NC = NO CO NNECT

Figure 37. Typical Applications Circuit for Fixed Output Voltage Options (ADP2105/ADP2106/ADP2107-xx)

0.1F

FB

16 15 14 13

FB PWIN1

GND

EN

GND

ADP2105/
ADP2106/
ADP2107

GND

GND

SS

COMP

5 6 7 8

COMP

C

SS

OFF

ON

R

C

1

2

3

4

COMP

Figure 38. Typical Applications Circuit for Adjustable Output Voltage Option (ADP2105/ADP2106/ADP2107-ADJ)

V

IN

10

PWIN1INGND

LX2

PGND

LX1

PWIN2

NC

10

IN

LX2

PGND

LX1

PWIN2

AGND

NC

NC = NO CONNECT

INPUT VOLTAGE = 2.7V TO 5.5V

12

11

10

V

9

VININPUT VOLTAGE = 2.7V TO 5.5V

12

11

10

into account when calculating resistor values. The FB bias
current can be ignored for a higher divider string current, but
this degrades efficiency at very light loads.

To limit output voltage accuracy degradation due to FB bias
current to less than 0.05% (0.5% maximum), ensure that the
divider string current is greater than 20 A. To calculate the
desired resistor values, first determine the value of the bottom
divider string resistor, R

V

I

STRING

FB

R =

BOT

, by

BOT

where:

= 0.8 V, the internal reference.

V

FB

I

is the resistor divider string current.

STRING

C

IN1

OUTPUT VOLTAGE = 1.2V, 1.5V, 1.8V, 3.3

L

IN

C

IN2

C

IN1

L

V

IN

9

FB

C

IN2

V

C

OUT

OUTPUT VOLTAGE
=0.8VTOV

R

TOP

R

BOT

OUT

LOAD

06079-065

IN

C

OUT

LOAD

06079-038

Rev. A | Page 16 of 32

ADP2105/ADP2106/ADP2107

I

Once R
R

TOP

The ADP2105/ADP2106/ADP2107-xx (where xx represents
the fixed output voltage) include the resistive voltage divider
internally, reducing the external circuitry required. Connect the
OUT_SENSE to the output voltage as close as possible to the
load for improved load regulation.

INDUCTOR SELECTION

The high switching frequency of ADP2105/ADP2106/ADP2107
allows for minimal output voltage ripple even with small inductors.
The sizing of the inductor is a trade-off between efficiency and
transient response. A small inductor leads to larger inductor
current ripple that provides excellent transient response but
degrades efficiency. Due to the high switching frequency of
ADP2105/ADP2106/ADP2107, shielded ferrite core inductors
are recommended for their low core losses and low EMI.

As a guideline, the inductor peak-to-peak current ripple, I
is typically set to 1/3 of the maximum load current for optimal
transient response and efficiency.

where fSW is the switching frequency (1.2 MHz).

The ADP2105/ADP2106/ADP2107 use slope compensation in
the current control loop to prevent subharmonic oscillations
when operating beyond 50% duty cycle. The fixed slope compen­sation limits the minimum inductor value as a function of
output voltage.

For the ADP2105:

For the ADP2106:

For the ADP2107:

Also, 4.7 µH or larger inductors are not recommended because
they may cause instability in discontinuous conduction mode
under light load conditions.

Finally, it is important that the inductor be capable of handling
the maximum peak inductor current, I
following equation:

is determined, calculate the value of the top resistor,

BOT

, by

⎡

=

RR

BOTTOP

⎢
⎣

OUT

I ≈

=Δ

L

L

IDEAL

IN

=⇒

L > (1.12 µH/V) × V

L > (0.83 µH/V) × V

L > (0.66 µH/V) × V

II

PK

)(LMAXLOAD

⎤

−

VV

FBOUT

⎥

V

FB

⎦

)(

VVV

−×

IN

OUT

LfV

××

SW

VVV

−××

×

OUT

OUT

OUT

Δ

2

I

IV

LOAD

IN

⎞
⎟
⎠

OUT

IN

⎛

+=

⎜
⎝

)(MAXLOAD

3

)(5.2

OUT

)(MAX

H

, determined by the

PK

Ensure that the maximum rms current of the inductor is greater
than the maximum load current, and the saturation current of
the inductor is greater than the peak current limit of the converter
used in the application.

Table 5. Minimum Inductor Value for Common Output
Voltage Options for the ADP2105 (1 A)

VIN

V

OUT

2.7 V 3.6 V 4.2 V 5.5 V

1.2 V 1.67 µH 2.00 µH 2.14 µH 2.35 µH

1.5 V 1.68 µH 2.19 µH 2.41 µH 2.73 µH

1.8 V 2.02 µH 2.25 µH 2.57 µH 3.03 µH

2.5 V 2.80 µH 2.80 µH 2.80 µH 3.41 µH

3.3 V 3.70 µH 3.70 µH 3.70 µH 3.70 µH

Table 6. Minimum Inductor Value for Common Output
Voltage Options for the ADP2106 (1.5 A)

VIN

V

OUT

,

L

1.2 V 1.11 µH 2.33 µH 2.43 µH 1.56 µH

1.5 V 1.25 µH 1.46 µH 1.61 µH 1.82 µH

2.7 V 3.6 V 4.2 V 5.5 V

1.8 V 1.49 µH 1.50 µH 1.71 µH 2.02 µH

2.5 V 2.08 µH 2.08 µH 2.08 µH 2.27 µH

3.3 V 2.74 µH 2.74 µH 2.74 µH 2.74 µH

Table 7. Minimum Inductor Value for Common Output
Voltage Options for the ADP2107 (2 A)

VIN

V

OUT

2.7 V 3.6 V 4.2 V 5.5 V

1.2 V 0.83 µH 1.00 µH 1.07 µH 1.17 µH

1.5 V 0.99 µH 1.09 µH 1.21 µH 1.36 µH

1.8 V 1.19 µH 1.19 µH 1.29 µH 1.51 µH

2.5 V 1.65 µH 1.65 µH 1.65 µH 1.70 µH

3.3 V 2.18 µH 2.18 µH 2.18 µH 2.18 µH

Table 8. Inductor Recommendations for the ADP2105/
ADP2106/ADP2107

Vendo r

Sumida

Toko

Small-Sized Inductors
( < 5 mm × 5 mm)

CDRH2D14, 3D16,
3D28

1069AS-DB3018,
1098AS-DE2812,

Large-Sized Inductors
( > 5 mm × 5 mm)

CDRH4D18, 4D22,
4D28, 5D18, 6D12

D52LC, D518LC,
D62LCB

1070AS-DB3020

Coilcraft

LPS3015, LPS4012,

DO1605T

DO3314

Cooper
Bussmann

SD3110, SD3112,
SD3114, SD3118,

SD10, SD12, SD14, SD52

SD3812, SD3814

Rev. A | Page 17 of 32

ADP2105/ADP2106/ADP2107

OUTPUT CAPACITOR SELECTION

The output capacitor selection affects both the output voltage
ripple and the loop dynamics of the converter. For a given loop
crossover frequency (the frequency at which the loop gain
drops to 0 dB), the maximum voltage transient excursion
(overshoot) is inversely proportional to the value of the output
capacitor. Therefore, larger output capacitors result in improved
load transient response. To minimize the effects of the dc-to-dc
converter switching, the crossover frequency of the compensation
loop should be less than 1/10 of the switching frequency. Higher
crossover frequency leads to faster settling time for a load transient
response, but it can also cause ringing due to poor phase
margin. Lower crossover frequency helps to provide stable
operation but needs large output capacitors to achieve competitive
overshoot specifications. Therefore, the optimal crossover
frequency for the control loop of ADP2105/ADP2106/ADP2107
is 80 kHz, 1/15 of the switching frequency. For a crossover
frequency of 80 kHz,
voltage excursion during a 1A load transient, as the product of
the output voltage and the output capacitor is varied. Choose
the output capacitor based on the desired load transient
response and target output voltage.

18
17
16
15
14
13
12
11
10

9
8
7
6
5
4
3

% OVERSHOOT OF OUTPUT VOLTAGE

2
1
0

15 70

Figure 39. % Overshoot for a 1 A Load Transient Response vs.

For example, if the desired 1 A load transient response (overshoot)
is 5% for an output voltage of 2.5 V, then from

Output Capacitor × Output Voltage = 50 C

The ADP2105/ADP2106/ADP2107 have been designed for
operation with small ceramic output capacitors that have low
ESR and ESL, thus are comfortably able to meet tight output
voltage ripple specifications. X5R or X7R dialectrics are
recommended with a voltage rating of 6.3 V or 10 V. Y5V and
Z5U dialectrics are not recommended, due to their poor
temperature and dc bias characteristics.
recommended MLCC capacitors from Murata and Taiyo Yuden.

Figure 39 shows the maximum output

20 25 30 35 40 45 50 55 60 65

OUTPUT CAPACIT OR × OUTPUT VOLT AGE (C)

Output Capacitor × Output Voltage

Figure 39

C50

≈=⇒ CapacitorOutput

5.2

F20

Tabl e 9 shows a list of

06079-070

When choosing output capacitors it is also important to account
for the loss of capacitance due to output voltage dc bias.

Figure 40
shows the loss of capacitance due to output voltage dc bias for a
few X5R MLCC capacitors from Murata.

20

0

–20

–40

–60

CAPACITANCE CHANGE (%)

–80

1

4.7µF 0805 X5R MURATA GRM21BR61A475K

2

10µF 0805 X5R MURATA G RM21BR61A106K

3

22µF 0805 X5R MURATA G RM21BR60J226M

–100

0

Figure 40. % Drop-In Capacitance vs. DC Bias for Ceramic Capacitors

(Information Provided by Murata Corporation)

246

VOLTAGE (VDC)

3

1

2

06079-060

For example, to get 20 µF output capacitance at an output voltage
of 2.5 V, based on

Figure 40, as well as give some margin for
temperature variance, it is suggested that a 22 F and a 10 F
capacitor be used in parallel to ensure that the output capacitance
is sufficient under all conditions for stable behavior.

Table 9. Recommended Input and Output Capacitor Selection
for the ADP2105/ADP2106/ADP2107

Vendo r

Capacitor

4.7 µF 10 V

Murata Taiyo Yuden

GRM21BR61A475K LMK212BJ475KG

X5R 0805
10 F 10 V

GRM21BR61A106K LMK212BJ106KG

X5R 0805
22 F 6.3 V

GRM21BR60J226M JMK212BJ226MG

X5R 0805

INPUT CAPACITOR SELECTION

The input capacitor reduces input voltage ripple caused by the
switch currents on the PWIN pins. Place the input capacitors as
close as possible to the PWIN pins. Select an input capacitor
capable of withstanding the rms input current for the maximum
load current in your application.

For the ADP2105, it is recommended that each PWIN pin be
bypassed with a 4.7 F or larger input capacitor. For the ADP2106,
bypass the PWIN pins with a 10 F and a 4.7 F capacitor, and
for the ADP2107, bypass each PWIN pin with a 10 F capacitor.

As with the output capacitor, a low ESR ceramic capacitor is
recommended to minimize input voltage ripple. X5R or X7R
dialectrics are recommended, with a voltage rating of 6.3 V or
10 V. Y5V and Z5U dialectrics are not recommended, due to
their poor temperature and dc bias characteristics. Refer to
Tabl e 9 for input capacitor recommendations.

Rev. A | Page 18 of 32

ADP2105/ADP2106/ADP2107

INPUT FILTER

The IN pin is the power source for the ADP2105/ADP2106/
ADP2107 internal circuitry, including the voltage reference and
current sense amplifier that are sensitive to power supply noise.
To prevent high frequency switching noise on the PWIN pins from
corrupting the internal circuitry of the ADP2105/ADP2106/
ADP2107, a low-pass RC filter should be placed between the IN
pin and the PWIN1 pin. The suggested input filter consists of
a small 0.1 F ceramic capacitor placed between IN and AGND
and a 10  resistor placed between IN and PWIN1. This forms
a 150 kHz low-pass filter between PWIN1 and IN that prevents
any high frequency noise on PWIN1 from coupling into the
IN pin.

SOFT START

The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect a soft start capacitor
(C

) from SS to AGND. The soft start period varies linearly

SS

with the size of the soft start capacitor, as shown in the
following equation:

= CSS × 109 ms

T

SS

To get a soft start period of 1 ms, a 1 nF capacitor must be
connected between SS and AGND.

LOOP COMPENSATION

The ADP2105/ADP2106/ADP2107 utilize a transconductance
error amplifier to compensate the external voltage loop. The
open loop transfer function at angular frequency, s, is given by

⎞

⎛

COMP

OUT

⎞

sZ

V

REF

⎟

⎜

⎟

⎟

⎜

⎟

V

OUT

⎠

⎝

⎠

⎛
⎜

GGsH)()(

=

m

CS

⎜

sC

⎝

where:

is the internal reference voltage (0.8 V).

V

REF

V

is the nominal output voltage.

OUT

(s) is the impedance of the compensation network at the

Z

COMP

angular frequency, s.

is the output capacitor.

C

OUT

G

is the transconductance of the error amplifier (50 A/V

m

nominal).

is the effective transconductance of the current loop.

G

CS

= 1.875 A/V for the ADP2105.

G

CS

G

= 2.8125 A/V for the ADP2106.

CS

= 3.625 A/V for the ADP2107.

G

CS

The transconductance error amplifier drives the compensation
network that consists of a resistor (R

) and capacitor (C

COMP

COMP

)
connected in series to form a pole and a zero, as shown in the
following equation:

⎛

sC

1

COMP

⎞

+

1

⎜

⎟

=

⎜

⎟
⎠

sC

COMP

⎝

⎛
⎜

+=

RsZ

)(

COMPCOMP

⎜
⎝

⎞

CsR

COMPCOMP

⎟
⎟
⎠

At the crossover frequency, the gain of the open loop transfer
function is unity. This yields the following equation for the
compensation network impedance at the crossover frequency:

⎛

π

⎜

=

)(

CROSSCOMP

⎜
⎝

m

FFZ)2(

CROSS

GG

⎞

⎛

⎟

⎜
⎜

⎟

CS

⎝

⎠

⎞

VC

OUTOUT

⎟
⎟

V

REF

⎠

where:

= 80 kHz, the crossover frequency of the loop.

F

CROSS

C

is determined from the Output Capacitor Selection

OUTVOUT

section.

To ensure that there is sufficient phase margin at the crossover
frequency, place the Compensator Zero at 1/4 of the crossover
frequency, as shown in the following equation:

F

⎞

⎛

CROSS

)π2( =

⎜

4

⎝

CR

⎟
⎠

COMPCOMP

1

Solving the three equations above simultaneously yields the
value for the compensation resistor and compensation
capacitor, as shown in the following equation:

COMP

COMP

⎛
⎜

8.0

=

⎜
⎝

=

2

RFCπ

FR)π2(

m

COMPCROSS

CROSS

GG

CS

⎞

⎛

⎟

⎜
⎜

⎟

V

⎝

⎠

VC

REF

⎞

OUTOUT

⎟
⎟
⎠

Rev. A | Page 19 of 32

ADP2105/ADP2106/ADP2107

BODE PLOTS

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAG E = 1. 8V

–10

INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A

–20

INDUCTOR = 2.2µH (L PS4012)
OUTPUT CAPACI TOR = 22µF + 22µ F

–30

COMPENSATION RESISTOR = 180k
COMPENSATI ON CAPACIT OR = 56pF

–40

1 300

Figure 41. ADP2106 Bode Plot at V

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAGE = 1.8V

–10

INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A

–20

INDUCTOR = 2.2µH (L PS4012)
OUTPUT CAPACITOR = 22µF + 22µF

–30

COMPENSATION RESISTOR = 180k
COMPENSATI ON CAPACIT OR = 56pF

–40

1 300

Figure 42. ADP2106 Bode Plot at V

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAGE = 1.2V

–10

INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A

–20

INDUCTOR = 3.3µH (S D3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF

–30

COMPENSATION RESISTOR = 267k
COMPENSATI ON CAPACIT OR = 39p F

–40

1 300

Figure 43. ADP2105 Bode Plot at VIN = 3.6 V, V

LOOP GAIN

PHASE

MARGIN = 48°

LOOP PHASE

CROSSOVER

FREQUENCY = 87kHz

10 100

NOTES

1. EXTE RNAL COMPO NENTS W ERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LO AD TRANSIENT.

LOOP GAIN

LOOP PHASE

NOTES

1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.

LOOP GAIN

LOOP PHASE

NOTES

1. EXTE RNAL COMPO NENTS W ERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LO AD TRANSIENT.

(kHz)

= 5.5 V, V

IN

CROSSOVER

FREQUENCY = 83kHz

10 100

(kHz)

= 3.6 V, V

IN

CROSSOVER

FREQUENCY = 71kHz

10 100

(kHz)

= 1.8 V and Load = 1 A

OUT

PHASE

MARGIN = 52°

= 1.8 V, and Load = 1 A

OUT

PHASE

MARGIN = 51°

= 1.2 V, and Load = 1 A

OUT

ADP2106

ADP2106

ADP2105

0

45

90

135

180

0

45

90

135

180

0

45

90

135

180

LOOP P HASE (Degrees)

06079-055

LOOP PHASE (Degrees)

06079-056

LOOP P HASE (Degrees)

06079-057

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAGE = 1.2V

–10

INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A

–20

INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF

–30

COMPENSATION RESI STOR = 267k
COMPENSATI ON CAPACIT OR = 39p F

–40

1 300

Figure 44. ADP2105 Bode Plot at V

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAG E = 2. 5V

–10

INPUT VOLTAGE = 5V
LOAD CURRENT = 1A

–20

INDUCTOR = 2µH (D62LCB)
OUTPUT CAPACITOR = 10µF + 4.7µF

–30

COMPENSATION RESISTOR = 70k
COMPENSATI ON CAPACIT OR = 120pF

–40

1 300

Figure 45. ADP2107 Bode Plot at V

60

50

40

30

20

10

0

LOOP GAIN (dB)

OUTPUT VOLTAG E = 3. 3V

–10

INPUT VOLTAGE = 5V
LOAD CURRENT = 1A

–20

INDUCTOR = 2.5µH (CDRH5D28)
OUTPUT CAPACI TOR = 10µF + 4. 7µF

–30

COMPENSATION RESISTOR = 70k
COMPENSATION CAPACITOR = 120pF

–40

1 300

Figure 46. ADP2107 Bode Plot at VIN = 5 V, V

LOOP GAIN

PHASE

MARGIN = 49°

LOOP PHASE

CROSSOVER

FREQUENCY = 79kHz

10 100

NOTES

1. EXTE RNAL COMPO NENTS W ERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LO AD TRANSIENT.

LOOP GAIN

LOOP PHASE

NOTES

1. EXTE RNAL COMPO NENTS W ERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LO AD TRANSIENT .

LOOP GAIN

LOOP PHASE

NOTES

1. EXTE RNAL COMPO NENTS W ERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LO AD TRANSIENT .

(kHz)

= 5.5 V, V

IN

CROSSOVER

FREQUENCY = 76kHz

10 100

(kHz)

= 5 V, V

IN

CROSSOVER

FREQUENCY = 67kHz

10 100

(kHz)

= 1.2 V and Load = 1 A

OUT

PHASE

MARGIN = 65°

= 2.5 V and Load = 1 A

OUT

PHASE

MARGIN = 70°

= 3.3 V, and Load = 1 A

OUT

ADP2105

ADP2107

ADP2107

0

45

90

135

180

0

45

90

135

180

0

45

90

135

180

LOOP P HASE (Degrees)

06079-058

LOOP P HASE (Degrees)

06079-059

LOOP PHAS E (Degrees)

06079-069

Rev. A | Page 20 of 32

ADP2105/ADP2106/ADP2107

LOAD TRANSIENT RESPONSE

T

3

OUTPUT CURRENT

T

3

OUTPUT CURRENT

OUTPUT VO LTAGE (AC-COUPLED)

2

1

CH3 1.00A 

OUTPUT CAPACI TOR: 22µ F + 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESIS TOR: 270k
COMPENSATI ON CAPACITO R: 39pF

LX NODE (SWITCH NODE)

CH2 100mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

Figure 47. 1 A Load Transient Response for ADP2105-1.2

with External Components Chosen for 5% Overshoot

T

3

2

OUTPUT CURRENT

OUTPUT VO LTAGE (AC-COUPLED)

OUTPUT VO LTAGE ( AC-COUPLED)

2

1

LX NODE (SW ITCH NODE)

CH3 1.00A 

OUTPUT CAPACI TOR: 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 135k

06079-087

COMPENSATI ON CAPACITOR: 82pF

CH2 100mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

06079-090

Figure 50. 1 A Load Transient Response for ADP2105-1.2

with External Components Chosen for 10% Overshoot

T

3

OUTPUT VO LTAGE ( AC-COUPLED)

2

OUTPUT CURRENT

1

CH3 1.00A 

OUTPUT CAPACI TOR: 22µ F + 22µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESIS TOR: 270k
COMPENSATI ON CAPACITO R: 39pF

LX NODE (SWITCH NODE)

CH2 100mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

Figure 48. 1 A Load Transient Response for ADP2105-1.8

with External Components Chosen for 5% Overshoot

T

3

OUTPUT VO LTAGE ( AC-COUPLED)

2

1

CH3 1.00A 

OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: CDRH5D18, 4. 1µH
COMPENSATION RESISTOR: 270k
COMPENSATION CAPACITOR: 39pF

OUTPUT CURRENT

LX NODE (SW ITCH NODE)

CH2 200mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

Figure 49. 1 A Load Transient Response for ADP2105-3.3

with External Components Chosen for 5% Overshoot

06079-088

06079-089

Rev. A | Page 21 of 32

1

CH3 1.00A 

OUTPUT CAPACI TOR: 10µF + 10µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 135k
COMPENSATI ON CAPACITOR: 82pF

LX NODE (SW ITCH NODE)

CH2 100mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

Figure 51. 1 A Load Transient Response for ADP2105-1.8

with External Components Chosen for 10% Overshoot

T

3

OUTPUT VO LTAGE (AC-COUPLED)

2

1

CH3 1.00A 

OUTPUT CAPACI TOR: 10µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 135k
COMPENSATI ON CAPACITOR: 82pF

OUTPUT CURRENT

LX NOD E (SWI TCH NO DE)

CH2 200mV~CH1 2.00V

M 20.0µs A CH3 700mA

T 10.00%

Figure 52. 1 A Load Transient Response for ADP2105-3.3

with External Components Chosen for 10% Overshoot

06079-091

06079-092

ADP2105/ADP2106/ADP2107

EFFICIENCY CONSIDERATIONS

Efficiency is defined as the ratio of output power to input power.
The high efficiency of the ADP2105/ADP2106/ADP2107 has
two distinct advantages. First, only a small amount of power is
lost in the dc-to-dc converter package that reduces thermal
constraints. In addition, high efficiency delivers the maximum
output power for the given input power, extending battery life
in portable applications.

There are four major sources of power loss in dc-to-dc
converters like the ADP2105/ADP2106/ADP2107:

• Power switch conduction losses

•

Inductor losses

•

Switching losses

•

Transition losses

Power Switch Conduction Losses

Power switch conduction losses are caused by the flow of output
current through the P-channel power switch and the N-channel
synchronous rectifier, which have internal resistances (R
associated with them. The amount of power loss can be approxi­mated by

P

SW − COND

where D = V

= [R

OUT/VIN

DS(ON) − P

.

× D + R

DS(ON) − N

× (1 − D)] × I

The internal resistance of the power switches increases with
temperature but decreases with higher input voltage.

Figure 20 show the change in RDS(ON) vs. input voltage,

and
whereas

Figure 28 and Figure 29 show the change in R

Figure 19

DS(ON)

temperature for both power devices.

Inductor Losses

Inductor conduction losses are caused by the flow of current
through the inductor, which has an internal resistance (DCR)
associated with it. Larger sized inductors have smaller DCR,
which can improve inductor conduction losses.

Inductor core losses are related to the magnetic permeability of
the core material. Because the ADP2105/ADP2106/ADP2107
are high switching frequency dc-to-dc converters, shielded ferrite
core material is recommended for its low core losses and low EMI.

The total amount of inductor power loss can be calculated by

= DCR × I

P

L

2

+ Core Losses

OUT

Switching Losses

Switching losses are associated with the current drawn by the
driver to turn on and turn off the power devices at the
switching frequency. Each time a power device gate is turned on
and turned off, the driver transfers a charge Q from the input
supply to the gate and then from the gate to ground.

The amount of power loss can by calculated by

P

SW

= (C

GATE − P

+ C

GATE − N

) × V

IN

2

× fSW

where:
(C

f

+ C

GATE − P

= 1.2 MHz, the switching frequency.

SW

GATE − N

) ≈ 600 pF.

)

DS(ON)

2

OUT

vs.

Rev. A | Page 22 of 32

Transition Losses

Transition losses occur because the P-channel MOSFET power
switch cannot turn on or turn off instantaneously. At the middle of
an LX node transition, the power switch is providing all the
inductor current, while the source to drain voltage of the power
switch is half the input voltage, resulting in power loss.
Transition losses increase with load current and input voltage
and occur twice for each switching cycle.

The amount of power loss can be calculated by

V

P OFFON ×+××= )(

TRAN

where t

ON

and t

IN

OUT

2

are the rise time and fall time of the LX node,

OFF

fttI

SW

and are both approximately 3 ns.

THERMAL CONSIDERATIONS

In most applications, the ADP2105/ADP2106/ADP2107 do not
dissipate a lot of heat due to their high efficiency. However, in
applications with high ambient temperature, low supply voltage,
and high duty cycle, the heat dissipated in the package is large
enough that it can cause the junction temperature of the die to
exceed the maximum junction temperature of 125°C. Once the
junction temperature exceeds 140°C, the converter goes into
thermal shutdown. It recovers only after the junction temperature
has decreased below 100°C to prevent any permanent damage.
Therefore, thermal analysis for the chosen application solution
is very important to guarantee reliable performance over all
conditions.

The junction temperature of the die is the sum of the ambient
temperature of the environment and the temperature rise of the
package due to the power dissipation, as shown in the following
equation:

T

= TA + TR

J

where:

T

is the junction temperature.

J

T

is the ambient temperature.

A

is the rise in temperature of the package due to power

T

R

dissipation in it.

The rise in temperature of the package is directly proportional
to the power dissipation in the package. The proportionality
constant for this relationship is defined as the thermal
resistance from the junction of the die to the ambient
temperature, as shown in the following equation:

= θJA × PD

T

R

where:

T

is the rise in temperature of the package.

R

P

is the power dissipation in the package.

D

is the thermal resistance from the junction of the die to the

θ

JA

ambient temperature of the package.

For example, consider an application where the ADP2107-1.8
is used with an input voltage of 3.6 V and a load current of 2 A.
Also, assume that the maximum ambient temperature is 85

°C.

ADP2105/ADP2106/ADP2107

−××

At a load current of 2 A, the most significant contributor of
power dissipation in the dc-to-dc converter package is the
conduction loss of the power switches. Using the graph of
switch resistance vs. temperature (see
equation of power loss given in the

section, the power dissipation in the package can be

Losses

Figure 29), as well as the

Power Switch Conduction

calculated by

P

SW − COND

= [R

DS(ON) − P

× D + R

DS(ON) − N

× (1 − D)] × I

[109 m × 0.5 + 90 m × 0.5] × (2 A)

The

θ

for the LFCSP_VQ package is 40°C/W, as shown in

JA

2

≈ 400 mW

OUT

2

=

Tabl e 3 . Thus, the rise in temperature of the package due to
power dissipation is

= θJA × PD = 40°C/W × 0.40 W = 16°C

T

R

The junction temperature of the converter is

T

= TA + TR = 85°C + 16°C = 101°C

J

which is below the maximum junction temperature of 125°C. Thus,
this application operates reliably from a thermal point of view.

DESIGN EXAMPLE

Consider an application with the following specifications:

Input Voltage = 3.6 V to 4.2 V.
Output Voltage = 2 V.
Typical Output Current = 600 mA.
Maximum Output Current = 1.2 A.
Soft Start Time = 2 ms.
Overshoot ≤ 100 mV under all load transient conditions.

Choose the dc-to-dc converter that satisfies the maximum

1.
output current requirement. Because the maximum output
current for this application is 1.2 A, the ADP2106 with a
maximum output current of 1.5 A is ideal for this
application.

See whether the output voltage desired is available as a

2.
fixed output voltage option. Because 2 V is not one of the
fixed output voltage options available, choose the adjustable
version of ADP2106.

3. The first step in external component selection for an

adjustable version converter is to calculate the resistance of
the resistive voltage divider that sets the output voltage.

OUT

V8.0

20

AI

⎤

−

VV

FB

⎥

V

FB

⎦

OUT

Ω=== k40

⎡

k40

×Ω=

⎢
⎢

⎣

⎤

V8.0V2

−
⎥

V8.0

⎥

⎦

Rev. A | Page 23 of 32

V

R

BOT

FB

STRING

⎡

= k60

RR

⎢

BOTTOP

⎣

Calculate the minimum inductor value as follows:

For the ADP2106:

L > (0.83 H/V) × V

L > 0.83 H/V × 2 V

⇒

L > 1.66 H

⇒

Ω=

Next, calculate the ideal inductor value that sets the
inductor peak-to-peak current ripple, I

, to 1/3 of the

L

maximum load current at the maximum input voltage.

−××

)(5.2

VVV

IN

= H

L

IDEAL

OUT

×

IV

IN

LOAD

)22.4(25.2

2.12.4

×

=

OUT

)(MAX

H2.18H

=

The closest standard inductor value is 2.2 H. The
maximum rms current of the inductor should be greater
than 1.2 A, and the saturation current of the inductor
should be greater than 2 A. One inductor that meets these
criteria is the LPS4012-2.2 H from Coilcraft.

4. Choose the output capacitor based on the transient

response requirements. The worst-case load transient is

1.2 A, for which the overshoot must be less than 100 mV,
which is 5% of the output voltage. Therefore, for a 1 A load
transient, the overshoot must be less than 4% of the output
voltage. For these conditions,

Figure 39 gives

Output Capacitor × Output Voltage = 60 C

C60

V0.2

F30

≈=⇒ CapacitorOutput

Next, taking into account the loss of capacitance due to dc
bias, as shown in

Figure 40, two 22 F X5R MLCC capacitors
from Murata (GRM21BR60J226M) are sufficient for this
application.

Because the ADP2106 is being used in this application, the

5.
input capacitors are 10 F and 4.7 F X5R Murata capacitors
(GRM21BR61A106K and GRM21BR61A475K).

The input filter consists of a small 0.1 F ceramic capacitor

6.
placed between IN and AGND and a 10  resistor placed
between IN and PWIN1.

Choose a soft start capacitor of 2 nF to achieve a soft start

7.
time of 2 ms.

Finally, the compensation resistor and capacitor can be

8.
calculated as

m

COMPCROSS

FR)π2(

CROSS

GG

kHz80)π2(

⎞

⎛

⎟

⎜
⎜

⎟

CS

V

⎝

⎠

⎞

⎛

⎟

⎜

⎟

⎜

V/A8125.2V/A50

⎠

⎝

==

VC

REF

⎞

OUTOUT

⎟

=

⎟
⎠

⎞

V2F30

×

⎟
⎟

V8.0

⎠

2

××

k215kHz80π

k215

Ω=

=

pF39

COMP

⎛
⎜

8.0

⎜
⎝

C

COMP

⎜
⎝

×
×

2

π

RF

⎛
⎜

8.0

=

ADP2105/ADP2106/ADP2107

EXTERNAL COMPONENT RECOMMENDATIONS

Table 10. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
10% Overshoot for a 1 A Load Transient (Refer to

Part V

(V) C

OUT

1

(μF) C

IN1

ADP2105-ADJ 0.9 4.7 4.7 22 + 10 2.0 135 82 5 40
ADP2105-ADJ 1.2 4.7 4.7 22 + 4.7 2.5 135 82 20 40
ADP2105-ADJ 1.5 4.7 4.7 10 + 10 3.0 135 82 35 40
ADP2105-ADJ 1.8 4.7 4.7 10 + 10 3.3 135 82 50 40
ADP2105-ADJ 2.5 4.7 4.7 10 + 4.7 3.6 135 82 85 40
ADP2105-ADJ 3.3 4.7 4.7 10 + 4.7 4.1 135 82 125 40
ADP2106-ADJ 0.9 4.7 10 22 + 10 1.5 90 100 5 40
ADP2106-ADJ 1.2 4.7 10 22 + 4.7 1.8 90 100 20 40
ADP2106-ADJ 1.5 4.7 10 10 + 10 2.0 90 100 35 40
ADP2106-ADJ 1.8 4.7 10 10 + 10 2.2 90 100 50 40
ADP2106-ADJ 2.5 4.7 10 10 + 4.7 2.5 90 100 85 40
ADP2106-ADJ 3.3 4.7 10 10 + 4.7 3.0 90 100 125 40
ADP2107-ADJ 0.9 10 10 22 + 10 1.2 70 120 5 40
ADP2107-ADJ 1.2 10 10 22 + 4.7 1.5 70 120 20 40
ADP2107-ADJ 1.5 10 10 10 + 10 1.5 70 120 35 40
ADP2107-ADJ 1.8 10 10 10 + 10 1.8 70 120 50 40
ADP2107-ADJ 2.5 10 10 10 + 4.7 1.8 70 120 85 40
ADP2107-ADJ 3.3 10 10 10 + 4.7 2.5 70 120 125 40
ADP2105-1.2 1.2 4.7 4.7 22 + 4.7 2.5 135 82 – –
ADP2105-1.5 1.5 4.7 4.7 10 + 10 3.0 135 82 – –
ADP2105-1.8 1.8 4.7 4.7 10 + 10 3.3 135 82 – –
ADP2105-3.3 3.3 4.7 4.7 10 + 4.7 4.1 135 82 – –
ADP2106-1.2 1.2 4.7 10 22 + 4.7 1.8 90 100 – –
ADP2106-1.5 1.5 4.7 10 10 + 10 2.0 90 100 – –
ADP2106-1.8 1.8 4.7 10 10 + 10 2.2 90 100 – –
ADP2106-3.3 3.3 4.7 10 10 + 4.7 3.0 90 100 – –
ADP2107-1.2 1.2 10 10 22 + 4.7 1.5 70 120 – –
ADP2107-1.5 1.5 10 10 10 + 10 1.5 70 120 – –
ADP2107-1.8 1.8 10 10 10 + 10 1.8 70 120 – –
ADP2107-3.3 3.3 10 10 10 + 4.7 2.5 70 120 – –

1

4.7 F 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 F 0805 X5R 10 V Murata–GRM21BR61A106KE19L.

2

4.7 F 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 F 0805 X5R 10 V Murata–GRM21BR61A106KE19L.

3

4.7 F 0805 X5R 10 V Murata–GRM21BR61A475KA73L. 10 F 0805 X5R 10 V Murata–GRM21BR61A106KE19L. 22 F 0805 X5R 6.3 V Murata–GRM21BR60J226ME39L.

4

0.5% accuracy resistor.

5

0.5% accuracy resistor.

Figure 37 and Figure 38)

2

(μF) C

IN2

OUT

3

(μF) L (μH) R

(kΩ) C

COMP

(pF) R

COMP

4

(kΩ) R

TOP

BOT

5

(kΩ)

Rev. A | Page 24 of 32

ADP2105/ADP2106/ADP2107

Table 11. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
5% Overshoot for a 1 A Load Transient (Refer to Figure 37 and Figure 38)

Part V

(V) C

OUT

1

(μF) C

IN1

ADP2105-ADJ 0.9 4.7 4.7 22 + 22 + 22 2.0 270 39 5 40
ADP2105-ADJ 1.2 4.7 4.7 22 + 22 + 4.7 2.5 270 39 20 40
ADP2105-ADJ 1.5 4.7 4.7 22 + 22 3.0 270 39 35 40
ADP2105-ADJ 1.8 4.7 4.7 22 + 22 3.3 270 39 50 40
ADP2105-ADJ 2.5 4.7 4.7 22 + 10 3.6 270 39 85 40
ADP2105-ADJ 3.3 4.7 4.7 22 + 4.7 4.1 270 39 125 40
ADP2106-ADJ 0.9 4.7 10 22 + 22 + 22 1.5 180 56 5 40
ADP2106-ADJ 1.2 4.7 10 22 + 22 + 4.7 1.8 180 56 20 40
ADP2106-ADJ 1.5 4.7 10 22 + 22 2.0 180 56 35 40
ADP2106-ADJ 1.8 4.7 10 22 + 22 2.2 180 56 50 40
ADP2106-ADJ 2.5 4.7 10 22 + 10 2.5 180 56 85 40
ADP2106-ADJ 3.3 4.7 10 22 + 4.7 3.0 180 56 125 40
ADP2107-ADJ 0.9 10 10 22 + 22 + 22 1.2 140 68 5 40
ADP2107-ADJ 1.2 10 10 22 + 22 + 4.7 1.5 140 68 20 40
ADP2107-ADJ 1.5 10 10 22 + 22 1.5 140 68 35 40
ADP2107-ADJ 1.8 10 10 22 + 22 1.8 140 68 50 40
ADP2107-ADJ 2.5 10 10 22 + 10 1.8 140 68 85 40
ADP2107-ADJ 3.3 10 10 22 + 4.7 2.5 140 68 125 40
ADP2105-1.2 1.2 4.7 4.7 22 + 22 + 4.7 2.5 270 39 – –
ADP2105-1.5 1.5 4.7 4.7 22 + 22 3.0 270 39 – –
ADP2105-1.8 1.8 4.7 4.7 22 + 22 3.3 270 39 – –
ADP2105-3.3 3.3 4.7 4.7 22 + 4.7 4.1 270 39 – –
ADP2106-1.2 1.2 4.7 10 22 + 22 + 4.7 1.8 180 56 – –
ADP2106-1.5 1.5 4.7 10 22 + 22 2.0 180 56 – –
ADP2106-1.8 1.8 4.7 10 22 + 22 2.2 180 56 – –
ADP2106-3.3 3.3 4.7 10 22 + 4.7 3.0 180 56 – –
ADP2107-1.2 1.2 10 10 22 + 22 + 4.7 1.5 140 68 – –
ADP2107-1.5 1.5 10 10 22 + 22 1.5 140 68 – –
ADP2107-1.8 1.8 10 10 22 + 22 1.8 140 68 – –
ADP2107-3.3 3.3 10 10 22 + 4.7 2.5 140 68 – –

1

4.7μF 0805 X5R 10V Murata–GRM21BR61A475KA73L. 10μF 0805 X5R 10V Murata–GRM21BR61A106KE19L.

2

4.7μF 0805 X5R 10V Murata–GRM21BR61A475KA73L. 10μF 0805 X5R 10V Murata–GRM21BR61A106KE19L.

3

4.7μF 0805 X5R 10V Murata–GRM21BR61A475KA73L. 10μF 0805 X5R 10V Murata–GRM21BR61A106KE19L. 22μF 0805 X5R 6.3V Murata–GRM21BR60J226ME39L

4

0.5% accuracy resistor.

5

0.5% accuracy resistor.

2

(μF) C

IN2

3

(μF) L (μH) R

OUT

COMP

(kΩ) C

(pF) R

COMP

4

(kΩ) R

TOP

BOT

5

(kΩ)

Rev. A | Page 25 of 32

ADP2105/ADP2106/ADP2107

CIRCUIT BOARD LAYOUT RECOMMENDATIONS

Make the high current path from the PGND pin of the

Good circuit board layout is essential to obtaining the best
performance from the ADP2105/ADP2106/ADP2107. Poor
circuit layout degrades the output ripple, as well as the
electromagnetic interference (EMI) and electromagnetic
compatibility (EMC) performance.

Figure 54 and Figure 55 show the ideal circuit board layout for
the ADP2105/ADP2106/ADP2107. Use this layout to achieve
the highest performance. Refer to the following guidelines if
adjustments to the suggested layout are needed:

•

Use separate analog and power ground planes. Connect

the ground reference of sensitive analog circuitry (such as
compensation and output voltage divider components) to
analog ground; connect the ground reference of power
components (such as input and output capacitors) to power
ground. In addition, connect both the ground planes to the
exposed pad of the ADP2105/ADP2106/ADP2107.

• For each PWIN pin, place an input capacitor as close to the

PWIN pin as possible and connect the other end to the closest
power ground plane.

•

Place the 0.1 F, 10  low-pass input filter between the IN

pin and the PWIN1 pin, as close to the IN pin as possible.

Ensure that the high current loops are as short and as wide

•

as possible. Make the high current path from C

, and the PGND plane back to CIN as short as possible.

L, C

OUT

through

IN

To accomplish this, ensure that the input and output
capacitors share a common PGND plane.

•

ADP2105/ADP2106/ADP2107 through L and C

back to

OUT

the PGND plane as short as possible. To do this, ensure
that the PGND pin of the ADP2105/ADP2106/ ADP2107
is tied to the PGND plane as close as possible to the input
and output capacitors.

•

Place the feedback resistor divider network as close as

possible to the FB pin to prevent noise pickup. Try to
minimize the length of trace connecting the top of the
feedback resistor divider to the output while keeping away
from the high current traces and the switch node (LX) that
can lead to noise pickup. To reduce noise pickup, place an
analog ground plane on either side of the FB trace. For the
low fixed voltage options (1.2 V and 1.5 V), poor routing
of the OUT_SENSE trace can lead to noise pickup, adversely
affecting load regulation. This can be fixed by placing a 1 nF
bypass capacitor close to the OUT_SENSE pin.

•

The placement and routing of the compensation components

are critical for proper behavior of the ADP2105/ADP2106/
ADP2107. The compensation components should be placed
as close to the COMP pin as possible. It is advisable to use
0402-sized compensation components for closer placement,
leading to smaller parasitics. Surround the compensation
components with analog ground plane to prevent noise
pickup. Also, ensure that the metal layer under the
compensation components is the analog ground plane.

Rev. A | Page 26 of 32

ADP2105/ADP2106/ADP2107

EVALUATION BOARD

EVALUATION BOARD SCHEMATIC (ADP2107-1.8V)

C7

0.1µF

VCC

OUT

EN

100k

R2

J1 U1

16 15 14 13

OUT_SENSE

1

EN

2

GND

ADP2107-1.8

3

GND

4

GND

SS

AGNDCOMP

5 6 7 8

R1

140k

68pF

C6

C5
1nF

Figure 53. Evaluation Board Schematic of the ADP2107-1.8 (Bold Traces are High Current Paths)

VCC

R3

10

10µF

PWIN1INGND

12

LX2

11

PGND

10

LX1

9

PWIN2

NCPADDLE

NC = NO CONNECT

INPUT VOLTAGE = 2.7V TO 5.5V

C1

1

VCC

C2
10µF

VIN

GND

2

L1
2µH

21

R4

NS

0

R5

OUT

1

1

MURATA X5R 0805

10F: GRM21BR61A106KE19L
22F: GRM21BR60J226ME39L

2

2H INDUCTOR D62LCB TO KO

OUTPUT VOLTAGE = 1.8V, 2A

22µF

C3

1

C4
22µF

1

V

OUT

GND

06079-044

RECOMMENDED PCB BOARD LAYOUT (EVALUATION BOARD LAYOUT)

JUMPER TO ENABLE

ENABLE

100k PULL-DOWN

INPUT CAPACI TOR

PLACE THE FEEDBACK RESISTORS AS

CLOSE TO THE FB PI N AS POSSIBLE.

R

TOPRBOT

ADP2105/ADP2106/ADP2107

R

COMP

C

COMP

PLACE THE COMPENSATION
COMPONENTS AS CLOSE TO
THE COMP PIN AS POSSIBLE.

ANALOG GROUND PLANE

CONNECT THE G ROUND RETURN OF ALL
SENSITIVE ANALOG CIRCUITRY SUCH AS
COMPENSATION AND OUTPUT VOLTAGE
DIVIDER TO THE ANALOG GROUND PLANE.

V

IN

INPUT

C

SS

INPUT CAPACITOR

C

IN

LX

PGND

LX

C

IN

POWER GROUND

GROUND

POWER GRO UND

PLANE

C

OUT

INDUCTOR (L)

C

OUT

CONNECT THE G ROUND RETURN OF
ALL POWER COMPO NENTS SUCH AS
INPUT AND OUTPUT CAPACITO RS TO
THE POWER GROUND PLANE.

OUTPUT CAP ACITOR

OUTPUT CAPACITOR

GROUND

OUTPUT

V

OUT

6079-045

Figure 54. Recommended Layout of Top Layer of ADP2105/ADP2106/ADP2107

Rev. A | Page 27 of 32

ADP2105/ADP2106/ADP2107

ENABLE

ANALOG GROUND PLANE

CONNECT THE EX POSED PAD OF
THE ADP2105/ADP2106/ADP2107
TO A LARGE GROUND PLANE TO
AID POWER DISSIPATION.

V

IN

V

IN

GND

POWER GROUND PLANE

INPUT VOLTAGE PLANE
CONNECTING THE TW O
PWIN PI NS AS CLOSE
AS POSSIBLE.

CONNECT THE PGND PIN
TO THE POWER GROUND
PLANE AS CLO SE TO THE
ADP2105/ADP2106/ADP2107
AS POSSIBLE.

GND

V

OUT

FEEDBACK TRACE: T HIS TRACE CONNECTS THE TOP OF THE
RESISTIV E VOLTAGE DIVIDER O N THE FB PIN TO THE OUTPUT.
PLACE THI S TRACE AS FAR AWAY FROM T HE LX NO DE AND HIGH
CURRENT TRACES AS POSSI BLE TO PREV ENT NOISE PICKUP.

Figure 55. Recommended Layout of Bottom Layer of ADP2105/ADP2106/ADP2107

06079-046

Rev. A | Page 28 of 32

ADP2105/ADP2106/ADP2107

APPLICATION CIRCUITS

0.1F

V

OUT

16 15 14 13

OUT_SENSE

ON

1

OFF

EN

2

GND

ADP2107-3.3

3

GND

4

GND

SS

COMP

70k

AGND

5 6 7 8

1nF

120pF

Figure 56. Application Circuit—V

0.1F

V

OUT

16 15 14 13

OUT_SENSE

ON

1

OFF

EN

2

GND

ADP2107-1.5

3

GND

4

GND

SS

COMP

140k

AGND

5 6 7 8

1nF

68pF

Figure 57. Application Circuit—V

0.1F

V

OUT

16 15 14 13

OUT_SENSE

ON

1

OFF

EN

2

GND

ADP2105-1.8

3

GND

4

GND

SS

COMP

270k

AGND

5 6 7 8

1nF

39pF

Figure 58. Application Circuit—V

VININPUT VOLTAGE = 5V

10

1

10F

PWIN1INGND

12

LX2

2.5H

11

PGND

10

LX1

V

IN

9

PWIN2

NC

VININPUT VOLTAGE = 3.6V

10

10F

= 5 V, V

IN

10F

1

1

PWIN1INGND

12

LX2

1.5H

11

PGND

10

LX1

V

IN

9

PWIN2

NC

VININPUT VOLTAGE = 2.7V TO 4.2V

10

1

10F

= 3.6 V, V

IN

4.7F

1

PWIN1INGND

12

LX2

2.7H

11

PGND

10

LX1

V

IN

9

PWIN2

NC

IN

1

4.7F

= Li-Ion Battery, V

Rev. A | Page 29 of 32

2

V

OUTPUT VOLTAGE = 3.3V

OUT

1

10F

1

MURATA X5R 0805
10F: GRM21BR61A106KE19L

4.7F: GRM21BR61A475KA73L

2

SUMIDA CD RH5D28: 2.5 H

NOTES

1. NC = NO CO NNECT.

2. EXTERNAL CO MPONENTS WE RE
CHOSEN FOR A 10% OVERSHOOT
FORA1ALOADTRANSIENT.

= 3.3 V, Load = 0 A to 2 A

OUT

2

1

22F

1

MURATA X5R 0805
10F: GRM21BR61A106KE19L
22F: GRM21BR60J226ME39L

2

TOKO D62LCB OR COI LCRAFT LPS4012

NOTES

1. NC = NO CO NNECT.

2. EXTERNAL CO MPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FORA1ALOADTRANSIENT.

= 1.5 V, Load = 0 A to 2 A

OUT

2

1

22F

1

MURATA X5R 0805

4.7F: GRM21BR61A475KA73L
22F: GRM21BR60J226ME39L

2

TOKO 1098AS-DE2812: 2. 7H

NOTES

1. NC = NO CO NNECT.

2. EXTERNAL CO MPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FORA1ALOADTRANSIENT.

OUT

1

4.7F

V

OUTPUT VOLTAGE = 1.5V

OUT

1

22F

V

OUTPUT VOLTAGE = 1.8V

OUT

1

22F

LOAD
0A TO 2A

LOAD
0A TO 2A

LOAD
0A TO 1A

= 1.8 V, Load = 0 A to 1 A

06079-047

06079-048

06079-049

ADP2105/ADP2106/ADP2107

0.1F

V

OUT

16 15 14 13

OUT_SENSE

ON

1

OFF

EN

2

GND

ADP2105-1.2

3

GND

4

GND

SS

COMP

135k

AGND

5 6 7 8

1nF

82pF

Figure 59. Application Circuit—V

0.1F

FB

16 15 14 13

OFF

ON

FB PWIN1

1

EN

2

GND

ADP2106-ADJ

3

GND

4

GND

COMP

5 6 7 8

180k

56pF

Figure 60. Application Circuit—V

GND

SS

IN

AGND

1nF

VININPUT VOLTAGE = 2.7V TO 4.2V

10

4.7F

PWIN1INGND

12

LX2

11

PGND

10

LX1

V

IN

9

PWIN2

NC

VININPUT VOLTAGE = 5V

10

LX2

4.7F

= Li-Ion Battery, V

IN

1

10F

12

2.5H

11

PGND

10

LX1

V

IN

9

PWIN2

NC

4.7F

= 5 V, V

IN

1

1

2

2.4H

1

V

OUTPUT VOLTAGE = 1.2V

OUT

1

22F

1

MURATA X5R 0805

4.7F: GRM21BR61A475KA73L
22F: GRM21BR60J226ME39L

2

TOKO 1069AS-DB3018HCT OR
TOKO 1070AS-DB3020HCT

NOTES

1. NC = NO CONNECT.

2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSI ENT.

OUT

2

85k

1

4.7F

= 1.2 V, Load = 0 A to 1 A

OUTPUT VOLTAGE = 2.5V

10F122F

FB

40k

1

MURATA X5R 08 05

4.7F: G RM21BR61A475KA73L

10F: GRM21BR61A106KE 19L
22F: GRM21BR60J226ME39L

2

COILTRONICS SD14: 2.5H

NOTES

1. NC = NO CONNECT.

2. EXTE RNAL COMPO NENTS W ERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSI ENT.

= 2.5 V, Load = 0 A to 1.5 A

OUT

LOAD
0A TO 1A

1

LOAD
0A TO 1.5A

06079-050

06079-051

Rev. A | Page 30 of 32

ADP2105/ADP2106/ADP2107

OUTLINE DIMENSIONS

PIN 1

INDICATOR

1.00

0.85

0.80

12° MAX

SEATING
PLANE

4.00

BSC SQ

TOP

VIEW

0.80 MAX

0.65 TYP

0.35

0.30

0.25

3.75

BSC SQ

0.20 REF

0.60 MAX

0.65 BSC

0.05 MAX

0.02 NOM

COPLANARITY

0.75

0.60

0.50

0.08

0.60 MAX

(BOTTOM VIEW)

13

12

9

8

16

5

1.95 BSC

PIN 1
INDICATOR

1

4

5

2

.

2

S

0

1

.

2

9

.

1

5

0.25 MIN

Q

ORDERING GUIDE

Model

ADP2105ACPZ-1.2-R7
ADP2105ACPZ-1.5-R7
ADP2105ACPZ-1.8-R7
ADP2105ACPZ-3.3-R7
ADP2105ACPZ-R7
ADP2106ACPZ-1.2-R7
ADP2106ACPZ-1.5-R7
ADP2106ACPZ-1.8-R7
ADP2106ACPZ-3.3-R7
ADP2106ACPZ-R7
ADP2107ACPZ-1.2-R7
ADP2107ACPZ-1.5-R7
ADP2107ACPZ-1.8-R7
ADP2107ACPZ-3.3-R7
ADP2107ACPZ-R7
ADP2105-1.8-EVALZ
ADP2105-EVALZ
ADP2106-1.8-EVALZ
ADP2106-EVALZ
ADP2107-1.8-EVALZ
ADP2107-EVALZ

1

Z = RoHS Compliant Part.

1

1

1

1

1

1

1

1

1

COMPLIANT TO JEDEC STANDARDS MO-220-VGG C

021207-A

Figure 61. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]

4 mm × 4 mm Body, Very Thin Quad

(CP-16-4)

Dimensions shown in millimeters

Output
Current

1

1 A −40°C to +125°C 1.2 V 16-Lead LFCSP_VQ CP-16-4

1

1 A −40°C to +125°C 1.5 V 16-Lead LFCSP_VQ CP-16-4

1

1 A −40°C to +125°C 1.8 V 16-Lead LFCSP_VQ CP-16-4

1

1 A −40°C to +125°C 3.3 V 16-Lead LFCSP_VQ CP-16-4

Temperature
Range

Output Voltage Package Description Package Option

1 A −40°C to +125°C ADJ 16-Lead LFCSP_VQ CP-16-4

1

1.5 A −40°C to +125°C 1.2 V 16-Lead LFCSP_VQ CP-16-4

1

1.5 A −40°C to +125°C 1.5 V 16-Lead LFCSP_VQ CP-16-4

1

1.5 A −40°C to +125°C 1.8 V 16-Lead LFCSP_VQ CP-16-4

1

1.5 A −40°C to +125°C 3.3 V 16-Lead LFCSP_VQ CP-16-4

1.5 A −40°C to +125°C ADJ 16-Lead LFCSP_VQ CP-16-4

1

2 A −40°C to +125°C 1.2 V 16-Lead LFCSP_VQ CP-16-4

1

2 A −40°C to +125°C 1.5 V 16-Lead LFCSP_VQ CP-16-4

1

2 A −40°C to +125°C 1.8 V 16-Lead LFCSP_VQ CP-16-4

1

2 A −40°C to +125°C 3.3 V 16-Lead LFCSP_VQ CP-16-4
2 A −40°C to +125°C ADJ 16-Lead LFCSP_VQ CP-16-4

1.8 V Evaluation Board
Adjustable, but set to 2.5 V Evaluation Board

1.8 V Evaluation Board
Adjustable, but set to 2.5 V Evaluation Board

1.8 V Evaluation Board
Adjustable, but set to 2.5 V Evaluation Board

Rev. A | Page 31 of 32

ADP2105/ADP2106/ADP2107

NOTES

©2006–2007 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06079-0-3/07(A)

Rev. A | Page 32 of 32