Datasheet ADP1110 Datasheet (Analog Devices)

Micropower, Step-Up/Step-Down Switching

a

Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V

FEATURES
Operates at Supply Voltages From 1.0 V to 30 V
Step-Up or Step-Down Mode
Minimal External Components Required
Low-Battery Detector
User-Adjustable Current Limiting
Fixed or Adjustable Output Voltage Versions
8-Pin DIP or SO-8 Package

APPLICATIONS
Cellular Telephones
Single-Cell to 5 V Converters
Laptop and Palmtop Computers
Pagers
Cameras
Battery Backup Supplies
Portable Instruments
Laser Diode Drivers
Hand-Held Inventory Computers

ADP1110

FUNCTIONAL BLOCK DIAGRAMS

SET

ADP1110

V

IN

220mV

REFERENCE

R1

GND

A2

GAIN BLOCK/
ERROR AMP

A1

COMPARATOR

R2

300kΩ

SENSE

OSCILLATOR

DRIVER

ADP1110 Block Diagram—Fixed Output Version

SET

A0

I

SW1

Q1

SW2

LIM

GENERAL DESCRIPTION

The ADP1110 is part of a family of step-up/step-down switch­ing regulators that operate from an input voltage supply as little
as 1.0 V. This very low input voltage allows the ADP1110 to be
used in applications that use a single cell as the primary power
source.

The ADP1110 can be configured to operate in either step-up or
step-down mode, but for input voltages greater than 3 V, the
ADP1111 would be a more effective solution.

An auxiliary gain amplifier can serve as a low battery detector or
as a linear regulator.

The quiescent current of 300 µA makes the ADP1110 useful in
remote or battery powered applications.

ADP1110

V

IN

220mV

REFERENCE

GND

A2

GAIN BLOCK/
ERROR AMP

A1

COMPARATOR

FB

OSCILLATOR

DRIVER

Q1

SW2

A0

I

LIM

SW1

ADP1110 Block Diagram—Adjustable Output Version

The 70 kHz frequency operation also allows for the use of
surface-mount external capacitors and inductors.

Battery protection circuitry limits the effect of reverse current to
safe levels at reverse voltages up to 1.6 V.

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 1996

ADP1110–SPECIFICATIONS

(08C to +708C, V

Parameter Conditions V

QUIESCENT CURRENT Switch Off I

INPUT VOLTAGE Step-Up Mode V

= 1.5 V unless otherwise noted)

IN

S

Q

IN

Min Typ Max Units

300 µA

1.15 12.6 V

Step-Down Mode 30 V

COMPARATOR TRIP POINT VOLTAGE ADP1110

OUTPUT SENSE VOLTAGE ADP1110-3.3

ADP1110-5
ADP1110-12

1

2

2

2

V

OUT

210 220 230 mV

3.13 3.30 3.47 V

4.75 5.00 5.25 V

11.4 12.00 12.6 V

COMPARATOR HYSTERESIS ADP1110 4 8 mV

OUTPUT HYSTERESIS ADP1110-3.3 66 130 mV

ADP1110-5 90 180 mV
ADP1110-12 200 400 mV

OSCILLATOR FREQUENCY f

DUTY CYCLE Full Load (VFB < V

)DC626978%

REF

SWITCH ON TIME t

FEEDBACK PIN BIAS CURRENT ADP1110 VFB = 0 V I

SET PIN BIAS CURRENT V

A0 OUTPUT LOW I

REFERENCE LINE REGULATION 1.0 V ≤ V

= V

SET

REF

= 300 µAV

AO

V

= 150 mV

SET

≤ 1.5 V 0.35 %/V

IN

OSC

ON

FB

I

SET

AO

52 70 90 kHz

7.5 10 12.5 µs

150 240 nA

300 500 nA

0.15 0.4 V

1.5 V ≤ VIN ≤ 12 V 0.05 0.1 %/V

SWITCH SATURATION VOLTAGE V

STEP-UP MODE T

A2 ERROR AMP GAIN R

REVERSE BATTERY CURRENT T

CURRENT LIMIT TEMPERATURE V

= 1.5 V, I

IN

to T

MIN

V

= 1.5 V, I

IN

T

to T

MIN

VIN = 5 V, I

= 100 kΩ

L

= +25°C

A

= +25°C –0.3 %/°C

IN, TA

= 400 mA, +25°CV

SW

MAX

= 500 mA, +25°C 400 650 mV

SW

MAX

= 1 A, +25°C 700 1000 mV

SW

3

4

CESAT

A

V

I

REV

1000 5000 V/V

300 500 mV

600 mV

750 mV

750 mA

COEFFICIENT

SWITCH OFF LEAKAGE CURRENT Measured at SW1 Pin, I

T

= +25°C

A

MAXIMUM EXCURSION BELOW GND I

NOTES

1

This specification guarantees that both the high and low trip point of the comparator fall within the 210 mV to 230 mV range.

2

This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a saw­tooth shape due to the comparator hysteresis.

3

100 kΩ resistor connected between a 5 V source and the AO pin.

4

The ADP1110 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while V

5

All limits at temperature extremes are guaranteed via correlation using standard statistical quality control methods.

Specifications subject to change without notice.

≤ 10 µA, Switch Off V

SW1

T

= +25°C

A

LEAK

SW2

, I

IN

110µA

–400 –350 mV

, and SW1 pins are grounded.

LIM

–2–

REV. 0

ADP1110

WARNING!

ESD SENSITIVE DEVICE

A0

I

LIM

SW1

GND

V

IN

SW2

*FIXED VERSIONS

FB (SENSE)*
SET

1
2
3
4

8
7
6
5

ADP1110

TOP VIEW

(Not to Scale)

T

JMAX

= 90o, θ

JA

= 130oC/W

ABSOLUTE MAXIMUM RATINGS

Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V

Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V

SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V

SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . –0.5 V to V

IN

Feedback Pin Voltage (ADP1110) . . . . . . . . . . . . . . . . . . 5.5 V

Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A

Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW

Operating Temperature Range . . . . . . . . . . . . . 0°C to +70°C

Storage Temperature Range . . . . . . . . . . . . . –65°C to 150°C

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300°C

TYPICAL APPLICATION

47µH

1 2

V

I

IN

LIM

1.5V
AA CELL*

OPERATES WITH CELL VOLTAGE ≥1.0V
*ADD 10µF DECOUPLING CAPACITOR IF BATTERY IS

*MORE THAN 2' AWAY FROM ADP1110.

ADP1110-5

GND

5

SW1

SENSE

SW2

4

3

8

5V

15µF
TANTALUM

Figure 1. 1.5 V to 5 V Converter

ORDERING GUIDE

Model Output Voltage Package

ADP1110AN ADJ N-8
ADP1110AR ADJ SO-8
ADP1110AN-3.3 3.3 V N-8
ADP1110AR-3.3 3.3 V SO-8
ADP1110AN-5 5 V N-8
ADP1110AR-5 5 V SO-8
ADP1110AN-12 12 V N-8
ADP1110AR-12 12 V SO-8

PIN CONFIGURATIONS

8-Lead Plastic DIP

(N-8)

8-Lead SOIC

I

1

LIM

ADP1110

2

V

IN

TOP VIEW

3

SW1
SW2

(Not to Scale)

4

= 90o, θ

T

JMAX

*FIXED VERSIONS

(SO-8)

= 150oC/W

JA

8

FB (SENSE)*

7

SET

6

A0

5

GND

PIN DESCRIPTION

Mnemonic Function

I

LIM

For normal conditions this pin is connected to
V

. When lower current is required, a resistor

IN

should be connected between I

and VIN.

LIM

Limiting the switch current to 400 mA is
achieved by connecting a 220 Ω resistor.

V

IN

Input Voltage.

SW1 Collector Node of Power Transistor. For step-

down configuration, connect to V

. For step-

IN

up configuration, connect to an inductor/diode.

SW2 Emitter Node of Power Transistor. For step-

down configuration, connect to inductor/diode.
For step-up configuration, connect to ground.
Do not allow this pin to go more than a diode

drop below ground.
GND Ground.
AO Auxiliary Gain (GB) Output. The open collec-

tor can sink 300 µA. It can be left open if unused.
SET Gain Amplifier Input. The amplifier has posi-

tive input connected to SET pin and negative

input connected to 220 mV reference. It can be

left open if unused.
FB/SENSE On the ADP1110 (adjustable) version this pin

is connected to the comparator input. On the

ADP1110-3.3, ADP1110-5 and ADP1110-12,

the pin goes directly to the internal application

resistor that set output voltage.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1110 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

REV. 0

–3–

ADP1110-Typical Characteristics

R

LIM

– Ω

1.9

1.7

0.1
1 100010 100

1.5

1.3

0.5

1.1

0.9

0.7

0.3

SWITCH CURRENT – A

STEP-DOWN WITH
V = +12V

R

LIM

– Ω

0.1
1 100010 100

1.5

1.3

0.5

1.1

0.9

0.7

0.3

STEP-UP MODE
WITH V ≤ +5V

SWITCH CURRENT – A

1.4

V

1.2

V

I

SWITCH

IN

= +1.5V

CURRENT – A

1

V

= +1.2V

0.8

0.6

0.4

SATURATION VOLTAGE – V

0.2

0

0.1 1.40.2 0.4 0.5 0.6 0.8 1 1.2 1.25

IN

VIN = +1V

IN

= +2V

V

IN

V

= +5V

IN

= +3V

76

74

72

70

68

66

64

OSCILLATOR FREQUENCY – kHz

62

60

2304 6 8 1012151821 2427

OSCILLATOR FREQUENCY

INPUT VOLTAGE – V

Figure 2. Saturation Voltage vs. I
Mode

2

1.8

1.6

1.4

1.2

1

0.8

ON VOLTAGE – V

0.6

0.4

0.2
0

0.1 0.90.2 0.4 0.6 0.8

Figure 3. Switch ON Voltage vs. I
Down Mode

1800

1600

1400

1200

1000

800

600

QUIESCENT CURRENT – µA

400

200

0

1303 6 9 12 15 18 21 24 27

Figure 4. Quiescent Current vs. Input Voltage

SWITCH

VIN = +12V

I

CURRENT – A

SWITCH

SWITCH

QUIESCENT CURRENT

INPUT VOLTAGE – V

Current in Step-Up

Current In Step-

–4–

Figure 5. Oscillator Frequency vs. Input Voltage

Figure 6. Maximum Switch Current vs. R

Figure 7. Maximum Switch Current vs. R

LIM

LIM

REV. 0

ADP1110

TEMPERATURE – 8C

160

120

07025

145

140

135

125

155

150

BIAS CURRENT

BIAS CURRENT – nA

130

78

76

74

72

70

68

OSCILLATOR FREQUENCY – kHz

66

64

07025

OSCILLATOR FREQUENCY

TEMPERATURE – 8C

Figure 8. Oscillator Frequency vs. Temperature

9.0

8.9

8.8

8.7

8.6

8.5

ON TIME – µs

8.4

8.3

8.2
07025

TEMPERATURE – 8C

SWITCH ON TIME

0.54

0.53

0.52

0.51

– V

0.5

CE(SAT)

V

0.49

0.48

0.47
07025

VIN = +1.5 @ I

TEMPERATURE – 8C

SWITCH

= +0.5A

Figure 11. Switch ON Voltage Step-Down vs. Temperature

350

300

QUIESCENT CURRENT

250

200

150

100

QUIESCENT CURRENT – µA

50

0

07025

TEMPERATURE – 8C

Figure 9. Switch ON Time vs. Temperature

66

65

64

63

62

DUTY CYCLE – %

61

60

59

07025

DUTY CYCLE

TEMPERATURE – 8C

Figure 10. Duty Cycle vs. Temperature

Figure 12. Quiescent Current vs. Temperature

Figure 13. FB Pin Bias Current vs. Temperature

REV. 0

–5–

ADP1110

V

REF

– mV

TEMPERATURE – 8C

220

219

211

07025

215

214

213

212

217

216

218

REFERENCE VOLTAGE

400

350

300

250

200

150

BIAS CURRENT – nA

100

50

0

07025

BIAS CURRENT

TEMPERATURE – 8C

Figure 14. Set Pin Bias Current vs. Temperature

THEORY OF OPERATION

The ADP1110 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance with
low quiescent current.

A functional block diagram of the ADP1110 is shown on the
first page. The internal 220 mV reference is connected to one
input of the comparator, while the other input is externally
connected (via the FB pin) to a feedback network connected to
the regulated output. When the voltage at the FB pin falls below
220 mV, the 70 kHz oscillator turns on. A driver amplifier provides
base drive to the internal power switch, and the switching action
raises the output voltage. When the voltage at the FB pin exceeds
220 mV, the oscillator is shut off. While the oscillator is off, the
ADP1110 quiescent current is only 300 µA. The comparator
includes a small amount of hysteresis, which ensures loop
stability without requiring external components for frequency
compensation.

The maximum current in the internal power switch can be set
by connecting a resistor between V

and the I

IN

pin. When the

LIM

maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 800 ns. If an
external resistor is not used, connect I
tion on I

is included in the “Applications” section of this data

LIM

to VIN. Further informa-

LIM

sheet.
The ADP1110 internal oscillator provides 10 µs ON and 5 µs

OFF times, which is ideal for applications where the ratio between
V

and V

IN

is roughly a factor of three (such as generating +5 V

OUT

from a single 1.5 V cell). Wider range conversions, as well as
step-down converters, can also be accomplished with a slight
loss in the maximum output power that can be obtained.

An uncommitted gain block on the ADP1110 can be connected
as a low–battery detector. The inverting input of the gain block
is internally connected to the 220 mV reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between V
pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300 µA.

The ADP1110 provides external connections for both the
collector and emitter of its internal power switch, which permits

and GND with the junction connected to the SET

IN

Figure 15. Reference Voltage vs. Temperature

both step-up and step-down modes of operation. For the step­up mode, the emitter (Pin SW2) is connected to GND and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to V

.

IN

The output voltage of the ADP1110 is set with two external
resistors. Three fixed-voltage models are also available:
ADP1110–3.3 (+3.3 V), ADP1110–5 (+5 V) and ADP1110-12
(+12 V). The fixed-voltage models are identical to the
ADP1110 except that laser-trimmed voltage-setting resistors are
included on the chip. Only three external components are
required to form a +3.3 V, +5 V or +12 V converter. On the
fixed-voltage models of the ADP1110, simply connect the
SENSE pin (Pin 8) directly to the output voltage.

COMPONENT SELECTION
General Notes on Inductor Selection

When the ADP1110 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is then
transferred to the load when the switch turns off. Because both
the collector and the emitter of the switch transistor are
accessible on the ADP1110, the output voltage can be higher,
lower, or of opposite polarity than the input voltage.

To specify an inductor for the ADP1110, the proper values of
inductance, saturation current, and DC resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both V

IN

and
switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1110 applications, an inductor of
15 µH to 100 µH with a saturation current rating of 300 mA to
1A and dc resistance <0.4 Ω is suitable. Ferrite-core inductors
that meet these specifications are available in small, surface­mount packages.

To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.

–6–

REV. 0

ADP1110

CALCULATING THE INDUCTOR VALUE

Selecting the proper inductor value is a simple three-step
process:

1. Define the operating parameters: minimum input voltage,
maximum input voltage, output voltage and output current.

2. Select the appropriate conversion topology (step-up, step­down, or inverting).

3. Calculate the inductor value, using the equations in the
following sections.

INDUCTOR SELECTION–STEP-UP CONVERTER

In a step-up or boost converter (Figure 19), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:

PL= V

where V

OUT+VD

()

is the diode forward voltage (≈ 0.5 V for a 1N5818

D

−V

IN MIN

()

•I

()

OUT

(Equation 1)

Schottky). Because energy is only stored in the inductor while
the ADP1110 switch is ON, the energy stored in the inductor
on each switching cycle must be must be equal to or greater
than:

P

L

f

OSC

(Equation 2)

in order for the ADP1110 to regulate the output voltage.
When the internal power switch turns ON, current flow in the

inductor increases at the rate of:

IL(t)=



V

IN



R'



1−e



L





(Equation 3)

–R't

where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8 Ω at +25°C) and the dc resistance of
the inductor. If the voltage drop across the switch is small
compared to V

IL(t)=

, a simpler equation can be used:

IN

V

IN

t

L

(Equation 4)

Replacing ‘t’ in the above equation with the ON time of the
ADP1110 (10 µs, typical) will define the peak current for a
given inductor value and input voltage. At this point, the
inductor energy can be calculated as follows:

1
2

L •I

2

PEAK

must be greater than PL/f

L

(Equation 5)

so

OSC

EL=

As previously mentioned, E
that the ADP1110 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current also
increases output ripple. As a general rule, keep peak current as low
as possible to minimize losses in the switch, inductor and diode.

In practice, the inductor value is easily selected using the equations
above. For example, consider a supply that will generate 12 V at
120 mA from a 4.5 V to 8 V source. The inductor power required
is from Equation 1:

PL= 12V + 0.5V − 4.5 V

()

•120 mA = 960 mW

On each switching cycle, the inductor must supply:

P

960 mW

L

=

f

70 kHz

OSC

=13.7µJ

Assuming a peak current of 1 A as a starting point, (Equation 4)
can be rearranged to recommend an inductor value:

L =

V

I

L(MAX )

4.5V

IN

t =

10 µs =45µH

1 A

Substituting a standard inductor value of 47 µH with 0.2 Ω dc
resistance will produce a peak switch current of:

–1.0 Ω•10 µs

I

PEAK

=

4.5V

1. 0 Ω






1 − e

47 µH



=862 mA




Once the peak current is known, the inductor energy can be
calculated from Equation 5:

1

EL=

47 µH

()

2

•862 mA

()

Since the inductor energy of 17.5 µJ is greater than the PL/f

2

=17.5µJ

OSC

requirement of 13.7 µJ, the 47 µH inductor will work in this
application. By substituting other inductor values into the same
equations, the optimum inductor value can be determined.
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A.

The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when V
maximum value of V

is at its minimum, the 1.5 A limit may be exceeded at the

IN

. In this case, the ADP1110’s current limit

IN

feature can be used to limit switch current. Simply select a resistor
(using Figure 7) that will limit the maximum switch current to the
I

value calculated for the minimum value of VIN. This will

PEAK

improve efficiency by producing a constant I

as VIN increases.

PEAK

See the “Limiting the Switch Current” section of this data sheet for
more information.

Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.

INDUCTOR SELECTION–STEP-DOWN CONVERTER

The step-down mode of operation is shown in Figure 20.
Unlike the step-up mode, the ADP1110’s power switch does not

saturate when operating in the step-down mode; therefore,
switch current should be limited to 800 mA in this mode. If the
input voltage will vary over a wide range, the I

pin can be

LIM

used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 22.

The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:

I

PEAK



OUT



VIN–VSW+V



V

OUT+VD

2 I

=

DC






D

(Equation 6)

where: DC = duty cycle (0.69 for the ADP1110)

= voltage drop across the switch

V

SW

= diode drop (0.5 V for a 1N5818)

V

D

= output current

I

OUT

= the output voltage

V

OUT

= the minimum input voltage

V

IN

REV. 0

–7–

ADP1110

As previously mentioned, the switch voltage is higher in step­down mode than in step-up mode. V
current and is therefore a function of V
For most applications, a V

value of 1.5 V is recommended.

SW

is a function of switch

SW

, L, time and V

IN

OUT

.

The inductor value can now be calculated:

V

IN(MIN )–VSW–VOUT

L =

where: t

I

PEAK

= Switch ON time (10 µs)

ON

•t

ON

(Equation 7)

If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an R
should be selected from Figure 6. The R

LIM

resistor

LIM

resistor will keep
switch current constant as the input voltage rises. Note that
there are separate R

values for step-up and step-down modes

LIM

of operation.
For example, assume that +5 V at 250 mA is required from a

+9 V to +18 V source. Deriving the peak current from Equation
6 yields:

I

PEAK

2•250 mA

=

0.69



5 +0.5



9 −1. 5 + 0.5





= 498 mA





Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:

9–1.5–5

L =

498 mA

•10 µs = 50 µs

Since 50 µH is not a standard value, the next lower standard
value of 47 µH would be specified.

To avoid exceeding the maximum switch current when the
input voltage is at +18 V, an R

resistor should be specified.

LIM

Using the step-down curve of Figure 6, a value of 560 Ω will
limit the switch current to 500 mA.

INDUCTOR SELECTION—POSITIVE-TO-NEGATIVE
CONVERTER

The configuration for a positive-to-negative converter using the
ADP1110 is shown in Figure 23. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:

P = I

VV

+

•

()

L OUT

OUT D

()

(Equation 8)

The ADP1110 power switch does not saturate in positive-to­negative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω
resistor. When the switch turns on, inductor current will rise at
a rate determined by:

–R't



V

R'

L





IL(t)=

where: R' = 0.65 Ω + R

1−e



L





L(DC)

(Equation 9)

VL = VIN – 0.75 V

For example, assume that a –5 V output at 75 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:

PL= |– 5V|+0.5V

()

•75mA

()

=413 mW

During each switching cycle, the inductor must supply the
following energy:

P

413 mW

L

=

f

70 kHz

OSC

=5. 9 µJ

Using a standard inductor value of 56 µH with 0.2 Ω dc
resistance will produce a peak switch current of:

–0.85 Ω•10µs

I

PEAK

4.5V –0.75V

=

0.65 Ω+0.2 Ω



1−e




56 µH



=621 mA




Once the peak current is known, the inductor energy can be
calculated from Equation 9:

1

EL=

56 µH

()

2

•621mA

()

Since the inductor energy of 10.8 µJ is greater than the PL/f

2

=10.8µJ

OSC

requirement of 5.9 µJ, the 56 µH inductor will work in this
application.

The input voltage only varies between 4.5 V and 5.5 V in this
example. Therefore, the peak current will not change enough to
require an R
directly to V

resistor and the I

LIM

. Care should be taken, of course, to ensure that

IN

pin can be connected

LIM

the peak current does not exceed 800 mA.

CAPACITOR SELECTION

For optimum performance, the ADP1110’s output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.

Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.

The effects of capacitor selection on output ripple are demon­strated in Figures 16, 17 and 18. These figures show the output
of the same ADP1110 converter, that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100 µF. Figure 16
shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18 Ω. In Figure 17,
the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps

–8–

REV. 0

ADP1110

I

PEAK

≅

V

IN–VCE(SAT )

L

•10 µs

I

LIM

V

IN

SW1

SW2

FB

GND

ADP1110

L1

C

1

R1

R2

D1

8

1 2

3

4

5

V

IN

V

OUT

R3*

*OPTIONAL

about 30 mV, which indicates an ESR of 0.06 Ω. Figure 18
shows an OS-CON 16–volt capacitor in the same circuit, and
ESR is only 0.02 Ω.

50mV

100

90

10

0%

1µs

Figure 16. Aluminum Electrolytic

50mV

100

90

10

0%

1µs

Figure 17. Tantalum Electrolytic

50mV

100

90

important in low-current applications, where the leakage can be
a significant percentage of the total quiescent current.

For most circuits, the 1N5818 is a suitable companion to the
ADP1110. This diode has a V

of 0.5 V at 1 A, 4 µA to 10 µA

F

leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130LT3, is also available.

For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a V

of 0.8 V at 100 mA and leakage less

F

than 1 µA. A similar device, the BAT54, is available in a SOT23
package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.

General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1110 circuits. These devices, which have turn-on times
of 10 µs or more, are too slow for switching power supply
applications. Using such a diode “just to get started” will result
in wasted time and effort. Even if an ADP1110 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.

CIRCUIT OPERATION, STEP-UP (BOOST) MODE

In boost mode, the ADP1110 produces an output voltage that is
higher than the input voltage. For example, +5 V can be derived
from one alkaline cell (+1.5 V), or +12 V can be generated from
a +5 V logic power supply.

Figure 19 shows an ADP1110 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to V
V

and current begins to flow through L1. This current

CE(SAT),

–

IN

reaches a final value (ignoring second-order effects) of:

10

0%

1µs

Figure 18. OS-CON Capacitor

If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.

DIODE SELECTION

In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1110 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so V
Schottky diodes are recommended. Leakage current is especially

REV. 0

must also be minimized. Again,

F

–9–

where 10 µs is the ADP1110 switch’s “on” time.

Figure 19. Step-Up Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.

The output voltage is fed back to the ADP1110 via resistors R1
and R2. When the voltage at pin FB falls below 220 mV, SW1

ADP1110

I

LIM

V

IN

SW1

SW2

FB

GNDSETAO

ADP1110

NC NC

L1

C

L

R1

R2

D1

1N5821

OUTPUT

C

INPUT

R

LIM

INPUT

6

7

8

1

2

45

3

R4

220Ω

R3

330Ω

MJE210

0.3Ω

R5

turns “on” again, and the cycle repeats. The output voltage is
therefore set by the formula:





V

OUT

=220 mV • 1+

R1





R2





The circuit of Figure 19 shows a direct current path from VIN to
V

, via the inductor and D1. Therefore, the boost converter

OUT

is not protected if the output is short circuited to ground.

CIRCUIT OPERATION, STEP-DOWN (BUCK) MODE

The ADP1110’s step-down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.

A typical configuration for step-down operation of the ADP1110 is
shown in Figure 20. In this case, the collector of the internal
power switch is connected to V

and the emitter drives the

IN

inductor. When the switch turns on, SW2 is pulled up towards
V

. This forces a voltage across L1 equal to VIN – VCE – V

IN

and causes current to flow in L1. This current reaches a final
value of:

V

I

PEAK

IN−VCE−VOUT

≅

L

•10 µs

where 10 µs is the ADP1110 switch’s “on” time.

V

IN

R

LIM

C

100Ω

2

2

1 3

V

I

IN

LIM

ADP1110

SETAO

75

6

NC

NC

SW1

GND

SW2

FB

8

4

L1

D1
1N5818

V

OUT

R1

C

1

R2

Figure 20. Step-Down Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1110’s SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. Using a silicon diode in this application will generate
forward voltages above 0.5 V that will cause potentially
damaging power dissipation within the ADP1110.

The output voltage of the buck regulator is fed back to the
ADP1110’s FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 220 mV, the internal power switch turns
“on” again and the cycle repeats. The output voltage is set by
the formula:





V

=220 mV • 1+

OUT

R1





R2





When operating the ADP1110 in step-down mode, the output
voltage is impressed across the internal power switch’s emitter-

OUT

base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2, as shown in Figure 21.

INPUT

C

INPUT

R

LIM

1 2 3

VINSW1

I

LIM

ADP1110

SETAO

7

6

NC

NC

GND

4

SW2

8

FB

5

L1

D1
1N5818

Figure 21. Step-Down Mode, V

OUT

R1

R2

> 6.2 V

C

L

OUTPUT

If the input voltage to the ADP1110 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maxi­mum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the “Limiting the Switch
Current” section of this data sheet for specific recommendations.

INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR

Unlike the boost configuration, the ADP1110’s internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1110 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure

22). In this circuit, the ADP1110 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1110’s internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1110, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 22, the switch voltage can be calculated from the
formula:

VSW=VR5+V

≅0.6V + 0.4 V ≅1V

Q1( SAT )

Figure 22. High Current Step-Down Operation

–10–

REV. 0

ADP1110

10

0%

100

90

10µs10mV

200mA/div.

POSITIVE-TO-NEGATIVE CONVERSION

The ADP1110 can convert a positive input voltage to a negative
output voltage as shown in Figure 23. This circuit is essentially
identical to the step-down application of Figure 19, except that
the “output” side of the inductor is connected to power ground.
When the ADP1110’s internal power switch turns off, current
flowing in the inductor forces the output (–V

) to a negative

OUT

potential. The ADP1110 will continue to turn the switch on
until its FB pin is 220 mV above its GND pin, so the output
voltage is determined by the formula:





R1





R2





L1

R1

D1

R2

OUTPUTC

C

L

NEGATIVE
OUTPUT

INPUT

INPUT

R

LIM

V

NC

V

=220 mV • 1+

OUT

1 2

I

LIM

IN

ADP1110

7

6

NC

SW1

SW2

GNDSETAO

FB

3

5

4

8

1N5818

Figure 23. A Positive-to-Negative Converter

The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 21.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1110.

NEGATIVE-TO-POSITIVE CONVERSION

The circuit of Figure 24 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 19, except the current
through feedback resistor R1 is level-shifted below ground by a
PNP transistor. The voltage across R1 is V

OUT

– V

. However,

BEQ1

diode D2 level-shifts the base of Q1 about 0.6 V below ground
thereby cancelling the V

of Q1. The addition of D2 also reduces

BE

the circuit’s output voltage sensitivity to temperature, which other­wise would be dominated by the –2 mV V

contribution of Q1.

BE

The output voltage for this circuit is determined by the formula:

LIMITING THE SWITCH CURRENT

The ADP1110’s R

pin permits the switch current to be

LIM

limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If V

rises to 4 V, however, the switch current will

IN

exceed 1.6 A. The ADP1110 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if V

increases. The relationship between R

IN

LIM

and maximum switch current is shown in Figure 6.
The I

feature is also valuable for controlling inductor current

LIM

when the ADP1110 goes into continuous-conduction mode.
This occurs in the step-up mode when the following condition is
met:



V

OUT+VD IODE



VIN–V



SW



<





1

1–DC

where DC is the ADP1110’s duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 25).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the I

LIM

resistor, output ripple current can be maintained at the design
values. Figure 26 illustrates the action of the I

circuit.

LIM





V

OUT

=220 mV •

R1





R2





Unlike the positive step-up converter, the negative-to-positive
converter’s output voltage can be either higher or lower than the
input voltage.

D1

L1

3

8

R1

Q1

2N3906

R2

10K

D2

1N4148

R

LIM

2

1

V

C

INPUT

NEGATIVE

INPUT

I

NC

LIM

ADP1110

7

6

NC

IN

SW1

FB

SW2

GNDSETAO

45

POSITIVE
OUTPUT

C

L

Figure 24. A Negative-to-Positive Converter

REV. 0

–11–

Figure 25. I

100

90

200mA/div.

0%

Figure 26. I

Operation—IL Characteristic

LIM

10

Operation—IL Characteristic

LIM

10µs10mV

ADP1110

ADP1110

GND

AO

R

L

R1

V

BAT

SET

R2

33kΩ

220V
V

REF

R

HYS

V

LOGIC

I

LIMVIN

SW1

SW2

FB

GND

SET

AO

ADP1110

L1

C

L

R1

R2

OUTPUT

C

INPUT

INPUT

6

7

8

1 2

4

5

3

1000µF

300kΩ

13.8kΩ

270kΩ

10µF

D1

CTX15-4

1N5818

15µH

The internal structure of the I

circuit is shown in Figure 27.

LIM

Q1 is the ADP1110’s internal power switch, that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that I
internal 80 Ω resistor and through the R

is 0.5% of IQ1. Current flows to Q2 through an

Q2

resistor. These two

LIM

resistors parallel the base-emitter junction of the oscillator­disable transistor, Q3. When the voltage across R1 and R

LIM

exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80 Ω internal resistor is used (i.e., the I
connected directly to V

1.5 A. Figure 6 gives R

V

IN

ADP1110

), the maximum switch current will be

IN

values for lower current-limit values.

LIM

R

LIM

(EXTERNAL)

R1

I

LIM

80Ω
(INTERNAL)

I

Q1

SW1

200

Q2

SW2

V

IN

72kHz
OSC

Q3

DRIVER

pin is

LIM

Q1
POWER

SWITCH

Figure 27. ADP1110 Current Limit Operation

The delay through the current limiting circuit is approximately
800 ns. If the switch ON time is reduced to less than 3 µs,
accuracy of the current trip-point is reduced. Attempting to
program a switch ON time of 800 ns or less will produce
spurious responses in the switch ON time; however, the
ADP1110 will still provide a properly-regulated output voltage.

PROGRAMMING THE GAIN BLOCK

The gain block of the ADP1110 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1110’s 220 mV reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300 µA.

Figure 28 shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 kΩ for R2 is a
good compromise. The value for R1 is then calculated from the
formula:

Figure 28. Setting the Low Battery Detector Trip Point

The circuit of Figure 28 may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit. Resistor R

HYS

, with a

value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition
of R

will change the trip point slightly, so the new value for

HYS

R1 will be:

R1=

V

220 mV

R2

LOBATT

–

– 220mV



V

– 220mV

L



R



L+RHYS






where VL is the logic power supply voltage, RL is the pull-up
resistor, and R

creates the hysteresis.

HYS

The gain block can also be used as a control element to reduce
output ripple. The ADP3000 is normally recommended for low­ripple applications, but its minimum input voltage is 2 V. The
gain-block technique using the ADP1110 can be useful for step­up converters operating down to 1 V.

A step-up converter using this technique is shown in Figure 29.
This configuration uses the gain block to sense the output
voltage and control the comparator. The result is that the
comparator hysteresis is reduced by the open loop gain of the
gain block. Output ripple can be reduced to only a few millivolts
with this technique, versus a typical value of 90 mV for a +5 V
converter using just the comparator. For best results, a large
output capacitor (1000 µF or more) should be specified. This
technique can also be used for step-down or inverting applica­tions, but the ADP3000 is usually a more appropriate choice.
See the ADP3000 data sheet for further details.

LOBATT

220 mV

– 220mV

R2

where V

V

R1=

is the desired low battery trip point. Since the

LOBATT

gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.

Figure 29. Using the Gain Block to Reduce Output Ripple

–12–

REV. 0

ADP1110

I

LIM

V

IN

SW1

SW2

SENSE

GNDSETAO

ADP1110-5

NC

NC

6

7

8

1

2

45

3

1.5V

L16
8µH

4.7µF

ONE

ALKALINE

CELL

4.7µF

+5V
3mA

NOTE: ALL DIODES 1N5818

–5V
3mA

4.7µF

CTX68-4

I

LIM

V

IN

SW1

SW2

SENSE

GNDSETAO

ADP1110-12

NC

NC

6

7

8

1

2

45

3

L1

50µH

10µF

C

L

LOGIC1 = PROGRAM
LOGIC0 = SHUTDOWN

D1

1N5818

CTX50-4MMBT4403

1kΩ

MMBF170

V

PP

+12V
120mA

+5V

10kΩ

APPLICATION CIRCUITS
All-Surface-Mount, Single-Cell to 5 V Converter

This is a very simple, compact, low-part-count circuit that takes
a single alkaline 1.5 V cell input and produces a 5 V output.
The output current should be kept to 10 mA or less to conserve
battery life.

D1

1N5817

ONE

ALKALINE

CELL

1.5V

L1

50µH

CTX50-4

2

1

I

V

LIM

IN

3

SW1

+5V
10mA

ADP1110-5

8

SENSE

SW2

NC

GNDSETAO

7

6

45

NC

C

L

15µF

Figure 30. All-Surface-Mount, Single-Cell to 5 V Converter

All-Surface-Mount, 3 V to 5 V Step-Up Converter

Similar to the previous circuit, this circuit takes a 3-volt input
and provides a 5 V output at 40 mA. As in the single-cell version,
the circuit is compact and uses only four external components.

D1

1N5817

3

8

C

L

10µF

+5V
40mA

TWO

ALKALINE

CELLS

220Ω

I

3V

NC

2

1

V

LIM

ADP1110-5

IN

SENSE

GNDSETAO

7

6

NC

L1

50µH

CTX50-4

SW1

SW2

45

1.5 V to 65 V Dual-Output Step-Up Converter

This circuit works from a single 1.5 V cell and provides simulta­neous outputs of +5 V and –5 V. The accuracy of the negative
output suffers slightly because of the extra diode drop of around

0.4 V.

Figure 33. 1.5 V to ±5 V Dual-Output Step-Up
Converter

All-Surface-Mount Flash Memory VPP Generator

Figure 34 shows a circuit that can generate the programming
voltage, VPP to program flash memory. The key components
are the MOSFET and the bipolar transistor. These two devices
form a switch that, when ON, allows the ADP1110 to power-up
and function as a step-up converter. The output is +12 V at 120
mA. When the MOSFET switch is OFF, the output of the
circuit drops to just under +5 V thereby disabling the program­ming capability.

Care should be taken so there is no short-circuit-current limiting
in the circuit in either operating mode.

All-Surface-Mount, 9 V to 5 V Step-Down Converter

Featuring the same low parts count of the step-up design, this
circuit is the complement to the preceding one. The 220 Ω
resistor programs the current limit to around 600 mA.

REV. 0

Figure 31. All-Surface-Mount, 3 V to 5 V Step-Up
Converter

R

LIM

220Ω

BATTERY

9V

2

1

I

LIM

3

SW1

V

IN

SW2

4

ADP1110-5

8

SENSE

GNDSET

AO

NC

Figure 32. All-Surface-Mount, 9 V to 5 V Step-Down
Converter

6

5

7

NC

1N5817

D1

L1

50µH

CTX50-4

C

L

10µF

+5V
40mA

Figure 34. All Surface-Mount Flash Memory VPP Generator

–13–

ADP1110

1.5 V to +5 V, +10 V Dual Output Step-Up Converter

The circuit of Figure 35 illustrates a way to get outputs of
+10 V and +5 V from the same converter. The main 5 V output
is derived from the feedback provided by the 487 kΩ and 11 kΩ
resistors. Capacitor C1 should be a multilayer ceramic variety
for best performance, but a good quality tantalum capacitor will
also give good performance at lower cost.

ONE

ALKALINE

CELL

1.5V

220Ω

I

2

1

V

LIM

ADP1110

GNDSETAO

7

6

NC

NC

IN

L1

50µH

CTX50-4

SW1

FB

SW2

45

3

8

D1

487kΩ

D2

D3

4.7µF

NOTE: ALL DIODES 1N5818

11kΩ

4.7µF

+10V
3mA

+5V
3mA

Figure 35. 1.5 V to +5 V, +10 V Dual Output Step-Up
Converter

5.1kΩ0.022µF

1.5V

1N4148

1kΩ

2N3906

1

I

LIM

ADP1110

AO

6

2

V

IN

8

SW1

SET

SW2FB GND

220Ω

10Ω

3

7

45

1kΩ

1.5 V-Powered Laser Diode Driver

Figure 36 shows a circuit suitable for driving many laser diodes
that incorporate a photodiode to monitor the laser diode
current, this circuit makes use of the gain block and current­limit functions to provide a feedback system based on the
average laser diode current. This current must be controlled
very closely or permanent damage to the laser diode is likely to
be the result.

To ensure that the laser is operating at the proper power level,
the actual optical power from the laser should be monitored
with a calibrated photodiode or optical power meter. In
addition, the actual diode current should also be monitored, and
R1 can be adjusted to give the correct output power.

NOTES

1. All inductors referenced are Coiltronics CTX-series except
where noted.

2. If the source of power is more than an inch or so from the
converter, the input to the converter should be bypassed with
approximately 10 µF of capacitance. This capacitor should be
a good quality tantalum or aluminum electrolytic.

TOSHIBA
TOLD-9321

1µF

2Ω

2.2µH

100µF

OS-Con

MJE210

1N5818

Figure 36. 1.5 V-Powered Laser Diode Driver

–14–

REV. 0

0.210 (5.33)
MAX

0.160 (4.06)

0.115 (2.93)

0.022 (0.558)

0.014 (0.356)

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

8-Lead Plastic DIP

(N-8)

0.430 (10.92)

0.348 (8.84)

8

14

PIN 1

0.100
(2.54)

BSC

5

0.280 (7.11)

0.240 (6.10)

0.060 (1.52)

0.015 (0.38)

0.070 (1.77)

0.045 (1.15)

0.130
(3.30)
MIN

SEATING
PLANE

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

8-Lead SOIC

(SO-8)

0.1968 (5.00)

0.1890 (4.80)

ADP1110

0.195 (4.95)

0.115 (2.93)

0.1574 (4.00)

0.1497 (3.80)

PIN 1

0.0098 (0.25)

0.0040 (0.10)

SEATING

PLANE

8

0.0500
(1.27)

BSC

5

0.2440 (6.20)

41

0.2284 (5.80)

0.0688 (1.75)

0.0532 (1.35)

0.0192 (0.49)

0.0138 (0.35)

0.0098 (0.25)

0.0075 (0.19)

0.0196 (0.50)

0.0099 (0.25)

8°
0°

0.0500 (1.27)

0.0160 (0.41)

x 45°

REV. 0

–15–

C2212–12–10/96

–16–

PRINTED IN U.S.A.