Page 1
12-Bit, 300 MSPS
AVDD ACOM
REFIO
FSADJ
PORT1
I
OUTA
I
OUTB
AD9753
DVDD
DCOM
LATCH
LATCH
CLK+
CLK–
CLKVDD
PLLVDD
CLKCOM
RESET LPF DIV0 DIV1 PLLLOCK
PLL
CLOCK
MULTIPLIER
DAC LATCH
DAC
REFERENCE
MUX
PORT2
High Speed TxDAC+
®
D/A Converter
FEATURES
12-Bit Dual Muxed Port DAC
300 MSPS Output Update Rate
Excellent SFDR and IMD Performance
SFDR to Nyquist @ 25 MHz Output: 69 dB
Internal Clock Doubling PLL
Differential or Single-Ended Clock Input
On-Chip 1.2 V Reference
Single 3.3 V Supply Operation
Power Dissipation: 155 mW @ 3.3 V
48-Lead LQFP
APPLICATIONS
Communications: LMDS, LMCS, MMDS
Base Stations
Digital Synthesis
QAM and OFDM
GENERAL DESCRIPTION
The AD9753 is a dual, muxed port, ultrahigh speed, singlechannel, 12-bit CMOS DAC. It integrates a high quality 12-bit
TxDAC+ core, a voltage reference, and digital interface circuitry
into a small 48-lead LQFP package. The AD9753 offers exceptional ac and dc performance while supporting update rates up
to 300 MSPS.
The AD9753 has been optimized for ultrahigh speed applications up to 300 MSPS where data rates exceed those possible on
a single data interface port DAC. The digital interface consists
of two buffered latches as well as control logic. These latches
can be time multiplexed to the high speed DAC in several ways.
This PLL drives the DAC latch at twice the speed of the externally applied clock and is able to interleave the data from the
two input channels. The resulting output data rate is twice that
of the two input channels. With the PLL disabled, an external
2× clock may be supplied and divided by two internally.
The CLK inputs (CLK+/CLK–) can be driven either differentially or single-ended, with a signal swing as low as 1 V p-p.
AD9753
*
FUNCTIONAL BLOCK DIAGRAM
The DAC utilizes a segmented current source architecture
combined with a proprietary switching technique to reduce
glitch energy and to maximize dynamic accuracy. Differential
current outputs support single-ended or differential applications. The differential outputs each provide a nominal full-scale
current from 2 mA to 20 mA.
The AD9753 is manufactured on an advanced low cost 0.35 µm
CMOS process. It operates from a single supply of 3.0 V to 3.6 V
and consumes 155 mW of power.
PRODUCT HIGHLIGHTS
1. The AD9753 is a member of a pin compatible family of high
speed TxDAC+s providing 10-, 12-, and 14-bit resolution.
2. Ultrahigh Speed 300 MSPS Conversion Rate.
3. Dual 12-Bit Latched, Multiplexed Input Ports. The AD9753
features a flexible digital interface allowing high speed data
conversion through either a single or dual port input.
4. Low Power. Complete CMOS DAC function operates on
155 mW from a 3.0 V to 3.6 V single supply. The DAC fullscale current can be reduced for lower power operation.
5. On-Chip Voltage Reference. The AD9753 includes a 1.20 V
temperature-compensated band gap voltage reference.
*Protected by U.S. Patent numbers 5450084, 5568145, 5689257, and
5703519. Other patents pending.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties that
may result from its use. No license is granted by implication or otherwise
under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective companies.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 www.analog.com
Fax: 781/326-8703 © 2003 Analog Devices, Inc. All rights reserved.
Page 2
AD9753–SPECIFICATIONS
(T
to T
, AVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 3.3 V, CLKVDD = 3.3 V, I
MAX
DC SPECIFICATIONS
MIN
otherwise noted.)
Parameter Min Typ Max Unit
RESOLUTION 12 Bits
DC ACCURACY
1
Integral Linearity Error (INL) –1.5 ± 0.5 +1.5 LSB
Differential Nonlinearity (DNL) –1 ±0.4 +1 LSB
ANALOG OUTPUT
Offset Error –0.025 ±0.01 +0.025 % of FSR
Gain Error (Without Internal Reference) –2 ±0.5 +2 % of FSR
Gain Error (With Internal Reference) –2 ±0.25 +2 % of FSR
Full-Scale Output Current
2
2.0 20.0 mA
Output Compliance Range –1.0 +1.25 V
Output Resistance 100 kΩ
Output Capacitance 5 pF
REFERENCE OUTPUT
Reference Voltage 1.14 1.20 1.26 V
Reference Output Current
3
100 nA
REFERENCE INPUT
Input Compliance Range 0.1 1.25 V
Reference Input Resistance 1 MΩ
TEMPERATURE COEFFICIENTS
Offset Drift 0 ppm of FSR/°C
Gain Drift (Without Internal Reference) ±50 ppm of FSR/°C
Gain Drift (With Internal Reference) ±100 ppm of FSR/°C
Reference Voltage Drift ±50 ppm/°C
POWER SUPPLY
Supply Voltages
AVDD 3.0 3.3 3.6 V
DVDD 3.0 3.3 3.6 V
PLLVDD 3.0 3.3 3.6 V
CLKVDD 3.0 3.3 3.6 V
Analog Supply Current (I
Digital Supply Current (I
PLL Supply Current (I
Clock Supply Current (I
Power Dissipation
Power Dissipation
4
(3 V, I
5
(3 V, I
PLLVDD
Power Supply Rejection Ratio
4
)
AVDD
4
)
DVDD
4
)
4
)
CLKVDD
= 20 mA) 155 165 mW
OUTFS
= 20 mA) 216 mW
OUTFS
6
—AVDD –1 +1 % of FSR/V
33 36 mA
3.5 4.5 mA
4.5 5.1 mA
10.0 11.5 mA
Power Supply Rejection Ratio6—DVDD –0.04 +0.04 % of FSR/V
OPERATING RANGE –40 +85 °C
NOTES
1
Measured at I
2
Nominal full-scale current, I
3
An external buffer amplifier is recommended to drive any external load.
4
100 MSPS f
5
300 MSPS f
6
±5% power supply variation.
Specifications subject to change without notice.
, driving a virtual ground.
OUTA
with PLL on, f
DAC
.
DAC
OUTFS
, is 32× the I
= 1 MHz, all supplies = 3.0 V.
OUT
current.
REF
= 20 mA, unless
OUTFS
–2–
REV.B
Page 3
(T
to T
DYNAMIC SPECIFICATIONS
MIN
, AVDD = 3.3 V, DVDD = 3.3 V, PLLVDD = 0 V, CLKVDD = 3.3 V, I
MAX
Differential Transformer-Coupled Output, 50 V Doubly Terminated, unless otherwise noted.)
OUTFS
= 20 mA,
Parameter Min Typ Max Unit
DYNAMIC PERFORMANCE
Maximum Output Update Rate (f
Output Settling Time (t
Output Propagation Delay (t
Glitch Impulse
1
) (to 0.1%)
ST
PD
Output Rise Time (10% to 90%)
Output Fall Time (10% to 90%)
Output Noise (I
Output Noise (I
= 20 mA) 50 pA/√Hz
OUTFS
= 2 mA) 30 pA/√Hz
OUTFS
) 300 MSPS
DAC
1
1
)
1
1
11 ns
1ns
5 pV-s
2.5 ns
2.5 ns
AC LINEARITY
Spurious-Free Dynamic Range to Nyquist
= 100 MSPS; f
f
DAC
= 1.00 MHz
OUT
0 dBFS Output 72 82 dBc
–6 dBFS Output 76 dBc
–12 dBFS Output 76 dBc
= 65 MSPS; f
f
DATA
= 65 MSPS; f
f
DATA
= 65 MSPS; f
f
DATA
f
= 65 MSPS; f
DATA
= 65 MSPS; f
f
DATA
= 200 MSPS; f
f
DAC
f
= 200 MSPS; f
DAC
= 200 MSPS; f
f
DAC
= 200 MSPS; f
f
DAC
f
= 200 MSPS; f
DAC
= 300 MSPS; f
f
DAC
= 300 MSPS; f
f
DAC
f
= 300 MSPS; f
DAC
= 300 MSPS; f
f
DAC
= 300 MSPS; f
f
DAC
= 1.1 MHz
OUT
= 5.1 MHz
OUT
= 10.1 MHz
OUT
= 20.1 MHz
OUT
= 30.1 MHz
OUT
= 1.1 MHz 78 dBc
OUT
= 11.1 MHz 75 dBc
OUT
= 31.1 MHz 70 dBc
OUT
= 51.1 MHz 70 dBc
OUT
= 71.1 MHz 67 dBc
OUT
= 1.1 MHz 78 dBc
OUT
= 26.1 MHz 69 dBc
OUT
= 51.1 MHz 65 dBc
OUT
= 101.1 MHz 59 dBc
OUT
= 141.1 MHz 58 dBc
OUT
2
2
2
2
2
77 dBc
77 dBc
76 dBc
72 dBc
68 dBc
Spurious-Free Dynamic Range within a Window
= 100 MSPS; f
f
DAC
= 1 MHz; 2 MHz Span
OUT
0 dBFS Output 82.5 92 dBc
= 65 MSPS; f
f
DAC
= 150 MSPS; f
f
DAC
= 5.02 MHz; 2 MHz Span 85 dBc
OUT
= 5.04 MHz; 4 MHz Span 85 dBc
OUT
Total Harmonic Distortion
= 100 MSPS; f
f
DAC
= 1.00 MHz
OUT
0 dBFS –82 –71 dBc
= 65 MHz; f
f
DAC
f
= 160 MHz; f
DAC
= 2.00 MHz –76 dBc
OUT
= 2.00 MHz –76 dBc
OUT
Multitone Power Ratio (Eight Tones at 110 kHz Spacing)
= 65 MSPS; f
f
DAC
= 2.00 MHz to 2.77 MHz
OUT
0 dBFS Output 73 dBc
–6 dBFS Output 71 dBc
–12 dBFS Output 69 dBc
NOTES
1
Measured single-ended into 50 Ω load.
2
Single-Port Mode (PLL disabled, DIV0 = 1, DIV1 = 0, data on Port 1).
Specifications subject to change without notice.
AD9753
REV. B
–3–
Page 4
AD9753
DIGITAL SPECIFICATIONS
(T
to T
MIN
, AVDD = DVDD = PLLVDD = CLKVDD = 3.3 V, I
MAX
= 20 mA, unless otherwise noted.)
OUTFS
Parameter Min Typ Max Unit
DIGITAL INPUTS
Logic 1 2.1 3 V
Logic 0 0 0.9 V
Logic 1 Current –10 +10 µA
Logic 0 Current –10 +10 µA
Input Capacitance 5 pF
Input Setup Time (t
), TA = 25°C 1.0 0.5 ns
S
Input Hold Time (tH), TA = 25°C 1.0 0.5 ns
Latch Pulsewidth (t
Input Setup Time (t
), TA = 25°C 1.5 ns
LPW
, PLLVDD = 0 V), TA = 25°C –1.0 –1.5 ns
S
Input Hold Time (tH, PLLVDD = 0 V), TA = 25°C 2.5 1.7 ns
CLK to PLLLOCK Delay (t
Latch Pulsewidth (t
LPW
, PLLVDD = 0 V), TA = 25°C 3.5 4.0 ns
D
PLLVDD = 0 V), TA = 25°C 1.5 ns
PLLOCK (VOH) 3.0 V
PLLOCK (VOL) 0.3 V
CLK INPUTS
Input Voltage Range 0 3 V
Common-Mode Voltage 0.75 1.5 2.25 V
Differential Voltage 0.5 1.5 V
Min CLK Frequency* 6.25 MHz
*Min CLK Frequency applies only when using internal PLL. When PLL is disabled, there is no minimum CLK frequency.
Specifications subject to change without notice.
–4–
REV. B
Page 5
AD9753
ABSOLUTE MAXIMUM RATINGS*
Parameter With Respect to Min Max Unit
AVDD, DVDD, CLKVDD, PLLVDD ACOM, DCOM, CLKCOM, PLLCOM –0.3 +3.9 V
AVDD, DVDD, CLKVDD, PLLVDD AVDD, DVDD, CLKVDD, PLLVDD –3.9 +3.9 V
ACOM, DCOM, CLKCOM, PLLCOM ACOM, DCOM, CLKCOM, PLLCOM –0.3 +0.3 V
REFIO, REFLO, FSADJ ACOM –0.3 AVDD + 0.3 V
, I
I
OUTA
OUTB
Digital Data Inputs (DB13 to DB0) DCOM –0.3 DVDD + 0.3 V
CLK+/CLK–, PLLLOCK CLKCOM –0.3 CLKVDD + 0.3 V
DIV0, DIV1, RESET CLKCOM –0.3 CLKVDD + 0.3 V
LPF PLLCOM –0.3 PLLVDD + 0.3 V
Junction Temperature 150 °C
Storage Temperature –65 +150 °C
Lead Temperature (10 sec) 300 °C
*Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device
at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended
periods may affect device reliability.
ACOM –1.0 AVDD + 0.3 V
ORDERING GUIDE
Temperature Package Package
PORT 1
DATA IN
PORT 2
t
S
DATA X
DATA Y
t
H
Model Range Description Option
AD9753AST –40°C to +85°C 48-Lead LQFP ST-48
AD9753ASTRL –40°C to +85°C 48-Lead LQFP ST-48
INPUT CLK
(PLL ENABLED)
I
OR I
OUTA
OUTB
t
LPW
t
PD
DATA X
t
PD
DATA Y
AD9753-EB Evaluation
THERMAL CHARACTERISTIC
Thermal Resistance
48-Lead LQFP
= 91°C/W
JA
Figure 1. I/O Timing
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although the
AD9753 features proprietary ESD protection circuitry, permanent damage may occur on devices
subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended
to avoid performance degradation or loss of functionality.
Board
REV. B
–5–
Page 6
AD9753
PIN CONFIGURATION
LPF
CLKCOM
ACOM
AD9753
TOP VIEW
(Not to Scale)
P1B3
P1B2
P1B1
OUTAIOUTB
I
AVDD
LSB–P1B0
RESERVED
RESERVED
FSADJ
REFIO
DVDD
DCOM
DIV1
DIV0
36
RESERVED
35
RESERVED
34
P2B0–LSB
33
P2B1
32
P2B2
31
P2B3
30
P2B4
29
P2B5
28
P2B6
27
P2B7
26
P2B8
25
P2B9
RESERVED = NO
USER CONNECTIONS
P2B10
MSB–P2B11
RESET
CLK+
CLK–
DCOM
DVDD
PLLLOCK
MSB–P1B11
P1B10
P1B9
P1B8
P1B7
P1B6
CLKVDD
PLLVDD
48 47 46 45 44 39 38 3743 42 41 40
1
PIN 1
2
IDENTIFIER
3
4
5
6
7
8
9
10
11
12
13 14 15 16 17 18 19 20 21 22 23 24
P1B5
P1B4
PIN FUNCTION DESCRIPTIONS
Pin No. Mnemonic Description
1 RESET Internal Clock Divider Reset
2 CLK+ Differential Clock Input
3 CLK– Differential Clock Input
4, 22 DCOM Digital Common
5, 21 DVDD Digital Supply Voltage
6 PLLLOCK Phase-Locked Loop Lock Indicator Output
7–18 P1B11–P1B0 Data Bits DB11 to DB0, Port 1
19–20, 35–36 RESERVED
23–34 P2B11–P2B0 Data Bits DB11 to DB0, Port 2
37, 38 DIV0, DIV1 Control Inputs for PLL and Input Port Selector Mode. See Tables I and II for details.
39 REFIO Reference Input/Output
40 FSADJ Full-Scale Current Output Adjust
41 AVDD Analog Supply Voltage
42 I
43 I
OUTB
OUTA
Differential DAC Current Output
Differential DAC Current Output
44 ACOM Analog Common
45 CLKCOM Clock and Phase-Locked Loop Common
46 LPF Phase-Locked Loop Filter
47 PLLVDD Phase-Locked Loop Supply Voltage
48 CLKVDD Clock Supply Voltage
–6–
REV. B
Page 7
AD9753
TERMINOLOGY
Linearity Error (Also Called Integral Nonlinearity or INL)
Linearity error is defined as the maximum deviation of the actual
analog output from the ideal output, determined by a straight
line drawn from zero to full scale.
Differential Nonlinearity (DNL)
DNL is the measure of the variation in analog value, normalized
to full scale, associated with a 1 LSB change in digital input code.
Monotonicity
A D/A converter is monotonic if the output either increases or
remains constant as the digital input increases.
Offset Error
The deviation of the output current from the ideal of zero is
called offset error. For I
inputs are all 0s. For I
, 0 mA output is expected when the
OUTA
, 0 mA output is expected when all
OUTB
inputs are set to 1s.
Gain Error
The difference between the actual and ideal output span. The
actual span is determined by the output when all inputs are set
to 1s, minus the output when all inputs are set to 0s.
Output Compliance Range
The range of allowable voltage at the output of a current-output
DAC. Operation beyond the maximum compliance limits may
cause either output stage saturation or breakdown, resulting in
nonlinear performance.
Temperature Drift
Specified as the maximum change from the ambient (25°C)
value to the value at either T
MIN
or T
. For offset and gain
MAX
drift, the drift is reported in ppm of full-scale range (FSR) per
degree Celsius. For reference drift, the drift is reported in ppm
per degree Celsius.
3.0V TO 3.6V
AVDD
PMOS CURRENT
SOURCE ARRAY
PORT 1 LATCH
DB0 – DB11
SEGMENTED
SWITCHES FOR
DB0 TO DB11
DAC LATCH
2–1 MUX
PORT 2 LATCH
DIGITAL DATA INPUTS
TEKTRONIX DG2020
AWG2021 w/OPTION 4
DB0 – DB11
OR
DAC
0.1F
R
SET
2k⍀
REFIO
FSADJ
DCOM
1.2V REF
DVDD
AD9753
ACOM
Power Supply Rejection
The maximum change in the full-scale output as the supplies
are varied from minimum to maximum specified voltages.
Settling Time
The time required for the output to reach and remain within a
specified error band about its final value, measured from the
start of the output transition.
Glitch Impulse
Asymmetrical switching times in a DAC give rise to undesired
output transients that are quantified by a glitch impulse. It is
specified as the net area of the glitch in pV-s.
Spurious-Free Dynamic Range
The difference, in dB, between the rms amplitude of the output
signal and the peak spurious signal over the specified bandwidth.
Total Harmonic Distortion (THD)
THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured fundamental. It is
expressed as a percentage or in decibels (dB).
Signal-to-Noise Ratio (SNR)
SNR is the ratio of the rms value of the measured output signal
to the rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in decibels.
Adjacent Channel Power Ratio (ACPR)
A ratio in dBc between the measured power within a channel
relative to its adjacent channel.
PLL
CIRCUITRY
CLK+ PLLLOCK
CLK–
MINI
CIRCUITS
T1-1T
1k⍀
I
OUTA
I
OUTB
PLLVDD
CLKVDD
RESET
LPF
CLKCOM
DIV0
DIV1
1k⍀
50⍀
50⍀
3.0V TO 3.6V
MINI
CIRCUITS
T1-1T
TO ROHDE &
SCHWARZ
FSEA30
SPECTRUM
ANALYZER
REV. B
LECROY 9210
PULSE GENERATOR
(FOR DATA RETIMING)
PLL ENABLEDPLL DISABLED
HP8644
SIGNAL
GENERATOR
Figure 2. Basic AC Characterization Test Setup
–7–
Page 8
AD9753–Typical Performance Characteristics
f
OUT
(MHz)
90
70
40
1000
SFDR (dBc)
80
60
50
20 40 60 80 120 140 160
0dBmFS
–6dBmFS
–12dBmFS
f
OUT
(MHz)
90
70
40
1000
SFDR (dBc)
80
60
50
20 40 60 80 120 140 160
SFDR CLOSE TO CARRIERS
(2F1-F2, 2F2-F1)
SFDR OVER NYQUIST BAND
90
80
70
–6dBmFS
60
SFDR (dBc)
50
40
0dBmFS
–12dBmFS
10 15 20 25 30
f
(MHz)
OUT
TPC 1. Single-Tone SFDR vs. f
f
= 65 MSPS, Single-Port Mode
DAC
90
80
200MSPS
70
60
SFDR (dBc)
65MSPS
300MSPS
50
OUT
3550
@
90
80
70
60
SFDR (dBc)
50
40
0dBmFS
20 30 40 50 60 70 80 90
f
OUT
–6dBmFS
–12dBmFS
(MHz)
TPC 2. Single-Tone SFDR vs.
f
OUT
90
80
70
60
SFDR (dBc)
50
@ f
NYQUIST BAND
= 200 MSPS
DAC
SFDR OVER
SFDR NEAR CARRIERS
(2F1-F2, 2F2-F1)
100100
TPC 3. Single-Tone SFDR vs.
f
OUT
@ f
= 300 MSPS
DAC
40
20 40 60 80 120 140
f
OUT
TPC 4. SFDR vs. f
90
11.82MHz @ 130MSPS
80
70
60
SFDR (dBc)
50
40
27.27MHz @ 300MSPS
–14 –12 –10 –8 –4 –2 0
A
OUT
TPC 7. Single-Tone SFDR vs.
A
OUT
@ f
OUT
= f
1000
(MHz)
@ 0 dBFS
OUT
18.18MHz @ 200MSPS
–6–16
(dB)
/11
DAC
40
20 30 40 50 60 70 80 90
f
(MHz)
OUT
TPC 5. Two-Tone IMD vs. f
f
= 200 MSPS, 1 MHz Spacing
DAC
between Tones, 0 dBFS
90
26MHz @ 130MSPS
80
70
60
SFDR (dBc)
50
40
–14 –12 –10 –8 –4 –2 0
40MHz @ 200MSPS
60MHz @ 300MSPS
–6–16
A
(dBm)
OUT
TPC 8. Single-Tone SFDR vs.
A
@ f
= f
OUT
OUT
DAC
/5
OUT
@
100100
TPC 6. Two-Tone IMD vs. f
f
= 300 MSPS, 1 MHz Spacing
DAC
OUT
@
between Tones, 0 dBFS
90
80
11.82MHz/12.82MHz
@ 130MSPS
70
18.18MHz/19.18MHz
–14 –12 –10 –8 –4 –2 0–18–20
@ 200MSPS
A
(dBm)
OUT
–6–16
60
SFDR (dBc)
27.27MHz/28.27MHz
50
40
@ 300MSPS
TPC 9. Two-Tone IMD (Third Order
Products) vs. A
OUT
@ f
OUT
= f
DAC
/11
–8–
REV. B
Page 9
AD9753
)
FREQUENCY (MHz)
–10
–30
–80
200
AMPLITUDE (dBm)
–20
–40
–75
40 60 100 120
–95
–100
140
0
–50
–60
80
f
DAC
= 300MSPS
f
OUT1
= 24MHz
f
OUT2
= 25MHz
f
OUT3
= 26MHz
f
OUT4
= 27MHz
f
OUT5
= 28MHz
f
OUT6
= 29MHz
f
OUT7
= 30MHz
f
OUT8
= 31MHz
SFDR = 58dBc
MAGNITUDE = 0dBFS
0
90
11.82MHz/12.82MHz
@ 130MSPS
80
70
60
SFDR (dBc)
50
40
18.18MHz/19.18MHz
@ 200MSPS
27.27MHz/28.27MHz
@ 300MSPS
–14 –12 –10 –8 –4 –2 0–18–20
A
(dBm)
OUT
–6–16
TPC 10. Two-Tone IMD (to Nyquist)
vs. A
@ f
OUT
90
85
80
75
70
65
SINAD (dBm)
60
55
50
= f
OUT
100 150 200 250
f
DAC
TPC 13. SINAD vs. f
f
= 10 MHz, 0 dBFS
OUT
DAC
(MHz)
/11
DAC
30050
@
90
40MHz/41MHz
80
70
60
SFDR (dBc)
50
40
26MHz/27MHz
@ 130MSPS
–14 –12 –10 –8 –4 –2 0–18–20
A
OUT
@ 200MSPS
60MHz/61MHz
@ 300MSPS
–6–16
(dBm)
TPC 11. Two-Tone IMD (Third Order
Products) vs. A
75
70
65
60
55
SFDR (dBc)
I
= 5mA
OUTFS
50
45
40
TPC 14. SFDR vs. I
@ f
I
OUTFS
(MHz)
= f
OUT
= 10mA
OUTFS
OUT
I
= 20mA
OUTFS
40 60 80 100 120
f
OUT
DAC
, f
140
/5
DAC
160200
=
300 MSPS @ 0 dBFS
90
26MHz/27MHz
80
70
60
SFDR (dBc)
50
40
40MHz/41MHz
@ 200MSPS
–14 –12 –10 –8 –4 –2 0–18–20
@ 130MSPS
60MHz/61MHz
@ 300MSPS
A
(dBm)
OUT
–6–16
TPC 12. Two-Tone IMD (to Nyquist)
vs. A
OUT
80
75
70
65
60
SFDR (dBc)
55
50
45
40
@ f
= f
OUT
10MHz
40MHz
80MHz
120MHz
–10 10 30 50
TEMPERATURE (ⴗC
DAC
/5
90–30–50
70
TPC 15. SFDR vs. Temperature,
f
= 300 MSPS @ 0 dBFS
DAC
0.1
0.6
0.4
0.2
0
INL (LSB)
–0.2
–0.4
–0.6
1023 1535 2047 3071
TPC 16. Typical INL
REV. B
CODE
2559
3583
0.54
0.50
0.46
0.42
0.38
0.34
0.30
0.26
0.22
DNL (LSB)
0.18
0.14
0.10
0.06
0.02
40955110
–0.02
1023 1535 2047 3071
2559
CODE
TPC 17. Typical DNL
3583
40955110
TPC 18. Eight-Tone SFDR @ f
f
DAC
/11, f
= 300 MSPS
DAC
OUT
≈
–9–
Page 10
AD9753
0.1F
R
2k⍀
SET
REFIO
FSADJ
DCOM
1.2V REF
ACOM
3.0V TO 3.6V
DVDD
PMOS CURRENT
SOURCE ARRAY
AD9753
AVDD
SEGMENTED
SWITCHES FOR
DB0 TO DB11
DAC LATCH
2–1 MUX
PORT 1 LATCH
DB0 – DB11
DIGITAL DATA INPUTS
DAC
PORT 2 LATCH
DB0 – DB11
CIRCUITRY
DIV0
Figure 3. Simplified Block Diagram
PLL
DIV1
PLLLOCK
V
I
OUTA
I
OUTB
PLLVDD
CLKVDD
CLK+
CLK–
CLKCOM
RESET
LPF
DIFF
= V
OUT
V
R
50⍀
A – V
OUT
LOAD
B
OUT
B R
V
50⍀
OUT
LOAD
A
FUNCTIONAL DESCRIPTION
Figure 3 shows a simplified block diagram of the AD9753. The
AD9753 consists of a PMOS current source array capable of
providing up to 20 mA of full-scale current, I
OUTFS
. The
array is divided into 31 equal sources that make up the five
most significant bits (MSBs). The next four bits, or middle bits,
consist of 15 equal current sources whose value is 1/16th of an
MSB current source. The remaining LSBs are a binary weighted
fraction of the middle bit current sources. Implementing the
middle and lower bits with current sources, instead of an R-2R
ladder, enhances dynamic performance for multitone or low
amplitude signals and helps maintain the DAC’s high output
impedance (i.e., >100 kΩ).
All of the current sources are switched to one of the two
outputs (i.e., I
OUTA
or I
) via PMOS differential current
OUTB
switches. The switches are based on a new architecture that
drastically improves distortion performance. This new switch
architecture reduces various timing errors and provides matching
complementary drive signals to the inputs of the differential
current switches.
The analog and digital sections of the AD9753 have separate
power supply inputs (i.e., AVDD and DVDD) that can operate
independently over a 3.0 V to 3.6 V range. The digital section,
which is capable of operating at a 300 MSPS clock rate, consists
of edge-triggered latches and segment decoding logic circuitry.
The analog section includes the PMOS current sources, the
associated differential switches, a 1.20 V band gap voltage reference, and a reference control amplifier.
The full-scale output current is regulated by the reference control
amplifier and can be set from 2 mA to 20 mA via an external
resistor, R
the reference control amplifier and voltage reference V
the reference current I
. The external resistor, in combination with both
SET
, which is replicated to the segmented
REF
REFIO
, sets
current sources with the proper scaling factor. The full-scale
current, I
, is 32 times the value of I
OUTFS
REF
.
REFERENCE OPERATION
The AD9753 contains an internal 1.20 V band gap reference.
This can easily be overdriven by an external reference with no
effect on performance. REFIO serves as either an input or output,
depending on whether the internal or an external reference is
used. To use the internal reference, simply decouple the REFIO
pin to ACOM with a 0.1 µF capacitor. The internal reference
voltage will be present at REFIO. If the voltage at REFIO is to
be used elsewhere in the circuit, an external buffer amplifier
with an input bias current less than 100 nA should be used. An
example of the use of the internal reference is given in Figure 4.
A low impedance external reference can be applied to REFIO,
as shown in Figure 5. The external reference may provide either
a fixed reference voltage to enhance accuracy and drift performance or a varying reference voltage for gain control. Note
that the 0.1 µF compensation capacitor is not required since
the internal reference is overdriven, and the relatively high input
impedance of REFIO minimizes any loading of the external
reference.
ADDITIONAL
EXTERNAL
LOAD
OPTIONAL
EXTERNAL
REFERENCE
BUFFER
0.1F
I
REF
2k⍀
REFIO
FSADJ
AD9753
REFERENCE
SECTION
1.2V REF
AVDD
CURRENT
SOURCE
ARRAY
Figure 4. Internal Reference Configuration
AVDD
CURRENT
SOURCE
ARRAY
AVDD
EXTERNAL
REFERENCE
I
REF
2k⍀
REFIO
FSADJ
AD9753
REFERENCE
SECTION
1.2V REF
–10–
Figure 5. External Reference Configuration
REV. B
Page 11
REFERENCE CONTROL AMPLIFIER
The AD9753 also contains an internal control amplifier that is
used to regulate the DAC’s full-scale output current, I
OUTFS
.
The control amplifier is configured as a voltage-to-current
converter as shown in Figure 4, so that its current output, I
determined by the ratio of V
as stated in Equation 4. I
REF
sources with the proper scaling factor to set I
and an external resistor, R
REFIO
is applied to the segmented current
, as stated in
OUTFS
REF
, is
SET
,
Equation 3.
The control amplifier allows a wide (10:1) adjustment span of
I
over a 2 mA to 20 mA range by setting I
OUTFS
62.5 µA and 625 µA. The wide adjustment span of I
between
REF
OUTFS
provides several application benefits. The first benefit relates
directly to the power dissipation of the AD9753, which is
proportional to I
(refer to the Power Dissipation section).
OUTFS
The second benefit relates to the 20 dB adjustment, which is
useful for system gain control purposes.
The small signal bandwidth of the reference control amplifier is
approximately 500 kHz and can be used for low frequency,
small signal multiplying applications.
PLL CLOCK MULTIPLIER OPERATION
The Phase-Locked Loop (PLL) is intrinsic to the operation of
the AD9753 in that it produces the necessary internally synchronized 2× clock for the edge-triggered latches, multiplexer,
and DAC.
With PLLVDD connected to its supply voltage, the AD9753 is
in PLL mode. Figure 6 shows a functional block diagram of the
AD9753 clock control circuitry with PLL active. The circuitry
consists of a phase detector, charge pump, voltage controlled
oscillator (VCO), input data rate range control, clock logic
circuitry, and control input/outputs. The ÷2 logic in the feedback loop allows the PLL to generate the 2× clock needed for
the DAC output latch.
1.0F
CLKVDD
(3.0V TO 3.6V)
PLLLOCK
LPF
392⍀
PLLVDD
3.0V TO
3.6V
AD9753
t
t
H
S
PORT 1
DATA IN
PORT 2
CLK
I
OR I
OUTA
OUTB
Figure 7a. DAC Input Timing Requirements with
PLL Active, Single Clock Cycle
PORT 1
DATA IN
PORT 2
OR I
I
OUTA
OUTB
Figure 7b. DAC Input Timing Requirements with
PLL Active, Multiple Clock Cycles
Typically, the VCO can generate outputs of 100 MHz to
400 MHz. The range control is used to keep the VCO operating
within its designed range, while allowing input clocks as low as
6.25 MHz. With the PLL active, logic levels at DIV0 and DIV1
determine the divide (prescaler) ratio of the range controller.
Table I gives the frequency range of the input clock for the
different states of DIV0 and DIV1.
Table I. CLK Rates for DIV0, DIV1 Levels with PLL Active
CLK
DATA X
DATA Y
t
LPW
1/2 CYCLE +
DATA W
DATA X DATA Z
XXX
t
DATA W
PD
DATA Y
DATA X
DATA X
t
DATA Y
PD
DATA Y
DATA Z
CLK+
CLK–
DIFFERENTIAL-
TO-
SINGLE-ENDED
AMP
DETECTOR
TO INPUT
LATCHES
AD9753
PHASE
CHARGE
PUMP
ⴜ2
TO DAC
LATCH
VCO
RANGE
CONTROL
(ⴜ 1, 2, 4, 8)
CLKCOM
DIV0
DIV1
Figure 6. Clock Circuitry with PLL Active
Figure 7 defines the input and output timing for the AD9753
with the PLL active. CLK in Figure 7 represents the clock
that is generated external to the AD9753. The input data at
both Ports 1 and 2 is latched on the same CLK rising edge.
CLK may be applied as a single-ended signal by tying CLK– to
midsupply and applying CLK to CLK+, or as a differential
signal applied to CLK+ and CLK–.
RESET has no purpose when using the internal PLL and should
be grounded. When the AD9753 is in PLL mode, PLLLOCK
is the output of the internal phase detector. When locked, the
lock output in this mode will be a Logic 1.
REV. B
–11–
CLK Frequency DIV1 DIV0 Range Controller
50 MHz–150 MHz 0 0 ÷1
25 MHz–100 MHz 0 1 ÷2
12.5 MHz–50 MHz 1 0 ÷4
6.25 MHz–25 MHz 1 1 ÷8
A 392 Ω resistor and 1.0 µF capacitor connected in series from
LPF to PLLVDD are required to optimize the phase noise versus
the settling/acquisition time characteristics of the PLL. To
obtain optimum noise and distortion performance, PLLVDD
should be set to a voltage level similar to DVDD and
CLKVDD.
In general, the best phase noise performance for any PLL range
control setting is achieved with the VCO operating near its maximum output frequency of 400 MHz.
As stated earlier, applications requiring input data rates below
6.25 MSPS must disable the PLL clock multiplier and provide
an external 2× reference clock. At higher data rates however,
applications already containing a low phase noise (i.e., jitter)
Page 12
AD9753
)
reference clock that is twice the input data rate should consider
disabling the PLL clock multiplier to achieve the best SNR
performance from the AD9753. Note, the SFDR performance
of the AD9753 remains unaffected with or without the PLL
clock multiplier enabled.
The effects of phase noise on the AD9753’s SNR performance
become more noticeable at higher reconstructed output frequencies and signal levels. Figure 8 compares the phase noise of a
full-scale sine wave at exactly f
/4 at different data rates
DATA
(thus carrier frequency) with the optimum DIV1, DIV0 setting.
SNR is partly a function of the jitter generated by the clock
circuitry. As a result, any noise on PLLVDD or CLKVDD may
decrease the SNR at the output of the DAC. To minimize this
potential problem, PLLVDD and CLKVDD can be connected
to DVDD using an LC filter network similar to the one shown
in Figure 9.
0
–10
–20
–30
–40
–50
–60
–70
–80
NOISE DENSITY (dBm/Hz)
–90
–100
–110
PLL OFF, f
DATA
= 50MSPS
PLL ON, f
234
FREQUENCY OFFSET (MHz
DATA
= 150MSPS
510
Figure 8. Phase Noise of PLL Clock Multiplier at
f
OUT
= f
/4 at Different f
DATA
Settings with DIV0/DIV1
DATA
Optimized, Using R&S FSEA30 Spectrum Analyzer
Following the rising edge of CLK at a time equal to half of its
period, the data in the Port 1 latch will be written to the DAC
output latch, again with a corresponding change in the DAC
output. Due to the internal PLL, the time at which the data in
the Port 1 and Port 2 input latches is written to the DAC latch
is independent of the duty cycle of CLK. When using the PLL,
the external clock can be operated at any duty cycle that meets
the specified input pulsewidth.
On the next rising edge of CLK, the cycle begins again with the
two input port latches being updated, and the DAC output latch
being updated with the current data in the Port 2 input latch.
PLL DISABLED MODE
When PLLVDD is grounded, the PLL is disabled. An external
clock must now drive the CLK inputs at the desired DAC output update rate. The speed and timing of the data present at
input Ports 1 and 2 are now dependent on whether or not the
AD9753 is interleaving the digital input data or only responding
to data on a single port. Figure 10 is a functional block diagram
of the AD9753 clock control circuitry with the PLL disabled.
PLLLOCK
TO DAC
LATCH
CLOCK
LOGIC
(ⴜ1 OR ⴜ2)
RESET DIV0 DIV1
TO INPUT
LATCHES
TO
INTERNAL
MUX
PLLVDD
CLKIN+
CLKIN–
AD9753
DIFFERENTIAL-
TO-
SINGLE-ENDED
AMP
Figure 10. Clock Circuitry with PLL Disabled
DIV0 and DIV1 no longer control the PLL but are used to set
the control on the input mux for either interleaving or noninterleaving the input data. The different modes for states of
DIV0 and DIV1 are given in Table II.
FERRITE
TTL/CMOS
LOGIC
CIRCUITS
3.3V POWER SUPPLY
BEADS
100F
ELECT.
10F
TANT.
0.1F
CER.
CLKVDD
PLLVDD
CLKCOM
Figure 9. LC Network for Power Filtering
DAC TIMING WITH PLL ACTIVE
As described in Figure 7, in PLL active mode, Port 1 and
Port 2 input latches are updated on the rising edge of CLK. On
the same rising edge, data previously present in the input Port 2
latch is written to the DAC output latch. The DAC output will
update after a short propagation delay (t
PD
).
–12–
Table II. Input Mode for DIV0,
DIV1 Levels with PLL Disabled
Input Mode DIV1 DIV0
Interleaved (2×)0 0
Noninterleaved
Port 1 Selected 0 1
Port 2 Selected 1 0
Not Allowed 1 1
REV. B
Page 13
AD9753
t
H
t
S
t
LPW
t
PD
DATA OUT
PORT 1 OR
PORT 2
1ⴛ CLOCK
I
OUTA
OR I
OUTB
XX
DATA IN
PORT 1 OR
PORT 2
INTERLEAVED (2ⴛ) MODE WITH PLL DISABLED
The relationship between the internal and external clocks in this
mode is shown in Figure 11. A clock at the output update data
rate (2× the input data rate) must be applied to the CLK inputs.
Internal dividers then create the internal 1× clock necessary for
the input latches. Although the input latches are updated on the
rising edge of the delayed internal 1× clock, the setup-and-hold
times given in the Digital Specifications table are with respect to
the rising edge of the external 2× clock. With the PLL disabled,
a load-dependent delayed version of the 1× clock is present at
the PLLLOCK pin. This signal can be used to synchronize the
external data.
tHt
S
PORT 1
DATA IN
PORT 2
EXTERNAL
2ⴛ CLK
DELAYED
INTERNAL
1ⴛ CLK
EXTERNAL
1ⴛ CLK
@ PLLLOCK
DATA X
DATA Y
I
OUTA
t
LPW
t
D
OR I
OUTB
DATA ENTERS
INPUT LATCHES
ON THIS EDGE
t
PD
DATA X
t
PD
DATA Y
Figure 11. Timing Requirements, Interleaved (2×)
Mode with PLL Disabled
Updates to the data at input Ports 1 and 2 should be synchronized to the specific rising edge of the external 2× clock that
corresponds to the rising edge of the 1× internal clock, as shown
in Figure 11. To ensure synchronization, a Logic 1 must be
momentarily applied to the RESET pin. Doing this and returning RESET to Logic 0 brings the 1× clock at PLLLOCK to a
Logic 1. On the next rising edge of the 2× clock, the 1× clock
will go to Logic 0. On the second rising edge of the 2× clock,
the 1× clock (PLLLOCK) will again go to Logic 1, as well as
update the data in both of the input latches. The details of this
are shown in Figure 12.
DATA ENTERS
INPUT LATCHES
ON THESE EDGES
RESET
PLLLOCK
EXTERNAL
2ⴛ CLOCK
t
= 1.2ns
t
RS
= 0.2ns
RH
Figure 12. RESET Function Timing with PLL Disabled
For proper synchronization, sufficient delay must be present
between the time RESET goes low and the rising edge of the 2×
clock. RESET going low must occur either at least t
the rising edge of the 2× clock, or t
ns afterwards. In the
RH
ns before
RS
first case, the immediately occurring CLK rising edge will
cause PLLLOCK to go low. In the second case, the next
CLK rising edge will toggle PLLLOCK.
REV. B
–13–
NONINTERLEAVED MODE WITH PLL DISABLED
If the data at only one port is required, the AD9753 interface can
operate as a simple double buffered latch with no interleaving.
On the rising edge of the 1× clock, input latch 1 or 2 is updated
with the present input data (depending on the state of DIV0/
DIV1). On the next rising edge, the DAC latch is updated and a
time t
later, the DAC output reflects this change. Figure 13
PD
represents the AD9753 timing in this mode.
Figure 13. Timing Requirements, Noninterleaved
Mode with PLL Disabled
DAC TRANSFER FUNCTION
The AD9753 provides complementary current outputs, I
and I
I
OUTFS
I
OUTB
current output appearing at I
both the input code and I
. I
OUTB
will provide a near full-scale current output,
OUTA
, when all bits are high (i.e., DAC CODE = 4095), while
, the complementary output, provides no current. The
OUTA
and can be expressed as
OUTFS
I
= (DAC CODE/4096) × I
OUTA
= (4095 – DAC CODE)/4096 × I
I
OUTB
and I
is a function of
OUTB
OUTFS
OUTFS
OUTA
(1)
(2)
where DAC CODE = 0 to 4095 (i.e., decimal representation).
As mentioned previously, I
current, I
V
REFIO
I
, which is nominally set by a reference voltage,
REF
, and an external resistor R
= 32 × I
OUTFS
REF
is a function of the reference
OUTFS
. It can be expressed as
SET
(3)
where
I
REF
= V
REFIO/RSET
(4)
The two current outputs typically drive a resistive load directly
or via a transformer. If dc coupling is required, I
should be directly connected to matching resistive loads, R
that are tied to analog common, ACOM. Note that R
represent the equivalent load resistance seen by I
OUTA
OUTA
and I
LOAD
or I
OUTB
LOAD
may
OUTB
,
as would be the case in a doubly terminated 50 Ω or 75 Ω cable.
The single-ended voltage output appearing at the I
I
nodes is simply
OUTB
V
= I
OUTA
= I
V
OUTB
Note that the full-scale values of V
OUTA
OUTB
× R
× R
LOAD
LOAD
OUTA
and V
OUTB
and
OUTA
should not
(5)
(6)
exceed the specified output compliance range to maintain
specified distortion and linearity performance.
V
DIFF
= (I
OUTA
– I
OUTB
) × R
LOAD
(7)
Page 14
AD9753
Substituting the values of I
OUTA
, I
OUTB,
and I
REF
, V
DIFF
can be
expressed as
V
= {(2 DAC CODE – 4095)/4096} ×
DIFF
(32 R
LOAD/RSET
) × V
REFIO
(8)
These last two equations highlight some of the advantages of
operating the AD9753 differentially. First, the differential operation will help cancel common-mode error sources associated
with I
OUTA
and I
such as noise, distortion, and dc offsets.
OUTB
Second, the differential code-dependent current and subsequent
voltage, V
output (i.e., V
, is twice the value of the single-ended voltage
DIFF
OUTA
or V
), thus providing twice the signal
OUTB
power to the load.
Note that the gain drift temperature performance for a singleended (V
OUTA
and V
) or differential output (V
OUTB
DIFF
) of the
AD9753 can be enhanced by selecting temperature tracking
resistors for R
LOAD
and R
due to their ratiometric relation-
SET
ship, as shown in Equation 8.
ANALOG OUTPUTS
The AD9753 produces two complementary current outputs,
I
and I
OUTA
differential operation. I
complementary single-ended voltage outputs, V
via a load resistor, R
, that may be configured for single-ended or
OUTB
LOAD
OUTA
and I
can be converted into
OUTB
OUTA
and V
, as described by Equations 5 through
OUTB
,
8 in the DAC Transfer Function section. The differential voltage,
, existing between V
V
DIFF
OUTA
and V
, can also be con-
OUTB
verted to a single-ended voltage via a transformer or differential
amplifier configuration. The ac performance of the AD9753 is
optimum and specified using a differential transformer-coupled
output in which the voltage swing at I
OUTA
and I
to ±0.5 V. If a single-ended unipolar output is desirable, I
should be selected as the output, with I
grounded.
OUTB
OUTB
is limited
OUTA
The distortion and noise performance of the AD9753 can be
enhanced when it is configured for differential operation. The
common-mode error sources of both I
OUTA
and I
OUTB
can be
significantly reduced by the common-mode rejection of a transformer or differential amplifier. These common-mode error
sources include even-order distortion products and noise. The
enhancement in distortion performance becomes more significant
as the frequency content of the reconstructed waveform increases.
This is due to the first order cancellation of various dynamic
common-mode distortion mechanisms, digital feedthrough,
and noise.
Performing a differential-to-single-ended conversion via a transformer also provides the ability to deliver twice the reconstructed
signal power to the load (i.e., assuming no source termination).
Since the output currents of I
OUTA
and I
are complemen-
OUTB
tary, they become additive when processed differentially. A
properly selected transformer will allow the AD9753 to provide
the required power and voltage levels to different loads. Refer to
the Applying the AD9753 Output Configurations section for
examples of various output configurations.
The output impedance of I
OUTA
and I
is determined by the
OUTB
equivalent parallel combination of the PMOS switches associated with the current sources and is typically 100 kΩ in parallel
with 5 pF. It is also slightly dependent on the output voltage
(i.e., V
OUTA
and V
) due to the nature of a PMOS device.
OUTB
As a result, maintaining I
OUTA
and/or I
at a virtual ground
OUTB
via an I–V op amp configuration will result in the optimum dc
linearity. Note that the INL/DNL specifications for the AD9753
are measured with I
OUTA
and I
maintained at virtual ground
OUTB
via an op amp.
I
OUTA
and I
also have a negative and positive voltage com-
OUTB
pliance range that must be adhered to in order to achieve optimum
performance. The negative output compliance range of –1.0 V is
set by the breakdown limits of the CMOS process. Operation
beyond this maximum limit may result in a breakdown of the
output stage and affect the reliability of the AD9753.
The positive output compliance range is slightly dependent on
the full-scale output current, I
nominal 1.25 V for an I
OUTFS
. It degrades slightly from its
OUTFS
= 20 mA to 1.00 V for an I
OUTFS
= 2 mA. The optimum distortion performance for a singleended or differential output is achieved when the maximum
full-scale signal at I
OUTA
and I
Applications requiring the AD9753’s output (i.e., V
or V
R
) to extend its output compliance range should size
OUTB
accordingly. Operation beyond this compliance range
LOAD
does not exceed 0.5 V.
OUTB
OUTA
and/
will adversely affect the AD9753’s linearity performance and
subsequently degrade its distortion performance.
DIGITAL INPUTS
The AD9753’s digital inputs consist of two channels of 14 data
input pins each and a pair of differential clock input pins. The
12-bit parallel data inputs follow standard straight binary coding
where DB13 is the most significant bit (MSB) and DB0 is the
least significant bit (LSB). I
current when all data bits are at Logic 1. I
produces a full-scale output
OUTA
produces a
OUTB
complementary output with the full-scale current split between
the two outputs as a function of the input code.
The digital interface is implemented using an edge-triggered
master slave latch. With the PLL active or disabled, the DAC
output is updated twice for every input latch rising edge, as
shown in Figures 7 and 11. The AD9753 is designed to support
an input data rate as high as 150 MSPS, giving a DAC output
update rate of 300 MSPS. The setup-and-hold times can also
be varied within the clock cycle as long as the specified minimum times are met. Best performance is typically achieved
when the input data transitions on the falling edge of a 50%
duty cycle clock.
The digital inputs are CMOS compatible with logic thresholds,
V
THRESHOLD
, set to approximately half the digital positive supply
(DVDD) or
V
THRESHOLD
= DVDD/2 (± 20%)
The internal digital circuitry of the AD9753 is capable of operating over a digital supply range of 3.0 V to 3.6 V. As a result,
the digital inputs can also accommodate TTL levels when DVDD
is set to accommodate the maximum high level voltage of the
TTL drivers V
(max). A DVDD of 3.0 V to 3.6 V typically
OH
ensures proper compatibility with most TTL logic families.
Figure 14 shows the equivalent digital input circuit for the data
and clock inputs.
–14–
REV. B
Page 15
AD9753
TIME OF DATA TRANSITION RELATIVE TO PLACEMENT OF
CLK RISING EDGE (ns), f
OUT
= 10MHz, f
DAC
= 300MHz
80
40
0
30–3
SNR (dBc)
60
20
70
30
50
10
–2 –1 1 2
DVDD
DIGITAL
INPUT
Figure 14. Equivalent Digital Input
The AD9753 features a flexible differential clock input operating from separate supplies (i.e., CLKVDD, CLKCOM) to
achieve optimum jitter performance. The two clock inputs,
CLK+ and CLK–, can be driven from a single-ended or differential clock source. For single-ended operation, CLK+ should
be driven by a logic source while CLK– should be set to the
threshold voltage of the logic source. This can be done via a
resistor divider/capacitor network, as shown in Figure 15a. For
differential operation, both CLK+ and CLK– should be biased to
CLKVDD/2 via a resistor divider network, as shown in Figure 15b.
AD9753
CLK+
CLKVDD
CLK–
CLKCOM
V
THRESHOLD
0.1F
R
SERIES
Figure 15a. Single-Ended Clock Interface
0.1F
0.1F
0.1F
AD9753
CLK+
CLKVDD
CLK–
Note that the clock input could also be driven via a sine wave
that is centered around the digital threshold (i.e., DVDD/2) and
meets the min/max logic threshold. This typically results in a
slight degradation in the phase noise, which becomes more
noticeable at higher sampling rates and output frequencies. Also,
at higher sampling rates, the 20% tolerance of the digital logic
threshold should be considered since it will affect the effective
clock duty cycle and, subsequently, cut into the required data
setup-and-hold times.
INPUT CLOCK AND DATA TIMING RELATIONSHIP
SNR in a DAC is dependent on the relationship between the
position of the clock edges and the point in time at which the
input data changes. The AD9753 is rising edge triggered, and
so exhibits SNR sensitivity when the data transition is close to
this edge. In general, the goal when applying the AD9753 is to
make the data transition close to the falling clock edge. This
becomes more important as the sample rate increases. Figure 16
shows the relationship of SNR to clock placement with different
sample rates. Note that the setup-and-hold times implied in
Figure 16 appear to violate the maximums stated in the Digital
Specifications of this data sheet. The variation in Figure 16 is
due to the skew present between data bits inherent in the digital
data generator used to perform these tests. Figure 16 is presented
to show the effects of violating setup-and-hold times and to
show the insensitivity of the AD9753 to clock placement when
data transitions fall outside of the so-called “bad window.” The
setup-and-hold times stated in the Digital Specifications table
were measured on a bit-by-bit basis, therefore eliminating the
skew present in the digital data generator. At higher data
rates, it becomes very important to account for the skew in
the input digital data when defining timing specifications.
Figure 15b. Differential Clock Interface
Because the output of the AD9753 can be updated at up to
300 MSPS, the quality of the clock and data input signals is
important in achieving the optimum performance. The drivers
of the digital data interface circuitry should be specified to
meet the minimum setup-and-hold times of the AD9753 as
well as its required min/max input logic level thresholds.
Digital signal paths should be kept short and run lengths matched
to avoid propagation delay mismatch. Inserting a low value resistor network (i.e., 20 Ω to 100 Ω) between the AD9753 digital
inputs and driver outputs may be helpful in reducing any overshooting and ringing at the digital inputs that contribute to data
feedthrough. For longer run lengths and high data update rates,
strip line techniques with proper termination resistors should be
considered to maintain “clean” digital inputs.
The external clock driver circuitry should provide the AD9753
with a low jitter clock input meeting the min/max logic levels
while providing fast edges. Fast clock edges help minimize any
jitter that will manifest itself as phase noise on a reconstructed
waveform. Thus, the clock input should be driven by the fastest
logic family suitable for the application.
REV. B
CLKCOM
Figure 16. SNR vs. Time of Data Transition
Relative to Clock Rising Edge
POWER DISSIPATION
The power dissipation, PD, of the AD9753 is dependent on several
factors that include the power supply voltages (AVDD and
DVDD), the full-scale current output I
, and the reconstructed digital input waveform. The
f
CLOCK
, the update rate
OUTFS
power dissipation is directly proportional to the analog supply current, I
is directly proportional to I
I
AVDD
, and the digital supply current, I
AVDD
, as shown in Figure 17,
OUTFS
DVDD
–15–
.
Page 16
AD9753
and is insensitive to f
both the digital input waveform, f
DVDD. Figure 18 shows I
f
) for various update rates. In addition, Figure 19 shows the
DAC
effect that the speed of f
. Conversely, I
CLOCK
CLOCK
as a function of the ratio (f
DVDD
has on the PLLVDD current, given
DAC
is dependent on
DVDD
, and digital supply,
the PLL divider ratio.
40
35
30
25
(mA)
20
AVDD
I
15
10
5
0
20
18
16
14
12
(mA)
10
DVDD
I
8
6
4
2
0
10
DIV SETTING 11
9
8
7
6
(mA)
5
DD
4
PLL_V
3
2
1
0
2.5 5.0 7.5 12.5 15.0 17.5
Figure 17. I
Figure 18. I
50 100 200 250
Figure 19. PLLVDD vs. f
I
(mA)
OUTFS
AVDD
300MSPS
200MSPS
100MSPS
50MSPS
25MSPS
RATIO
(f
OUT
vs. f
DVDD
DIV SETTING 10
f
(MHz)
DAC
vs. I
OUTFS
0.1
/
f
)
DAC
OUT/fDAC
17525 75 125 225 275
Ratio
DIV SETTING 01
DIV SETTING 00
DAC
20.010.00
3001500
10.010.001
OUT
APPLYING THE AD9753 OUTPUT CONFIGURATIONS
/
The following sections illustrate some typical output configurations for the AD9753. Unless otherwise noted, it is assumed
that I
is set to a nominal 20 mA. For applications requir-
OUTFS
ing the optimum dynamic performance, a differential output
configuration is suggested. A differential output configuration
may consist of either an RF transformer or a differential op amp
configuration. The transformer configuration provides the optimum high frequency performance and is recommended for any
application allowing for ac coupling. The differential op amp
configuration is suitable for applications requiring dc coupling,
a bipolar output, signal gain, and/or level shifting, within the
bandwidth of the chosen op amp.
A single-ended output is suitable for applications requiring a
unipolar voltage output. A positive unipolar output voltage will
result if I
OUTA
and/or I
sized load resistor, R
is connected to an appropriately
OUTB
, referred to ACOM. This configu-
LOAD
ration may be more suitable for a single-supply system requiring
a dc-coupled, ground referred output voltage. Alternatively, an
amplifier could be configured as an I-V converter, thus converting I
OUTA
or I
configuration provides the best dc linearity since I
is maintained at a virtual ground. Note that I
slightly better performance than I
into a negative unipolar voltage. This
OUTB
OUTA
OUTA
OUTB
.
provides
DIFFERENTIAL COUPLING USING A TRANSFORMER
An RF transformer can be used to perform a differential-tosingle-ended signal conversion, as shown in Figure 20. A
differentially-coupled transformer output provides the optimum
distortion performance for output signals whose spectral content
lies within the transformer’s pass band. An RF transformer such
as the Mini-Circuits T1-1T provides excellent rejection of
common-mode distortion (i.e., even-order harmonics) and noise
over a wide frequency range. When I
OUTA
and I
OUTB
are termi-
nated to ground with 50 Ω, this configuration provides 0 dBm
power to a 50 Ω load on the secondary with a DAC full-scale
current of 20 mA. A 2:1 transformer, such as the Coilcraft
WB2040-PC, can also be used in a configuration in which I
and I
are terminated to ground with 75 Ω. This configura-
OUTB
tion improves load matching and increases power to 2 dBm into
a 50 Ω load on the secondary. Transformers with different impedance ratios may also be used for impedance matching purposes.
Note that the transformer provides ac coupling only.
AD9753
I
OUTA
I
OUTB
MINI-CIRCUITS
T1-1T
R
LOAD
Figure 20. Differential Output Using a Transformer
The center tap on the primary side of the transformer must
be connected to ACOM to provide the necessary dc current
path for both I
appearing at I
OUTA
OUTA
and I
and I
. The complementary voltages
OUTB
OUTB
(i.e., V
OUTA
and V
OUTB
symmetrically around ACOM and should be maintained with
the specified output compliance range of the AD9753. A differential resistor, R
output of the transformer is connected to the load, R
, may be inserted in applications where the
DIFF
LOAD
–16–
or I
OUTB
OUTA
) swing
, via a
REV. B
Page 17
AD9753
AD9753
I
OUTA
I
OUTB
C
OPT
200
⍀
V
OUT
= I
OUTFS
ⴛ R
FB
R
FB
200
⍀
passive reconstruction filter or cable. R
is determined by the
DIFF
transformer’s impedance ratio and provides the proper source
termination that results in a low VSWR.
DIFFERENTIAL COUPLING USING AN OP AMP
An op amp can also be used to perform a differential-tosingle-ended conversion, as shown in Figure 21. The AD9753 is
configured with two equal load resistors, R
differential voltage developed across I
OUTA
, of 25 Ω. The
LOAD
and I
OUTB
is converted to a single-ended signal via the differential op amp
configuration. An optional capacitor can be installed across
I
OUTA
and I
, forming a real pole in a low-pass filter. The
OUTB
addition of this capacitor also enhances the op amp’s distortion performance by preventing the DAC’s high slewing output
from overloading the op amp’s input.
⍀
500
AD9753
I
I
OUTA
OUTB
C
OPT
⍀
25
⍀
225
⍀
225
500
⍀
25
AD8047
⍀
Figure 21. DC Differential Coupling Using an Op Amp
The common-mode rejection of this configuration is typically
determined by the resistor matching. In this circuit, the differential op amp circuit using the AD8047 is configured to
provide some additional signal gain. The op amp must operate
from a dual supply since its output is approximately ±1.0 V.
A high speed amplifier capable of preserving the differential
performance of the AD9753, while meeting other system level
objectives (i.e., cost, power), should be selected. The op amp’s
differential gain, its gain setting resistor values, and full-scale
output swing capabilities should all be considered when optimizing this circuit.
The differential circuit shown in Figure 22 provides the necessary level-shifting required in a single-supply system. In this
case, AVDD, which is the positive analog supply for both the
AD9753 and the op amp, is also used to level-shift the differential output of the AD9753 to midsupply (i.e., AVDD/2). The
AD8041 is a suitable op amp for this application.
R
represents the equivalent load resistance seen by I
LOAD
. The unused output (I
I
OUTB
ACOM directly or via a matching R
I
OUTFS
and R
can be selected as long as the positive compli-
LOAD
OUTA
or I
) can be connected to
OUTB
. Different values of
LOAD
OUTA
or
ance range is adhered to. One additional consideration in this
mode is the integral nonlinearity (INL), as discussed in the Analog
Outputs section. For optimum INL performance, the singleended, buffered voltage output configuration is suggested.
AD9753
I
OUTA
I
OUTB
I
OUTFS
= 20mA
25⍀
50⍀
V
OUTA
= 0V TO 0.5V
50⍀
Figure 23. 0 V to 0.5 V Unbuffered Voltage Output
SINGLE-ENDED BUFFERED VOLTAGE OUTPUT
Figure 24 shows a buffered single-ended output configuration in
which the op amp performs an I–V conversion on the AD9753
output current. The op amp maintains I
OUTA
(or I
OUTB
) at a
virtual ground, thus minimizing the nonlinear output impedance effect on the DAC’s INL performance as discussed in the
Analog Outputs section. Although this single-ended configuration typically provides the best dc linearity performance, its ac
distortion performance at higher DAC update rates may be
limited by the op amp’s slewing capabilities. The op amp provides a negative unipolar output voltage and its full-scale output
voltage is simply the product of R
FB
and I
. The full-scale
OUTFS
output should be set within the op amp’s voltage output swing
capabilities by scaling I
distortion performance may result with a reduced I
and/or RFB. An improvement in ac
OUTFS
OUTFS
, since
the signal current the op amp will be required to sink will
subsequently be reduced.
500⍀
AD9753
I
I
OUTA
OUTB
225⍀
500⍀25⍀25⍀
AD8041
1k⍀
AVDD
225⍀
C
OPT
Figure 22. Single-Supply DC Differential Coupled Circuit
of 25 Ω. In this case,
LOAD
OUTFS
SINGLE-ENDED UNBUFFERED VOLTAGE OUTPUT
Figure 23 shows the AD9753 configured to provide a unipolar
output range of approximately 0 V to 0.5 V for a doubly terminated 50 Ω cable since the nominal full-scale current, I
20 mA flows through the equivalent R
REV. B
, of
Figure 24. Unipolar Buffered Voltage Output
POWER AND GROUNDING CONSIDERATIONS, POWER SUPPLY REJECTION
Many applications seek high speed and high performance under
less than ideal operating conditions. In these applications, the
implementation and construction of the printed circuit board is
as important as the circuit design. Proper RF techniques must
be used for device selection, placement, and routing, as well as
power supply bypassing and grounding, to ensure optimum
performance. Figures 34 to 41 illustrate the recommended
printed circuit board ground, power, and signal plane layouts
that are implemented on the AD9753 evaluation board.
One factor that can measurably affect system performance is the
ability of the DAC output to reject dc variations or ac noise
superimposed on the analog or digital dc power distribution.
–17–
Page 18
AD9753
)
This is referred to as the Power Supply Rejection Ratio. For dc
variations of the power supply, the resulting performance of the
DAC directly corresponds to a gain error associated with the
DAC’s full-scale current, I
. AC noise on the dc supplies is
OUTFS
common in applications where the power distribution is generated by a switching power supply. Typically, switching power
supply noise will occur over the spectrum from tens of kHz to
several MHz. The PSRR versus the frequency of the AD9753
AVDD supply over this frequency range is shown in Figure 25.
85
80
75
70
65
60
PSRR (dB)
55
50
45
40
24 810
FREQUENCY (MHz
1260
Figure 25. Power Supply Rejection Ratio
Note that the units in Figure 25 are given in units of (amps out/
volts in). Noise on the analog power supply has the effect of
modulating the internal switches, and therefore the output
current. The voltage noise on AVDD will thus be added in a
nonlinear manner to the desired I
. Due to the relative
OUT
different size of these switches, PSRR is very code-dependent.
This can produce a mixing effect that can modulate low frequency power supply noise to higher frequencies. Worst-case
PSRR for either one of the differential DAC outputs will occur
when the full-scale current is directed toward that output. As a
result, the PSRR measurement in Figure 25 represents a worstcase condition in which the digital inputs remain static and the
full-scale output current of 20 mA is directed to the DAC output being measured.
An example serves to illustrate the effect of supply noise on the
analog supply. Suppose a switching regulator with a switching
frequency of 250 kHz produces 10 mV rms of noise and, for
simplicity sake (i.e., ignore harmonics), all of this noise is concentrated at 250 kHz. To calculate how much of this undesired
noise will appear as current noise superimposed on the DAC’s
full-scale current, I
, one must determine the PSRR in dB
OUTFS
using Figure 25 at 250 kHz. To calculate the PSRR for a given
, such that the units of PSRR are converted from A/V to
R
LOAD
V/V, adjust the curve in Figure 25 by the scaling factor 20 × Log
). For instance, if R
(R
LOAD
is 50 Ω, the PSRR is reduced
LOAD
by 34 dB, i.e., PSRR of the DAC at 250 kHz, which is 85 dB in
Figure 25, becomes 51 dB V
OUT/VIN
.
Proper grounding and decoupling should be a primary objective
in any high speed, high resolution system. The AD9753 features
separate analog and digital supply and ground pins to optimize
the management of analog and digital ground currents in a system.
In general, AVDD, the analog supply, should be decoupled to
ACOM, the analog common, as close to the chip as physically
possible. Similarly, DVDD, the digital supply, should be
decoupled to DCOM as close to the chip as physically possible.
For those applications that require a single 3.3 V supply for
both the analog and digital supplies, a clean analog supply may
be generated using the circuit shown in Figure 26. The circuit
consists of a differential LC filter with separate power supply
and return lines. Lower noise can be attained by using low ESR
type electrolytic and tantalum capacitors.
FERRITE
TTL/CMOS
LOGIC
CIRCUITS
3.3V
POWER SUPPLY
BEADS
100F
ELECT.
10F
TANT.
0.1F
CER.
AVDD
ACOM
Figure 26. Differential LC Filter for a Single 3.3 V Application
APPLICATIONS
QAM/PSK Synthesis
Quadrature modulation (QAM or PSK) consists of two baseband PAM (Pulse Amplitude Modulated) data channels. Both
channels are modulated by a common frequency carrier. However, the carriers for each channel are phase-shifted 90° from
each other. This orthogonality allows twice the spectral efficiency
(data for a given bandwidth) of digital data transmitted via AM.
Receivers can be designed to selectively choose the “in phase” and
“quadrature” carriers, and then recombine the data. The recombination of the QAM data can be mapped as points representing
digital words in a two dimensional constellation as shown in
Figure 27. Each point, or symbol, represents the transmission of
multiple bits in one symbol period.
0100 0101 0001 0000
0110 0111 0011 0010
1110 1111 1011 1010
1100 1101 1001 1000
Figure 27. 16 QAM Constellation, Gray Coded (Two
4-Level PAM Signals with Orthogonal Carriers)
Typically, the I and Q data channels are quadrature-modulated
in the digital domain. The high data rate of the AD9753 allows
extremely wideband (>10 MHz) quadrature carriers to be synthesized. Figure 28 shows an example of a 25 MSymbol/S QAM
signal, oversampled by 8 at a data rate of 200 MSPS, modulated onto a 25 MHz carrier and reconstructed using the
AD9753. The power in the reconstructed signal is measured
to be –11.92 dBm. In the first adjacent band, the power is
–76.86 dBm, while in the second adjacent band, the power is
–80.96 dBm.
–18–
REV. B
Page 19
AD9753
MARKER 1 [T1] RBW 5kHz RF ATT 0dB
–74.34dBm VBW 50kHz
–30
–40
–50
–60
–70
–80
–90
REF LV1 (dBm)
–100
–110
–120
–130
COMMENT A: 25 MSYMBOL, 64 QAM, CARRIER = 25MHz
9.71442886MHz SWT 12.5 s UNIT dBm
1
C11
START 100kHz
1 [T1]
CH PWR
ACP UP
ACP LOW
C11
C0
12.49MHz/ STOP 125MHz
Cu1
C0
+9.71442886MHz
–74.34dBm
–76.86dBm
–80.96dBm
–11.92dBm
Cu1
1RM
Figure 28. Reconstructed 64-QAM Signal at a 25 MHz IF
A figure of merit for wideband signal synthesis is the ratio of
signal power in the transmitted band to the power in an adjacent channel. In Figure 28, the adjacent channel power ratio
(ACPR) at the output of the AD9753 is measured to be 65 dB.
The limitation on making a measurement of this type is often
not the DAC but the noise inherent in creating the digital data
record using computer tools. To find how much this is limiting
the perceived DAC performance, the signal amplitude can be
reduced, as is shown in Figure 29. The noise contributed by the
DAC will remain constant as the signal amplitude is reduced.
When the signal amplitude is reduced to the level where the
noise floor drops below that of the spectrum analyzer, the
ACPR will fall off at the same rate that the signal level is being
reduced. Under the conditions measured in Figure 28, this
point occurs in Figure 29 at –10 dBFS. This shows that the
data record is actually degrading the measured ACPR by up to
10 dB.
A single-channel active mixer such as the Analog Devices AD8343
can then be used for the hop to the transmit frequency. Figure 30
shows an applications circuit using the AD9753 and the AD8343.
The AD8343 is capable of mixing carriers from dc to 2.5 GHz.
Figure 31 shows the result of mixing the signal in Figure 28 up
to a carrier frequency of 800 MHz. ACPR measured at the
output of the AD8343 is shown in Figure 31 to be 59 dB.
80
70
60
ACPR (dB)
50
40
–15 –5
AMPLITUDE (dBFS)
0–10–20
Figure 29. ACPR vs. Amplitude for QAM Carrier
DVD D AV DD
PORT 1
PORT 2
RSET2
1.9k⍀
DATA
INPUT
DATA
INPUT
FSADJ
CLK+ CLK–
0.1F
PLLLOCK
PLL/DIVIDER
INPUT
LATCHES
DAC
DAC
INPUT
LATCHES
LATCHES
AD9753
REFIO ACOM1 ACOM DCOM
I
OUTA
I
OUTB
50⍀
50⍀
0.1F
0.1F
68⍀ 68⍀
INPP
INPM
AD8343 ACTIVE MIXER
0.1F
LOINPUT
0.1F
M/A-COM ETC-1-1-13 WIDEBAND BALUN
Figure 30. QAM Transmitter Architecture Using AD9753 and AD8343 Active Mixer
OUTP
OUTM
LOIM
LOIP
REV. B
–19–
Page 20
AD9753
,
MARKER 1 [T2] RBW 10kHz RF ATT 0dB
–99.88dBm VBW 10kHz
–20
–30
–40
–50
–60
–70
–80
REF LV1 (dBm)
–90
–100
–110
–120
COMMENT A: 25 MSYMBOL
859.91983968MHz SWT 2.8 s UNIT dBm
1
2
C11
CENTER 860MHz
1 [T2]
CH PWR
ACP UP
ACP LOW
1 [T2]
2 [T2]
1
C11
C0
11MHz/ SPAN 110MHz
64 QAM CARRIER @ 825MHz
+859.91983968MHz
–49.91983968MHz
–49.91983968MHz
Cu1
C0
–99.88bBm,
–65.67dBm
–65.15dBm
–7.05dBm
33.10dB
33.10dB
Cu1
2MA
Figure 31. Signal of Figure 28 Mixed to Carrier
Frequency of 800 MHz
Effects of Noise and Distortion on Bit Error Rate (BER)
Textbook analyses of Bit Error Rate (BER) performance are
generally stated in terms of E (energy in watts-per-symbol or
watts-per-bit) and NO (spectral noise density in watts/Hz).
For QAM signals, this performance is shown graphically in
Figure 32. M represents the number of levels in each quadrature PAM signal (i.e., M = 8 for 64 QAM, M = 16 for 256 QAM).
Figure 32 implies gray coding in the QAM constellation, as well
as the use of matched filters at the receiver, which is typical.
The horizontal axis of Figure 32 can be converted to units of
energy/symbol by adding to the horizontal axis 10 log of the
number of bits in the desired curve. For instance, to achieve a
BER of 1e-6 with 64 QAM, an energy per bit of 20 dB is necessary. To calculate energy per symbol, we add 10 log(6), or
7.8 dB. 64 QAM with a BER of 1e-6 (assuming no source or
channel coding) can therefore theoretically be achieved with
an energy/symbol-to-noise (E/NO) ratio of 27.8 dB. Due to the
loss and interferers inherent in the wireless path, this signal-tonoise ratio must be realized at the receiver to achieve the given
bit error rate.
Distortion effects on BER are much more difficult to determine
accurately. Most often in simulation, the energies of the strongest
distortion components are root-sum-squared with the noise, and
the result is treated as if it were all noise. That being said, if the
example above of 64 QAM with the BER of 1e-6, using the
E/NO ratio is much greater than the worst-case SFDR, the noise
will dominate the BER calculation.
The AD9753 has a worst-case in-band SFDR of 47 dB at the
upper end of its frequency spectrum (see TPCs 4 and 7). When
used to synthesize high level QAM signals as described above,
noise, as opposed to distortion, will dominate its performance in
these applications.
1E–0
1E–1
1E–2
1E–3
1E–4
SYMBOL ERROR PROBABILITY
1E–5
1E–6
4 QAM
16 QAM
SNR/ BIT (dB)
64 QAM
10 15
2050
20
Figure 32. Probability of a Symbol Error for QAM
Pseudo Zero Stuffing/IF Mode
The excellent dynamic range of the AD9753 allows its use in
applications where synthesis of multiple carriers is desired. In
addition, the AD9753 can be used in a pseudo zero stuffing
mode that improves dynamic range at IF frequencies. In this
mode, data from the two input channels is interleaved to the
DAC, which is running at twice the speed of either of the input
ports. However, the data at Port 2 is held constant at midscale.
The effect of this is shown in Figure 33. The IF signal is the
image, with respect to the input data rate, of the fundamental. Normally, the sinx/x response of the DAC will attenuate
this image. Zero stuffing improves the pass-band flatness so that
the image amplitude is closer to that of the fundamental signal. Zero stuffing can be an especially useful technique in the
synthesis of IF signals.
0
–10
AMPLITUDE
–20
–30
EFFECT OF SINX/X ROLL-OFF
–40
–50
AMPLITUDE
OF IMAGE
WITHOUT
ZERO STUFFING
FREQUENCY (Normalized to Input Data Rate)
OF IMAGE
USING
ZERO STUFFING
1 1.5
20.50
Figure 33. Effects of Pseudo Zero Stuffing on
Spectrum of AD9753
–20–
REV. B
Page 21
AD9753
EVALUATION BOARD
The AD9753-EB is an evaluation board for the AD9753 TxDAC.
Careful attention to layout and circuit design, combined with
prototyping area, allows the user to easily and effectively evaluate the AD9753 in different modes of operation.
Referring to Figures 34 and 35, the AD9753’s performance can
be evaluated differentially or single-ended either using a transformer, or directly coupling the output. To evaluate the output
differentially using the transformer, it is recommended that
either the Mini-Circuits T1-1T (through-hole) or the Coilcraft
TTWB-1-B (SMT) be placed in the position of T1 on the evaluation board. To evaluate the output either single-ended or directcoupled, remove the transformer and bridge either BL1 or BL2.
The digital data to the AD9753 comes from two ribbon cables that
interface to the 40-lead IDC connectors P1 and P2. Proper termination or voltage scaling can be accomplished by installing the
resistor pack networks RN1–RN12. RN1, 4, 7, and 10 are 22 Ω
DIP resistor packs and should be installed as they help reduce the
digital edge rates and therefore peak current on the inputs.
A single-ended clock can be applied via J3. By setting the SE/
DIFF labeled jumpers J2, 3, 4, and 6, the input clock can be
directed to the CLK+/CLK– inputs of the AD9753 in either a
single-ended or differential manner. If a differentially applied
clock is desired, a Mini-Circuits T1-1T transformer should be
used in the position of T2. Note that with a single-ended square
wave clock input, T2 must be removed. A clock can also be
applied via the ribbon cable on Port 1 (P1), Pin 33. By inserting
the EDGE jumper (JP1), this clock will be applied to the CLK+
input of the AD9753. JP3 should be set in its SE position in this
application to bias CLK– to 1/2 the supply voltage.
The AD9753’s PLL clock multiplier can be enabled by inserting
JP7 in the IN position. As described in the Typical Performance
Characteristics and Functional Description sections, with the PLL
enabled, a clock at 1/2 the output data rate should be applied as
described in the last paragraph. The PLL takes care of the internal
2× frequency multiplication and all internal timing requirements.
In this application, the PLLLOCK output indicates when lock
is achieved on the PLL. With the PLL enabled, the DIV0 and
DIV1 jumpers (JP8 and JP9) provide the PLL divider ratio as
described in Table I.
The PLL is disabled when JP7 is in the EX setting. In this mode, a
clock at the speed of the output data rate must be applied to the
clock inputs. Internally, the clock is divided by 2. For data
synchronization, a 1⫻ clock is provided on the PLLLOCK pin
in this application. Care should be taken to read the timing
requirements described earlier in the data sheet for optimum
performance. With the PLL disabled, the DIV0 and DIV1 jumpers
define the mode (interleaved, noninterleaved) as described in
Table II.
REV. B
–21–
Page 22
AD9753
RN2
P1B13
P1B12
P1B11
P1B10
P1B09
P1B08
P1B07
P1B06
P1B05
P1B04
P1B03
P1B02
P1B01
P1B00
OUT15
OUT16
P2B13
P2B05
P2B04
P2B03
P2B02
P2B01
P2B00
P2OUT15
P2OUT16
VALUE
1
2
3
4
5
6
7
8
9
10
RN5
VALUE
1
2
3
4
5
6
7
8
9
10
RN8
VALUE
1
2
3
4
5
6
7
8
9
10
RN11
VALUE
1
2
3
4
5
6
7
8
9
10
1O15
JP10
2OUT15
2OUT16
RN1
VALUE
1B13
2
P1
P1
4
P1
P1
6
P1
P1
8
P1
P1
10
P1
P1
12
P1
P1
14
P1
P1
16
P1
P1
18
P1
P1
20
P1
P1
22
P1
P1
24
P1
P1
26
P1
P1
28
P1
P1
30
P1
P1
32
P1
P1
P1
34
P1
P1
36
P1
P1
38
P1
P1
40
P1
2
P2
P2
4
P2
P2
6
P2
P2
8
P2
P2
10
P2
P2
12
P2
P2
14
P2
P2
16
P2
P2
P2
P2
18
P2
20
P2
P2
P2
22
P2
24
P2
P2
P2
26
28
P2
P2
P2
P2
30
P2
P2
32
34
P2
P2
P2
36
P2
P2
38
P2
P2
40
P2
116
1
1B12
215
3
1B11
314
5
1B10
413
7
1B09
512
9
1B08
611
11
1B07
710
13
1B06
89
15
RN4
VALUE
1B05
116
17
1B04
215
19
1B03
314
21
1B02
413
23
1B01
512
25
1B00
611
27
1O17
710
29
89
31
33
1O15
35
1O16
37
39
11
13
15
17
19
21
23
25
27
29
31
RN7
VALUE
2B13
116
1
2B12 P2B12
215
3
2B11 P2B11
314
5
2B10 P2B10
413
7
2B09 P2B09
512
9
2B08 P2B08
611
2B07 P2B07
710
2B06 P2B06
89
RN10
VALUE
2B05
116
2B04
215
2B03
314
2B02
413
2B01
512
2B00
611
710
89
33
35
37
39
1B13
1B12
1B11
1B10
1B09
1B08
1B07
1B06
1B05
1B04
1B03
1B02
1B01
1B00
2B13
2B12
2B11
2B10
2B09
2B08
2B07
2B06
2B05
2B04
2B03
2B02
2B01
2B00
1O16
1O17
RN3
VALUE
1
2
3
4
5
6
7
8
9
10
RN6
VALUE
1
2
3
4
5
6
7
8
9
10
RN9
VALUE
1
2
3
4
5
6
7
8
9
10
RN12
VALUE
1
2
3
4
5
6
7
8
9
10
J1
1
DVDD
P1B13 MSB
P1B12
P1B11
P1B10
P1B09
P1B08
P1B07
P1B06
P1B05
P1B04
P1B03
P1B02
P1B01
P1B00 LSB
PLANE
MSB
P2B13
P2B12
P2B11
P2B10
P2B09
P2B08
DGND: 3,4,5
13
14
15
16
17
18
19
20
21
22
23
24
2
12 11 10 9 8 3 2 176 54
AD9751/AD9753/AD9755
25
26 27 28 29 30 31 32 33 34 35 36
P2B07
P2B06
P2B05
P2B04
P2B03
P2B02
P2B01
P2B00 LSB
NOTES
1. ALL DIGITAL INPUTS FROM RN1–RN12
MUST BE OF EQUAL LENGTH.
2. ALL DECOUPLING CAPS TO BE LOCATED
AS CLOSE AS POSSIBLE TO DUT,
PREFERABLY UNDER DUT ON BOTTOM
SIGNAL LAYER.
3. CONNECT GNDS UNDER DUT USING
BOTTOM SIGNAL LAYER.
4. CREATE PLANE CAPACITOR WITH 0.007"
DIELECTRIC BETWEEN LAYERS 2 AND 3.
TP4
TP5
BLK
TP6
BLK
DVDD PLANE
U1
TP7
BLK
CLK–
BLK
CLK+
TP8
RESET
48
47
46
45
44
43
42
41
40
39
38
37
BLK
CLKVDD
LPF
IA
IB
AVDD PLANE
1
DIV1
123
DIV0
TP9
BLK
EDGE
EXT
PLLVDD
50⍀
10pF
FSADJ
REFIO
JP8
AB
2
JP9
AB
TP10
BLK
R3
C10
P
OUT16
2
PLANE
TP1
TP2
3
TP12
A
B
WHT
WHT
1
3
R4
50⍀
P
C11
1.0F
R5
392⍀
BLK
JP5
R10
OPT
RESET
TP3
WHT
NOTE:
SHIELD AROUND
R5 AND C11 ARE
CONNECTED TO
PLLVDD
R2
50⍀
C9
10pF
3
2
1
R1
1.91k⍀
C12
0.1F
AVDD_PLANE
BL1
T1
S
BL2
PLANE
4
6
P
I
OUT
J5
1
2
Figure 34. Evaluation Board Circuitry
–22–
REV. B
Page 23
OUT15
AD9753
DVDD
J8
DGND
J9
AVDD
J10
AGND
J11
CLKVDD
J12
CLKGND
J13
1
1
1
1
1
1
CLK+
CLK–
L1
FBEAD
12
L2
FBEAD
12
L3
FBEAD
12
P
3
C13
10F
10V
C14
10F
10V
C15
10F
10V
B
2
JP6
A
R8
50⍀
1
TP13
TP14
TP15
TP16
TP17
TP11
P
RED
BLK
RED
BLK
RED
BLK
SE
DF
JP7
A
2
B
CKLVDD
1
JP3
3
DVDD PLANE
AVDD PLANE
1
A
2
B
3
JP1EDGE
SE
DF
R9
1k⍀
R7
C16
1k⍀
0.1F
P
CLKVDD
PLLVDD PLANE
2
1
A
B
3
JP2
T2
3
2
1
S
DVDD PLANE
JP4
4
6
P
P
CLK
1
DF
J3
2
PGND: 3, 4, 5
P
U1 BYPASS CAPS
PINS 5, 6
C1
0.1F
PINS 41, 44
C5
0.1F
PINS 45, 47
C7
0.1F
P
PINS 21, 22
C2
1FC30.1FC41F
AVDD PLANE
C6
1F
CLKVDD
C8
1F
Figure 35. Evaluation Board Clock Circuitry
REV. B
–23–
Page 24
AD9753
Figure 36. Evaluation Board, Assembly—Top
Figure 37. Evaluation Board, Assembly—Bottom
–24–
REV. B
Page 25
AD9753
Figure 38. Evaluation Board, Top Layer
REV. B
Figure 39. Evaluation Board, Layer 2, Ground Plane
–25–
Page 26
AD9753
Figure 40. Evaluation Board, Layer 3, Power Plane
Figure 41. Evaluation Board, Bottom Layer
–26–
REV. B
Page 27
OUTLINE DIMENSIONS
48-Lead Low Profile Quad Flat Package [LQFP]
(ST-48)
Dimensions shown in millimeters
AD9753
1.45
1.40
1.35
0.15
0.05
10ⴗ
6ⴗ
2ⴗ
SEATING
PLANE
ROTATED 90ⴗ CCW
VIEW A
0.08 MAX
COPLANARITY
0.75
0.60
0.45
SEATING
PLANE
0.20
0.09
7ⴗ
3.5ⴗ
0ⴗ
COMPLIANT TO JEDEC STANDARDS MS-026BBC
1.60
MAX
VIEW A
1
12
0.50
BSC
48
13
9.00 BSC
PIN 1
TOP VIEW
(PINS DOWN)
SQ
37
24
0.27
0.22
0.17
36
25
7.00
BSC SQ
REV. B
–27–
Page 28
AD9753
Revision History
Location Page
9/03—Data Sheet changed from REV. A to REV. B.
Changes to GENERAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Changes to PRODUCT HIGHLIGHTS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Changes to DIGITAL SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Changes to ORDERING GUIDE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Changes to Figure 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Changes to Figure 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Changes to FUNCTIONAL DESCRIPTION section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Changes to Figure 6 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Changes to Figure 9 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Changes to Figure DIGITAL INPUTS section . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Changes to Figure 34 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
1/03—Data Sheet changed from REV. 0 to REV. A.
Changes to DIGITAL SPECIFICATIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
Changes to Figure 1 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Changes to Figure 2 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Changes to Figure 11 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Updated OUTLINE DIMENSIONS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
C02251–0–9/03(B)
REV. B–28–
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