Datasheet AD9260 Datasheet (Analog Devices)

Page 1

High-Speed Oversampling CMOS

ADC with 16-Bit Resolution

FEATURES Monolithic 16-Bit, Oversampled A/D Converter 8 Oversampling Mode, 20 MSPS Clock

2.5 MHz Output Word Rate

1.01 MHz Signal Passband w/0.004 dB Ripple Signal-to-Noise Ratio: 88.5 dB Total Harmonic Distortion: –96 dB Spurious Free Dynamic Range: 100 dB Input Referred Noise: 0.6 LSB Selectable Oversampling Ratio: 1, 2, 4, 8 Selectable Power Dissipation: 150 mW to 585 mW 85 dB Stopband Attenuation

0.004 dB Passband Ripple Linear Phase Single +5 V Analog Supply, +5 V/+3 V Digital Supply Synchronize Capability for Parallel ADC Interface Twos-Complement Output Data 44-Lead MQFP

at a 2.5 MHz Output Word Rate

AD9260

FUNCTIONAL BLOCK DIAGRAM

VINA

VINB

REF TOP

REF

BOTTOM

COMMON

MODE

VREF

SENSE

REFCOM

AVSS

AVDD

MULTIBIT

SIGMA-DELTA

MODULATOR

AD9260

REFERENCE

BUFFER

BANDGAP

REFERENCE

AVDD

AVSS

RESET/

AVSS

AVDD

12-BIT: 20MHz

16-BIT: 10MHz

16-BIT: 5MHz

16-BIT: 2.5MHz

BIAS

CIRCUIT

SYNC

DVSS DVDD

DIGITAL

DEMODULATOR

STAGE 1:2X

DECIMATION

FILTER

STAGE 2:2X

DECIMATION

FILTER

STAGE 3:2X

DECIMATION

FILTER

CLOCK

BUFFER

MODE

DRVSS

DRVDD

OUTPUT REGISTER

OUTPUT MODE MULTIPLEXER

OTR

BIT1–BIT16

DAV

READ

PRODUCT DESCRIPTION

The AD9260 is a 16-bit, high-speed oversampled analog-todigital converter (ADC) that offers exceptional dynamic range over a wide bandwidth. The AD9260 is manufactured on an advanced CMOS process. High dynamic range is achieved with an oversampling ratio of 8× through the use of a proprietary technique that combines the advantages of sigma-delta and pipeline converter technologies.

The AD9260 is a switched-capacitor ADC with a nominal fullscale input range of 4 V. It offers a differential input with 60 dB of common-mode rejection of common-mode signals. The signal range of each differential input is ±1 V centered on a 2.0 V common-mode level.

The on-chip decimation filter is configured for maximum performance and flexibility. A series of three half-band FIR filter stages provide 8× decimation filtering with 85 dB of stopband attenuation and 0.004 dB of passband ripple. An onboard digital multiplexer allows the user to access data from the various stages of the decimation filter.

The on-chip programmable reference and reference buffer amplifier are configured for maximum accuracy and flexibility. An external reference can also be chosen to suit the user’s specific dc accuracy and drift requirements.

MODECLKBIAS ADJUST

The AD9260 operates on a single +5 V supply, typically consuming 585 mW of power. A power scaling circuit is provided allowing the AD9260 to operate at power consumption levels as low as 150 mW at reduced clock and data rates. The AD9260 is available in a 44-lead MQFP package and is specified to operate over the industrial temperature range.

PRODUCT HIGHLIGHTS

The AD9260 is fabricated on a very cost effective CMOS process. High-speed, precision mixed-signal analog circuits are combined with high-density digital filter circuits.

The AD9260 offers a complete single-chip 16-bit sampling ADC with a 2.5 MHz output data rate in a 44-lead MQFP.

Selectable Internal Decimation Filtering—The AD9260 provides a high-performance decimation filter with 0.004 dB passband ripple and 85 dB of stopband attenuation. The filter is configurable with options for 1×, 2×, 4×, and 8× decimation.

Power Scaling—The AD9260 consumes a low 585 mW of power at 16-bit resolution and 2.5 MHz output data rate. Its power can be scaled down to as low as 150 mW at reduced clock rates.

Single Supply— Both of the analog and digital portions of the AD9260 can operate off of a single +5 V supply simplifying system power supply design. The digital logic will also accommodate a single +3 V supply for reduced power.

REV. B

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 2000

Page 2

AD9260–SPECIFICATIONS

CLOCK INPUT FREQUENCY RANGE

Parameter—Decimation Factor (N) AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units

CLOCK INPUT (Modulator Sample Rate, f

)1111kHz min

CLOCK

20 20 20 20 MHz max

OUTPUT WORD RATE (FS = f

/N) 0.125 0.250 0.500 1 kHz min

CLOCK

2.5 5 10 20 MHz max

Specifications subject to change without notice

(AVDD = +5 V, DVDD = +3 V, DRVDD = +3 V, f

DC SPECIFICATIONS

arameter—Decimation Factor (N) AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units

unless otherwise noted, R

BIAS

= 2 k)

= 20 MSPS, V

CLOCK

= +2.5 V, Input CML = 2.0 V T

REF

MIN

to T

RESOLUTION 16 16 16 12 Bits min

INPUT REFERRED NOISE (TYP)

1.0 V Reference 1.40 2.4 6.0 1.3 LSB rms typ

2.5 V Reference

0.68 (90.6) 1.2 (86) 3.7 (76) 1.0 (63.2) LSB rms typ (dB typ)

ACCURACY

Integral Nonlinearity (INL) ± 0.75 ± 0.75 ± 0.75 ± 0.3 LSB typ Differential Nonlinearity (DNL) ±0.50 ± 0.50 ± 0.50 ± 0.25 LSB typ No Missing Codes 16 16 16 12 Bits Guaranteed Offset Error 0.9 (0.5) (0.5) (0.5) (0.5) % FSR max (typ @ +25°C) Gain Error Gain Error

2.75 (0.66) (0.66) (0.66) (0.66) % FSR max (typ @ +25°C)

1.35 (0.7) (0.7) (0.7) (0.7) % FSR max (typ @ +25°C)

TEMPERATURE DRIFT

Offset Error 2.5 2.5 2.5 2.5 ppm/°C typ Gain Error Gain Error

22 22 22 22 ppm/°C typ

7.0 7.0 7.0 7.0 ppm/°C typ

MAX

POWER SUPPLY REJECTION

AVDD, DVDD, DRVDD (+5 V ± 0.25 V) 0.06 0.06 0.06 0.06 % FSR max

ANALOG INPUT

Input Span

= 1.0 V 1.6 1.6 1.6 1.6 V p-p Diff. max

REF

= 2.5 V 4.0 4.0 4.0 4.0 V p-p Diff. max

REF

Input (VINA or VINB) Range +0.5 +0.5 +0.5 +0.5 V min

+AVDD – 0.5 +AVDD – 0.5 +AVDD – 0.5 +AVDD – 0.5 V max

Input Capacitance 10.2 10.2 10.2 10.2 pF typ

INTERNAL VOLTAGE REFERENCE

Output Voltage (1 V Mode) 1111 V typ Output Voltage Error (1 V Mode) ± 14 ±14 ± 14 ±14 mV max Output Voltage (2.5 V Mode) 2.5 2.5 2.5 2.5 V typ Output Voltage Error (2.5 V Mode) ± 35 ± 35 ± 35 ± 35 mV max Load Regulation

1 V REF 0.5 0.5 0.5 0.5 mV max

2.5 V REF 2.0 2.0 2.0 2.0 mV max

REFERENCE INPUT RESISTANCE 8888 kΩ

–2–

REV. B

Page 3

arameter—Decimation Factor (N) AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units

POWER SUPPLIES

Supply Voltages

AVDD +5 +5 +5 +5 V (± 5%) DVDD and DRVDD +5.5 +5.5 +5.5 +5.5 V max

+2.7 +2.7 +2.7 +2.7 V min

Supply Current

IAVDD 115 115 115 115 mA typ

134 mA max

IDVDD 12.5 10.3 6.5 2.4 mA typ

3.5 mA max

IDRVDD 0.450 0.850 1.7 2.6 mA typ

POWER CONSUMPTION 613 608 600 585 mW typ

630 mW max

NOTES

VINA and VINB Connect to DUT CML.

Including Internal 2.5 V reference.

Excluding Internal 2.5 V reference.

Load regulation with 1 mA load Current (in addition to that required by AD9260).

Specifications subject to change without notice.

AC SPECIFICATIONS

(AVDD = +5 V, DVDD = +3 V, DRVDD = +3 V, f unless otherwise noted, R

= 2 k)

BIAS

= 20 MSPS, V

CLOCK

= +2.5 V, Input CML = 2.0 V T

REF

Parameter—Decimation Factor (N) AD9260(8) AD9260(4) AD9260(2) AD9260(1) Units

DYNAMIC PERFORMANCE

INPUT TEST FREQUENCY: 100 kHz (typ)

Signal-to-Noise Ratio (SNR)

Input Amplitude = –0.5 dBFS 88.5 82 74 63 dB typ Input Amplitude = –6.0 dBFS 82.5 78 68 58 dB typ

SNR and Distortion (SINAD)

Input Amplitude = –0.5 dBFS 87.5 82 74 63 dB typ Input Amplitude = –6.0 dBFS 82 77.5 69 58 dB typ

Total Harmonic Distortion (THD)

Input Amplitude = –0.5 dBFS –96 –96 –97 –98 dB typ Input Amplitude = –6.0 dBFS –93 –98 –96 –98 dB typ

Spurious Free Dynamic Range (SFDR)

Input Amplitude = –0.5 dBFS 100 98 98 88 dB typ Input Amplitude = –6.0 dBFS 94 100 94 84 dB typ

INPUT TEST FREQUENCY: 500 kHz

Signal-to-Noise Ratio (SNR)

Input Amplitude = –0.5 dBFS 86.5 82 74 63 dB typ

80.5 dB min

Input Amplitude = –6.0 dBFS 82.5 77 68 58 dB typ

SNR and Distortion (SINAD)

Input Amplitude = –0.5 dBFS 86.0 81 74 63 dB typ

80.0 dB min

Input Amplitude = –6.0 dBFS 82.0 77 68 58 dB typ

Total Harmonic Distortion (THD)

Input Amplitude = –0.5 dBFS –97.0 –92 –89 –86 dB typ

–90.0 dB max

Input Amplitude = –6.0 dBFS –95.5 –96 –89 –86 dB typ

Spurious Free Dynamic Range (SFDR)

Input Amplitude = –0.5 dBFS 99.0 92 91 88 dB typ

90.0 dB max

Input Amplitude = –6.0 dBFS 98 100 91 82 dB typ

AD9260

to T

MIN

MAX

–3–REV. B

Page 4

AD9260–SPECIFICATIONS

AC SPECIFICATIONS (Continued)

Parameter—Decimation Factor (N) AD9260 (8) AD9260 (4) AD9260 (2) AD9260 (1) Units

DYNAMIC PERFORMANCE (Continued)

INPUT TEST FREQUENCY: 1.0 MHz (typ)

Signal-to-Noise Ratio (SNR)

Input Amplitude = –0.5 dBFS 85 82 74 63 dB typ Input Amplitude = –6.0 dBFS 80 76 68 58 dB typ

SNR and Distortion (SINAD)

Input Amplitude = –0.5 dBFS 84.5 81 74 63 dB typ Input Amplitude = –6.0 dBFS 80 76 69 58 dB typ

Total Harmonic Distortion (THD)

Input Amplitude = –0.5 dBFS –102 –96 –82 –79 dB typ Input Amplitude = –6.0 dBFS –96 –94 –84 –77 dB typ

Spurious Free Dynamic Range (SFDR)

Input Amplitude = –0.5 dBFS 105 98 83 80 dB typ Input Amplitude = –6.0 dBFS 98 96 87 80 dB typ

INPUT TEST FREQUENCY: 2.0 MHz (typ)

Signal-to-Noise Ratio (SNR)

Input Amplitude = –0.5 dBFS 82 74 63 dB typ Input Amplitude = –6.0 dBFS 76 68 58 dB typ

SNR and Distortion (SINAD)

Input Amplitude = –0.5 dBFS 81 73 62 dB typ Input Amplitude = –6.0 dBFS 76 69 58 dB typ

Total Harmonic Distortion (THD)

Input Amplitude = –0.5 dBFS –101 –80 –75 dB typ Input Amplitude = –6.0 dBFS –95 –80 –76 dB typ

Spurious Free Dynamic Range (SFDR)

Input Amplitude = –0.5 dBFS 104 80 78 dB typ Input Amplitude = –6.0 dBFS 100 83 79 dB typ

INPUT TEST FREQUENCY: 5.0 MHz (typ)

Signal-to-Noise Ratio (SNR)

Input Amplitude = –0.5 dBFS 59 dB typ Input Amplitude = –6.0 dBFS 57 dB typ

SNR and Distortion (SINAD)

Input Amplitude = –0.5 dBFS 58 dB typ Input Amplitude = –6.0 dBFS 57 dB typ

Total Harmonic Distortion (THD)

Input Amplitude = –0.5 dBFS –58 dB typ Input Amplitude = –6.0 dBFS –67 dB typ

Spurious Free Dynamic Range (SFDR)

Input Amplitude = –0.5 dBFS 59 dB typ Input Amplitude = –6.0 dBFS 70 dB typ

INTERMODULATION DISTORTION

1 = 475 kHz, fIN2 = 525 kHz –93 –91 –91 –83 dBFS typ

fIN1 = 950 kHz, fIN2 = 1.050 MHz –95 –86 –85 –83 dBFS typ

DYNAMIC CHARACTERISTICS

Full Power Bandwidth 75 75 75 75 MHz typ Small Signal Bandwidth (AIN = –20 dBFS) 75 75 75 75 MHz typ Aperture Jitter 2222ps rms typ

Specifications subject to change without notice.

–4–

REV. B

Page 5

AD9260

DIGITAL FILTER CHARACTERISTICS

Parameter AD9260 Units

8× DECIMATION (N = 8)

Passband Ripple 0.00125 dB max Stopband Attenuation 82.5 dB min Passband 0 MHz min

0.605 × (f

Stopband 1.870 × (f

18.130 × (f

Passband/Transition Band Frequency

(–0.1 dB Point) 0.807 × (f

(–3.0 dB Point) 1.136 × (f Absolute Group Delay Group Delay Variation 0 µs max Settling Time (to ±0.0007%)

13.55 × (20 MHz/f

24.2 × (20 MHz/f

4× DECIMATION (N = 4)

Passband Ripple 0.001 dB max Stopband Attenuation 82.5 dB min Passband 0 MHz min

1.24 × (f

Stopband 3.75 × (f

16.25 × (f

Passband/Transition Band Frequency

(–0.1 dB Point) 1.61 × (f

(–3.0 dB Point) 2.272 × (f Absolute Group Delay Group Delay Variation 0 µs max Settling Time (to ±0.0007%)

2.90 × (20 MHz/f

5.05 × (20 MHz/f

2× DECIMATION (N = 2)

Passband Ripple 0.0005 dB max Stopband Attenuation 85.5 dB min Passband 0 MHz min

2.491 × (f

Stopband 7.519 × (f

12.481 × (f

Passband/Transition Band Frequency

(–0.1 dB Point) 3.231 × (f

(–3.0 dB Point) 4.535 × (f Absolute Group Delay Group Delay Variation 0 µs max Settling Time (to ±0.0007%)

0.80 × (20 MHz/f

1.40 × (20 MHz/f

1× DECIMATION (N = 1)

Propagation Delay: t

PROP

13 ns max

Absolute Group Delay (225 × (20 MHz/f

NOTES

To determine “overall” Absolute Group Delay and/or Settling Time inclusive of delay from the sigma-delta modulator, add Absolute Group Delay and/or Settling Time pertaining to specific decimation mode to the Absolute Group Delay specified in 1 × decimation.

Specifications subject to change without notice.

/20 MHz) MHz max

CLOCK

/20 MHz) MHz min

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz min

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz min

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

/20 MHz) MHz max

CLOCK

) µs max

CLOCK

) µs max

CLOCK

) µs max

CLOCK

) µs max

CLOCK

) µs max

CLOCK

) µs max

CLOCK

)) + t

CLOCK

PROP

ns max

–5–REV. B

Page 6

)

AD9260

–Digital Filter Characteristics

–20

–40

–60

MAGNITUDE – dB

–80

–100

–120

0 1.0

0.2 0.4 0.6 0.8 FREQUENCY (NORMALIZED TO 

Figure 1a. 8× FIR Filter Frequency Response

–20

–40

–60

1.2

1.0

0.8

0.6

0.4

0.2

NORMALIZED OUTPUT RESPONSE

–0.2

–0.4

0 300200100

CLOCK PERIODS – RELATIVE TO CLK

400 500

Figure 1b. 8× FIR Filter Impulse Response

1.0

0.8

0.6

0.4

MAGNITUDE – dB

–80

–100

–120

0.2 0.4 0.6 0.8

0 1.0

FREQUENCY (NORMALIZED TO )

Figure 2a. 4× FIR Filter Frequency Response

–20

–40

–60

MAGNITUDE – dB

–80

–100

–120

0.2 0.4 0.6 0.8

0 1.0

FREQUENCY (NORMALIZED TO )

Figure 3a. 2× FIR Filter Frequency Response

1.2

0.2

NORMALIZED OUTPUT RESPONSE

–0.2

10 100 11020 30 40 50 60 70 80 90

CLOCK PERIODS – RELATIVE TO CLK

Figure 2b. 4× FIR Filter Impulse Response

1.0

0.8

0.6

0.4

0.2

NORMALIZED OUTPUT RESPONSE

–0.2

CLOCK PERIODS – RELATIVE TO CLK

Figure 3b. 2× FIR Filter Impulse Response

201510

–6–

REV. B

Page 7

AD9260

Table I. Integer Filter Coefficients for First Stage Decimation Filter (23-Tap Halfband FIR Filter)

Lower Upper Integer Coefficient Coefficient Value

H(1) H(23) –1 H(2) H(22) 0 H(3) H(21) 13 H(4) H(20) 0 H(5) H(19) –66 H(6) H(18) 0 H(7) H(17) 224 H(8) H(16) 0 H(9) H(15) –642 H(10) H(14) 0 H(11) H(13) 2496 H(12) 4048

Table II. Integer Filter Coefficients for Second Stage Decimation Filter (43-Tap Halfband FIR Filter)

Lower Upper Integer Coefficient Coefficient Value

H(1) H(43) 3 H(2) H(42) 0 H(3) H(41) –12 H(4) H(40) 0 H(5) H(39) 35 H(6) H(38) 0 H(7) H(37) –83 H(8) H(36) 0 H(9) H(35) 172 H(10) H(34) 0 H(11) H(33) –324 H(12) H(32) 0 H(13) H(31) 572 H(14) H(30) 0 H(15) H(29) –976 H(16) H(28) 0 H(17) H(27) 1680 H(18) H(26) 0 H(19) H(25) –3204 H(20) H(24) 0 H(21) H(23) 10274 H(22) 16274

NOTE: The composite filter undecimated coefficients (i.e., impulse response) in the 4× decimation mode can be determined by convolving the first stage filter taps with a “zero stuffed” version of the second stage filter taps (i.e., insert one zero between samples). Similarly, the composite filter coefficients in the 8× decimation mode can be determined by convolving the taps of the composite 4× decimation mode (as previously determined) with a “zero stuffed” version of the third stage filter taps (i.e., insert three zeros between samples).

Table III. Integer Filter Coefficients for Third Stage Decimation Filter (107-Tap Halfband FIR Filter)

Lower Upper Integer Coefficient Coefficient Value

H(1) H(107) –1 H(2) H(106) 0 H(3) H(105) 2 H(4) H(104) 0 H(5) H(103) –2 H(6) H(102) 0 H(7) H(101) 3 H(8) H(100) 0 H(9) H(99) –3 H(10) H(98) 0 H(11) H(97) 1 H(12) H(96) 0 H(13) H(95) 3 H(14) H(94) 0 H(15) H(93) –12 H(16) H(92) 0 H(17) H(91) 27 H(18) H(90) 0 H(19) H(89) –50 H(20) H(88) 0 H(21) H(87) 85 H(22) H(86) 0 H(23) H(85) –135 H(24) H(84) 0 H(25) H(83) 204 H(26) H(82) 0 H(27) H(81) –297 H(28) H(80) 0 H(29) H(79) 420 H(30) H(78) 0 H(31) H(77) –579 H(32) H(76) 0 H(33) H(75) 784 H(34) H(74) 0 H(35) H(73) –1044 H(36) H(72) 0 H(37) H(71) 1376 H(38) H(70) 0 H(39) H(69) –1797 H(40) H(68) 0 H(41) H(67) 2344 H(42) H(66) 0 H(43) H(65) –3072 H(44) H(64) 0 H(45) H(63) 4089 H(46) H(62) 0 H(47) H(61) –5624 H(48) H(60) 0 H(49) H(59) 8280 H(50) H(58) 0 H(51) H(57) –14268 H(52) H(56) 0 H(53) H(55) 43520 H(54) 68508

–7–REV. B

Page 8

AD9260–SPECIFICATIONS

DIGITAL SPECIFICATIONS

(AVDD = +5 V, DVDD = +5 V, T

MIN

to T

unless otherwise noted)

MAX

Parameter AD9260 Units

CLOCK

AND LOGIC INPUTS

High-Level Input Voltage

(DVDD = +5 V) +3.5 V min (DVDD = +3 V) +2.1 V max

Low-Level Input Voltage

(DVDD = +5 V) +1.0 V min

(DVDD = +3 V) +0.9 V max High-Level Input Current (V Low-Level Input Current (V

= DVDD) ± 10 µA max

= 0 V) ± 10 µA max

Input Capacitance 5 pF typ

LOGIC OUTPUTS (with DRVDD = 5 V)

High-Level Output Voltage (I High-Level Output Voltage (I Low-Level Output Voltage Low-Level Output Voltage (I

= 50 µA) +4.5 V min

= 0.5 mA) +2.4 V min

(IOL = 0.3 mA) +0.4 V max

= 50 µA) +0.1 V max

Output Capacitance 5 pF typ

LOGIC OUTPUTS (with DRVDD = 3 V)

High-Level Output Voltage (I

= 50 µA) +2.4 V min

Low-Level Output Voltage (IOL = 50 µA) +0.7 V max

NOTES

Since CLK is referenced to AVDD, +5 V logic input levels only apply.

The AD9260 is not guaranteed to meet VOL = 0.4 V max for standard TTL load of IOL = 1.6 mA.

Specifications subject to change without notice.

ANALOG INPUT

INPUT CLOCK

DATA OUTPUT

DAV

READ

INPUT CLOCK

RESET

DAV

Figure 4a. Timing Diagram

RES-DAV

CLK-DAV

Figure 4b.

RESET

Timing Diagram

–8–

REV. B

Page 9

AD9260

WARNING!

ESD SENSITIVE DEVICE

SWITCHING SPECIFICATIONS

(AVDD = +5 V, DVDD = +5 V, CL = 20 pF, T

MIN

to T

unless otherwise noted)

MAX

Parameters Symbol AD9260 Units

Clock Period t Data Available (DAV) Period t Data Invalid t Data Setup Time t Clock Pulsewidth High t Clock Pulsewidth Low t Data Hold Time t RESET to DAV Delay t CLOCK to DAV Delay t Three-State Output Disable Time t Three-State Output Enable Time t

Specifications subject to change without notice.

DAV

RES–DAV

CLK–DAV

ABSOLUTE MAXIMUM RATINGS*

With Respect

Parameter to Min Max Units

AVDD AVSS –0.3 +6.5 V DVDD DVSS –0.3 +6.5 V AVSS DVSS –0.3 +0.3 V AVDD DVDD –6.5 +6.5 V DRVDD DRVSS –0.3 +6.5 V DRVSS AVSS –0.3 +0.3 V REFCOM AVSS –0.3 +0.3 V CLK, MODE, READ,

Model Range Description Option*

AD9260AS –40°C to +85°C 44-Lead MQFP S-44 AD9260EB Evaluation Board

*S = Metric Quad Flatpack.

THERMAL CHARACTERISTICS

Thermal Resistance 44-Lead MQFP

= 53.2°C/W

= 19°C/W

50 ns min tC × Mode ns min 40% t t

–tH–t

DAV

ns max ns min

22.5 ns min

3.5 ns min 10 ns typ 15 ns typ 8 ns typ 45 ns typ

ORDERING GUIDE

Temperature Package Package

CS, RESET DVSS –0.3 DVDD + 0.3 V

Digital Outputs DRVSS –0.3 DRVDD

+ 0.3 V

VINA, VINB,

CML, BIAS AVSS –0.3 AVDD VREF AVSS –0.3 AVDD SENSE AVSS –0.3 AVDD

+ 0.3 V + 0.3 V

+ 0.3 V CAPB, CAPT AVSS –0.3 AVDD + 0.3 V Junction Temperature +150 °C Storage Temperature –65 +150 °C Lead Temperature

(10 sec) +300 °C

*Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. Exposure to absolute maximum ratings for extended periods may effect device reliability.

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although the AD9260 features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.

–9–REV. B

Page 10

AD9260

DEFINITIONS OF SPECIFICATION

INTEGRAL NONLINEARITY (INL)

INL refers to the deviation of each individual code from a line drawn from “negative full scale” through “positive full scale.” The point used as “negative full scale” occurs 1/2 LSB before the first code transition. “Positive full scale” is defined as a level 1 1/2 LSB beyond the last code transition. The deviation is measured from the middle of each particular code to the true straight line.

DIFFERENTIAL NONLINEARITY (DNL, NO MISSING CODES)

An ideal ADC exhibits code transitions that are exactly 1 LSB apart. DNL is the deviation from this ideal value. Guaranteed no missing codes to 14-bit resolution indicates that all 16384 codes, respectively, must be present over all operating ranges.

NOTE: Conventional INL and DNL measurements don’t really apply to Σ∆ converters: the DNL looks continually better if longer data records are taken. For the AD9260, INL and DNL numbers are given as representative.

ZERO ERROR

The major carry transition should occur for an analog value 1/2 LSB below VINA = VINB. Zero error is defined as the deviation of the actual transition from that point.

GAIN ERROR

The first code transition should occur at an analog value 1/2 LSB above negative full scale. The last transition should occur at an analog value 1 1/2 LSB below the nominal full scale. Gain error is the deviation of the actual difference between first and last code transitions and the ideal difference between first and last code transitions.

TEMPERATURE DRIFT

The temperature drift for zero error and gain error specifies the maximum change from the initial (+25°C) value to the value at

or T

MIN

POWER SUPPLY REJECTION

MAX

The specification shows the maximum change in full scale from the value with the supply at the minimum limit to the value with the supply at its maximum limit.

APERTURE JITTER

Aperture jitter is the variation in aperture delay for successive samples and is manifested as noise on the input to the A/D.

SIGNAL-TO-NOISE AND DISTORTION (S/N+D, SINAD) RATIO

S/N+D is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, including harmonics but excluding dc. The value for S/N+D is expressed in decibels.

EFFECTIVE NUMBER OF BITS (ENOB)

For a sine wave, SINAD can be expressed in terms of the number of bits. Using the following formula,

N = (SINAD – 1.76)/6.02

it is possible to get a measure of performance expressed as N, the effective number of bits.

Thus, effective number of bits for a device for sine wave inputs at a given input frequency can be calculated directly from its measured SINAD.

TOTAL HARMONIC DISTORTION (THD)

THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is expressed as a percentage or in decibels.

SIGNAL-TO-NOISE RATIO (SNR)

SNR is the ratio of the rms value of the measured input signal to the rms sum of all other spectral components below the Nyquist frequency, excluding the first six harmonics and dc. The value for SNR is expressed in decibels.

SPURIOUS FREE DYNAMIC RANGE (SFDR)

SFDR is the difference in dB between the rms amplitude of the input signal and the peak spurious signal.

TWO-TONE SFDR

The ratio of the rms value of either input tone to the rms value of the peak spurious component. The peak spurious component may or may not be an IMD product. May be reported in dBc (i.e., degrades as signal level is lowered), or in dBFS (always related back to converter full scale).

–10–

REV. B

Page 11

PIN CONFIGURATION

AD9260

VINB

BIT11

VINA

CML

40 39 3841

AD9260

TOP VIEW

(Not to Scale)

BIT8

BIT9

BIT10

AVSS

BIT7

CAPT

BIT6

CAPB

BIT5

BIAS

BIT4

MODE

BIT3

REFCOM

VREF

SENSE

RESET

AVSS

AVDD

DAV

OTR

BIT1 (MSB)

BIT2

DVSS

AVSS

DVDD

AVDD

DRVSS

DRVDD

CLK

READ

(LSB) BIT16

BIT15

BIT14

AVDD

4344 36 35 3437

PIN 1 IDENTIFIER

12 13 14 15 16 17 18 19 20 21 2 2

BIT12

BIT13

NC = NO CONNECT

PIN FUNCTION DESCRIPTIONS

Pin No. Name Description

1 DVSS Digital Ground. 2, 29, 38 AVSS Analog Ground. 3 DVDD +3 V to +5 V Digital Supply. 4, 28, 44 AVDD +5 V Analog Supply. 5 DRVSS Digital Output Driver Ground. 6 DRVDD +3 V to +5 V Digital Output Driver Supply. 7 CLK Clock Input. 8 READ Part of DSP Interface—Pull Low to Disable Output Bits. 9 BIT16 Least Significant Data Bit (LSB). 10–23 BIT15–BIT2 Data Output Bit. 24 BIT1 Most Significant Data Bit (MSB). 25 OTR Out of Range—Set When Converter or Filter Overflows. 26 DAV Data Available. 27 CS Chip Select (CS): Active LOW. 30 RESET RESET: Active LOW. 31 SENSE Reference Amplifier SENSE: Selects REF Level. 32 VREF Input Span Select Reference I/O. 33 REFCOM Reference Common. 34 MODE Mode Select—Selects Decimation Mode. 35 BIAS Power Bias. 36 CAPB Noise Reduction Pin—Decouples Reference Level. 37 CAPT Noise Reduction Pin—Decouples Reference Level. 39 CML Common-Mode Level (AVDD/2.5). 40, 43 NC No Connect (Ground for Shielding Purposes). 41 VINA Analog Input Pin (+). 42 VINB Analog Input Pin (–).

–11–REV. B

Page 12

AD9260

–Typical Performance Characteristics

(AVDD = DVDD = DRVDD = +5.0 V, 4 V Input Span, Differential DC Coupled Input with CML = 2.0 V, f

= 20 MSPS, Full Bias)

CLOCK

100kHz INPUT

–20

–40

–60

–80

dB BELOW FULL SCALE

–100

–120

0.4 0.6 0.8 FREQUENCY – MHz

20MHz CLOCK 8 DECIMATION

THD: –96dB

1.0

1.20.2

Figure 5. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, 8

–20

–40

–60

–80

dB BELOW FULL SCALE

–100

–120

OSR (2.5 MHz Output Data Rate)

100kHz INPUT

20MHz CLOCK 4 DECIMATION

THD: –98dB

0.5

1 1.5 2 2.5

FREQUENCY – MHz

Figure 6. Spectral Plot of the AD9260 at 100 kHz Input,

20 MHz Clock, 4

OSR (5 MHz Output Data Rate)

–20

–40

–60

–80

dB BELOW FULL SCALE

–100

–120

3456 78910

FREQUENCY – MHz

100kHz INPUT

20MHz CLOCK 1 DECIMATION

THD: –98dB

Figure 8. Spectral Plot of the AD9260 at 100 kHz Input, 20 MHz Clock, Undecimated (20 MHz Output Data Rate)

110

106

102

WORST CASE SPUR – dBFS

–12dBFS/TONE

–6.5dBFS/TONE

–26dBFS/TONE

0.20

–46dBFS/TONE

0.4 0.6 1

FREQUENCY – MHz

0.8

Figure 9. Dual Tone SFDR vs. Input Frequency (F1 = F2,

– F2, Span = 10% Center Frequency, Mode = 8×)

–20

–40

–60

–80

dB BELOW FULL SCALE

–100

–120

1 1.5 2 2.5 3 3.5 4 4.5

FREQUENCY – MHz

100kHz INPUT

20MHz CLOCK 2 DECIMATION

THD: –98dB

50.5

Figure 7. Spectral Plot of the AD9260 at 100 kHz Input,

20 MHz Clock, 2

OSR (10 MHz Output Data Rate)

–12–

DUAL-TONE TEST

f1 = 1.0MHz

–20

f2 = 975kHz

20MHz CLOCK

–40

8 DECIMATION

IM3: –94dB

–60

–80

dB BELOW FULL SCALE

–100

–120

0.4 0.6 0.8 1 FREQUENCY – MHz

1.20.2

Figure 10. Two-Tone Spectral Performance of the AD9260 Given Inputs at 975 kHz and 1.0 MHz, 20 MHz

Clock, 8

Decimation

REV. B

Page 13

Typical AC Characterization Curves vs. Decimation Mode

INPUT FREQUENCY – MHz

0.1 10

SINAD – dBFS

1 MODE

2 MODE

4 MODE

8 MODE

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, AIN = 0.5 dBFS Full Bias)

AD9260

SINAD – dBFS

0.1 10 INPUT FREQUENCY – MHz

8 MODE

4 MODE

Figure 11. SINAD vs. Input Frequency (f

–50

–60

–70

–80

THD – dBFS

–90

–100

–110

0.1 10

8 MODE

INPUT FREQUENCY – MHz

2 MODE

4 MODE

Figure 12. THD vs. Input Frequency (f

2 MODE

1 MODE

= 20 MSPS)

CLOCK

1 MODE

= 20 MSPS)

CLOCK

Figure 14. SINAD vs. Input Frequency (f

–70

–75

–80

–85

–90

–95

–100

THD – dBFS

–105

–110

–115

–120

0.1 10

8 MODE

INPUT FREQUENCY – MHz

Figure 15. THD vs. Input Frequency (f

4 MODE

CLOCK

1 MODE

2 MODE

CLOCK

= 10 MSPS)

–50

–60

–70

–80

SFDR – dBFS

–90

–100

–110

8 MODE

0.1 10 INPUT FREQUENCY – MHz

Figure 13. SFDR vs. Input Frequency (f

8× SINAD performance limited by noise contribution of input differential op amp driver.

–70

1 MODE

2 MODE

4 MODE

= 20 MSPS)

CLOCK

–75

–80

–85

–90

–95

–100

SFDR – dBFS

–105

–110

–115

–120

0.1 10

8 MODE

INPUT FREQUENCY – MHz

2 MODE

4 MODE

Figure 16. SFDR vs. Input Frequency (f

1 MODE

CLOCK

= 10 MSPS)

–13–REV. B

Page 14

AD9260

Typical AC Characterization Curves for 8 Mode

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias)

–0.5dBFS

SINAD – dB

–6.0dBFS

–20dBFS

0.1 INPUT FREQUENCY – MHz

Figure 17. SINAD vs. Input Frequency (f

–70

–75

–80

–85

–90

THD – dB

–95

–100

–105

–110

–20dBFS

–0.5dBFS

–6.0dBFS

0.1 INPUT FREQUENCY – MHz

Figure 18. THD vs. Input Frequency (f

= 20 MSPS)

CLOCK

= 20 MSPS)

CLOCK

SINAD – dB

–70

–75

–80

–85

–90

THD – dB

–95

–100

–105

0.1 INPUT FREQUENCY – MHz

Figure 20. SINAD vs. Input Frequency (f

Figure 21. THD vs. Input Frequency (f

–0.5dBFS

–6.0dBFS

–20dBFS

= 10 MSPS)

CLOCK

–20dBFS

–6.0dBFS

–0.5dBFS

= 10 MSPS)

CLOCK

105

100

–0.5dBFS

–6.0dBFS

–20dBFS

INPUT FREQUENCY – MHz

= 20 MSPS)

CLOCK

SFDR – dBc

0.1

Figure 19. SFDR vs. Input Frequency (f

SINAD performance limited by noise contribution of input differential op amp driver.

–14–

105

100

–6.0dBFS

–0.5dBFS

SFDR – dBc

0.1 INPUT FREQUENCY – MHz

–20dBFS

Figure 22. SFDR vs. Input Frequency (f

= 10 MSPS)

CLOCK

REV. B

Page 15

Typical AC Characterization Curves for 4 Mode

INPUT FREQUENCY – MHz

0.1

SFDR – dBc

–0.5dBFS

–6.0dBFS

–20dBFS

110

105

100

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias)

AD9260

SINAD – dB

0.1 10

–0.5dBFS

–6.0dBFS

–20dBFS

INPUT FREQUENCY – MHz

Figure 23. SINAD vs. Input Frequency (f

–70

–75

–80

–85

–90

THD – dB

–95

–100

–105

–110

0.1 10

Figure 24. THD vs. Input Frequency (f

–20dBFS

–0.5dBFS

–6.0dBFS

INPUT FREQUENCY – MHz

CLOCK

= 20 MSPS)

CLOCK

= 20 MSPS)

–0.5dBFS

–6.0dBFS

SINAD – dB

–20dBFS

0.1 INPUT FREQUENCY – MHz

Figure 26. SINAD vs. Input Frequency (f

–70

–75

–80

–85

–90

THD – dB

–95

–6.0dBFS

–100

–105

–110

0.1 INPUT FREQUENCY – MHz

Figure 27. THD vs. Input Frequency (f

–0.5dBFS

CLOCK

= 10 MSPS)

CLOCK

–20dBFS

= 10 MSPS)

110

105

100

SFDR – dBc

0.1 10

Figure 25. SFDR vs. Input Frequency (f

INPUT FREQUENCY – MHz

–0.5dBFS

–6.0dBFS

–20dBFS

CLOCK

= 20 MSPS)

Figure 28. SFDR vs. Input Frequency (f

–15–REV. B

= 10 MSPS)

CLOCK

Page 16

AD9260

Typical AC Characterization Curves for 2 Mode

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias)

SINAD – dB

0.1 10 INPUT FREQUENCY – MHz

Figure 29. SINAD vs. Input Frequency (f

–60

–65

–70

–0.5dBFS

THD – dB

–75

–80

–85

–90

–95

–6.0dBFS

–5.0dBFS

–6.0dBFS

–20dBFS

CLOCK

–20dBFS

= 20 MSPS)

SINAD – dB

0.1 10 INPUT FREQUENCY – MHz

–0.5dBFS

–6.0dBFS

–20dBFS

Figure 32. SINAD vs. Input Frequency (f

–60

–65

–70

–75

THD – dB

–80

–85

–90

–0.5dBFS

–95

–20dBFS

–6.0dBFS

= 10 MSPS)

CLOCK

–100

0.1 10 INPUT FREQUENCY – MHz

1.0

Figure 30. THD vs. Input Frequency (f

100

–6.0dBFS

SFDR – dBc

0.1 10

–0.5dBFS

INPUT FREQUENCY – MHz

Figure 31. SFDR vs. Input Frequency (f

= 20 MSPS)

CLOCK

–20dBFS

= 20 MSPS)

CLOCK

–100

0.1 10 INPUT FREQUENCY – MHz

Figure 33. THD vs. Input Frequency (f

100

–6.0dBFS

–0.5dBFS

SFDR – dBc

0.1 10 INPUT FREQUENCY – MHz

–20dBFS

Figure 34. SFDR vs. Input Frequency (f

= 10 MSPS)

CLOCK

= 10 MSPS)

CLOCK

–16–

REV. B

Page 17

Typical AC Characterization Curves for 1 Mode

INPUT FREQUENCY – MHz

0.1 10

SINAD – dB

–0.5dBFS

–6.0dBFS

–20dBFS

INPUT FREQUENCY – MHz

0.1 10

THD – dBc

–0.5dBFS

–6.0dBFS

–20dB

–60

–70

–80

–85

–90

–95

–100

–75

–65

–55

INPUT FREQUENCY – MHz

0.1 10

SFDR – dBc

–6.0dBFS

–20dBFS

–0.5dBFS

100

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, Differential DC Coupled Input with CML = 2 V, Full Bias)

–0.5dBFS

AD9260

SINAD – dB

0.1 10

–6.0dBFS

–20dBFS

INPUT FREQUENCY – MHz

Figure 35. SINAD vs. Input Frequency (f

–55

–60

–65

–70

–75

–80

THD – dB

–85

–90

–95

–100

0.1 10

–20dBFS

INPUT FREQUENCY – MHz

–0.5dBFS

Figure 36. THD vs. Input Frequency (f

= 20 MSPS)

CLOCK

–6.0dBFS

= 20 MSPS)

CLOCK

Figure 38. SINAD vs. Input Frequency (f

Figure 39. THD vs. Input Frequency (f

CLOCK

= 10 MSPS)

CLOCK

= 10 MSPS)

100

SDFR – dBc

0.1 10

Figure 37. SFDR vs. Input Frequency (f

–0.5dBFS

–6.0dBFS

–20dBFS

INPUT FREQUENCY – MHz

= 20 MSPS)

CLOCK

Figure 40. SFDR vs. Input Frequency (f

= 10 MSPS)

CLOCK

–17–REV. B

Page 18

AD9260

Typical AC Characterization Curves

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, AIN = –0.5 dBFS, Differential DC Coupled Input with CML = 2 V)

100

SFDR – dBFS

QUARTER BIAS

CLOCK FREQUENCY – MHz

10 15 20

FULL BIAS

HALF BIAS

Figure 41. SFDR vs. Clock Rate (fIN = 100 kHz in 8× Mode)

100

FULL BIAS

HALF BIAS

–60

–65

–70

–75

–80

THD – dBc

–85

–90

–95

–100

1.0

1.2

1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 3.0 COMMON MODE INPUT LEVEL – Volts

Figure 44. THD vs. Common-Mode Input Level (CML)

–40

–50

–60

FS = 10MHz

= 1MHz, 2 MODE

= 100kHz, 8 MODE

FS = 20MHz

SFDR – dBFS

10 15 25

CLOCK FREQUENCY – MHz

QUARTER BIAS

Figure 42. SFDR vs. Clock Rate (fIN = 500 kHz in 4× Mode)

100

FULL BIAS

SFDR – dBFS

HALF BIAS

QUARTER BIAS

10 15 25

CLOCK FREQUENCY – MHz

Figure 43. SFDR vs. Clock Rate (fIN = 1.0 MHz in 2× Mode)

CMR – dB

–70

–80

–90

Figure 45. CMR vs. Input Frequency (V

100k 10M1M10k1k

INPUT FREQUENCY – Hz

CML

Mode)

100

4V SPAN SNR-8 MODE

SFDR – dBFS

0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 FREQUENCY – MHz

4V SPAN SFDR-2 MODE

1.6V SPAN SNR-8 MODE

1.6V SPAN SFDR-2 MODE

Figure 46. 4 V vs. 1.6 V Span SNR/SFDR (f

FS = 5MHz

100M

= 2 V p-p, 1

= 20 MSPS)

CLOCK

–18–

REV. B

Page 19

AD9260

20MSPS-dBFS FULL BIAS

AIN – dBFS

WORST SPUR – dBc and dBFS

–60

100

110

120

–50 –40 –30 –20 –10 0

10MSPS-dBc HALF BIAS

10MSPS-dBFS HALF BIAS

20MSPS-dBc FULL BIAS

FULL BIAS-dBFS

AIN – dBFS

WORST SPUR – dBc and dBFS

–60

100

110

120

–50 –40 –30 –20 –10 0

HALF BIAS-dBFS

HALF BIAS-dBc

FULL BIAS-dBc

AIN – dBFS

WORST SPUR – dBc and dBFS

–60

100

110

120

–50 –40 –30 –20 –10 0

dBc

dBFS

Additional AC Characterization Curves

(AVDD = DVDD = DRVDD = +5 V, 4 V Input Span, AIN = –0.5 dBFS, Differential DC Coupled Input with CML = 2 V, Full Bias, unless otherwise noted)

120

115

110

105

100

SFDR – dBFS

–50

–45 –40 –35 –30 –25 –20 –15 –10 –5 0

20 MSPS FULL BIAS

AIN – dBFS

20 MSPS HALF BIAS

10 MSPS HALF BIAS

10 MSPS FULL BIAS

Figure 47. Single-Tone SFDR vs. Amplitude (fIN =100 kHz,

Mode)

110

105

100

SFDR – dBFS

20 MSPS FULL BIAS

10 MSPS HALF BIAS

10 MSPS FULL BIAS

Figure 50. Two-Tone SFDR (F1 = 475 kHz, F2 = 525 MHz,

Mode)

–50

–45 –40 –35 –30 –25 –20 –15 –10 –5 0

AIN – dBFS

Figure 48. Single-Tone SFDR vs. Amplitude (fIN =1.0 MHz,

Mode)

110

10 MSPS

105

100

SFDR – dBFS

–50

Figure 49. Single-Tone SFDR vs. Amplitude (fIN = 500 kHz, 2

Mode)

HALF BIAS

20 MSPS FULL BIAS

–45 –40 –35 –30 –25 –20 –15 –10 –5 0

AIN – dBFS

10 MSPS FULL BIAS

Figure 51. Two-Tone SFDR (F1 = 0.95 kHz, F2 = 1.05 MHz,

Mode 20 MSPS)

Figure 52. Two-Tone SFDR (F1 = 1.9 MHz, F2 = 2.1 MHz, 4

Mode 20 MSPS)

–19–REV. B

Page 20

AD9260

–

DAC1

INT1

–

DAC2

INT2

SHUFFLE

ADC

DAC

CONTROL/TEST

LOGIC

BANDGAP

REFERENCE

BUFFER

OUT

Figure 53. Simplified Block Diagram

THEORY OF OPERATION

The AD9260 utilizes a new analog-to-digital converter architecture to combine sigma-delta techniques with a high-speed, pipelined A/D converter. This topology allows the AD9260 to offer the high dynamic range associated with sigma-delta converters while maintaining very wide input signal bandwidth (1.25 MHz) at a very modest 8× oversampling ratio. Figure 53 provides a block diagram of the AD9260. The differential analog input is fed into a second order, multibit sigma-delta modulator. This modulator features a 5-bit flash quantizer and 5-bit feedback. In addition, a 12-bit pipelined A/D quantizes the input to the 5-bit flash to greater accuracy. A special digital modulation loop combines the output of the 12-bit pipelined A/D with the delayed output of the 5-bit flash to produce the equivalent response of a second order loop with a 12-bit quantizer and 12-bit feedback. The combination of a second order loop and multibit feedback provides inherent stability: the AD9260 is not prone to idle tones or full-scale idiosyncracies sometimes associated with higher order single bit sigmadelta modulators.

The output of this 12-bit modulator is fed into the digital decimation filter. The voltage level on the MODE pin establishes the configuration for the digital filter. The user may bring the data out undecimated (at the clock rate), or at a decimation factor of 2×, 4×, or a full 8×. The spectra for these four cases are shown in Figures 5, 6, 7 and 8, all for a 100 kHz full-scale input and 20 MHz clock. The spectra of the undecimated output clearly shows the second order shaping characteristic of the quantization noise as it rises at frequencies above 1.25 MHz.

The on-chip decimation filter provides excellent stopband rejection to suppress any stray input signal between 1.25 MHz and

18.75 MHz, substantially easing the requirements on any antialiasing filter for the analog input path. The decimation filters are integrated with symmetric FIR filter structures, providing a linear phase response and excellent passband flatness.

ADC3BDAC

–D

HALF-BAND

DECIMATION FILTER STAGE 1

HALF-BAND

DECIMATION FILTER STAGE 2

HALF-BAND

DECIMATION FILTER STAGE 3

OUTPUT BITS

–

ADC3BDAC

PIPELINE CORRECTION LOGIC

DIFFERENTIATOR

OUT

LSB

8 LSBs

–

ADC

The digital output driver register of the AD9260 features both READ and CHIP SELECT pins to allow easy interfacing. The digital supply of the AD9260 is designed to operate over a

2.7 V to 5.25 V supply range, though 3 V supplies are recommended to minimize digital noise on the board. A DATA AVAILABLE pin allows the user to easily synchronize to the converter’s decimated output data rate. OUT-OF-RANGE (OTR) indication is given for an overflow in the pipelined A/D converter or digital filters. A RESETB function is provided to synchronize the converter’s decimated data and clear any overflow condition in the analog integrators.

An on-chip reference and reference buffer are included on the AD9260. The reference can be configured in either a 2.5 V mode (providing a 4 V pk-pk differential input full scale), a 1 V mode (providing a 1.6 V pk-pk differential input full scale), or programmed with an external resistor divider to provide any voltage level between 1 V and 2.5 V. However, optimum noise and

distortion performance for the AD9260 can only be achieved with a

2.5 V reference as shown in Figure 46.

For users wishing to operate the part at reduced clock frequencies, the bias current of the AD9260 is designed to be scalable. This scaling is accomplished through use of the proper external resistor tied to the BIAS pin: the power can be reduced roughly proportionately to clock frequency by as much as 75% (for clock rates of 5 MHz). Refer to Figures 41–43 and 47–51 for characterization curves showing performance tradeoffs.

ANALOG INPUT AND REFERENCE OVERVIEW

Figure 54, a simplified model of the AD9260, highlights the relationship between the analog inputs, VINA, VINB and the reference voltage VREF. Like the voltage applied to the top of the resistor ladder in a flash A/D converter, the value VREF defines the maximum input voltage to the A/D converter. An internal reference buffer in the AD9260 scales the reference voltage VREF before it is applied internally to the AD9260

–20–

REV. B

Page 21

AD9260

A/D core. The scale factor of this reference buffer is 0.8. Consequently, the maximum input voltage to the A/D core is +0.8 × VREF. The minimum input voltage to the A/D core is automatically defined to be –0.8 × VREF. With this scale factor, the maximum differential input span of 4 V p-p is obtained with a VREF voltage of 2.5 V. A smaller differential input span may be obtained by using a VREF voltage of less than 2.5 V at the expense of ac performance (refer to Figure 46).

+0.8VREF

VINA

VINB



–

A/D CORE

–0.8VREF

Figure 54. Simplified Input Model

INPUT SPAN

The AD9260 is implemented with a differential input structure. This structure allows the common-mode level (average voltage of the two input pins) of the input signal to be varied independently of the input span of the converter over a wide range, as shown in Figure 44. Specifically, the input to the A/D core is the difference of the voltages applied at the VINA and VINB input pins. Therefore, the equation,

VCORE = VINA–VINB (1)

defines the output of the differential input stage and provides the input to the A/D core.

The voltage, VCORE, must satisfy the condition,

–0.8 × VREF ≤ VCORE ≤ +0.8 × VREF (2)

where VREF is the voltage at the VREF pin.

INPUT COMPLIANCE RANGE

In addition to the limitations on the differential span of the input signal indicated in Equation 2, an additional limitation is placed on the inputs by the analog input structure of the AD9260. The analog input structure bounds the valid operating range for VINA and VINB. The condition,

AVSS +0.5 V < VINA < AVDD – 0.5 V

AVSS +0.5 V < VINB < AVDD + 0.5 V

(3)

where AVSS is nominally 0 V and AVDD is nominally +5 V, defines this requirement. Thus the valid inputs for VINA and VINB are any combination that satisfies both Equations 2 and

3. Note, the clock clamping method used in the differential driver circuit shown in Figure 57 is sufficient for protecting the AD9260 in an undervoltage condition.

For additional information showing the relationships between VINA, VINB, VREF and the digital output of the AD9260, see Table V.

Refer to Table IV for a summary of the various analog input and reference configurations.

ANALOG INPUT OPERATION

The analog input structure of the AD9260 is optimized to meet the performance requirements for some of the most demanding communication and data acquisition applications. This input structure is composed of a switched-capacitor network that samples the input signal applied to pins VINA and VINB on every rising edge of the CLK pin. The input switched capacitors are charged to the input voltage during each period of CLK. The resulting charge, q, on these capacitors is equal to C × V

, where C is the input capacitor. The change in charge

on these capacitors, delta q, as the capacitors are charged from a previous sample of the input signal to the next sample, is approximated in the following equation,

delta q ~ C × deltaV

where V

N–2

represents the present sample of the input signal and

represents the sample taken two clock cycles earlier. The

= C × (VN – V

) (4)

N–2

average current flow into the input (provided from an external source) is given in the following equation,

I = delta q/T ~ C × (V

where T represents the period of CLK and f

– V

N–2

) × f

CLOCK

represents the

CLOCK

(5)

frequency of CLK. Equations 4 and 5 provide simplifying approximations of the operation of the analog input structure of the AD9260. A more exact, detailed description and analysis of the input operation is provided below.

SS3

SS1

VINA

CPA1

SS2

VINB

CPA2

CPB1

CPB2

CS1

CS2

SH3

SH4

SH1

SS4

SH2

ANALOG

MODULATOR

Figure 55. Detailed Analog Input Structure

Figure 55 illustrates the analog input structure of the AD9260. For the moment, ignore the presence of the parasitic capacitors CPA and CPB. The effects of these parasitic capacitors will be discussed near the end of this section. The switched capacitors, CS1 and CS2, sample the input voltages applied on pins VINA and VINB. These capacitors are connected to input pins VINA and VINB when CLK is low. When CLK rises, a sample of the input signal is taken on capacitors CS1 and CS2. When CLK is high, capacitors CS1 and CS2 are connected to the Analog Modulator. The modulator precharges capacitors CS1 and CS2 to minimize the amount of charge required from any circuit used in combination with the AD9260 to drive input pins VINA and VINB. This reduces the input drive requirements of the analog circuitry driving pins VINA and VINB. The Analog Modulator precharges the voltages across capacitors CS1 and CS2, approximately equal to a delayed version of the input signal. When capacitors CS1 and CS2 are connected to input pins VINA and VINB, the differential charge, Q(n), on these capacitors is given in the following equation,

Q(n) = q1 – q2 = CS × VCORE (6)

–21–REV. B

Page 22

AD9260

where q1 and q2 are the individual charges stored on capacitors CS1 and CS2 respectively, and CS is the capacitance value of CS1 and CS2. When capacitors CS1 and CS2 are connected to the Analog Modulator during the preceding “precharge” clock phase, the capacitors are precharged equal to an approximation of a previous sample of the input signal. Consequently the differential charge on these capacitors while CLK is high is given in the following equation,

Q(n–1) = CS × VCORE(delay) + CS × Vdelta (7)

where VCORE(delay) is the value of VCORE sampled during a previous period of CLK, and Vdelta is the sigma-delta error voltage left on the capacitors. Vdelta is a natural artifact of the sigma-delta feedback techniques utilized in the Analog Modulator of the AD9260. It is a small random voltage term that changes every clock period and varies from 0 to ±0.05 × VREF.

The analog circuitry used to drive the input pins of the AD9260 must respond to the charge glitch that occurs when capacitors CS1 and CS2 are connected to input pins VINA and VINB. This circuitry must provide additional charge, qdelta, to capacitors CS1 and CS2, which is the difference between the precharged value, Q(n–1), and the new value, Q(n), as given in the following equation,

Qdelta = Q(n) – Q(n–1) (8)

Qdelta = CS × [VCORE–VCORE(delay) + Vdelta] (9)

DRIVING THE INPUT Transient Response

The charge glitch occurs once at the beginning of every period of the input CLK (falling edge), and the sample is taken on capacitors CS1 and CS2 exactly one-half period later (rising edge). Figure 56 presents a typical input waveform applied to input Pins VINA and VINB of the AD9260.

TRACK SAMPLE TRACK SAMPLE TRACK SAMPLE TRACK SAMPLE

CLOCK

VINA-VINB

Figure 56. Typical Input Waveform

Figure 56 illustrates the effect of the charge glitch when a source with nonzero output impedance is used to drive the input pins. This source must be capable of settling from the charge glitch in one-half period of the CLK. Unfortunately, the MOS switches used in any CMOS-switched capacitor circuit (including those in the AD9260) include nonlinear parasitic junction capacitances connected to their terminals. Figure 55 also illustrates the parasitic capacitances, Cpa1, Cpb1, Cpa2 and Cpb2, associated with the input switches.

Parasitic capacitor Cpa1 and Cpa2 are always connected to Pins VINA and VINB and therefore do not contribute to the glitch energy. Parasitic capacitors Cpb1 and Cpb2, on the other hand, cause a charge glitch that adds to that of input capacitors CS1

and CS2 when they are connected to input Pins VINA and VINB. The nonlinear junction capacitance of Cpb1 and Cpb2 cause charge glitch energy that is nonlinearily related to the input signal. Therefore, linear settling is difficult to achieve unless the input source completely settles during one-half period of CLK. A portion of the glitch impulse energy “kicked” back at the source is not linearly related to the input signal. Therefore, the best way to ensure that the input signal settles linearly is to use wide bandwidth circuitry, which settles as completely as possible from the glitch during one-half period of the CLK.

The AD9260 utilizes a proprietary clock-boosted boot-strapping technique to reduce the nonlinear parasitic capacitances of the internal CMOS switches. This technique improves the linearity of the input switches and reduces the nonlinear parasitic capacitance. Thus, this technique reduces the nonlinear glitch energy. The capacitance values for the input capacitors and parasitic capacitors for the input structure of the AD9260, as illustrated in Figure 55, are listed as follows.

CS = 3.2 pF, Cpa = 6 pF, Cpb = 1 pF (where CS is the capacitance value of capacitors CS1 and CS2, Cpa is the value of capacitors Cpa1 and Cpa2, and Cpb is the value of capacitors Cpb1 and Cpb2). The total capacitance at each input pin is

= CS + Cpa + Cpb = 10.2 pF.

Input Driver Considerations

The optimum noise and distortion performance of the AD9260 can ONLY be achieved when the AD9260 is driven differentially with a 4 V input span . Since not all applications have a signal precon-

ditioned for differential operation, there is often a need to perform a single-ended-to-differential conversion. In the case of the AD9260, a single-ended-to-differential conversion is best realized using a differential op amp driver. Although a transformer will perform a similar function for ac signals, its usefulness is precluded by its inability to directly drive the AD9260 and thus the additional requirement of an active low noise, low distortion buffer stage.

Single-Ended-to-Differential Op Amp Driver

There are two single-ended-to-differential op amp driver circuits useful for driving the AD9260. The first circuit, shown in Figure 57, uses the AD8138 and represents the best choice in most applications. The AD8138 is a low-distortion differential ADC driver designed to convert a ground-referenced singleended input signal to a differential output signal with a specified common-mode level for dc-coupling applications. It is capable of maintaining the typical THD and SFDR performance of the AD9260 with only a slight degradation in its noise performance in the 8× mode (i.e., SNR of 85 dB–86 dB).

In this application, the AD8138 is configured for unity gain and its common-mode output level is set to 2.5 V (i.e., VREF of the AD9260) to maximize its output headroom while operating from a single supply. Note, single-supply operation has the benefit of not requiring an input protection network for the AD9260 in dc-coupled applications. A simple R-C network at the output is used to filter out high-frequency noise from the AD8138. Recall, the AD9260’s small signal bandwidth is 75 MHz, hence any noise falling within the baseband bandwidth of the AD9260 defined by its sample and decimation rate, as well as “images” of its baseband response occurring at multiples of the sample rate, will degrade its overall noise performance.

–22–

REV. B

Page 23

AD9260

499

499 50

+5V

100pF

AD8138

50

499

100pF

VINA

AD9260

VINB

VREF

0.1F10F

VIN

499

Figure 57. AD8138 Single-Ended Differential ADC Driver

The second driver circuit, shown in Figure 58, can provide slightly enhanced noise performance relative to the AD8138, assuming low-noise, high-speed op amps are used. This differential op amp driver circuit is configured to convert and level-shift a 2 V p-p single-ended, ground-referenced signal to a 4 V p-p differential signal centered at the common-mode level of the AD9260. The circuit is based on two op amps that are configured as matched unity gain difference amplifiers. The single-ended input signal is applied to opposing inputs of the difference amplifiers, thus providing differential outputs. The common-mode offset voltage is applied to the noninverting resistor leg of each difference amplifier providing the required offset voltage. This offset voltage is derived from the common-mode level (CML) pin of the AD9260 via a low output impedance buffer amplifier capable of driving a 1 µF capacitive load. The common-mode offset can be varied over a 1.8 V to 2.5 V span without any serious degradation in distortion performance as shown in Figure 44, thus providing some flexibility in improving output compression distortion from some ± 5 op amps with limited positive voltage swing.

To protect the AD9260 from an undervoltage fault condition from op amps specified for ±5 V operation, two 50 Ω series resistors and a diode to AGND are inserted between each op amp output and the AD9260 inputs. The AD9260 will inherently be protected against any overvoltage condition if the op amps share the same positive power supply (i.e., AVDD) as the AD9260. Note, the gain accuracy and common-mode rejection of each difference amplifier in this driver circuit can be enhanced by using a matched thin-film resistor network (i.e., Ohmtek ORNA5000F) for the op amps. Resistor values should be 500 Ω or less to maintain the lowest possible noise.

The noise performance of each unity gain differential driver circuit is limited by its inherent noise gain of two. For unity gain op amps ONLY, the noise gain can be reduced from two to one

-VIN

CML

50

100pF

-VIN

CML

100pF

50

100pF

0.1F

50

AD817

1.0F

VINA

AD9260

VINB

CML

VIN

Figure 58. DC-Coupled Differential Driver with Level-Shifting

beyond the input signals passband by adding a shunt capacitor, C

, across each op amp’s feedback resistor. This will essentially

establish a low-pass filter which reduces the noise gain to one beyond the filter’s f input signal to f

–3 dB

while simultaneously bandlimiting the

–3 dB

. Note, the pole established by this filter can also be used as the real pole of an antialiasing filter. Since the noise contribution of two op amps from the same product family are typically equal but uncorrelated, the total output-referred noise of each op amp will add root-sum square leading to a further 3 dB degradation in the circuit’s noise performance. Further out-of-band noise reduction can be realized with the addition of single-ended and differential capacitors, C

and CD.

The distortion and noise performance of the two op amps within the signal path are critical in achieving the AD9260’s optimum performance. Low noise op amps capable of providing greater than 85 dB THD at 1 MHz while swinging over a 1 V to 3 V range are a rare commodity, yet should only be considered. The AD9632 op amp was found to provide superb distortion performance in this circuit due to its ability to maintain excellent distortion performance over a wide bandwidth while swinging over a 1 V to 3 V range. Since the AD9632 is gain-of-two or greater stable, the use of the noise reduction shunt capacitors discussed above was prohibited thus degrading its noise performance slightly (1 dB–2 dB) when compared to the OPA642. Note, the majority of the AD9260 test and characterization data presented in this data sheet was taken using the AD9632 op amp in this dc coupled driver circuit. This driver circuit is also provided on the AD9260 evaluation board since the AD8138 was unreleased at that time.

Table IV. Reference Configuration Summary

Reference Input Span (VINA–VINB) Required VREF Operating Mode (V p-p) (V) Connect To

INTERNAL 1.6 1 SENSE VREF INTERNAL 4.0 2.5 SENSE REFCOM INTERNAL 1.6 ≤ SPAN ≤ 4.0 and 1 ≤ VREF ≤ 2.5 and R1 VREF and SENSE

SPAN = 1.6 × VREF VREF = (1+R1/R2) R2 SENSE and REFCOM

EXTERNAL 1.6 ≤ SPAN ≤ 4.0 1 ≤ VREF ≤ 2.5 SENSE AVDD

VREF EXT. REF.

–23–REV. B

Page 24

AD9260

The outputs of each op amp are ac coupled via a small series resistor and capacitor (i.e., 50 Ω and 0.1 µF) to the respective inputs of the AD9260. Similar to the dc coupled driver, further out-of-band noise reduction can be realized with the addition of 100 pF single-ended and differential capacitors, CS and CD. The lower-cutoff frequency of this ac coupled circuit is determined by R level pin, CML, of the AD9260 for proper biasing of the inputs. Although the OPA642 was found to provide the lowest overall noise and distortion performance (i.e., 88.8 dB and 96 dB THD @ 100 kHz), the AD8055 (or dual AD8056) suffered only a 0.5 dB to 1.5 dB degradation in overall performance. It is worth noting that given the high-level of performance attainable by the AD9260, special consideration must be given to both the quality of the test equipment and test setup in its evaluation.

Common-Mode Level

The CML pin is an internal analog bias point used internally by the AD9260. This pin must be decoupled to analog ground with at least a 0.1 µF capacitor as shown in Figure 59. The dc level of CML is approximately AVDD/2.5. This voltage should be buffered if it is to be used for any external biasing.

Note: the common-mode voltage of the input signal applied to the AD9260 need not be at the exact same level as CML. While this level is recommended for optimal performance, the AD9260 is tolerant of a range of input common-mode voltages around AVDD/2.5.

REFERENCE OPERATION

The AD9260 contains an onboard bandgap reference and internal reference buffer amplifier. The onboard reference provides a pin-strappable option to generate either a 1 V or 2.5 V output. With the addition of two external resistors, the user can generate reference voltages other than 1 V and 2.5 V. Another alternative is to use an external reference for designs requiring enhanced accuracy and/or drift performance. See Table IV for a summary of the pin-strapping options for the AD9260 reference configurations. Note, the optimum noise and distortion can only be achieved with a 2.5 V reference.

Figure 60 shows a simplified model of the internal voltage reference of the AD9260. A pin-strappable reference amplifier buffers a 1 V fixed reference. The output from the reference amplifier, A1, appears on the VREF pin and MUST be decoupled with 0.1 µF and 10 µF capacitor to REFCOM. The voltage on the VREF pin determines the full-scale input span of the A/D. This input span equals:

The voltage appearing at the VREF pin, as well as the state of the internal reference amplifier, A1, are determined by the voltage appearing at the SENSE pin. The logic circuitry contains two comparators that monitor the voltage at the SENSE pin. The comparator with the lowest set point (approximately 0.3 V)

and CC in which RC is tied to the common-mode

0.1F

CML

AD9260

Figure 59. CML Decoupling

Full-Scale Input Span = 1.6 × VREF

TO A/D

6.25k

DISABLE

–

AD9260

DISABLE

5k

LOGIC

CAPT

CAPB

VREF

7.5k7.5k

SENSE

5k

REFCOM

Figure 60. Simplified Reference

controls the position of the switch within the feedback path of A1. If the SENSE pin is tied to REFCOM, the switch is connected to the internal resistor network, thus providing a VREF of 2.5 V. If the SENSE pin is tied to the VREF pin via a short or resistor, the switch is connected to the SENSE pin. A short will provide a VREF of 1.0 V while an external resistor network will provide an alternative VREF SPAN between 1.0 V and

2.5 V. The external resistor network may, for example, be implemented as a resistor divider circuit. This divider circuit could consist of a resistor (R1) connected between VREF and SENSE and another resistor (R2) connected between SENSE and REFCOM. The other comparator controls internal circuitry that will disable the reference amplifier if the SENSE pin is tied to AVDD. Disabling the reference amplifier allows the VREF pin to be driven by an external voltage reference.

The reference buffer circuit, level shifts the reference to an appropriate common-mode voltage for use by the internal circuitry. The on-chip buffer provides the low impedance necessary for driving the internal switched capacitor circuits and eliminates the need for an external buffer op amp.

The actual reference voltages used by the internal circuitry of the AD9260 appear on the CAPT and CAPB pins. If VREF is configured for 2.5 V, thus providing a 4 V full-scale input span, the voltages appear at CAPT and CAPB are 3.0 V and 1.0 V respectively. For proper operation when using the internal or an external reference, it is necessary to add a capacitor network to decouple the CAPT and CAPB pins. Figure 61 shows the recommended decoupling network. This capacitive network performs the following three functions: (1) along with the reference amplifier, A2, it provides a low source impedance over a large frequency range to drive the A/D internal circuitry, (2) it provides the necessary compensation for A2, and (3) it bandlimits the noise contribution from the reference. The turn-on time of the reference voltage appearing between CAPT and CAPB is approximately 15 ms and should be evaluated in any powerdown mode of operation.

–24–

REV. B

Page 25

AD9260



0.1F

AD9260

REF

10F

SENSE

REFCOM

CAPT

0.1F

CAPB

0.1F

10F

0.1

Figure 61. Recommended Reference Decoupling Network

DIGITAL INPUTS AND OUTPUTS Digital Outputs

The AD9260 output data is presented in a twos complement format. Table V indicates the output data formats for various input ranges and decimation modes. A straight binary output data format can be created by inverting the MSB.

Table V. Output Data Format

Input (V) Condition (V) Digital Output

8 Decimation Mode

VINA–VINB < –0.8 × VREF 1000 0000 0000 0000 VINA–VINB = –0.8 × VREF 1000 0000 0000 0000 VINA–VINB = 0 0000 0000 0000 0000 VINA–VINB = +0.8 × VREF – 1 LSB 0111 1111 1111 1111 VINA–VINB >= + 0.8 × VREF 0111 1111 1111 1111

4 Decimation Mode

VINA–VINB < –0.825 × VREF 1000 0001 0001 1100 VINA–VINB = –0.825 × VREF 1000 0001 0000 1100 VINA–VINB = 0 0000 0000 0000 0000 VINA–VINB = +0.825 × VREF – 1 LSB 0111 1110 1110 0011 VINA–VINB >= + 0.825 × VREF 0111 1110 1110 0011

2 Decimation Mode

VINA–VINB < –0.825 × VREF 1000 0000 0100 0001 VINA–VINB = –0.825 × VREF 1000 0000 0100 0001 VINA–VINB = 0 0000 0000 0000 0000 VINA–VINB = +0.825 × VREF – 1 LSB 0111 1111 1011 1110 VINA–VINB >= + 0.825 × VREF 0111 1111 1011 1110

The slight different ± full-scale input voltage conditions and their corresponding digital output code for the 4× and 2× decimation modes can be attributed to the different digital scaling factors applied to each of the AD9260’s FIR decimation stages for filter optimization purposes. Thus, a + full-scale reading of 0111 1111 1111 1111 and – full-scale reading of 1000 0000 0000 0000 is unachievable in the 2× and 4× decimation mode. As a result, a digital overrange condition can never exist in the 2× and 4× decimation mode and thus OTR being set high indicates an overrange condition in the analog modulator.

The output data format in 1× decimation differs from that in 2×, 4× and 8× decimation modes. In 1× decimation mode the output data remains in a twos complement format, but the digital numbers are scaled by a factor of 7/128. This factor of 7/128 is the product of an internal scale factor of 7/8 in the analog modulator and a 1/16 scale factor caused by LSB justification of the 12-bit modulator data.

CS AND READ PINS

The CS and READ pins control the state of the output data pins (BIT1–BIT16) on the AD9260. The CS pin is active low and the READ pin is active high. When CS and READ are both active the ADC data is driven on the output data pins, otherwise the output data pins are in a high-impedance (Hi-Z)

–25–REV. B

state. Table VI indicates the relationship between the CS and READ pins and the state of Pins Bit 1–Bit 16.

Table VI. CS and READ Pin Functionality

CS READ Condition of Data Output Pins

Low Low Data Output Pins in Hi-Z State Low High ADC Data on Output Pins High Low Data Output Pins in Hi-Z State High High Data Output Pins in Hi-Z State

DAV PIN

The DAV pin indicates when the output data of the AD9260 is valid. Digital output data is updated on the rising edge of DAV. The data hold time (t

) is dependent on the external loading of

DAV and the digital data output pins (BIT1–BIT16) as well as the particular decimation mode. The internal DAV driver is sized to be larger than the drivers pertaining to the digital data outputs to ensure that rising edge of DAV occurs before the data transitions under similar loading conditions (i.e., fanout) regardless of mode. Note that minimum data hold (t

) of 3.5 ns

is specified in the Figure 4 timing diagram from the 50% point of DAV’s rising edge to the 50% of data transition using a capacitive load of 20 pF for DAV and BIT1–BIT16. Applications interfacing to TTL logic and/or having larger capacitive loading for DAV than BIT1–BIT16 should consider latching data on the falling edge of DAV since the falling edge of DAV occurs well after the data has transitioned in the case of the 2×, 4× and 8× modes. The duty cycle of DAV is approximately 50% and it remains active independent of CS and READ.

RESET PIN

The RESET pin is an asynchronous digital input that is active low. Upon asserting RESET low, the clocks in the digital decimation filters are disabled, the DAV pin goes low and the data on the digital output data pins (Bit 1–Bit 16) is invalid. In addition, the analog modulator in the AD9260 and internal clock dividers used in the decimation filters are reset and will remain reset as long as RESET is maintained low. In the 2×, 4×, or 8× mode, the RESET must remain low for at least a clock period to ensure all the clock dividers and analog modulator are reset. Upon bringing RESET high, the internal clock dividers will begin to count again on the next falling edge of CLK and DAV will go high approximately 15 ns after this falling edge, resuming normal operation. Refer to Figure 4b for a timing diagram.

The state of the internal decimation filters in the AD9260 remains unchanged when RESET is asserted low. Consequently, when RESET is pulsed low, this resets the analog modulator but does not clear all the data in the digital filters. The data in the filters is corrupted by the effect of resetting the analog modulator (this causes an abrupt change at the input of the digital filter and this change is unrelated to the signal at the input of the A/D converter). Similarly, in multiplexed applications in which the input of the A/D converters sees an abrupt change, the data in the analog modulator and digital filter will be corrupted.

For this reason, following a pulse on the RESET pin, or change in channels (i.e., multiplexed applications only), the decimation filters must be flushed of their data. These filters have a memory length, hence delay, equal to the number of filter taps times the clock rate of the converter. This memory length may be

Page 26

AD9260

interpreted in terms of a number of samples stored in the decimation filter. For example, if the part is in 8× decimation mode, the delay is 321/f

. This corresponds to 321 samples

CLOCK

stored in the decimation filter. These 321 samples must be flushed from the AD9260 after RESET is pulsed high prior to reusing the data from the AD9260. That is, the AD9260 should be allowed to clock for 321 samples as the corrupted data is flushed from the filters. If the part is in 4× or 2× decimation mode, then the relatively smaller group delays of the 4× and 2× decimation filters result fewer samples that must be flushed from the filters (108 samples and 23 samples respectively).

In 2×, 4× or 8× mode, RESET may be used to synchronize multiple AD9260s clocked with the same clock. The decimation filters in the AD9260 are clocked with an internal clock divider. The state of this clock divider determines when the output data becomes available (relative to CLK). In order to synchronize multiple AD9260s clocked with the same clock, it is necessary that the clock dividers in each of the individual AD9260s are all reset to the same state. When RESET is asserted low, these clock dividers are cleared. On the next falling edge of CLK following the rising edge of RESET, the clock dividers begin counting and the clock is applied to the digital decimation filters.

OTR PIN

The OTR pin is a synchronous output that is updated each CLK period. It indicates that an overrange condition has occurred within the AD9260. Ideally, OTR should be latched on the falling edge of CLK to ensure proper setup-and-hold time. However, since an overrange condition typically extends well beyond one clock cycle (i.e., does not toggle at the CLK rate). OTR typically remains high for more than a clock cycle, allowing it to be successfully detected on the rising edge of CLK or monitored asynchronously.

An overrange condition must be carefully handled because of the group delays in the low-pass digital decimation filters in the output stages of the AD9260. When the input signal exceeds the full-scale range of the converter, this can have a variety of effects upon the operation of the AD9260, depending on the duration and amplitude of this overrange condition. A short duration overrange condition (<< filter group delay) may cause the analog modulator to briefly overrange without causing the data in the low pass digital filters to exceed full scale. The analog modulator is actually capable of processing signals slightly (3%) beyond the full-scale range of the AD9260 without internally clipping. A long duration overrange condition will cause the digital filter data to exceed full scale. For this reason, the OTR signal is generated using two separate internal out-ofrange detectors.

The first of these out-of-range detectors is placed at the output of the analog modulator and indicates whether the modulator output signal has extended 3% beyond the full-scale range of the converter. If the modulator output signal exceeds 3% beyond full scale, the digital data is hard-limited (i.e., clipped) to a number that is 3% larger than full scale. Due to the delay of the switched capacitor analog modulator, the OTR signal is delayed 3 1/2 clock cycles relative to the clock edge in which the overranged analog input signal was sampled.

The second out-of-range detector is placed at the output of the stage three decimation filter and detects whether the low pass filtered data has exceeded full scale. When this occurs, the filter output data is hard limited to full scale. The OTR signal is a logical OR function of the signals from these two internal outof-range detectors. If either of these detectors produces an outof-range signal, the OTR pin goes high and the data may be seriously corrupted.

If the AD9260 is used in a system that incorporates automatic gain control (AGC), the OTR signal may be used to indicate that the signal amplitude should be reduced. This may be particularly effective for use in maximizing the signal dynamic range if the signal includes high-frequency components that occasionally exceed full scale by a small amount. If, on the other hand, the signal includes large amplitude low frequency components that cause the digital filters to overrange, this may cause the low pass digital filter to overrange. In this case the data may become seriously corrupted and the digital filters may need to be flushed. See the RESET pin function description above for an explanation of the requirements for flushing the digital filters.

OTR should be sampled with the falling edge of CLK. This signal is invalid while CLK is HIGH.

MODE OPERATION

The Mode Select Pin (MODE) allows the user to select one of four available digital filter modes using a single pin. Each mode configures the internal decimation filter to decimate at: 1×, 2×, 4× or 8×. Refer to Table VII for mode pin ranges.

The mode selection is performed by using a set of internal comparators, as illustrated in Figure 62, so that each mode corresponds to a voltage range on the input of the MODE pin. The output of the comparators are fed into encoding logic where, on the falling edge of the clock, the encoded data is latched.

Table VII. Recommended Mode Pin Ranges and Configurations

Mode Pin Typical Decimation Range Mode Pin Mode

0 V–0.5 V GND 8×

0.5 V–1.5 V VREF/2 2×

1.5 V–3.0 V CML 4×

3.0 V–5.0 V AVDD 1×

BIAS PIN OPERATION

The Bias Select Pin (BIAS) gives the user, who is able to operate the AD9260 at a slower clock rate, the added flexibility of running the device in a lower, power consumption mode when it is clocked at less than 20 MHz.

This is accomplished by scaling the bias current of the AD9260 as illustrated in Figure 63. The bias amplifier drives a source follower and forces 1 V across R

, which sets the bias current.

EXT

This effectively adjusts the bias current in the modulator amplifiers and FLASH preamplifiers. When a large value of R

EXT

is used, a smaller bias current is available to the internal amplifier circuitry. As a result these amplifiers need more time to settle, thus dictating the use of a slower clock as the power is reduced. Refer to the characterization curves shown in Figures 41–48 revealing the performance tradeoffs.

–26–

REV. B

Page 27

The scaling is accomplished by properly attaching an external

SAMPLE RATE – MSPS

AVDD

– mA

110

130

10 15 20

FULL BIAS-2k

HALF BIAS-4k

QUARTER BIAS-8k

8

2

4

1

SAMPLE RATE – MSPS

DVDD

DRVDD

– mA

10 15 20

resistor to the BIAS pin of the AD9260 as shown in Table IX. R

is normally 2 kΩ for a clock speed of 20 MHz and scales

EXT

inversely with clock rate. Because BIAS is an external pin, minimization of capacitance to this pin is recommended in order to prevent instability of the bias pin amplifier.

AVDD

MODE PIN

AD9260

AVSS

LATCH

ENCODER

CLOCK

ENCODED MODE

Figure 64. I Mode 1

×–4×

vs. Sample Rate (AVDD = +5 V,

AVDD

)

Figure 62. Simplified Mode Pin Circuitry

BIAS CURRENT

BIAS PIN

REXT

Figure 63. Simplified Bias Pin Circuitry

POWER DISSIPATION CONSIDERATIONS

The power dissipation of the AD9260 is dependent on its

Figure 65a. I

= 1 MHz)

3 V, f

DVDD/IDRVDD

vs. Sample Rate (DVDD = DRVDD =

8

4

application-specific configuration and operating conditions. The analog power dissipation as shown in Figure 64 is primarily a function of its power bias setting and sample rate. It remains insensitive to the particular input waveform being digitized or digital filter MODE setting. The digital power dissipation is primarily a function of the digital supply setting (i.e., +3 V to

– mA

DRVDD

DVDD

1

2

+5 V), the sample rate and, to a lesser extent, the MODE setting and input waveform. Figures 65a and 65b show the total current dissipation of the “combined” digital (DVDD) and digital driver supply (DRVDD) for +3 V and +5 V supplies. Note, DVDD and DRVDD are typically derived from the same supply bus since no degradation in performance results. A 1 MHz fullscale sine wave was used to ensure maximum digital activity in the digital filters and the digital drivers had a fanout of one. Note also that a twofold decrease in digital supply current results when the digital supply is reduced form +5 V to +3 V.

Figure 65b. I = 5 V, f

DVDD/IDRVDD

= 1 MHz)

10 15 20

SAMPLE RATE – MSPS

vs. Sample Rate (DVDD = DRVDD

–27–REV. B

Page 28

AD9260

Digital Output Driver Considerations (DRVDD)

The AD9260 output drivers can be configured to interface with +5 V or 3.3 V logic families by setting DRVDD to +5 V or 3.3 V respectively. The AD9260 output drivers in each mode are appropriately sized to provide sufficient output current to drive a wide variety of logic families. However, large drive currents tend to cause glitches on the supplies and may affect SINAD performance. Applications requiring the AD9260 to drive large capacitive loads or large fanout may require additional decoupling capacitors on DRVDD. The addition of external buffers or latches helps reduce output loading while providing effective isolation from the databus.

Clock Input and Considerations

The AD9260 internal timing uses the two edges of the clock input to generate a variety of internal timing signals. The clock input must meet or exceed the minimum specified pulse width high and low (t

and tCL) specifications for the given A/D as

defined in the Switching Specifications at the beginning of the data sheet to meet the rated performance specifications. For example, the clock input to the AD9260 operating at 20 MSPS may have a duty cycle between 45% to 55% to meet this timing requirement since the minimum specified t

and tCL is 22.5 ns.

For clock rates below 20 MSPS, the duty cycle may deviate from this range to the extent that both t

and tCL are satisfied.

All high-speed high-resolution A/Ds are sensitive to the quality of the clock input. The degradation in SNR at a given full-scale input frequency (f

) due to only aperture jitter (tA) can be calcu-

lated with the following equation:

SNR = 20 log

In the equation, the rms aperture jitter, t

[1/(2 π fIN tA)]

, represents the root-

sum square of all the jitter sources which include the clock input, analog input signal, and A/D aperture jitter specification. For example, if a 500 kHz full-scale sine wave is sampled by an A/D with a total rms jitter of 15 ps, the SNR performance of the A/D will be limited to 86.5 dB.

The clock input should be treated as an analog signal in cases where aperture jitter may affect the dynamic range of the AD9260. In fact, the CLK input buffer is internally powered from the AD9260’s analog supply, AVDD. Thus the CLK logic high and low input voltage levels are +3.5 V and +1.0 V, respectively. Supplies for clock drivers should be separated from the A/D output driver supplies to avoid modulating the clock signal with digital noise. Low jitter crystal controlled oscillators make the best clock sources. If the clock is generated from another type of source (by gating, dividing, or other method), it should be retimed by the original clock at the last step.

GROUNDING AND DECOUPLING Analog and Digital Grounding

Proper grounding is essential in any high-speed, high-resolution system. Multilayer printed circuit boards (PCBs) are recommended to provide optimal grounding and power schemes. The use of ground and power planes offers distinct advantages:

1. The minimization of the loop area encompassed by a signal and its return path.

2. The minimization of the impedance associated with ground and power paths.

3. The inherent distributed capacitor formed by the power plane, PCB insulation, and ground plane.

These characteristics result in both a reduction of electromagnetic interference (EMI) and an overall improvement in performance.

It is important to design a layout that prevents noise from coupling onto the input signal. Digital signals should not be run in parallel with input signal traces and should be routed away from the input circuitry. While the AD9260 features separate analog and digital ground pins, it should be treated as an analog component.

The AVSS, DVSS and DRVSS pins must be joined together directly under the AD9260. A solid ground plane under the A/D is ac-

ceptable if the power and ground return currents are managed carefully. Alternatively, the ground plane under the A/D may contain serrations to steer currents in predictable directions where cross-coupling between analog and digital would otherwise be unavoidable. The AD9260/EB ground layout, shown in Figure 76, depicts the serrated type of arrangement. The analog and digital grounds are connected by a jumper below the A/D.

Analog and Digital Supply Decoupling

The AD9260 features separate analog, digital, and driver supply and ground pins, helping to minimize digital corruption of sensitive analog signals.

Figure 66 shows the power supply rejection ratio vs. frequency for a 200 mV p-p ripple applied to AVDD, DVDD, and DAVDD.

DVDD & DRVDD

PSRR – dBFS

AVDD

101

FREQUENCY – kHz

100

1000

10000

Figure 66. AD9260 PSRR vs. Frequency (8 × Mode)

In general, AVDD, the analog supply, should be decoupled to AVSS, the analog common, as close to the chip as physically possible. Figure 67 shows the recommended decoupling for the analog supplies; 0.1 µF ceramic chip capacitors should provide adequately low impedance over a wide frequency range. Note that the AVDD and AVSS pins are co-located on the AD9260

0.1F

AVDD

AVSS

AVDD

AVSS

AD9260

AVDD

AVSS

0.1F

Figure 67. Analog Supply Decoupling

–28–

REV. B

Page 29

AD9260

to simplify the layout of the decoupling capacitors and provide the shortest possible PCB trace lengths. The AD9260/EB power plane layout, shown in Figure 77 depicts a typical arrangement using a multilayer PCB.

The digital activity on the AD9260 chip falls into two general categories: digital logic, and output drivers. The internal digital logic draws surges of current, mainly during the clock transitions. The output drivers draw large current impulses while the output bits are changing. The size and duration of these currents are a function of the load on the output bits: large capacitive loads are to be avoided. Note that the digital logic of the AD9260 is referenced DVDD while the output drivers are referenced to DRVDD. Also note that the SNR performance of the AD9260 remains independent of the digital or driver supply setting.

The decoupling shown in Figure 68, a 0.1 µF ceramic chip capacitor, is appropriate for a reasonable capacitive load on the digital outputs (typically 20 pF on each pin). Applications involving greater digital loads should consider increasing the digital decoupling proportionally, and/or using external buffers/ latches.

0.1F

AD9260

DVSS

DRVDD

DRVSS

0.1F

DVDD

Figure 68. Digital Supply Decoupling

A complete decoupling scheme will also include large tantalum or electrolytic capacitors on the PCB to reduce low-frequency ripple to negligible levels. Refer to the AD9260/EB schematic and layouts in Figures 73–77 for more information regarding the placement of decoupling capacitors.

An alternative layout and decoupling scheme is shown in Figure

69. This layout and decoupling scheme is well suited for applications in which multiple AD9260s are located on the same PC board and/or the AD9260 is part of a multicard mixed signal system in which grounds are tied back at the system supplies (i.e., star ground configuration). In this case, the AD9260 is treated as an analog component in which its analog (i.e., AVDD) and digital (DVDD and DRVDD) supplies are derived from the systems +5 V analog supply and all of the AD9260’s ground pins are tied directly to the analog ground plane which resides directly underneath the IC.

Referring to Figure 69, each supply pin is directly decoupled to their respective ground pin or analog ground plane via a ceramic

0.1 µF chip capacitor. Surface mount ferrite beads are used to isolate the analog (AVDD), digital (DVDD), and driver supplies (DRVDD) of the AD9260 from the +5 V power buss. Properly selected ferrite beads can provide more than 40 dB of isolation from high-frequency switching transients originating from AD9260 supply pins. Further noise immunity from noise is provided by the inherent power-supply rejection of the AD9260 as shown in Figure 64. If digital operation at 3 V is desirable for power savings and or to provide for a 3 V digital logic interface, a 5 V to 3 V linear regulator can be used to drive DVDD and/or DRVDD. A more complete discussion on this layout and decoupling scheme can be found in Chapter 7, pages 7-27 through 7-55 of the High

Speed Design Techniques seminar book, which is available at www.analog.com/support/frames/lin_frameset.hml.

INSERT 5/3 VOLT LINEAR REGULATOR FOR 3 OR 3.3V DIGITAL OPERATION

10F

FERRITE

BEAD CORE*

0.1F

DVDD

DVSS

AVDD

AVSS

AVDD

AVSS

AVDD

AVSS

DRVDD

DRVSS

AD9260

BITS 1–16,

0.1F

DAV

CLK

SAMPLING CLOCK

GENERATOR

BUFFER

LATCH

Figure 69.

AD9260 EVALUATION BOARD

GENERAL DESCRIPTION

The AD9260 Evaluation Board is designed to provide an easy and flexible method of exercising the AD9260 and demonstrate its performance to data sheet specifications. The evaluation board is fabricated in four layers: the component layer; the ground layer; the power layer and the solder layer. The board is clearly labeled to provide easy identification of components. Ample space is provided near the analog and clock inputs to provide additional or alternate signal conditioning.

FEATURES AND USER CONTROL

• Jumper Controlled Mode/OSR Selection: The choice of Mode/OSR can easily be varied by jumping either JP1, JP2, JP3 or JP4 as illustrated in Figure 71 within the Mode/OSR Control Block. To obtain the desired mode refer to Table VIII.

Table VIII. AD9260 Evaluation Board Mode Select

Mode/OSR Connect Jumper

1× JP4 2× JP2 4× JP3 8× JP1

• Selectable Power Bias: The power consumption of the AD9260 can be scaled down if the user is able to operate the device at a lower clock frequency. As illustrated in Figure 71, pin cups are provided for the external resistor (R2) tied to the BIAS pin of the AD9260. Table IX defines the recommended resistance for a given clock speed to obtain the desired power consumption.

–29–REV. B

Page 30

AD9260

2.5/3V

VOUT

TRIM

AD780R

TEMP

GNDS

15k

10k

+VIN

R12

15k

R13

10k

C18

0.1F

AGND

JP10

C19

0.1F

R11

49.9

C17

10F

C15

0.1F

1KPOT

Figure 70. Evaluation Board External Reference Circuitry

Table IX. Evaluation Board Recommended Resistance Value for External Bias Resistor

Resistor Clock Speed Power Value (max) Consumption

2 kΩ 20 MHz 585 mW 4 kΩ 10 MHz 325 mW 8 kΩ 5 MHz 200 mW 16 kΩ 2.5 MHz 150 mW

• Data Interfacing Controls: The data interfacing controls (RESETB, CSB, READ, DAV) are all accessible via SMA connectors (J2–J5) as illustrated in Figure 71 within the data interfacing control block. The RESETB, CSB and READ connections are each supplied with two sets or resistor pin cups to allow the user to pull-up or pull-down each signal to a fixed state. R5, R6 and R30 will terminate to ground, while R7, R28 and R29 terminate to DRVDD. The DAV and OTR signals are also directly connected to the data output connector P1. All interfacing controls are buffered through the CMOS line driver 74HC541.

• Buffered Output Data: The twos complement output data is buffered through two CMOS noninverting bus transceivers (U2 and U3) and made available at pin connector P1 as illustrated in Figure 71 within the data output block.

• Jumper Controlled Reference Source: The choice of reference for the AD9260 can easily be varied between 1.0 V,

2.5 V or external, by using Jumpers JP5, JP6, JP7 and JP9 as illustrated in Figure 71 within the reference configuration block. To obtain the desired reference see Table X.

Table X. Evaluation Board Reference Pin Configuration

Reference Input Voltage Voltage Connect Jumper (pk-pk FS)

2.5 V JP7 4.0 V

1.0 V JP6 1.6 V

External JP5, JP9 and JP10 4.0 V

VCC2

R10 1k

C14

AGND

AD817R

VCC2

0.1F

1k

R8 390

AGND

Q1 2N2222

C12

0.1F

VREFEXT

C13 10F

The external reference circuitry, is illustrated in Figure 70. By connecting or disconnecting JP10, the external reference can be configured for either 1.0 V or 2.5 V. That is, by connecting JP10, the external reference will be configured to provide a 2.5 V reference. By leaving JP10 open, the external reference will be configured to provide a 1.0 V reference.

• Flexible DC or AC Coupled External Clock Inputs: As illustrated in Figure 71, the AD9260 Evaluation Board is designed to allow the user the flexibility of selecting how to connect the external clock source. It is also equipped with a playpen area for experimenting with optional clock drivers or crystals.

• Selecting DC or AC Coupled External Clock:

DC Coupled: To directly drive the clock externally via the CLKIN connector, connect JP11 and disconnect JP12. Note: 50 Ω terminated by R27.

AC Coupled: To ac couple the external clock and level shift it to midsupply, connect JP12 and disconnect JP11. Note: 50 Ω terminated by R27.

• Flexible Input Signal Configuration Circuitry: The AD9260 Evaluation Board’s Input Signal Configuration Block is illustrated in Figure 72. It is comprised of an input signal summing amplifier (U7), a variable input signal commonmode generator (U10) and a pair of amplifiers (U8 and U9) that configure the input into a differential signal and drive it, through a pair of isolation resistors, into the input pins of AD9260. The user can either input a signal or dual signal into the evaluation board via the two SMA connectors (J6 and J7) labeled IN-1 or IN-2.

The user should refer to the Driving the Input section of the data sheet for a detailed explanation of how the inputs are to be driven and what amplifier requirements are recommended.

• Selecting Single or Dual Signal Input: The input ampli- fier (U7) can either be configured as a dual input signal inverting summer or a single tone inverting buffer. This flexibility will allow for slightly better noise performance in the single tone mode due to the inherent noise gain difference in the two amplifier configurations. An optional feedback capacitor (C9) was added to allow the user additional out-of band filtering of the input signal if needed.

–30–

REV. B

Page 31

AD9260

For two-tone input signals: The user would leave jumpers (JP8) connected and use IN-1 and IN-2 (J7 and J6) as the connectors for the input signals.

For signal tone input signal: The user would remove jumper (JP8) and use only IN-1 as the input signal connector.

• Selectable Input Signal Common-Mode Level Source:

The input signal’s common-mode level (CML) can be set by U10.

To use the Input CML generated by U10: Disconnect jumper JP13 and Connect resistors RX3 and RX4. The CML generated by U10 is variable and adjustable using the 1 kΩ trimpot R35.

SHIPMENT CONFIGURATION AND QUICK SETUP

• The AD9260 Evaluation Board is configured as follows when shipped:

1. 2.5 V external reference/4.0 V differential full-scale input: JP5, JP9 and JP10 connected, JP6 and JP7 disconnected.

2. 8× Mode/OSR: JP1 connected, JP2, JP3, and JP4 disconnected.

3. Full Speed Power Bias: R2 = 2 kΩ and connected.

4. CSB pulled low: R6 = 49.9 Ω and connected, R29 disconnected.

5. RESETB pulled high: R7 = 10 kΩ and connected, R30 disconnected.

6. READ pulled high: R28 = 10 kΩ and connected, R5 disconnected.

7. Single Tone Input: JP8 removed, input applied via IN-1 (J7).

8. Input signal common-mode level set by Trimpot R35 to

2.0 V: Jumper JP12 is disconnected and resistors Rx4 and Rx3 are connected.

9. AC Coupled Clock: JP12 connected and JP11 disconnected. Note: 50 Ω terminated by R27.

QUICK SETUP

1. Connect the required power supplies to the Evaluation Board as illustrated in Figure 22:

⇒±5 VA supplies to P5—Analog Power ⇒ +5 VA supply to P4—Analog Power ⇒ +5 VD supply to P3—Digital Power ⇒ +5 VD supply to P2—Driver Power

2. Connect a Clock Source to CLKIN (J1): Note: 50 Ω terminated by R1.

3. Connect an Input Signal Source to the IN-1 (J7).

4. Turn On Power!

5. The AD9260 Evaluation Board is now ready for use.

APPLICATION TIPS

1. The ADC analog input should not be overdriven. Using a

signal amplitude slightly lower than FSR will allow a small amount of “headroom” so that noise or DC offset voltage will not overrange the ADC and “hard limit” on signal peaks.

2. Two-tone tests can produce signal envelopes that exceed

FSR. Set each test signal to slightly less than –6 dB to prevent “hard limiting” on peaks.

3. Bandpass filtering of test signal generators is absolutely necessary for SNR, THD and IMD tests. Note, a low noise signal generator along with a high Q bandpass filter is often necessary to achieve the attainable noise performance of the AD9260.

4. Test signal generators must have exceptional noise performance to achieve accurate SNR measurements. Good generators, together with fifth-order elliptical bandpass filters, are recommended for SNR tests. Narrow bandwidth crystal filters can also be used to filter generator broadband noise, but they should be carefully tested for operation at highsignal levels.

5. The analog inputs of the AD9260 should be terminated directly at the input pin sockets with the correct filter terminating impedance (50 Ω or 75 Ω), or it should be driven by a low output impedance buffer. Short leads are necessary to prevent digital noise pickup.

6. A low noise (jitter) clock signal generator is required for good ADC dynamic performance. A poor generator can seriously impair good SNR performance particularly at higher input frequencies. A high-frequency generator, based on a clock source (e.g., crystal source), is recommended. Frequency-synthesized clock generators should generally be avoided because they typically provide poor jitter performance. See Note 8 if a crystal-based clock generator is used during FFT testing.

A low jitter clock may be generated by using a high-frequency clock source and dividing this frequency down with a low noise clock divider to obtain the AD9260 input CLK. Maintaining a large amplitude clock signal may also be very beneficial in minimizing the effects of noise in the digital gates of the clock generation circuitry.

Finally, special care should be taken to avoid coupling noise into any digital gates preceding the AD9260 CLK pin. Short leads are necessary to preserve fast rise times and careful decoupling should be used with these digital gates and the supplies for these digital gates should be connected to the same supplies as that of the internal AD9260 clock circuitry (Pins 44 and 38).

7. Two-tone testing will require isolation between test signal generators to prevent IMD generation in the test generator output circuits.

8. A very low side-lobe window must be used for FFT calculations if generators cannot be phase-locked and set to exact frequencies.

9. A well designed, clean PC board layout will assure proper operation and clean spectral response. Proper grounding and bypassing, short lead lengths, separation of analog and digital signals, and the use of ground planes are particularly important for high-frequency circuits. Multilayer PC boards are recommended for best performance, but if carefully designed, a two-sided PC board with large heavy (20 oz. foil) ground planes can give excellent results.

10. Prototype “plug-boards” or wire-wrap boards will not be satisfactory.

–31–REV. B

Page 32

AD9260

TP7 TP9 TP11 TP12 TP13

P1 17

P1 19

P1 21

P1 23

P1 25

P1 27

P1 29

P1 31

P1 33

P1 38

P1 39

P1 37

P1 35

P1 2

P1 4

P1 6

P1 8

P1 10

P1 12

P1 14

P1 16

P1 18

P1 20

P1 22

P1 24

P1 26

P1 28

P1 30

P1 32

P1 34

P1 38

P1 40

P1 1

P1 3

P1 5

P1 7

P1 9

P1 11

P1 13

P1 15

12 34567891011

33 32 31 30 29 28 27 26 25 24 23

TP1:RD

TP8:OTR

DRVDD

VCC

GND

74HC541

74HC245

DIR

GND

VCC

OUT_EN

74HC245

DRVDD

TP10

JP15

CT1

CT2

CT3

CT4

CT5

CT6

CT7

CT8

CT9

CT10

CT11

CT12

CT13

CT14

CT15

CT16

CT18

CT17

DATA OUTPUT BLOCK

J3J4J5

RESET CS

READ DAV

DRVDD

49.9

R28

10k

49.9

R29

10k

R30

49.9

10k

DATA OUTPUT CONTROL BLOCK

JP5:EXT REF

JP6:1V REF

JP7:2.5V REF

JP9:EXT REF

MDAVDD

REFERENCE CONFIGURATION

BLOCK

C10

10F

C11

0.1F

TP6

MDAVDD

JP4:1

JP3:4

JP2:2

JP1:8

MODE/OSR

CONTROL BLOCK

TP2

TP3

TP4:REFB

TP5:REFT

0.1F

10F

0.1F

2k

0.1F

10F

0.1F

INVDD

DVDD

FLAVDD

CT20

C61

10F

C62

0.1F

DVDD

CT19

CLKIN

MODE

REFCOM

BIAS

CAPB

CAPT

AVSS

CML

VINA

VINB

AVDD

VREF

SENSE

RESET

AVSS

AVDD

DAV

OTR

BIT01(MSB)

BIT02

DVSS

AVSS

DVDD

AVDD

DRVSS

DRVDD

CLK

READ

BIT16(LSB)

BIT15

BIT14

BIT13

BIT12

BIT11

BIT10

BIT09

BIT08

BIT07

BIT06

BIT05

BIT04

BIT03

AD9260

CML

VINA

VINB

CML

VREFEXT

R27

49.9k

0.1F

R33

1k

JP13

JP11

R31

1k

RESETB

CSBBUE

TP15

AGND

DC COUPLED

AC COUPLED

SHIELDED_TRACE

NC = NO CONNECT

MDAVDD

DIR

GND

VCC

OUT_EN

Figure 71. Evaluation Board Top Level Schematic

–32–

REV. B

Page 33

IN-2

IN-1

AD9260

R18

390

C20

R32

390

RX4 XXX

0.1F

R19

390

R17

390

R23

390

R24

390

R25

390

C23 10F

TBD

AD817R

VCC2

AD9632

VCC2

390

VEE

U10

R22

CX4

XXX

57.6

R15

57.6

IKPOT

R21

JP8

390

R16

390

R14 50

R34

390

RX3 XXX

R35

1k

C25

0.1F

C22

0.1F

AD9632

VCC2

VEE

VCC2

VEE

C16

100pF

R20

390

R26

390

JP16

JP17

R46 50

R48 50

100pF

JP12

C24

C26

9260CML

R47

50

R49 50

VINA

VINB

Figure 72. Evaluation Board Input Configuration Block

R40

R42

R44

R38

R41

C38

0.1F

R43

C42

0.1F

R45

C46

0.1F

R39

C34

0.1F

P4:+5V

C36 10F

C37

0.1F

C40 10F

C41

0.1F

C44 10F

C45

C32 22F

0.1F

C33

0.1F

P3:D5

EVALUATION BOARD POWER SUPPLY CONFIGURATION

C39

0.01F

C43

0.01F

C47

0.01F

C35

0.01F

FLAVDD

MDAVDD

INVDD

DVDD

P2:VDD

C27 47F

C55

C56 47F

C28 47F

0.1F

C57

0.1F

C29

0.1F

P5:+5AUX

P5:–5AUX

VEE VEE VEE

C30

C31

0.1F

C64

0.1F

U7 U8 U9

VCC2 DRVDD DRVDD DRVDD

C51

C52

0.1F

C53

0.1F

R50

R52

R36

VCC2

VCC2 VCC2

C48

0.1F

U7 U8 U9

C54

0.1F

R51

R53

R37

C49

0.1F

VCC2

VEE

DRVDD

C50

0.1F

U10 U2 U3 U4

DEVICE SUPPLY DECOUPLING

Figure 73. Evaluation Board Power Supply Configuration and Coupling

–33–REV. B

Page 34

AD9260

Figure 74. Evaluation Board Component Side Layout (Not to Scale)

Figure 75. Evaluation Board Solder Side Layout (Not to Scale)

–34–

REV. B

Page 35

AD9260

Figure 76. Evaluation Board Ground Plane Layout (Not to Scale)

Figure 77. Evaluation Board Power Plane Layout (Not to Scale)

–35–REV. B

Page 36

AD9260

1.03 (0.041)

0.73 (0.029)

SEATING

PLANE

0.25 (0.01) MIN

0.23 (0.009)

0.13 (0.005)

OUTLINE DIMENSIONS

Dimensions shown in millimeters and (inches).

44-Lead MQFP

(S-44)

13.45 (0.529)

0.8 (0.031)

12.95 (0.510)

10.1 (0.398)

9.90 (0.390)

TOP VIEW

(PINS DOWN)

BSC

0.45 (0.018)

0.3 (0.012)

2.45 (0.096) MAX

2.1 (0.083)

1.95 (0.077)

0 MIN

8.45 (0.333)

8.3 (0.327)

C3197a–0–5/00 (rev. B) 00581

–36–

PRINTED IN U.S.A.

REV. B

Datasheet AD9260 Datasheet (Analog Devices)

Specifications and Main Features

Frequently Asked Questions

User Manual