FEATURES
Monolithic 12-Bit, 25 MSPS A/D Converter
Low Power Dissipation: 280 mW
Single +5 V Supply
No Missing Codes Guaranteed
Differential Nonlinearity Error: ⴞ0.4 LSB
Complete On-Chip Sample-and-Hold Amplifier and
Voltage Reference
Signal-to-Noise and Distortion Ratio: 71 dB
Spurious-Free Dynamic Range: –85 dB
Out-of-Range Indicator
Straight Binary Output Data
28-Lead SOIC
28-Lead SSOP
Compatible with 3 V Logic
PRODUCT DESCRIPTION
The AD9225 is a monolithic, single supply, 12-bit, 25 MSPS,
analog-to-digital converter with an on-chip, high performance
sample-and-hold amplifier and voltage reference. The AD9225
uses a multistage differential pipelined architecture with output
error correction logic to provide 12-bit accuracy at 25 MSPS
data rates, and guarantees no missing codes over the full operating temperature range.
The AD9225 combines a low cost high speed CMOS process
and a novel architecture to achieve the resolution and speed of
existing bipolar implementations at a fraction of the power
consumption and cost.
The input of the AD9225 allows for easy interfacing to both
imaging and communications systems. With a truly differential
input structure, the user can select a variety of input ranges and
offsets including single-ended applications. The dynamic performance is excellent.
The sample-and-hold (SHA) amplifier is well suited for both
multiplexed systems that switch full-scale voltage levels in successive channels and sampling single-channel inputs at frequencies up to and well beyond the Nyquist rate.
The AD9225’s wideband input combined with the power and
cost savings over previously available monolithics, is suitable for
applications in communications, imaging and medical ultrasound.
The AD9225 has an onboard programmable reference. An
external reference can also be chosen to suit the dc accuracy and
temperature drift requirements of the application.
Monolithic A/D Converter
AD9225
FUNCTIONAL BLOCK DIAGRAM
AVDDCLK
SHA
VINA
VINB
CAPT
CAPB
VREF
SENSE
MODE
SELECT
MDAC1
GAIN = 16
5
5
REFCOM
MDAC2
GAIN = 4
A/DA/D
3
DIGITAL CORRECTION LOGIC
OUTPUT BUFFERS
1V
AVSS
A single clock input is used to control all internal conversion
cycles. The digital output data is presented in straight binary
output format. An out-of-range signal indicates an overflow
condition that can be used with the most significant bit to determine low or high overflow.
PRODUCT HIGHLIGHTS
The AD9225 is fabricated on a very cost effective CMOS
process. High-speed precision analog circuits are now combined
with high density logic circuits.
The AD9225 offers a complete single-chip sampling 12-bit,
25 MSPS analog-to-digital conversion function in 28-lead SOIC
and SSOP packages.
Low Power—The AD9225 at 280 mW consumes a fraction of
the power of presently available in existing monolithic solutions.
On-Board Sample-and-Hold (SHA)—The versatile SHA
input can be configured for either single-ended or differential
inputs.
Out of Range (OTR)—The OTR output bit indicates when
the input signal is beyond the AD9225’s input range.
Single Supply—The AD9225 uses a single +5 V power supply
simplifying system power supply design. It also features a separate digital driven supply line to accommodate 3 V and 5 V logic
families.
Pin Compatibility—The AD9225 is pin compatible with the
AD9220, AD9221, AD9223 and AD9224 ADCs.
DRVDD
MDAC3
GAIN = 4
3
12
A/D
AD9225
DRVSS
3
3
CML
A/D
4
OTR
BIT 1
(MSB)
BIT 12
(LSB)
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
Junction Temperature+150°C
Storage Temperature–65+150°C
Lead Temperature (10 sec)+300°C
*Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum
ratings for extended periods may affect device reliability.
S2
t
C
t
CL
S3
S4
t
OD
DATA 1
+ 0.3V
+ 0.3V
+ 0.3V
PIN CONNECTION
28-Lead SOIC and SSOP
CLK
(LSB) BIT 12
BIT 11
BIT 10
BIT 9
BIT 8
BIT 7
BIT 6
BIT 5
BIT 4
BIT 3
BIT 2
(MSB) BIT 1
OTR
1
2
3
4
5
6
AD9225
7
TOP VIEW
(Not to Scale)
8
9
10
11
12
13
14
28
DRVDD
27
DRVSS
26
AVDD
25
AVSS
24
VINB
23
VINA
22
CML
21
CAPT
20
CAPB
19
REFCOM (AVSS)
18
VREF
17
SENSE
16
AVSS
15
AVDD
PIN FUNCTION DESCRIPTIONS
Pin
NumberNameDescription
1CLKClock Input Pin
2BIT 12Least Significant Data Bit (LSB)
3–12BIT 11–2Data Output Bit
13BIT 1Most Significant Data Bit (MSB)
14OTROut of Range
15, 26AVDD+5 V Analog Supply
16, 25AVSSAnalog Ground
17SENSEReference Select
18VREFInput Span Select (Reference I/O)
19REFCOMReference Common
(AVSS)
20CAPBNoise Reduction Pin
21CAPTNoise Reduction Pin
22CMLCommon-Mode Level (Midsupply)
23VINAAnalog Input Pin (+)
24VINBAnalog Input Pin (–)
27DRVSSDigital Output Driver Ground
28DRVDD+3 V to +5 V Digital Output
AD9225AR–40°C to +85°C28-Lead Wide Body Small OutlineR-28
AD9225ARS–40°C to +85°C28-Lead Shrink Small OutlineRS-28
AD9225-EBEvaluation Board
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD9225 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–4–
REV. A
Page 5
AD9225
DEFINITIONS OF SPECIFICATION
INTEGRAL NONLINEARITY (INL)
INL refers to the deviation of each individual code from a line
drawn from “negative full scale” through “positive full scale.”
The point used as “negative full scale” occurs 1/2 LSB before
the first code transition. “Positive full scale” is defined as a level
1 1/2 LSB beyond the last code transition. The deviation is
measured from the middle of each particular code to the true
straight line.
DIFFERENTIAL NONLINEARITY (DNL, NO MISSING
CODES)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 12-bit resolution indicates that all 4096
codes, respectively, must be present over all operating ranges.
ZERO ERROR
The major carry transition should occur for an analog value
1/2 LSB below VINA = VINB. Zero error is defined as the
deviation of the actual transition from that point.
GAIN ERROR
The first code transition should occur at an analog value
1/2 LSB above negative full scale. The last transition should
occur at an analog value 1 1/2 LSB below the nominal full scale.
Gain error is the deviation of the actual difference between first
and last code transitions and the ideal difference between first
and last code transitions.
TEMPERATURE DRIFT
The temperature drift for zero error and gain error specifies the
maximum change from the initial (+25°C) value to the value at
or T
T
MIN
POWER SUPPLY REJECTION
MAX
.
The specification shows the maximum change in full scale from
the value with the supply at the minimum limit to the value with
the supply at its maximum limit.
APERTURE DELAY
Aperture delay is a measure of the sample-and-hold amplifier
(SHA) performance and is measured from the rising edge of the
clock input to when the input signal is held for conversion.
SIGNAL-TO-NOISE AND DISTORTION (S/N+D, SINAD)
RATIO
S/N+D is the ratio of the rms value of the measured input signal
to the rms sum of all other spectral components below the
Nyquist frequency, including harmonics but excluding dc. The
value for S/N+D is expressed in decibels.
EFFECTIVE NUMBER OF BITS (ENOB)
For a sine wave, SINAD can be expressed in terms of the number of bits. Using the following formula,
N = (SINAD – 1.76)/6.02
it is possible to get a measure of performance expressed as N,
the effective number of bits.
Thus, effective number of bits for a device for sine wave inputs
at a given input frequency can be calculated directly from its
measured SINAD.
TOTAL HARMONIC DISTORTION (THD)
THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured input signal and is
expressed as a percentage or in decibels.
SIGNAL-TO-NOISE RATIO (SNR)
SNR is the ratio of the rms value of the measured input signal to
the rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in decibels.
SPURIOUS FREE DYNAMIC RANGE (SFDR)
SFDR is the difference in dB between the rms amplitude of the
input signal and the peak spurious signal.
APERTURE JITTER
Aperture jitter is the variation in aperture delay for successive
samples and is manifested as noise on the input to the A/D.
Figure 10. “Grounded-Input” Histogram (Input Span =
40 V p-p)
SNR – dB
60
55
220.0dB 2V INT REF
50
1
FREQUENCY – MHz
10
Figure 12. SNR vs. Input Frequency (Input
Span = 4.0 V p-p, V
= 2.5 V Single-Ended
CM
Input)
–7–REV. A
Page 8
AD9225
INTRODUCTION
The AD9225 is a high performance, complete single-supply 12bit ADC. The analog input range of the AD9225 is highly flexible allowing for both single-ended or differential inputs of
varying amplitudes that can be ac or dc coupled.
It utilizes a four-stage pipeline architecture with a wideband
input sample-and-hold amplifier (SHA) implemented on a costeffective CMOS process. Each stage of the pipeline, excluding
the last stage, consists of a low resolution flash A/D connected
to a switched capacitor DAC and interstage residue amplifier
(MDAC). The residue amplifier amplifies the difference between the reconstructed DAC output and the flash input for the
next stage in the pipeline. One bit of redundancy is used in each
of the stages to facilitate digital correction of flash errors. The
last stage simply consists of a flash A/D.
The pipeline architecture allows a greater throughput rate at the
expense of pipeline delay or latency. This means that while the
converter is capable of capturing a new input sample every clock
cycle, it actually takes three clock cycles for the conversion to be
fully processed and appear at the output. This latency is not a
concern in most applications. The digital output, together with
the out-of-range indicator (OTR), is latched into an output
buffer to drive the output pins. The output drivers of the
AD9225 can be configured to interface with +5 V or +3.3 V
logic families.
The AD9225 uses both edges of the clock in its internal timing
circuitry (see Figure 1 and specification page for exact timing
requirements). The A/D samples the analog input on the rising
edge of the clock input. During the clock low time (between the
falling edge and rising edge of the clock), the input SHA is in
the sample mode; during the clock high time it is in hold. System disturbances just prior to the rising edge of the clock and/or
excessive clock jitter may cause the input SHA to acquire the
wrong value, and should be minimized.
ANALOG INPUT AND REFERENCE OVERVIEW
Figure 13 is a simplified model of the AD9225. It highlights the
relationship between the analog inputs, VINA, VINB, and the
reference voltage, VREF. Like the voltage applied to the top of
the resistor ladder in a flash A/D converter, the value VREF
defines the maximum input voltage to the A/D core. The minimum input voltage to the A/D core is automatically defined to
be –VREF.
VINA
VINB
AD9225
V
CORE
+VREF
A/D
CORE
–VREF
12
difference of the voltages applied at the VINA and VINB input
pins. Therefore, the equation,
V
= VINA – VINB(1)
CORE
defines the output of the differential input stage and provides
the input to the A/D core.
The voltage, V
–VREF≤V
, must satisfy the condition,
CORE
≤VREF(2)
CORE
where VREF is the voltage at the VREF pin.
While an infinite combination of VINA and VINB inputs exist
that satisfy Equation 2, there is an additional limitation placed
on the inputs by the power supply voltages of the AD9225. The
power supplies bound the valid operating range for VINA and
VINB. The condition,
AVSS – 0.3 V < VINA < AVDD + 0.3 V
AVSS – 0.3 V < VINB < AVDD + 0.3 V
(3)
where AVSS is nominally 0 V and AVDD is nominally +5 V,
defines this requirement. The range of valid inputs for VINA
and VINB is any combination that satisfies both Equations 2
and 3.
For additional information showing the relationship between
VINA, VINB, VREF and the digital output of the AD9225, see
Table IV.
Refer to Table I and Table II at the end of this section for a
summary of both the various analog input and reference
configurations.
ANALOG INPUT OPERATION
Figure 14 shows the equivalent analog input of the AD9225
which consists of a differential sample-and-hold amplifier
(SHA). The differential input structure of the SHA is highly
flexible, allowing the devices to be easily configured for either a
differential or single-ended input. The dc offset, or commonmode voltage, of the input(s) can be set to accommodate either
single-supply or dual-supply systems. Also, note that the analog
inputs, VINA and VINB, are interchangeable with the exception
that reversing the inputs to the VINA and VINB pins results in a
polarity inversion.
C
H
Q
S2
Q
S2
C
H
VINA
VINB
+
C
PIN
Q
S1
C
PAR
Q
S1
–
C
PIN
C
PAR
C
S
Q
C
H1
S
Figure 13. Equivalent Functional Input Circuit
The addition of a differential input structure gives the user an
additional level of flexibility that is not possible with traditional
flash converters. The input stage allows the user to easily configure the inputs for either single-ended operation or differential
operation. The A/D’s input structure allows the dc offset of the
input signal to be varied independently of the input span of
the converter. Specifically, the input to the A/D core is the
–8–
Figure 14. Simplified Input Circuit
The AD9225 has a wide input range. The input peaks may be
moved to AVDD or AVSS before performance is compromised.
This allows for much greater flexibility when selecting singleended drive schemes. Op amps and ac coupling clamps can be
set to available reference levels rather than be dictated to what
the ADC “needs.”
REV. A
Page 9
Due to the high degree of symmetry within the SHA topology, a
significant improvement in distortion performance for differential input signals with frequencies up to and beyond Nyquist can
be realized. This inherent symmetry provides excellent cancellation of both common-mode distortion and noise. Also, the
required input signal voltage span is reduced by a half which
further reduces the degree of R
modulation and its effects on
ON
distortion.
The optimum noise and dc linearity performance for either
differential or single-ended inputs is achieved with the largest
input signal voltage span (i.e., 4 V input span) and matched
input impedance for VINA and VINB. Only a slight degradation in dc linearity performance exists between the 2 V and
4 V input spans.
Referring to Figure 14, the differential SHA is implemented
using a switched-capacitor topology. Its input impedance and
its switching effects on the input drive source should be considered in order to maximize the converter’s performance. The
combination of the pin capacitance, C
C
, and the sampling capacitance, CS, is typically less than
PAR
, parasitic capacitance
PIN
5 pF. When the SHA goes into track mode, the input source
must charge or discharge the voltage stored on C
input voltage. This action of charging and discharging C
to the new
S
, aver-
S
aged over a period of time and for a given sampling frequency,
, makes the input impedance appear to have a benign resistive
F
S
component. However, if this action is analyzed within a sampling period (i.e., T = 1/F
), the input impedance is dynamic
S
and hence certain precautions on the input drive source should be
observed.
The resistive component to the input impedance can be computed by calculating the average charge that gets drawn by C
H
from the input drive source. It can be shown that if CS is
allowed to fully charge up to the input voltage before switches
are opened, then the average current into the input is the
Q
S1
same as if there were a resistor of 1/(C
) ohms connected
S FS
between the inputs. This means that the input impedance is
inversely proportional to the converter’s sample rate. Since C
is
S
only 5 pF, this resistive component is typically much larger than
that of the drive source (i.e., 8 kΩ at F
= 25 MSPS).
S
The SHA’s input impedance over a sampling period appears as
a dynamic input impedance to the input drive source. When the
SHA goes into the track mode, the input source should ideally
provide the charging current through R
of switch QS1 in an
ON
exponential manner. The requirement of exponential charging
means that the most common input source, an op amp, must
exhibit a source impedance that is both low and resistive up to
and beyond the sampling frequency.
The output impedance of an op amp can be modeled with a
series inductor and resistor. When a capacitive load is switched
onto the output of the op amp, the output will momentarily
drop due to its effective output impedance. As the output recovers, ringing may occur. To remedy the situation, a series resistor
can be inserted between the op amp and the SHA input as
shown in Figure 15. The series resistance helps isolate the op
amp from the switched-capacitor load.
AD9225
V
CC
R
S
R
S
V
EE
10mF
0.1mF
Figure 15. Series Resistor Isolates Switched-Capacitor
SHA Input from Op Amp. Matching Resistors Improve
SNR Performance.
The optimum size of this resistor is dependent on several factors
which include the ADC sampling rate, the selected op amp, and
the particular application. In most applications, a 30 Ω to 100 Ω
resistor is sufficient. However, some applications may require a
larger resistor value to reduce the noise bandwidth or possibly
limit the fault current in an overvoltage condition. Other applications may require a larger resistor value as part of an antialiasing filter. In any case, since the THD performance is dependent
on the series resistance and the above mentioned factors, optimizing this resistor value for a given application is encouraged.
The source impedance driving VINA and VINB should be
matched. Failure to provide that matching will result in degradation of the AD9225’s superb SNR, THD and SFDR.
For noise sensitive applications, the very high bandwidth of the
AD9225 may be detrimental and the addition of a series resistor
and/or shunt capacitor can help limit the wideband noise at the
A/D’s input by forming a low-pass filter. Note, however, that
the combination of this series resistance with the equivalent
input capacitance of the AD9225 should be evaluated for those
time domain applications that are sensitive to the input signal’s
absolute settling time. In applications where harmonic distortion
is not a primary concern, the series resistance may be selected in
combination with the SHA’s nominal 10 pF of input capacitance to set the filter’s 3 dB cutoff frequency.
A better method of reducing the noise bandwidth, while possibly establishing a real pole for an antialiasing filter, is to add
some additional shunt capacitance between the input (i.e.,
VINA and/or VINB) and analog ground. Since this additional
shunt capacitance combines with the equivalent input capacitance of the AD9225, a lower series resistance can be selected to
establish the filter’s cutoff frequency while not degrading the
distortion performance of the device. The shunt capacitance
also acts like a charge reservoir, sinking or sourcing the additional charge required by the hold capacitor, C
ing current transients seen at the op amp’s output.
The effect of this increased capacitive load on the op amp driving the AD9225 should be evaluated. To optimize performance
when noise is the primary consideration, increase the shunt
capacitance as much as the transient response of the input signal
will allow. Increasing the capacitance too much may adversely
affect the op amp’s settling time, frequency response, and distortion performance.
AD9225
VINA
VINB
VREF
SENSE
REFCOM
, further reduc-
H
–9–REV. A
Page 10
AD9225
REFERENCE OPERATION
The AD9225 contains an onboard bandgap reference that provides a pin strappable option to generate either a 1 V or 2 V
output. With the addition of two external resistors, the user can
generate reference voltages other than 1 V and 2 V. Another
alternative is to use an external reference for designs requiring
enhanced accuracy and/or drift performance. See Table II for a
summary of the pin-strapping options for the AD9225 reference
configurations.
Figure 16 shows a simplified model of the internal voltage reference of the AD9225. A pin strappable reference amplifier
buffers a 1 V fixed reference. The output from the reference
amplifier, A1, appears on the VREF pin. The voltage on the
VREF pin determines the full-scale input span of the A/D. This
input span equals,
Full-ScaleInput Span = 2 × VREF
The voltage appearing at the VREF pin as well as the state of
the internal reference amplifier, A1, are determined by the voltage appearing at the SENSE pin. The logic circuitry contains
two comparators which monitor the voltage at the SENSE pin.
The comparator with the lowest set point (approximately 0.3 V)
controls the position of the switch within the feedback path of
A1. If the SENSE pin is tied to AVSS (AGND), the switch is
connected to the internal resistor network thus providing a
VREF of 2.0 V. If the SENSE pin is tied to the VREF pin via a
short or resistor, the switch will connect to the SENSE pin. This
short will provide a VREF of 1.0 V. An external resistor network
will provide an alternative VREF between 1.0 V and 2.0 V. The
other comparator controls internal circuitry which will disable
the reference amplifier if the SENSE pin is tied AVDD. Disabling the reference amplifier allows the VREF pin to be driven
by an external voltage reference.
5kV
A2
5kV
A1
LOGIC
AD9225
CAPT
CAPB
LOGIC
VREF
6.25kV
SENSE
6.25kV
REFCOM
TO
A/D
5kV
5kV
DISABLE
1V
DISABLE
A2
A1
Figure 16. Equivalent Reference Circuit
The actual reference voltages used by the internal circuitry of
the AD9225 appears on the CAPT and CAPB pins. For proper
operation when using the internal or an external reference, it is
necessary to add a capacitor network to decouple these pins.
Figure 17 shows the recommended decoupling network. This
capacitive network performs the following three functions: (1)
along with the reference amplifier, A2, it provides a low source
impedance over a large frequency range to drive the A/D internal circuitry, (2) it provides the necessary compensation for A2,
and (3) it bandlimits the noise contribution from the reference.
The turn-on time of the reference voltage appearing between
CAPT and CAPB is approximately 15 ms and should be evaluated in any power-down mode of operation.
The A/D’s input span may be varied dynamically by changing
the differential reference voltage appearing across CAPT and
CAPB symmetrically around 2.5 V (i.e., midsupply). To change
the reference at speeds beyond the capabilities of A2, it will be
necessary to drive CAPT and CAPB with two high speed, low
noise amplifiers. In this case, both internal amplifiers (i.e., A1
and A2) must be disabled by connecting SENSE to AVDD,
connecting VREF to AVSS and removing the capacitive decoupling network. The external voltages applied to CAPT and
CAPB must be 2.0 V + Input Span/4 and 2.0 V – Input Span/4
respectively in which the input span can be varied between 2 V
and 4 V. Note that those samples within the pipeline A/D during any reference transition will be corrupted and should be
discarded.
DRIVING THE ANALOG INPUTS
The AD9225 has a highly flexible input structure allowing it to
interface with single ended or differential input interface circuitry. The applications shown in sections “Driving the Analog
Inputs” and “Reference Configurations” along with the information presented in “Input and Reference Overview” of this
data sheet, give examples of both single-ended and differential
operation. Refer to Tables I and II for a list of the different
possible input and reference configurations and their associated
figures in the data sheet.
The optimum mode of operation, analog input range, and associated interface circuitry will be determined by the particular
applications performance requirements as well as power supply
options. For example, a dc coupled single-ended input would be
appropriate for most data acquisition and imaging applications.
Also, many communication applications which require a dc
coupled input for proper demodulation can take advantage of
the excellent single-ended distortion performance of the AD9225.
The input span should be configured such that the system’s
performance objectives and the headroom requirements of the
driving op amp are simultaneously met.
–10–
REV. A
Page 11
Table I. Analog Input Configuration Summary
AD9225
InputInput␣␣␣␣␣␣Input Range (V)Figure
Connection␣ Coupling Span (V)VINA
Single-Ended␣ DC20 to 2119, 20Best for stepped input response applications, requires ±5 V op amp.
2 × VREF0 toVREF19, 20Same as above but with improved noise performance due to
40 to 42.019, 20Optimum noise performance, excellent THD performance, often
2 × VREF2.0 – VREF2.030Optimum THD performance with VREF = 1. Single supply
Single-Ended␣ AC2 or0 to 1 or1 or VREF21, 22
2 × VREF0 to 2 × VREF
40.5 to 4.52.022Optimum noise performance, excellent THD performance, ability
Differential␣ AC/DC22 to 33 to 223, 24Optimum full-scale THD and SFDR performance well beyond
(via Transformer)the A/Ds Nyquist frequency. Preferred mode for undersampling
or Amplifierapplications.
2 × VREF2.0 – VREF/22.0 + VREF/223, 24Same as above with the exception that full-scale THD and SFDR
4.02.0 to 3.03.0 to 2.023, 24Optimum noise performance.
NOTE
1
VINA and VINB can be interchanged if signal inversion is required.
1
2 × VREFincrease in dynamic range. Headroom/settling time requirements
␣␣␣␣tooperation (i.e., +5 V) for many op amps.
2.0 + VREF
␣␣␣␣toAbility to use either +5 V or ±5 V op amp.
2.0 + VREF
␣␣␣␣to␣␣␣␣toperformance can be traded off for better noise performance.
2.0 + VREF/22.0 – VREF/2
VINB
1
#Comments
of ±5 op amp should be evaluated.
requires low distortion op amp with VCC > +5 V due to its headroom issues.
Differential modes of operation (ac- or dc-coupled input) provide the best THD and SFDR performance over a wide frequency range. Differential operation should be considered for themost demanding spectral-based applications (e.g., direct IF-todigital conversion). See Figures 23, 24 and section on Differential Mode of Operation. Differential input characterization was
performed for this data sheet using the configuration shown in
Figure 24.
Single-ended operation requires that VINA be ac or dc coupled
to the input signal source while VINB of the AD9225 be biased
to the appropriate voltage corresponding to a midscale code
transition. Note that signal inversion may be easily accomplished
by transposing VINA and VINB. Most of the single-ended specifications for the AD9225 are characterized using Figure 21 circuitry with input spans of 4 V and 2 V as well as VINB = 2.5 V.
Differential operation requires that VINA and VINB be simultaneously driven with two equal signals that are in and out of
phase versions of the input signal. Differential operation of the
AD9225 offers the following benefits: (1) Signal swings are
smaller and therefore linearity requirements placed on the input
signal source may be easier to achieve, (2) Signal swings are
smaller and therefore may allow the use of op amps which may
otherwise have been constrained by headroom limitations, (3)
Differential operation minimizes even-order harmonic products,
and (4) Differential operation offers noise immunity based on
the device’s common-mode rejection.
As is typical of most IC devices, exceeding the supply limits will
turn on internal parasitic diodes resulting in transient currents
within the device. Figure 18 shows a simple means of clamping
an ac or dc coupled single ended input with the addition of two
series resistors and two diodes. An optional capacitor is shown
for ac coupled applications. Note that a larger series resistor
could be used to limit the fault current through D1 and D2 but
should be evaluated since it can cause a degradation in overall
performance. A similar clamping circuit could also be used for
each input if a differential input signal is being applied. The
diodes might cause nonlinearities in the signal. Careful evaluation should be performed on the diodes used.
OPTIONAL
V
CC
V
EE
AC COUPLING
CAPACITOR
R
30V
AVDD
D2
S1
1N4148
D1
1N4148
R
20V
S2
AD9225
Figure 18. Simple Clamping Circuit
SINGLE-ENDED MODE OF OPERATION
The AD9225 can be configured for single ended operation using
dc or ac coupling. In either case, the input of the A/D must be
driven from an operational amplifier that will not degrade the
A/D’s performance. Because the A/D operates from a single
supply, it will be necessary to level shift ground-based bipolar
signals to comply with its input requirements. Both dc and ac
coupling provide this necessary function, but each method results
in different interface issues which may influence the system
design and performance.
Single-ended operation is often limited by the availability of
driving op amps. Very low distortion op amps that provide great
performance out to the Nyquist frequency of the converter are
hard to find. Compounding the problem, for dc-coupled, singleended applications, is the inability of many high performance
amplifiers to maintain low distortions as their outputs approach
their positive output voltage limit (i.e., 1 dB compression point).
For this reason, it is recommended that applications requiring high
performance dc coupling use the single-ended to differential
circuit shown in Figure 23.
DC COUPLING AND INTERFACE ISSUES
Many applications require the analog input signal to be dc coupled
to the AD9225. An operational amplifier can be configured to
rescale and level shift the input signal so that it is compatible
with the selected input range of the A/D. The input range to the
A/D should be selected on the basis of system performance
objectives as well as the analog power supply availability since
this will place certain constraints on the op amp selection.
Many of the new high performance op amps are specified for
only ±5 V operation and have limited input/output swing capa-
bilities. The selected input range of the AD9225 should be
considered with the headroom requirements of the particular op
amp to prevent clipping of the signal. Also, since the output of a
dual supply amplifier can swing below –0.3 V, clamping its
output should be considered in some applications.
In some applications, it may be advantageous to use an op amp
specified for single supply +5 V operation since it will inherently
limit its output swing to within the power supply rails. Amplifiers
like the AD8041, AD8011 and are useful for this purpose but
their low bandwidths will limit the AD9225’s performance. High
performance amplifiers such as the AD9631, AD9632, AD8056
or AD8055 allow the AD9225 to be configured for larger
input spans which will improve the ADC’s noise performance.
Op amp circuits using a noninverting and inverting topology are
discussed in the next section. Although not shown, the noninverting and inverting topologies can be easily configured as part
of an antialiasing filter by using a Sallen-Key or Multiple-Feedback topology. An additional R-C network can be inserted between the op amp’s output and the AD9225 input to provide a
filter pole.
Simple Op Amp Buffer
In the simplest case, the input signal to the AD9225 will already
be biased at levels in accordance with the selected input range.
It is simply necessary to provide an adequately low source impedance for the VINA and VINB analog pins of the A/D. Figure
19 shows the recommended configuration a single-ended drive
using an op amp. In this case, the op amp is shown in a noninverting unity gain configuration driving the VINA pin. The
internal reference drives the VINB pin. Note that the addi-
tion of a small series resistor of 30 Ω to 50 Ω connected to
VINA and VINB will be beneficial in nearly all cases. Refer to
section “Analog Input Operation” a discussion on resistor selection. Figure 19 shows the proper connection for a 0 V to 4 V
input range. Alternative single ended ranges of 0 V to 2 × VREF
can also be realized with the proper configuration of VREF
(refer to the section “Using the Internal Reference”). Headroom
limitations of the op amp must always be considered.
–12–
REV. A
Page 13
AD9225
VINA
VINB
AD9225
1kV
R
S
V
IN
C2
0.1mF
R
S
VCM
1kV
C2
0.1mF
C1
10mF
C1
10mF
C3
0.1mF
4V
0V
+V
R
2.0V
10mF
S
R
0.1mF
U1
–V
AD9225
VINA
S
VINB
VREF
SENSE
maintaining a low impedance over a wide frequency range. The
combination of the capacitor and the resistor form a high-pass
filter with a high-pass –3 dB frequency determined by the equation,
f
= 1/(2 × π×R× (C1 + C2))
–3 dB
The low impedance VREF voltage source biases both the VINB
input and provides the bias voltage for the VINA input. Figure
21 shows the VREF configured for 2.0 V thus the input range of
the A/D is 0 V to 4 V. Other input ranges could be selected by
Figure 19. Single-Ended AD9225 Op Amp Drive Circuit
changing VREF.
Op Amp with DC Level Shifting
AD9631
C1
10mF
C2
0.1mF
0.1mF
4.5
2.5
0.5
10mF
+V
+V
R
R
RR
AD9225
R
S
VINA
R
S
VINB
Figure 20 shows a dc-coupled level shifting circuit employing an
op amp, A1, to sum the input signal with the desired dc set.
Configuring the op amp in the inverting mode with the given
resistor values results in an ac signal gain of –1. If the signal
inversion is undesirable, interchange the VINA and VINB connections to reestablish the original signal polarity. The dc voltage at VREF sets the common-mode voltage of the AD9225.
For example, when VREF = 2.0 V, the input level from the op
+2V
–2V
+5V
0V
V
IN
–5V
amp will also be centered around 2.0 V. The use of ratio matched,
thin-film resistor networks will minimize gain and offset errors.
Also, an optional pull-up resistor, RP, may be used to reduce
the output load on VREF to less than its 1 mA maximum.
Alternative AC Interface
Figure 21. AC-Coupled Input
Figure 22 shows a flexible ac coupled circuit which can be con-
500V*
+V
CC
0.1mF
figured for different input spans. Since the common mode voltage of VINA and VINB are biased to midsupply independent of
VREF, VREF can be pin strapped or reconfigured to achieve
input spans between 2 V and 4 V p-p. The AD9225’s CMRR
+VREF
–VREF
+V
VREF
0V
RP**
*OPTIONAL RESISTOR NETWORK-OHMTEK ORNA500D
**OPTIONAL PULL-UP RESISTOR WHEN USING INTERNAL REFERENCE
DC
500V*
0.1mF
500V*
500V*
Figure 20. Single-Ended Input with DC-Coupled Level Shift
AC COUPLING AND INTERFACE ISSUES
For applications where ac coupling is appropriate, the op amp’s
output can be easily level-shifted via a coupling capacitor. This
NC
7
2
3
1
A1
5
4
NC
R
S
6
VINA
AD9225
R
S
VINB
along with the symmetrical coupling R-C networks will reject
both power supply variations and noise. The resistors, R, establish the common-mode voltage. They may have a high value
(e.g., 5 kΩ) to minimize power consumption and establish a low
cutoff frequency. The capacitors, C1 and C2, are typically a
0.1 µF ceramic and 10 µF tantalum capacitor in parallel to
achieve a low cutoff frequency while maintaining a low impedance over a wide frequency range. R
isolates the buffer ampli-
S
fier from the A/D input. The optimum performance is achieved
when VINA and VINB are driven via symmetrical networks.
The f
point can be approximated by the equation,
–3 dB
f
=
–3 dB
2 π×6K +(C1+C2)
1
has the advantage of allowing the op amp’s common-mode level
to be symmetrically biased to its midsupply level (i.e., (V
)/2). Op amps that operate symmetrically with respect to
V
EE
CC
+
their power supplies typically provide the best ac performance as
well as greatest input/output span. Various high speed/performance amplifiers which are restricted to +5 V/–5 V operation
and/or specified for +5 V single-supply operation can be easily
configured for the 4 V or 2 V input span of the AD9225. Note
that differential transformer coupling, which is another form of
ac coupling, should be considered for optimum ac performance.
Simple AC Interface
Figure 21 shows a typical example of an ac-coupled, singleended configuration. The bias voltage shifts the bipolar, groundreferenced input signal to approximately AVDD/2. The value
for C1 and C2 will depend on the size of the resistor, R. The
capacitors, C1 and C2, are a 0.1 µF ceramic and 10 µF tanta-
lum capacitor in parallel to achieve a low cutoff frequency while
–13–REV. A
Figure 22. AC-Coupled Input-Flexible Input Span,
V
= 2.5 V
CM
Page 14
AD9225
OP AMP SELECTION GUIDE
Op amp selection for the AD9225 is highly dependent on a
particular application. In general, the performance requirements
of any given application can be characterized by either time
domain or frequency domain parameters. In either case, one
should carefully select an op amp which preserves the performance of the A/D. This task becomes challenging when one
considers the AD9225’s high performance capabilities coupled
with other extraneous system level requirements such as power
consumption and cost.
The ability to select the optimal op amp may be further complicated by either limited power supply availability and/or limited
acceptable supplies for a desired op amp. Newer, high performance op amps typically have input and output range limitations in accordance with their lower supply voltages. As a result,
some op amps will be more appropriate in systems where accoupling is allowable. When dc-coupling is required, op amps
without headroom constraints such as rail-to-rail op amps or
ones where larger supplies can be used should be considered.
The following section describes some op amps currently available from Analog Devices. The system designer is always
encouraged to contact the factory or local sales office to be
updated on Analog Devices latest amplifier product offerings.
Highlights of the areas where the op amps excel and where they
may limit the performance of the AD9225 is also included.
AD8055: f
= 300 MHz.
–3 dB
Low cost. Best used for driving single-ended ac
coupled configuration.
Limit: THD is compromised when output is not
swinging about 0 V.
AD8056: Dual Version of above amp.
Perfect for single-ended to differential configuration
(see Figure 23). Harmonics cancel each other in
differential drive, making this amplifier highly recommended for a single-ended input signal source. Handles
input signals past the 20 MHz Nyquist frequency.
AD9631: f
= 250 MHz.
–3 dB
Moderate cost.
Good for single-ended drive applications when signal
is anywhere between 0 V and 3 V.
Limits: THD is compromised above 8 MHz.
DIFFERENTIAL MODE OF OPERATION
Since not all applications have a signal preconditioned for differential operation, there is often a need to perform a single-endedto-differential conversion. In systems that do not need to be dc
coupled, an RF transformer with a center tap is the best method
to generate differential inputs for the AD9225. It provides all
the benefits of operating the A/D in the differential mode
without contributing additional noise or distortion. An RF
transformer also has the added benefit of providing electrical
isolation between the signal source and the A/D.
An improvement in THD and SFDR performance can be realized
by operating the AD9225 in the differential mode. The performance enhancement between the differential and single-ended
mode is most noteworthy as the input frequency approaches
and goes beyond the Nyquist frequency (i.e., f
> FS/2).
IN
500V
VREF
0V
500V
500V
500V
500V
500V
500V
500V
*OPTIONAL
50V
50V
VINA
VINB
10mF
AD9225
REF
+V
R*
0.1mF
Figure 23. Direct Coupled Drive Circuit with AD8056
Dual Op Amp
The circuit shown in Figure 23 is an ideal method of applying a
differential dc drive to the AD9225. We have used this configuration to drive the AD9225 from 2 V to 4 V spans at frequencies
approaching Nyquist with performance numbers matching those
shown on pages 2 and 3 of this data sheet (gathered through a
transformer). The dc input is shifted to a dc point swinging
symmetrically about the reference voltage. The optional resistor
will provide additional current if more reference drive is required.
The driver circuit shown in Figure 23 is optimized for dc coupling applications requiring optimum distortion performance.
This differential op amp driver circuit is configured to convert
and level shift a 2 V p-p single-ended, ground referenced signal
to a 4 V p-p differential signal centered at the VREF level of the
ADC. The circuit is based on two op amps which are configured
as matched unity gain difference amplifiers. The single-ended
input signal is applied to opposing inputs of the difference amplifiers, thus providing differential drive. The common-mode
offset voltage is applied to the noninverting resistor leg of each
difference amplifier providing the required offset voltage. The
common-mode offset can be varied over a wide span without
any serious degradation in distortion performance as shown in
Figures 25a and 25b, thus providing some flexibility in improv-
ing output compression distortion from some ±5 V op amps
with limited positive voltage swing.
To protect the AD9225 from an undervoltage fault condition
from op amps specified for ±5 V operation, two diodes to AGND
can be inserted between each op amp output and the AD9225
inputs. The AD9225 will inherently be protected against any
overvoltage condition if the op amps share the same positive
power supply (i.e., AVDD) as the AD9225. Note, the gain
accuracy and common-mode rejection of each difference amplifier in this driver circuit can be enhanced by using a matched
thin-film resistor network (i.e., Ohmtek ORNA5000F) for the
op amps. Recall, the AD9225’s small signal bandwidth is
105 MHz, hence any noise falling within the baseband bandwidth of the AD9225 will degrade its overall noise performance.
The noise performance of each unity gain differential driver
circuit is limited by its inherent noise gain of two. For unity gain
op amps ONLY, the noise gain can be reduced from two to one
beyond the input signals passband by adding a shunt capacitor,
, across each op amp’s feedback resistor. This will essentially
C
F
establish a low-pass filter, which reduces the noise gain to one
beyond the filter’s f
input signal to f
while simultaneously bandlimiting the
–3 dB
. Note, the pole established by this filter can
–3 dB
also be used as the real pole of an antialiasing filter.
–14–
REV. A
Page 15
AD9225
FUND
2ND
3RD
4TH
–119.7
–110.0
–100.0
–90.0
–80.0
–70.0
–60.0
–50.0
–40.0
–30.0
–20.0
–10.0
0.0
2.0E+6 4.0E+6 6.0E+68.0E+6 10.0E+612.5E+6
FIN = 2.5MHz
F
S
= 25MHz
Figure 24 shows the schematic of the suggested transformer
circuit. The circuit uses a minicircuits RF transformer, model
#T4-1T, which has an impedance ratio of four (turns ratio of
2). The schematic assumes that the signal source has a 50 Ω
source impedance. The 1:4 impedance ratio requires the 200 Ω
secondary termination for optimum power transfer and VSWR.
The center tap of the transformer provides a convenient means
of level-shifting the input signal to a desired common-mode
voltage.
R
S
49.9V
MINICIRCUITS
T4-1T
200V
33V
R
33V
S
0.1mF
VINA
CML
AD9225
VINB
Figure 24. Transformer Coupled Input
This (Figure 24) configuration was used to gather the differential data on pages 2 and 3.
Transformers with other turns ratios may also be selected to
optimize the performance of a given application. For example, a
given input signal source or amplifier may realize an improvement in distortion performance at reduced output power levels
and signal swings. For example, selecting a transformer with a
higher impedance ratio (e.g., Minicircuits T16-6T with a 1:16
impedance ratio) effectively “steps up” the signal level thus
further reducing the driving requirements of the signal source.
Referring to Figure 24, a series resistors, R
, were inserted between the AD9225 and the secondary
tor, C
S
, and shunt capaci-
S
of the transformer. The value of 33 Ω was selected to specifi-
cally optimize both the THD and SNR performance of the A/D.
and CS help provide a low-pass filter to block high frequency
R
S
noise.
The AD9225 can be easily configured for either a 2 V p-p input
span or 4.0 V p-p input span by setting the internal reference
(see Table II). Other input spans can be realized with two external gain setting resistors as shown in Figure 28 of this data
sheet. Figures 25a and 25b demonstrate how both spans of the
AD9225 achieve the high degree of linearity and SFDR over a
wide range of amplitudes required by the most demanding communication applications.
Figures 25a and 25b demonstrate the flexibility of commonmode voltage (transformer center tap) with respect to THD.
–76
–78
–80
F
= 10MHz
THD – dB
–82
–84
–86
051
IN
FIN = 2.5MHz
234
COMMON-MODE VOLTAGE – V
Figure 25a. Common-Mode Voltage vs. THD
= 2 V Differential)
(A
IN
–76
–78
–80
THD – dB
–82
–84
–86
0.51.0
F
= 10MHz
IN
FIN = 2.5MHz
1.5
2.02.53.03.54.04.5
COMMON-MODE VOLTAGE – V
Figure 25b. Common-Mode Voltage vs. THD
(A
= 4 V Differential)
IN
Figure 25c. Single Tone Frequency Domain Plot
Common-Mode Voltage = 2.5 V (A
= 4 V
IN
Differential)
–15–REV. A
Page 16
AD9225
REFERENCE CONFIGURATIONS
The figures associated with this section on internal and external
reference operation do not show recommended matching series
resistors for VINA and VINB for the purpose of simplicity.
Please refer to section “Driving the Analog Inputs, Introduction” for a discussion of this topic. Also, the figures do not show
the decoupling network associated with the CAPT and CAPB
pins. Please refer to the section “Reference Operation” for a
discussion of the internal reference circuitry and the recommended decoupling network shown in Figure 17.
USING THE INTERNAL REFERENCE
Single-Ended Input with 0 to 2 ⴛ VREF Range
Figure 26 shows how to connect the AD9225 for a 0 V to 2 V or
0 V to 4 V input range via pin strapping the SENSE pin. An
intermediate input range of 0 to 2 × VREF can be established
using the resistor programmable configuration in Figure 28.
In either case, both the midscale voltage and input span are
directly dependent on the value of VREF. More specifically, the
midscale voltage is equal to VREF while the input span is equal
to 2 × VREF. Thus, the valid input range extends from 0 to 2 ×
VREF. When VINA is ≤ 0 V, the digital output will be 000 Hex;
when VINA is ≥ 2 × VREF, the digital output will be FFF Hex.
Shorting the VREF pin directly to the SENSE pin places the
internal reference amplifier in unity-gain mode and the resultant
VREF output is 1 V. Therefore, the valid input range is 0 V to
2 V. However, shorting the SENSE pin directly to the REFCOM
pin configures the internal reference amplifier for a gain of 2.0
and the resultant VREF output is 2.0 V. Thus, the valid input
range becomes 0 V to 4 V. The VREF pin should be bypassed
to the REFCOM pin with a 10 µF tantalum capacitor in parallel
with a low-inductance 0.1 µF ceramic capacitor.
2 3 VREF
0V
10mF
SHORT FOR 0V TO 2V
SHORT FOR 0V TO 4V
0.1mF
INPUT SPAN
INPUT SPAN
VINA
VINB
VREF
AD9225
SENSE
REFCOM
Figure 26. Internal Reference—2 V p-p Input Span,
= 1 V, or 4 V p-p Input Span
V
CM
Figure 27 shows the single-ended configuration that gives good
dynamic performance (SINAD, SFDR). To optimize dynamic
specifications, center the common mode voltage of the analog
input at approximately by 2.5 V by connecting VINB to a low
impedance 2.5 V source. As described above, shorting the VREF
pin directly to the SENSE pin results in a 1 V reference voltage
and a 2 V p-p input span. The valid range for input signals is
1.5 V to 3.5 V. The VREF pin should be bypassed to the REF-
COM pin with a 10 µF tantalum capacitor in parallel with a
low-inductance 0.1 µF ceramic capacitor.
This reference configuration could also be used for a differential
input in which VINA and VINB are driven via a transformer as
shown in Figure 24. In this case, the common-mode voltage,
VCM , is set at midsupply by connecting the transformer’s
center tap to CML of the AD9225. VREF can be configured
for 1.0 V or 2.0 V by connecting SENSE to either VREF or
REFCOM respectively. Note that the valid input range for
each of the differential inputs is one half of the single-ended
input and thus becomes V
3.5V
1.5V
10mF
– VREF/2 to VCM + VREF/2.
CM
VINA
VCM
AD9225
VINB
1V
VREF
0.1mF
SENSE
REFCOM
Figure 27. Internal Reference—2 V p-p Input Span,
= 2.5 V
V
CM
Resistor Programmable Reference
Figure 28 shows an example of how to generate a reference
voltage other than 1.0 V or 2.0 V with the addition of two external resistors and a bypass capacitor. Use the equation,
VREF = 1 V× (1 + R1/R2)
to determine appropriate values for R1 and R2. These resistors
should be in the 2 kΩ to 100 kΩ range. For the example shown,
R1 equals 2.5 kΩ and R2 equals 5 kΩ. From the equation above,
the resultant reference voltage on the VREF pin is 1.5 V. This
sets the input span to be 3 V p-p. To assure stability, place a
0.1 µF ceramic capacitor in parallel with R1.
The midscale voltage can be set to VREF by connecting VINB
to VREF to provide an input span of 0 to 2 × VREF. Alterna-
tively, the midscale voltage can be set to 2.5 V by connecting
VINB to a low impedance 2.5 V source. For the example
shown, the valid input single-ended range for VINA is 1 V to
4 V since VINB is set to an external, low impedance 2.5 V source.
The VREF pin should be bypassed to the REFCOM pin with a
10 µF tantalum capacitor in parallel with a low inductance
0.1 µF ceramic capacitor.
4V
1V
10mF
2.5V
0.1mF
R1
2.5kV
R2
5kV
1.5V
C1
0.1mF
VINA
AD9225
VINB
VREF
SENSE
REFCOM
Figure 28. Resistor Programmable Reference +3 V p-p
Input Span, V
= 2.5 V
CM
–16–
REV. A
Page 17
AD9225
2 3 REF
0V
+5V
10mF
VINA
VINB
VREF
SENSE
AD9225
+5V
0.1mF
VREF
0.1mF
0.1mF
1111 1111 1111
1111 1111 1111
1111 1111 1110
OTR
–FS
+FS
–FS+1/2 LSB
+FS –1/2 LSB–FS –1/2 LSB
+FS –1 1/2 LSB
0000 0000 0001
0000 0000 0000
0000 0000 0000
1
0
0
0
0
1
OTR DATA OUTPUTS
USING AN EXTERNAL REFERENCE
Using an external reference may enhance the dc performance
of the AD9225 by improving drift and accuracy. Figures 29 and
30 show examples of how to use an external reference with the
A/D. Table III is a list of suitable voltage references from Analog Devices. To use an external reference, the user must disable
the internal reference amplifier and drive the VREF pin. Connecting the SENSE pin to AVDD disables the internal reference
amplifier.
The AD9225 contains an internal reference buffer, A2 (see
Figure 16), that simplifies the drive requirements of an external
reference. The external reference must be able to drive about
5 kΩ (±20%) load. Note that the bandwidth of the reference
buffer is deliberately left small to minimize the reference noise
contribution. As a result, it is not possible to change the reference voltage rapidly in this mode.
2.5V+VREF
2.5V–VREF
+5V
0.1mF
2.5V
2.5V
REF
22mF
R1
0.1mF
VINA
AD9225
VINB
mum of a 10 µF capacitor in parallel with a 0.1 µF low inductance
ceramic capacitor decouple the reference output to ground.
Figure 30. Input Range = 0 V to 2 × VREF
DIGITAL INPUTS AND OUTPUTS
Digital Outputs
The AD9225 output data is presented in positive true straight
binary for all input ranges. Table IV indicates the output data
formats for various input ranges regardless of the selected input
range. A twos complement output data format can be created by
inverting the MSB.
Figure 29 shows an example of the AD9225 configured for an
input span of 2 × VREF centered at 2.5 V. An external 2.5 V
reference drives the VINB pin thus setting the common-mode
voltage at 2.5 V. The input span can be independently set by a
voltage divider consisting of R1 and R2 which generates the
VREF signal. A1 buffers this resistor network and drives
VREF. Choose this op amp based on accuracy requirements. It
is essential that a minimum of a 10 µF capacitor in parallel with
a 0.1 µF low inductance ceramic capacitor decouple A1’s output
to ground.
Single-Ended Input with 0 to 2 ⴛ VREF Range
Figure 30 shows an example of an external reference driving
both VINB and VREF. In this case, both the common-mode
voltage and input span are directly dependent on the value of
VREF. More specifically, the common-mode voltage is equal to
VREF while the input span is equal to 2 × VREF. Thus, the
valid input range extends from 0 to 2 × VREF. For example, if
the REF191, a 2.048 V external reference was selected, the valid
input range extends from 0 to 4.096 V. In this case, 1 LSB of
the AD9225 corresponds to 1 mV. It is essential that a mini-
–17–REV. A
Figure 31. Output Data Format
Out Of Range (OTR)
An out-of-range condition exists when the analog input voltage
is beyond the input range of the converter. OTR is a digital out
put that is updated along with the data output corresponding to
the particular sampled analog input voltage. Hence, OTR has
the same pipeline delay (latency) as the digital data. It is LOW
when the analog input voltage is within the analog input range.
It is HIGH when the analog input voltage exceeds the input
range as shown in Figure 32. OTR will remain HIGH until the
analog input returns within the input range and another conversion is completed. By logical ANDing OTR with the MSB and
its complement, overrange high or underrange low conditions
can be detected. Table V is a truth table for the over/underrange
circuit in Figure 33 which uses NAND gates. Systems requiring
programmable gain conditioning of the AD9225 input signal
can immediately detect an out-of-range condition, thus eliminating gain selection iterations. Also, OTR can be used for
digital offset and gain calibration.
Page 18
AD9225
Table V. Out-of-Range Truth Table
OTRMSBAnalog Input Is
00In Range
01In Range
10Underrange
11Overrange
MSB
OTR
MSB
OVER = “1”
UNDER = “1”
Figure 32. Overrange or Underrange Logic
Digital Output Driver Considerations (DRVDD)
The AD9225 output drivers can be configured to interface with
+5 V or 3.3 V logic families by setting DRVDD to +5 V or 3.3 V
respectively. The output drivers are sized to provide sufficient
output current to drive a wide variety of logic families. However,
large drive currents tend to cause glitches on the supplies and
may affect SINAD performance. Applications requiring the
ADC to drive large capacitive loads or large fanout may require
additional decoupling capacitors on DRVDD. In extreme cases,
external buffers or latches may be required.
Clock Input and Considerations
The AD9225 internal timing uses the two edges of the clock
input to generate a variety of internal timing signals. The clock
input must meet or exceed the minimum specified pulsewidth
high and low (t
and tCL) specifications for the given A/D as
CH
defined in the Switching Specifications at the beginning of the
data sheet to meet the rated performance specifications. For
example, the clock input to the AD9225 operating at 25 MSPS
may have a duty cycle between 45% to 55% to meet this timing
requirement since the minimum specified t
and tCL is 18 ns.
CH
For low clock rates, the duty cycle may deviate from this range
to the extent that both t
and tCL are satisfied.
CH
All high speed high resolution A/Ds are sensitive to the quality
of the clock input. The degradation in SNR at a given full-scale
input frequency (f
) due to only aperture jitter (tA) can be cal-
IN
culated with the following equation:
SNR =20 log
10
2π f
1
INtA
In the equation, the rms aperture jitter, tA, represents the rootsum square of all the jitter sources which include the clock input, analog input signal, and A/D aperture jitter specification.
Undersampling applications are particularly sensitive to jitter.
Clock input should be treated as an analog signal in cases where
aperture jitter may affect the dynamic range of the AD9225.
Power supplies for clock drivers should be separated from the
A/D output driver supplies to avoid modulating the clock signal
with digital noise. Low jitter crystal controlled oscillators make
the best clock sources. If the clock is generated from another
type of source (by gating, dividing, or other method), it should
be retimed by the original clock at the last step.
The clock input is referred to as the analog supply. Its logic
threshold is AVDD/2. If the clock is being generated by 3 V
logic, it will have to be level shifted into 5 V CMOS logic levels.
This can also be accomplished by ac-coupling and level-shifting
the clock signal.
–18–
The AD9225 has a clock tolerance of 5% at 25 MHz. One way
to obtain a 50% duty cycle clock is to divide down a clock of
higher frequency, as shown in Figure 33. This configuration will
also decrease the jitter of the source clock.
+5V
R
D
50MHz
+5V
Q
Q
S
25MHz
Figure 33. Divide-by-Two Clock Circuit
In this case a 50 MHz clock is divided by two to produce the
25 MHz clock input for the AD9225. In this configuration, the
duty cycle of the 50 MHz clock is irrelevant.
The input circuitry for the CLOCK pin is designed to accommodate CMOS inputs. The quality of the logic input, particularly the rising edge, is critical in realizing the best possible jitter
performance of the part: the faster the rising edge, the better the
jitter performance.
As a result, careful selection of the logic family for the clock
driver, as well as the fanout and capacitive load on the clock
line, is important. Jitter-induced errors become more predominant at higher frequency, large amplitude inputs, where the
input slew rate is greatest.
Most of the power dissipated by the AD9225 is from the analog
power supplies. However, lower clock speeds will reduce digital
current. Figure 34 shows the relationship between power and
clock rate.
380
360
340
320
300
280
260
POWER – mW
240
220
200
180
0355
2V
INTERNAL
REFERENCE
1015
SAMPLE RATE
1V
INTERNAL
REFERENCE
302025
Figure 34. Power Consumption vs. Clock Rate
Direct IF Down Conversion Using the AD9225
Sampling IF signals above an ADC’s baseband region (i.e., dc
to F
/2) is becoming increasingly popular in communication
S
applications. This process is often referred to as Direct IF Down
Conversion or Undersampling. There are several potential benefits in using the ADC to alias (i.e., or mix) down a narrowband
or wideband IF signal. First and foremost is the elimination of a
complete mixer stage with its associated baseband amplifiers
and filters, reducing cost and power dissipation. Second is the
ability to apply various DSP techniques to perform such functions as filtering, channel selection, quadrature demodulation,
data reduction, detection, etc. A detailed discussion on using
REV. A
Page 19
AD9225
this technique in digital receivers can be found in Analog
Devices Application Notes AN-301 and AN-302.
In Direct IF Down Conversion applications, one exploits the
inherent sampling process of an ADC in which an IF signal
lying outside the baseband region can be aliased back into the
baseband region in a similar manner that a mixer will downconvert an IF signal. Similar to the mixer topology, an image
rejection filter is required to limit other potential interfering
signals from also aliasing back into the ADC’s baseband region.
A trade-off exists between the complexity of this image rejection
filter and the ADC’s sample rate as well as dynamic range.
The AD9225 is well suited for various IF sampling applications.
The AD9225’s low distortion input SHA has a full-power bandwidth extending beyond 130 MHz thus encompassing many
popular IF frequencies. A DNL of ±0.4 LSB (typ) combined
with low thermal input referred noise allows the AD9225 in the
2 V span to provide 69 dB of SNR for a baseband input sine
wave. Also, its low aperture jitter of 0.8 ps rms ensures minimum SNR degradation at higher IF frequencies. In fact, the
AD9225 is capable of still maintaining 68 dB of SNR at an IF of
71 MHz with a 2 V input span. Note, although the AD9225 can
yield a 1 dB to 2 dB improvement in SNR when configured for
the larger 4 V span, the 2 V span achieves the optimum fullscale distortion performance at these higher input frequencies.
Also, the 2 V span reduces the performance requirements of the
input driver circuitry (i.e., IP3) and thus may also be more
attractive from a system implementation perspective.
Figure 35 shows a simplified schematic of the AD9225 configured in an IF sampling application. To reduce the complexity of
the digital demodulator in many quadrature demodulation applications, the IF frequency and/or sample rate are strategically
selected such that the bandlimited IF signal aliases back into the
center of the ADC’s baseband region (i.e., F
/4). This demodu-
S
lation technique typically reduces the complexity of the post
digital demodulator ASIC which follows the ADC.
HIGH
FROM
PREVIOUS
STAGES
MIXER
SAW
FILTER
LINEARITY
AMPLIFIER
RF2317
RF2312
RF
OPTIONAL
BANDPASS
FILTER
MINICIRCUITS
T4-6T
0.1mF
10mF
200V
0.1mF
20V
20V
AD9225
VINA
VINB
CML
VREF
SENSE
REFCOM
Figure 35. Example of AD9225 IF Sampling Circuit
To maximize its distortion performance, the AD9225 is configured in the differential mode with a 2 V span using a transformer.
The center-tap of the transformer is biased at midsupply via the
CML output of the AD9225. Preceding the AD9225 and transformer is an optional bandpass filter as well as a gain stage. A
low Q passive bandpass filter can be inserted to reduce outof-band distortion and noise which lies within the AD9225’s
130 MHz bandwidth. A large gain stage(s) is often required to
compensate for the high insertion losses of a SAW filter used for
channel selection and image rejection. The gain stage will also
provide adequate isolation for the SAW filter from the charge
“kick back” currents associated with the AD9225’s switched
capacitor input stage.
The distortion and noise performance of an ADC at the given
IF frequency is of particular concern when evaluating an ADC
for a narrowband IF sampling application. Both single tone and
dual tone SFDR vs. amplitude are very useful in assessing an
ADC’s dynamic and static nonlinearities. SNR vs. amplitude
performance at the given IF is useful in assessing the ADC’s
noise performance and noise contribution due to aperture jitter.
In any application, one is advised to test several units of the
same device under the same conditions to evaluate the given
applications sensitivity to that particular device.
Figures 36–39 combine the dual tone SFDR as well as single
tone SFDR and SNR performances at IF frequencies of 35 MHz,
45 MHz, 70 MHz, and 85 MHz. Note, the SFDR vs. amplitude
data is referenced to dBFS while the single tone SNR data is
referenced to dBc. The performance characteristics in these
figures are representative of the AD9225 without any preceding
gain stage. The AD9225 was operated in the differential mode
(via transformer) with a 2 V span and a sample rate between
28 MSPS and 36 MSPS. The analog supply (AVDD) and the
digital supply (DRVDD) were set to +5 V and 3.3. V respectively.
100
95
90
85
80
75
70
SNR/SFDR – dBFS
65
60
55
50
–15
SFDR
DUAL-TONE
(dBFS)
SINGLE-TONE
–10–5
AIN – dBFS
SINGLE-TONE
SNR
(dBc)
SFDR
(dBFS)
0
Figure 36. IF Undersampling at 35 MHz (F1 = 34.63 MHz,
= 35.43 MHz, CLOCK = 20 MHz)
F
2
100
95
90
85
80
75
70
SNR/SFDR – dBFS
65
60
55
50
–15
SFDR
DUAL-TONE
(dBFS)
SINGLE-TONE
–10–5
AIN – dBFS
SINGLE-TONE
SNR
(dBc)
SFDR
(dBFS)
0
Figure 37. IF Undersampling at 45 MHz (F1 = 44.81 MHz,
= 45.23 MHz, CLOCK = 20 MHz)
F
2
–19–REV. A
Page 20
AD9225
0.1mF
CML
AD9225
100
95
90
85
80
75
70
SNR/SFDR – dBFS
65
60
55
50
–15
SFDR
DUAL-TONE
(dBFS)
SFDR
SINGLE-TONE
(dBFS)
–10–5
AIN – dBFS
SNR
SINGLE-TONE
(dBc)
0
Figure 38. IF Undersampling at 70 MHz (F1 = 69.50 MHz,
= 70.11 MHz, CLOCK = 25 MHz)
F
2
100
95
90
85
80
75
70
SNR/SFDR – dBFS
65
60
55
50
–15
SFDR
DUAL-TONE
(dBFS)
–10–5
AIN – dBFS
SFDR
SINGLE-TONE
(dBFS)
SINGLE-TONE
SNR
(dBc)
0
Figure 39. IF Undersampling at 85 MHz (F1 = 84.81 MHz,
= 85.23 MHz, CLOCK = 20 MHz)
F
2
GROUNDING AND DECOUPLING
Analog and Digital Grounding
Proper grounding is essential in any high speed, high resolution
system. Multilayer printed circuit boards (PCBs) are recommended to provide optimal grounding and power schemes. The
use of ground and power planes offers distinct advantages:
1. The minimization of the loop area encompassed by a signal
and its return path.
2. The minimization of the impedance associated with ground
and power paths.
3. The inherent distributed capacitor formed by the power
plane, PCB insulation and ground plane.
These characteristics result in both a reduction of electromagnetic interference (EMI) and an overall improvement in
performance.
It is important to design a layout that prevents noise from coupling onto the input signal. Digital signals should not be run in
parallel with input signal traces and should be routed away from
the input circuitry. While the AD9225 features separate analog
and driver ground pins, it should be treated as an analog component. The AVSS and DRVSS pins must be joined together
directly under the AD9225. A solid ground plane under the A/D
is acceptable if the power and ground return currents are carefully managed. Alternatively, the ground plane under the A/D
may contain serrations to steer currents in predictable directions
where cross coupling between analog and digital would otherwise be unavoidable. The AD9225/AD9225EB ground layout,
shown in Figure 47, depicts the serrated type of arrangement.
The board is primarily built over a common ground plane. It has a
“slit” to route currents near the clock driver. Figure 40 illustrates
a general scheme of ground and power implementation, in and
around the AD9225.
LOGIC
DV
SUPPLY
D
DIGITAL
LOGIC
ICs
GND
D
V
IN
A
A
D
AA
ADC
IC
= ANALOG
= DIGITAL
AVDD
ANALOG
CIRCUITS
A
DVDD
C
STRAY
DIGITAL
CIRCUITS
C
STRAY
I
A
B
I
D
DVSSAVSS
AA
Figure 40. Ground and Power Consideration
Analog and Digital Driver Supply Decoupling
The AD9225 features separate analog and driver supply and
ground pins, helping to minimize digital corruption of sensitive
analog signals. In general, AVDD, the analog supply, should be
decoupled to AVSS, the analog common, as close to the chip as
physically possible. Figure 41 shows the recommended decou-
pling for the analog supplies; 0.1 µF ceramic chip and 10 µF
tantalum capacitors should provide adequately low impedance
over a wide frequency range. Note that the AVDD and AVSS
pins are co-located on the AD9225 to simplify the layout of the
decoupling capacitors and provide the shortest possible PCB
trace lengths. The AD9225/AD9225EB power plane layout,
shown in Figure 48 depicts a typical arrangement using a multilayer PCB.
10mF
0.1mF
AVDD
AD9225
AVSS
Figure 41. Analog Supply Decoupling
The CML is an internal analog bias point used internally by the
AD9225. This pin must be decoupled with at least a 0.1 µF
capacitor as shown in Figure 42. The dc level of CML is approximately AVDD/2. This voltage should be buffered if it is to
be used for any external biasing.
Figure 42. CML Decoupling
–20–
REV. A
Page 21
The digital activity on the AD9225 chip falls into two general
categories: correction logic, and output drivers. The internal
correction logic draws relatively small surges of current, mainly
during the clock transitions. The output drivers draw large
current impulses while the output bits are changing. The size
and duration of these currents are a function of the load on the
output bits: large capacitive loads are to be avoided. Note, the
internal correction logic of the AD9225 is referenced to AVDD
while the output drivers are referenced to DRVDD.
The decoupling shown in Figure 43, 0.1 µF ceramic chip capaci-
tors and a 10 µF tantalum are appropriate for a reasonable
capacitive load on the digital outputs (typically 20 pF on each
pin). Applications involving greater digital loads should consider
increasing the digital decoupling proportionally, and/or using
external buffers/latches.
AD9225
10mF
0.1mF
Figure 43. Digital Supply Decoupling
A complete decoupling scheme will also include large tantalum
or electrolytic capacitors on the PCB to reduce low frequency
ripple to negligible levels. Refer to the AD9225/AD9225EB
schematic and layouts in Figures 44–50 for more information
regarding the placement of decoupling capacitors.
DRVDD
AD9225
DRVSS
–21–REV. A
Page 22
AD9225
6
VOUT
1
R25
2.49kV
2
JP19
12
1
R26
4.99kV
2
JP20
12
1
R27
4.99kV
2
1
P4
1
P4
3
P4
1
P5
2
P5
J5
J2
U5
REF43
VIN
GND
4
VCCIN
AGND
J1
VEEIN
JP10
12
JP11
12
JP12
12
DRVDDIN
DGND
1
2
1
2
JP13
JP6
2
1
C30
0.1mF
2
1
C29
0.1mF
2
R28
50V
12
2
C26
0.1mF
1
L3
FBEAD
12
1
C48
+
22mF
20V
2
1
2
L4
FBEAD
12
1
C49
+
22mF
20V
2
U9
11
L7404
L5
FBEAD
12
1
C51
+
22mF
20V
2
TP39
1
1
AVDD
C13
0.1mF
1
1
R1
50V
2
U8
21
56
L7404
21
JP7
21
JP8
21
R24
50V
+
1
2
11
1
+
2
10
C8
10mF
10V
2
C2
10mF
10V
VEE
TP37
1
1
2
TP28
1
9
L7404
R35
50V
3
U8
L7404
1
C18
+
10mF
10V
2
C19
10mF
10V
C53
0.1mF
1
2
3
1
2
U9
1
C56
0.1mF
2
2
JP17
AB
2
TP10
1
10
3
IN
AD187
2
IN
2
+
1
VCC
T1
T4-6T
TP36
1
C54
0.1mF
TP6
1
8
5U96
L7404
DRVDD
1
C16
0.1mF
2
1
TP4
1
TP1
1
U8
1312
L7404
TP38
R34
50V
1
VCC
1
R31
820V
2
7
8
+V
N2
–V
4
VEE
VIN
OUT
N1
1
1
C31
0.1mF
2
4
5
6
TP7
1
U1
REF43
VOUT
GND
R19
4kV
1
U8
3
L7404
TP2
1
U8
1
L7404
TP27
1
4
JP14
2
6
1
0.1mF
12
1
1
C25
0.1mF
2
TP8
D11
+
2
R2
5kV
3
2
CCV
98
L7404
1
2
1
TP3
1
U4
1
+
C28
0.1mF
2
R29
1kV
2
C27
J3
2
1
2
13U912
DVDD
DECOUPLING
1
C9
10mF
10V
1
CW
2
U8
U9
2
L7404
AVDD
DECOUPLING
1
2
TP37
TP30
2
R30
316V
1
R23
200V
L7404
U9
1
2
1
2
3
1
R32
50V
2
JP18
1
P2
P2
1
C15
0.1mF
2
C17
0.1mF
R18
1kV
2
JP9
21
3
34
1
C10
0.1mF
2
U8
JP21
12
10kV
Q1
2N2222
1
2
1
2
JP16
1
A
CLK
2
B
A
JP32
U9
L7404
R3
1
2
C46
15pF
DVDDIN
DGND
JP4
JP1
JP5
B
2
1
C7
+
10mF
10V
2
1
1
+
2
1
2
1
2
21
21
21
3
C3
10mF
10V
TP31
1
R4
10kV
TP29
DUTAVDD
C34
0.1mF
1
1
C45
15pF
2
1
+
2
D10
D9
D8
D7
D6
D5
JP3
1
JP2
1
1
2
1
1
C6
22mF
20V
TP9
1
C57
0.1mF
C21
10mF
10V
12
R21
200V
R22
200V
FBEAD
1
2
2
1
19
OTR
1
2
3
4
5
6
7
8
L1
2
3
4
5
6
7
8
9
D4
D3
D2
D1
D0
2
2
1
2
10mF
G1
G2
A1
A2
A3
A4
A5
A6
A7
A8
CLR
CLK
A
B
C
D
ENP
GND
+
2
C20
10V
1
1
C44
15pF
2
2
74541
1
19
2
3
4
5
6
7
8
9
U2
74LS161
1
1
1
1
+
TP33
1
TP32
1
U6
G1
G2
A1
A2
A3
A4
A5
A6
A7
A8
JP22
JP23
JP24
JP25
1
2
1
2
LOAD
C35
0.1mF
C33
0.1mF
C14
0.1mF
VCC
GND
74541
VCC
RCO
QA
QB
QC
QD
ENT
2
2
2
2
2
1
C32
0.1mF
1
2
TP5
1
10V 10mF
2
Y1
Y2
Y3
Y4
Y5
Y6
Y7
Y8
U7
16
15
14
13
12
11
10
9
1
C24
0.1mF
2
C43
15pF
C12
0.1mF
12
C4
+
20
10
18
17
16
15
14
13
12
11
VCC
GND
Y1
Y2
Y3
Y4
Y5
Y6
Y7
Y8
1
C22
+
10mF
10V
2
2
1
1
C42
15pF
2
1
10V 10mF
20
10
18
17
16
15
14
13
12
11
JP28
12
JP30
12
R33
1kV
1
C39
0.001mF
1
C36
0.1mF
2
DUTDRVDD
DVDD
C11
0.1mF
12
C5
21
+
1
JP29
12
JP31
2
1
2
DVDD
1
2
DUTAVDD
JP15
B
A
2
21
J4
AVDD
JP26
12
15
AVDD2
16
AVSS2
17
SENSE
18
VREF
19
REFCOM
20
CAPB
21
CAPT
22
U3
CML
23
VINA
24
VINB
25
AVSS1
26
AVDD1
27
DRVSS
28
DRVDD
AD9225
1
C40
0.001mF
2
1
C41
0.001mF
2
TP40
1
R17 22V
1
R5 22V
1
R6 22V
1
R7 22V
1
R8 22V
1
R9 22V
1
R10 22V
1
R11 22V
1
R12 22V
1
R13 22V
1
R14 22V
1
R15 22V
1
R16
22V
1
3
R20
22V
12
1
C55
0.1mF
2
TP34
L2
FBEAD
1
12
1
C52
0.1mF
2
AVDD
14
OTR
13
D11
12
D10
11
D9
10
D8
9
D7
8
D6
7
D5
6
D4
5
D3
4
D2
3
D1
2
D0
1
CLK
1
C37
0.1mF
2
1
C38
0.1mF
2
L6
FBEAD
1
1
C59
0.1mF
2
TP25
1
2
TP24
1
2
TP23
1
2
TP22
1
2
TP21
1
2
TP20
1
2
TP19
1
2
TP26
1
2
TP18
1
2
TP17
1
2
TP16
1
2
TP15
1
2
TP14
1
2
TP13
1
TP12
1
TP11
1
DVDD
1
C50
+
10mF
10V
2
1
+
2
1
+
2
2
1
C1
10mF
10V
C23
10mF
10V
AVDDIN1
1
C47
+
22mF
20V
2
AGND
JP27
1
C58
+
22mF
20V
2
1P1
3P1
5P1
7P1
9P1
11 P1
13 P1
15
17
19
21
23
25
27
29
31
33
35
37
39
1
2
OTR
D11
D10
D9
D8
D7
D6
D5
D4
D3
D2
D1
D0
CLK
2
DRVDD
DUTAUDD
DUTAVDDIN
2
AGND
1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P1
P3
P3
P6
P6
2
4
6
8
10
12
14
16
18
20
22
24
26
28
30
32
34
36
38
40
Figure 44. Evaluation Board Schematic
–22–
REV. A
Page 23
AD9225
Figure 45. Evaluation Board Component Side Layout
(Not to Scale)
Figure 46. Evaluation Board Ground Plane Layout
(Not to Scale)
Figure 48. Evaluation Board Solder Side Layout
(Not to Scale)
Figure 49. Evaluation Board Power Plane Layout
Figure 47. Evaluation Board Component Side Silkscreen
(Not to Scale)
Figure 50. Evaluation Board Solder Side Silkscreen
(Not to Scale)
–23–REV. A
Page 24
AD9225
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Lead Wide Body Small Outline IC
(R-28)
0.7125 (18.10)
0.6969 (17.70)
2815
0.2992 (7.60)
0.2914 (7.40)
141
0.4193 (10.65)
0.3937 (10.00)
C3241a–1–7/98
0.0118 (0.30)
0.0040 (0.10)
0.311 (7.9)
0.301 (7.64)
0.078 (1.98)
0.068 (1.73)
0.008 (0.203)
0.002 (0.050)
PIN 1
0.0500
(1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
0.1043 (2.65)
0.0926 (2.35)
SEATING
PLANE
28-Lead Shrink Small Outline IC
(RS-28)
0.407 (10.34)
0.397 (10.08)
2815
141
SEATING
PLANE
0.07 (1.79)
0.066 (1.67)
0.009 (0.229)
0.005 (0.127)
PIN 1
0.0256
(0.65)
BSC
0.015 (0.38)
0.010 (0.25)
0.0125 (0.32)
0.0091 (0.23)
0.212 (5.38)
0.205 (5.21)
8°
0°
0.0291 (0.74)
0.0098 (0.25)
0.0500 (1.27)
88
08
0.0157 (0.40)
0.03 (0.762)
0.022 (0.558)
x 458
–24–
PRINTED IN U.S.A.
REV. A
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