Datasheet AD9148 Datasheet (ANALOG DEVICES)

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Page 1

Quad 16-Bit,1 GSPS,

Data Sheet

FEATURES

Single-carrier W-CDMA ACLR = 80 dBc at 150 MHz IF Channel-to-channel isolation > 90 dB Analog output

Adjustable 8.7 mA to 31.7 mA R

= 25 Ω to 50 Ω

Novel 2×, 4×, and 8× interpolator eases data interface On-chip fine complex NCO allows carrier placement

anywhere in DAC bandwidth High performance, low noise PLL clock multiplier Multiple chip synchronization interface Programmable digital inverse sinc filter Auxiliary DACs allow for offset control Gain DACs allow for I and Q gain matching Programmable I and Q phase compensation Digital gain control Flexible LVDS digital I/F supports 32- or 16-bit bus width 196-ball CSP_BGA, 12 mm × 12 mm

APPLICATIONS

Wireless infrastructure

LTE, TD-SCDMA, WiMAX, W-CDMA, CDMA2000, GSM MIMO/transmit diversity Digital high or low IF synthesis

TYPICAL SIGNAL CHAIN

COMPLEX BASEBAND COMPLEX IF RF

TxDAC+ Digital-to-Analog Converter

AD9148

GENERAL DESCRIPTION

The AD9148 is a quad, 16-bit, high dynamic range, digital-toanalog converter (DAC) that provides a sample rate of 1000 MSPS. This device includes features optimized for direct conversion transmit applications, including gain, phase, and offset compensation. The DAC outputs are optimized to interface seamlessly with analog quadrature modulators such as the ADL5371/ADL5372/

ADL5373/ADL5374/ADL5375. A serial peripheral interface (SPI)

is provided for programming of the internal device parameters. Full-scale output current can be programmed over a range of 8.7 mA to 31.7 mA. The device operates from 1.8 V and 3.3 V supplies for a total power consumption of 3 W at the maximum sample rate. The AD9148 is enclosed in a 196-ball chip scale package ball grid array with the option of an attached heat spreader.

PRODUCT HIGHLIGHTS

1. Low noise and intermodulation distortion (IMD) enable

high quality synthesis of wideband signals from baseband to high intermediate frequencies.

2. A proprietary DAC output switching technique enhances

dynamic performance.

3. The current outputs are easily configured for various

single-ended or differential circuit topologies.

4. The LVDS data input interface includes FIFO to ease input

timing.

DIGITAL INTERPOL ATION F ILTERS

↑2 ↑2 ↑2

FPGA/ASI C/DSP

↑2 ↑2 ↑2

NOTES

1. AQM = ANALO G QUADRATURE MO DULATOR.

Rev. A

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

LO ±

DAC1

POST DAC

ANALOG FI LTER

DAC2

DAC3

POST DAC

DAC4

Figure 1.

AQM

08910-001

Page 2

AD9148 Data Sheet

Features.............................................................................................. 1

Applications....................................................................................... 1

General Description ......................................................................... 1

Product Highlights ........................................................................... 1

Typical Signal Chain......................................................................... 1

Revision History ............................................................................... 3

Functional Block Diagram .............................................................. 4

Specifications..................................................................................... 5

DC Specifications ......................................................................... 5

Input/Output Signal Specifications............................................ 6

Digital Input Data Timing Specifications ................................. 7

AC Specifications.......................................................................... 8

Absolute Maximum Ratings............................................................ 9

Thermal Resistance ...................................................................... 9

Maximum Safe Power Dissipation............................................. 9

ESD Caution.................................................................................. 9

Pin Configurations and Function Descriptions ......................... 11

Typical Performance Characteristics ........................................... 15

Terminology .................................................................................... 21

Serial Peripheral Interface ............................................................. 22

General Operation of the Serial Interface............................... 22

Data Format ................................................................................22

SPI Pin Descriptions ..................................................................22

SPI Options .................................................................................23

SPI Register Map............................................................................. 24

SPI Register Descriptions.......................................................... 26

Input Data Ports.............................................................................. 40

Dual-Port Mode.......................................................................... 40

Single-Port Mode........................................................................ 40

Byte Mode.................................................................................... 41

Data Interface Options ..............................................................41

Recommended Frame Input Bias Circuitry............................ 41

FIFO Operation ..............................................................................42

Synchronizing and Resetting the FIFO ................................... 43

Monitoring the FIFO Status...................................................... 44

Device Synchronization................................................................. 45

Synchronizing Multiple Devices .............................................. 45

Synchronization with Clock Multiplication............................... 45

Synchronization with Direct Clocking.................................... 47

Additional Synchronization Features ...................................... 48

Interface Timing............................................................................. 49

Digital Data Path ............................................................................ 50

Premodulation............................................................................ 50

Programmable Inverse Sinc Filter............................................ 50

Interpolation Filters ................................................................... 51

Fine Modulation......................................................................... 54

Quadrature Phase Correction................................................... 55

DC Offset Correction ................................................................ 55

Digital Gain Control.................................................................. 55

Clock Generation ........................................................................... 56

DAC Input Clock Configurations............................................ 56

Driving the CLK_x and REFCLK_x Inputs............................ 56

Direct Clocking .......................................................................... 56

Clock Multiplication.................................................................. 57

Analog Outputs............................................................................... 59

Transmit DAC Operation.......................................................... 59

Auxiliary DAC Operation......................................................... 60

Interfacing to Modulators ......................................................... 61

Device Power Dissipation.............................................................. 63

Temperature Sensor ....................................................................... 65

Interrupt Request Operation ........................................................ 66

Interrupt Service Routine.......................................................... 66

Interface Timing Validation.......................................................... 67

SED Operation............................................................................ 67

SED Example .............................................................................. 67

Example Start-Up Routine ............................................................ 68

Derived PLL Settings ................................................................. 68

Derived NCO Settings ............................................................... 68

Start-Up Sequence...................................................................... 68

Device Verification Sequence ................................................... 68

Outline Dimensions....................................................................... 69

Ordering Guide .......................................................................... 70

Rev. A | Page 2 of 72

Page 3

Data Sheet AD9148

REVISION HISTORY

9/11—Rev. 0 to Rev. A

Changes to General Description Section .......................................1

Deleted Input High Voltage, V Low Voltage, V

or VIB Parameter, Table 2, and

or VIB Parameter, Table 2, Input

Note 2, Table 2; Renumbered Sequentially ....................................6

Added Input Voltage Range, V

or VIB Parameter, Table 2.........6

Changes to Table 10 ........................................................................13

Changes to Figure 41 and Figure 42 .............................................23

Changes to 0x1E Addr, Table 12....................................................24

Deleted 0x74 Row, Table 12 ...........................................................26

Changes to PLL Control 2, Table 13 .............................................29

Changes to HB3 Control, 1E, Bit 7 Row, Table 13 ......................32

Deleted LVDS Pad Ctrl Row, Table 13..........................................38

Deleted Frame Input Levels Section and Table 15......................41

Added Recommended Frame Input Bias Circuitry Section and

Figure 45; Renumbered Sequentially............................................41

Changes to Timing Optimization Section...................................48

Added Table 15 ................................................................................ 48

Changes to Filter Implementation Section ..................................50

Changes to Figure 74 ......................................................................57

Deleted Test Access Port Section, Table 27, Figure 92, and

Table 28.............................................................................................68

Changes to Start-Up Sequence Section........................................ 68

Deleted Figure 93 ............................................................................69

Deleted Table 29 ..............................................................................70

6/10—Revision 0: Initial Version

Rev. A | Page 3 of 72

Page 4

AD9148 Data Sheet

FUNCTIONAL BLOCK DIAGRAM

310MHz/620MHz 500MHz/1GHz 500MHz/1GHz

310MHz

–1

SINC

2× 2×

2× 2× 2×

–1

SINC

–1

SINC

2× 2×

–1

SINC

HB1_EN

HB1_CLK

INVSINE_EN

FILTER

COEFFI CIENT

INTERNA L CLOCK TIMING AND CON TRO L LOGI C

SERIAL

IN/OUT PORT

POWER-O N

RESET

1GHz

16-BIT

2×

COS

32-BIT

SIN

NCO

2×

HB3_EN

HB3_CLK

HB2_EN

HB2_CLK

MULTI-CHIP

SYNC

PHASE

CORRECTION

DAC_CLK

GAIN/ OFFSET_CTRL

PLL_CTRL

I OFFSET

Q OFFSET

I OFFSET

Q OFFSET

I GAIN

Q GAIN

I GAIN

Q GAIN

CLOCK

MULTIPLIER

(2× – 16×)

GAIN

16-BIT

GAIN

16-BIT

GAIN

16-BIT

GAIN

REFERENCE

BIAS

DAC1

DAC2

DAC3

DAC4

AUX1

AUX2

AUX3

AUX4

IOUT1_P

IOUT1_N

AUX1_P

AUX1_N

IOUT2_P

IOUT2_N

AUX2_P

AUX2_N

IOUT3_P

IOUT3_N

AUX3_P

AUX3_N

IOUT4_P

IOUT4_N

AUX4_P

AUX4_N

VREF

I120

CLK_P

CLK_N

REFCLK_P/ SYNC_P

REFCLK_N/ SYNC_N

DCIA_P/

DCIA_N

FRAMEA_P/

FRAMEA_N

A[15:0]_P/

A[15:0]_N

B[15:0]_P/

B[15:0]_N

FRAMEB_P/

FRAMEB_N

DCIB_P/

DCIB_N

1.2GHz

DATA RECEIVER

MODE

PROGRAMMING

REGISTERS

310MHz

MOD

FIFO

MOD

FIFO

MOD

PREMOD_EN

PREMOD_CLK

SDO

IRQ

SDIO

SCLK

RESET

08910-002

Figure 2.

Rev. A | Page 4 of 72

Page 5

Data Sheet AD9148

SPECIFICATIONS

DC SPECIFICATIONS

to T

MIN

otherwise noted.

Table 1.

Parameter Min Typ Max Unit

RESOLUTION 16 Bits ACCURACY

Differential Nonlinearity (DNL) ±2.1 LSB

Integral Nonlinearity (INL) ±3.7 LSB MAIN DAC OUTPUTS

Offset Error ±0.001 % FSR

Gain Error (with Internal Reference) ±2 % FSR

Full-Scale Output Current1 8.66 20.2 31.66 mA

Output Compliance Range −1.0 +1.0 V

Output Resistance 10 MΩ

Gain DAC Monotonicity Guaranteed

Settling Time to Within ±0.5 LSB 20 ns TEMPERATURE DRIFT

Main DAC Offset 0.04 ppm/°C

Main DAC Gain 100 ppm/°C

Reference Voltage 30 ppm/°C REFERENCE

Internal Reference Voltage 1.2 V

Output Resistance 5 kΩ ANALOG SUPPLY VOLTAGES

AVDD33 3.13 3.3 3.47 V

CVDD18 1.71 1.8 1.89 V DIGITAL SUPPLY VOLTAGES

IOVDD 1.71 1.8/3.3 3.47 V

DVDD18 1.71 1.8 1.89 V POWER CONSUMPTION (NCO OFF, PLL DISABLED, AND SINC−1 FILTER BYPASSED,

UNLESS OTHERWISE NOTED)

1 × Mode, f

2 × Mode, f

4 × Mode, f

8 × Mode, f

Power-Down Mode 1 12 mW OPERATING RANGE −40 +25 +85 °C

Based on a 10 k external resistor.

, AVDD33 = 3.3 V, IOVDD = 3.3 V, DVDD18 = 1.8 V, CVDD18 = 1.8 V, I

MAX

= 20 mA, maximum sample rate, unless

OUTFS

= 300 MSPS, f

DAC

= 500 MSPS, f

DAC

= 800 MSPS, f

DAC

= 800 MSPS, f

DAC

= 800 MSPS, f

DAC

= 800 MSPS, f

DAC

= 800 MSPS, f

DAC

= 600 MSPS 0.79 W

INTERFACE

= 500 MSPS 1.49 W

INTERFACE

= 400 MSPS 2.18 W

INTERFACE

= 400 MSPS, NCO On 2.47 W

INTERFACE

= 400 MSPS, PLL Enabled 2.26 W

INTERFACE

= 400 MSPS, Sinc−1 Filter Enabled 2.44 W

INTERFACE

= 200 MSPS 2.01 2.16 W

INTERFACE

AVDD33 368 373 mW CVDD18 261 280 mW IOVDD 0.8 1.6 mW DVDD18 1377 1504 mW

Rev. A | Page 5 of 72

Page 6

AD9148 Data Sheet

INPUT/OUTPUT SIGNAL SPECIFICATIONS

to T

MIN

otherwise noted. LVDS driver and receiver are compliant to the IEEE-1596 reduced range link, unless otherwise noted.

Table 2.

Parameter Min Typ Max Unit

CMOS INPUT LOGIC LEVEL (SCLK, SDIO, CS, RESET, TMS, TDI, TCK)

Input VIN Logic High (IOVDD = 1.8 V) 1.2 V Input VIN Logic High (IOVDD = 3.3 V) 2.0 V Input VIN Logic Low (IOVDD = 1.8 V) 0.6 V Input VIN Logic Low (IOVDD = 3.3 V) 0.8 V

CMOS OUTPUT LOGIC LEVEL (SDIO, SDO, IRQ, PLL_LOCK, TDO)

Output V Output V Output V Output V

LVDS RECEIVER INPUTS (A[15:0]_x, B[15:0]_x, DCIA_x, DCIB_x)

Input Voltage Range, VIA or VIB 825 1575 mV Input Differential Threshold, V Input Differential Hysteresis, V Receiver Differential Input Impedance, RIN 80 120 Ω LVDS Input Rate, f

LVDS RECEIVER INPUTS (FRAMEA_x, FRAMEB_x)

Input Voltage Range, VIA or VIB 825 1575 mV

DAC CLOCK INPUT (CLK_P, CLK_N)

Differential Peak-to-Peak Voltage 100 500 2000 mV Common-Mode Voltage (Self-Biasing, AC-Coupled) 1.25 V Maximum Clock Rate 1000 MSPS

REFERENCE CLOCK INPUT (REFCLK_x/SYNC_x)

Differential Peak-to-Peak Voltage 100 500 2000 mV Common-Mode Voltage (Self-Biasing, AC-Coupled) 1.25 V Maximum Clock Rate 500 MSPS Minimum Clock Rate (PLL Enabled)

SERIAL PERIPHERAL INTERFACE

Maximum Clock Rate (SCLK) 40 MHz Minimum Pulse Width High (t Minimum Pulse Width Low (t Set-Up Time, SDI to SCLK (tDS) 1.9 ns Hold Time, SDI to SCLK (tDH) 0.2 ns Data Valid, SDO to SCLK (tDV) 23 ns Setup time, CS to SCLK (t

, AVDD33 = 3.3 V, IOVDD = 3.3 V, DVDD18 = 1.8 V, CVDD18 = 1.8 V, I

MAX

= 20 mA, maximum sample rate, unless

OUTFS

Logic High (IOVDD = 1.8 V) 1.4 V

OUT

Logic High (IOVDD = 3.3 V) 2.4 V

OUT

Logic Low (IOVDD = 1.8 V) 0.4 V

OUT

Logic Low (IOVDD = 3.3 V) 0.4 V

OUT

−100 +100 mV

IDTH

to V

IDTHH

(See Table 4) 1200 MSPS

INTERFACE

20 mV

IDTHL

Loop Divider = /2 125 MSPS Loop Divider = /4 62.5 MSPS Loop Divider = /8 31.25 MSPS Loop Divider = /16 15.625 MSPS

) 12.5 ns

PWH

) 12.5 ns

PWL

DCSB

)

1.4 ns

Rev. A | Page 6 of 72

Page 7

Data Sheet AD9148

DIGITAL INPUT DATA TIMING SPECIFICATIONS

Table 3.

Parameter Min Typ Max Unit

LATENCY (DACCLK CYCLES)

1× Interpolation (with or Without Coarse Modulation) 64 Cycles

2× Interpolation (with or Without Coarse Modulation) 125 Cycles

4× Interpolation (with or Without Coarse Modulation) 254 Cycles

8× Interpolation (with or Without Coarse Modulation) 508 Cycles

Inverse Sinc (1× Interpolation) 10 Cycles

Inverse Sinc (2× Interpolation) 20 Cycles

Inverse Sinc (4× Interpolation) 40 Cycles

Inverse Sinc (8× Interpolation) 80 Cycles

Fine Modulation 12 Cycles

Power–Up Time 100 ms

Table 4. Maximum Rate

Interface Mode f

INTERFACE

Dual Port Mode 620 310 620 1000 1000 1000 Single Port Mode or Byte Mode 1200 300 600 1000 1000 1000

Maximum Rate (MSPS)

DATA

HB1

HB2

HB3

DAC

DAC1

DATA

PORT A

DCIA

DCIB

DATA

PORT B

INPUT

LATCH

ONE DCI

INPUT

LATCH

INTERFACE

DATA

ASSEMBLER

DATA

ASSEMBLER

WRITE PTR A

WRITE PTR B

32 32

INTERFACE

MODE

DATA

FIFO A

FIFO B

2×

A R T

P D A E R

B R T

P D A E R

2×

DATAPATH

CLK GENERATOR

AND DISTRIBUTO R

DATAPATH

2×

HB1

HB2

2×

AND

DAC2

DACCLK

DAC3

AND

DAC4

HB3

DAC

08910-003

Figure 3. Defining Maximum Rates

Rev. A | Page 7 of 72

Page 8

AD9148 Data Sheet

AC SPECIFICATIONS

to T

MIN

otherwise noted.

Table 5.

Parameter Min Typ Max Unit

SPURIOUS-FREE DYNAMIC RANGE (SFDR)

DAC

TWO-TONE INTERMODULATION DISTORTION (IMD)

DAC

NOISE SPECTRAL DENSITY (NSD) EIGHT-TONE, 500 kHz TONE SPACING

DAC

W-CDMA ADJACENT CHANNEL LEAKAGE RATIO (ACLR), SINGLE CARRIER

DAC

W-CDMA ALTERNATE CHANNEL LEAKAGE RATIO, SINGLE CARRIER

DAC

, AVDD33 = 3.3 V, IOVDD = 3.3 V, DVDD18 = 1.8 V, CVDD18 = 1.8 V, I

MAX

= 400 MSPS, f = 600 MSPS, f = 1000 MSPS, f

= 200 MSPS, f = 400 MSPS, f = 800 MSPS, f

= 737.28 MSPS, f = 737.28 MSPS, f = 737.28 MSPS, f = 737.28 MSPS, f

= 80 MHz 72 dBc

OUT

= 100 MHz 67 dBc

OUT

= 100 MHz 65 dBc

OUT

= 100 MHz 85 dBc

OUT

= 120 MHz 82 dBc

OUT

= 150 MHz 76 dBc

OUT

= 80 MHz −160 dBm/Hz

OUT

= 100 MHz −161 dBm/Hz

OUT

= 100 MHz −162.5 dBm/Hz

OUT

= 100 MHz, PLL Off −81 dBc

OUT

= 100 MHz, PLL On −78 dBc

OUT

= 200 MHz, PLL Off −79 dBc

OUT

= 200 MHz, PLL On −72.5 dBc

OUT

= 100 MHz, PLL Off −87 dBc

OUT

= 100 MHz, PLL On −83 dBc

OUT

= 200 MHz, PLL Off −84 dBc

OUT

= 200 MHz, PLL On −80.5 dBc

OUT

= 20 mA, maximum sample rate, unless

OUTFS

Rev. A | Page 8 of 72

Page 9

Data Sheet AD9148

(

)

(

)

ABSOLUTE MAXIMUM RATINGS

Table 7. Thermal Resistance Table 6.

With

Parameter

AVDD33, IOVDD

DVDD18, CVDD18

AGND

DGND

CGND

I120, VREF AGND −0.3 V to AVDD33 + 0.3 V IOUT1_P, IOUT1_N,

IOUT2_P, IOUT2_N, IOUT3_P, IOUT3_N, IOUT4_P, IOUT4_N

A15_P to A0_P, A15_N to A0_N, B15_P to B0_P, B15_N, B0_N

DCIA_P, DCIA_N, FRAMEA_P, FRAMEA_N, DCIB_P, DCIB_N, FRAMEB_P, FRAMEB_N

CLK_P, CLK_N, REFCLK_P, REFCLK_N

CSB, SCLK, SDIO, SDO, TDO, TDI, TCK, TMS, RESET, IRQ, PLL_LOCK

Junction Temperature 125°C Storage Temperature

Range

Respect To Ratin g

AGND, DGND, CGND

DGND, CGND

AGND, CGND

AGND, DGND

AGND −1.0 V to AVDD33 + 0.3 V

DGND −0.3 V to DVDD18 + 0.3 V

DGND −0.3 V to DVDD18+ 0.3 V

CGND −0.3 V to CVDD18 + 0.3 V

DGND −0.3 V to IOVDD + 0.3 V

−65°C to +150°C

−0.3 V to +3.6 V

−0.3 V to +2.10 V

−0.3 V to +0.3 V

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

THERMAL RESISTANCE

Typical θJA, θJB, and θJC are specified vs. the number of PCB layers in still air for each package offering. Airflow increases heat dissipation effectively reducing θ

and θJB.

Rev. A | Page 9 of 72

Package Type θ

196-Ball CSP_BGA 24.7 12.6 5.7 °C/W

19.2 10.9 5.3 °C/W

18.1 10.5 5.3 °C/W

18.0 10.5 5.3 °C/W

196-Ball BGA_ED 20.9 8.6 3.1 °C/W

16.2 7.7 3.1 °C/W

15.2 7.4 3.1 °C/W

15.0 7.4 3.1 °C/W

MAXIMUM SAFE POWER DISSIPATION

The maximum junction temperature for the AD9148 is 125°C. With the thermal resistance of the molded package (CSP_BGA) given for a 12 layer board, the maximum power that can be dissipated in this package can be calculated as

Power

MAX

To increase the maximum power, the AD9148 is available in a second package option (BGA_ED), which includes a heat spreader on top of the package. Also, an external heat sink can be attached to the top of the AD9148 CSP_BGA package. The adjusted maximum power for each of these conditions is shown in Tab le 8.

With the thermal resistance of the heat spreader package (BGA_ED) given for a 12-layer board, the maximum power that can be dissipated in this package can be calculated as

Power

MAX

To increase the maximum power, an external heat sink can be attached to the top of the AD9148 BGA_ED package. The adjusted maximum power for an external heat sink is shown in Tabl e 8.

To aid in the selection of package, the maximum f power dissipation over several operating conditions is shown in Tabl e 9. The maximum f Note that, if the programmable inverse sinc filter is enabled, the maximum f

rate specified in Ta b le 9 decreases.

DAC

ESD CAUTION

JB θJC

−

()

rate applies to all interpolation rates.

DAC

Unit Notes

85125=−

0.18

85125=−

0.15

22.2

67.2

DAC

4-layer board, 25 PCB vias

8-layer board, 25 PCB vias

10-layer board, 25 PCB vias

12-layer board, 25 PCB vias

4-layer board, 25 PCB vias

8-layer board, 25 PCB vias

10-layer board, 25 PCB vias

12-layer board, 25 PCB vias

rate for a given

Page 10

AD9148 Data Sheet

Table 8. Thermal Resistance and Maximum Power

PCB Package Type TA (°C) PCB Layers PCB Vias External Heat Sink1 Case TJ (°C) θJA (°C/W)

196-ball CSP_BGA 85 12 25 No CSP_BGA 125 18.0 2.22 196-ball CSP_BGA 85 12 25 Yes CSP_BGA 125 16.0 2.50 196-ball BGA_ED 85 12 25 No BGA_ED 125 15.0 2.67 196-ball BGA_ED 85 12 25 Yes BGA_ED 125 14.0 2.86

Heat sink is used in the thermal model: 13 mm × 13 mm, 15 mm tall.

Maximum Power (W)

Table 9. Power vs. f

Rate and Functionality

DAC

Maximum f

(MSPS)1

DAC

Coarse Modulation Fine Modulation (NCO) Maximum Power (W) Package Heat-Sink Combination2 PLL Off PLL On PLL Off PLL On

2.22 CSP_BGA No 820 740 695 630

2.50 CSP_BGA Yes 950 875 810 740

2.67 BGA_EP No 1000 945 870 810

2.86 BGA_EP Yes 1000 1000 940 870

Typical maximum f

Heat sink is used in the thermal model: 13 mm × 13 mm, 15 mm tall.

rate with inverse sinc filter off.

DAC

Rev. A | Page 10 of 72

Page 11

Data Sheet AD9148

PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS

IOUT2 IOUT2

CLK

REF

I120

REF

12 13

IOUT3 IOUT3

10 11

AVSS

NC REF

AUX3 AUX3

10 11

POSITIVE TERMINAL

CLKCLK

AUX4

12 13

NEGATIVE TERMINAL

IOUT4AUX4

IOUT4

08910-004

AUX1

CVDD18

AUX2 AUX2

IOUT1

AVDD33

CLK

Figure 4. Pin Configuration (Top View), Analog and Clock Domain Pins

Rev. A | Page 11 of 72

Page 12

AD9148 Data Sheet

SPI

INTERFACE

CSSDO

SCLK

SDIO

FrB

DCIB DCIA

DCIB

A0 A2

RESET IRQ

B0 B2

B3 B4B4B5

A5 A6A6A7

XXXXXX

B5 B6B6B7

B7 B8 B9 B10

A8 A9 A10 A11 A12 A1 3 A1 4 A15A1

10 11

B8 B9 B10

Trch Trc h

B11

12 13

TMS

TDONC PLL

TCK

B13

B12 B14 B15

DCIA

FrA

FrAFrB

TAP INTERFACE

TDI

A4A4A5

IOVDD

DVDD18

A7 A8 A9 A10 A 11 A12 A13 A14 A15

DVSS

10 11

+LVDS –LVDS

A15A15

12 13

Figure 5. Pin Configuration (Top View), Digital Domain Pins

Table 10. Pin Function Description

Pin No. Mnemonic Description

E6, E7, E8, E9 CVDD18 1.8 V Clock Supply. F5, F6, F7, F8, F9, F10 AVDD33 3.3 V Analog Supply. A1, A2, A5, A10, A13, A14, B1,

AVSS Analog Supply Ground. B2, B5, B10, B13, B14, C3, C4, C5, C6, C7, C8, C9, C10, C11, C12, D3, D4, D5, D6, D7, D8, D9, D10, D11, D12, E1, E2, E3, E4, E5, E10, E11, E12, E13, E14, F1, F2, F3, F4, F11, F12, F13, F14

G5, G6, G7, G8, G9, G10, H5, H6,

DVSS Digital Supply Ground. H7, H8, H9, H10

G3, G4 IOVDD

Supply for Serial Ports (SPI and TAP), RESET supplied to these pins.

J5, J6, J7, J8, J9, J10, K5, K6, K7,

DVDD18 1.8 V Digital Supply. K8, K9, K10

B7, B8, H11 NC No Connect. Do not connect to this pin. C1 IOUT1_N DAC 1 Complementary Output Current. D1 IOUT1_P DAC 1 Positive Output Current. A3 IOUT2_N DAC 2 Complementary Output Current. A4 IOUT2_P DAC 2 Positive Output Current. A11 IOUT3_P DAC 3 Positive Output Current. A12 IOUT3_N DAC 3 Complementary Output Current.

Rev. A | Page 12 of 72

08910-005

and IRQ. 1.8 V to 3.3 V can be

Page 13

Data Sheet AD9148

Pin No. Mnemonic Description

C14 IOUT4_N DAC 4 Complementary Output Current. D14 IOUT4_P DAC 4 Positive Output Current. C2 AUX1_N Auxiliary DAC 1 Complementary Output Current. D2 AUX1_P Auxiliary DAC 1 Positive Output Current. B3 AUX2_N Auxiliary DAC 2 Complementary Output Current. B4 AUX2_P Auxiliary DAC 2 Positive Output Current. B11 AUX3_P Auxiliary DAC 3 Positive Output Current. B12 AUX3_N Auxiliary DAC 3 Complementary Output Current. C13 AUX4_N Auxiliary DAC 4 Complementary Output Current. D13 AUX4_P Auxiliary DAC 4 Positive Output Current. A8 I120 Tie to analog ground via a 10 kΩ resistor to generate a 120 μA reference current. A7 VREF

B6, A6 CLK_P/CLK_N Positive/Negative DAC Clock Input (CLK). B9, A9

H3 G1 SDO Serial Data Output for SPI. G2 H1 SDIO Serial Data Input/Output for SPI. H2 SCLK Qualifying Clock Input for SPI. G11, G12 TRENCH Connect this pin to VSS. H12 PLL_LOCK Active High LVCMOS Output. It indicates the lock status of the PLL circuitry. G13 TMS Reserved for Future Use. Connect to DVSS. G14 TDI Reserved for Future Use. Connect to DVSS. H13 TCK Reserved for Future Use. Connect to DVSS. H14 TDO Reserved for Future Use. Leave unconnected. M1, L1 A0_P/A0_N LVDS Data Input Pair, Port A (LSB). P1, N1 A1_P/A1_N LVDS Data Input Pair, Port A. M2, L2 A2_P/A2_N LVDS Data Input Pair, Port A. P2, N2 A3_P/A3_N LVDS Data Input Pair, Port A. P3, N3 A4_P/A4_N LVDS Data Input Pair, Port A. P4, N4 A5_P/A5_N LVDS Data Input Pair, Port A. P5, N5 A6_P/A6_N LVDS Data Input Pair, Port A. P6, N6 A7_P/A7_N LVDS Data Input Pair, Port A. P7, N7 A8_P/A8_N LVDS Data Input Pair, Port A. P8, N8 A9_P/A9_N LVDS Data Input Pair, Port A. P9, N9 A10_P/A10_N LVDS Data Input Pair, Port A.

P10, N10 A11_P/A11_N LVDS Data Input Pair, Port A. P11, N11 A12_P/A12_N LVDS Data Input Pair, Port A. P12, N12 A13_P/A13_N LVDS Data Input Pair, Port A. P13, N13 A14_P/A14_N LVDS Data Input Pair, Port A. P14, N14 A15_P/A15_N LVDS Data Input Pair, Port A (MSB). K13, J13 DCIA_P/DCIA_N LVDS Data Clock Input Pair for Port A. K14, J14 FRAMEA_P/FRAMEA_N

K3, J3 B0_P/B0_N LVDS Data Input Pair, Port B (LSB). M3, L3 B1_P/B1_N LVDS Data Input Pair, Port B. K4, J4 B2_P/B2_N LVDS Data Input Pair, Port B. M4, L4 B3_P/B3_N LVDS Data Input Pair, Port B. M5, L5 B4_P/B4_N LVDS Data Input Pair, Port B M6, L6 B5_P/B5_N LVDS Data Input Pair, Port B.

REFCLK_P/REFCLK_N or SYNC_P/SYNC_N

Active Low Open-Drain Interrupt Request Output. Pull up to IOVDD with

IRQ

RESET

Band Gap Voltage Reference I/O. Decouple to analog ground via a 0.1 μF capacitor. Output impedance is approximately 5 kΩ.

PLL Reference Clock Input (REFCLK_x). This pin has a secondary function as a synchronization input (SYNC_x).

a 10 kΩ resistor. An active low LVCMOS input resets the device. Pull up to IOVDD.

Active Low Chip Select for SPI.

LVDS Frame Input for Port A. Tie to LVDS logic low if not used. Recommended external bias circuit is shown in Figure 49.

Rev. A | Page 13 of 72

Page 14

AD9148 Data Sheet

Pin No. Mnemonic Description

M7, L7 B6_P/B6_N LVDS Data Input Pair, Port B. M8, L8 B7_P/B7_N LVDS Data Input Pair, Port B. M9, L9 B8_P/B8_N LVDS Data Input Pair, Port B. M10, L10 B9_P/B9_N LVDS Data Input Pair, Port B. M11, L11 B10_P/B10_N LVDS Data Input Pair, Port B. K11, J11 B11_P/B11_N LVDS Data Input Pair, Port B. M12, L12 B12_P/B12_N LVDS Data Input Pair, Port B. K12, J12 B13_P/B13_N LVDS Data Input Pair, Port B. M13, L13 B14_P/B14_N LVDS Data Input Pair, Port B. M14, L14 B15_P/B15_N LVDS Data Input Pair, Port B (MSB). K2, J2 DCIB_P/DCIB_N LVDS Data Clock Input Pair for Port B. K1, J1 FRAMEB_P/FRAMEB_N

LVDS Frame Input for Port B. Tie to LVDS logic low if not used. Recommended external bias circuit is shown in Figure 49.

Rev. A | Page 14 of 72

Page 15

Data Sheet AD9148

–

TYPICAL PERFORMANCE CHARACTERISTICS

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

Figure 6. Harmonic Level vs. f

= 200MSPS, SE COND HARMONIC

DATA

= 200MSPS, T HIRD HARMONIC

DATA

= 310MSPS, SE COND HARMONIC

DATA

= 310MSPS, T HIRD HARMONIC

DATA

0 50 100 150 200 250 300

OUT

(MHz)

over f

, 2× Interpolation,

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

Figure 7. Harmonic Level vs. f

= 150MSPS, SE COND HARMONIC

DATA

= 150MSPS, T HIRD HARMONIC

DATA

= 250MSPS, SE COND HARMONIC

DATA

= 250MSPS, T HIRD HARMONIC

DATA

0 50 100 150 200 250 300 350 400 450 500

OUT

(MHz)

over f

, 4× Interpolation,

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

Figure 8. Harmonic Level vs. f

= 125MSPS, SE COND HARMONIC

DATA

= 125MSPS, T HIRD HARMONIC

DATA

0 50 100 150 200 250 300 350 400 450 500

(MHz)

OUT

, 8× Interpolation over f

OUT

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

= 125 MSPS,

DATA

08910-006

08910-007

08910-008

–35

–40

–45

–50

–55

–60

–65

–70

SPUR LEVEL (dBc)

–75

–80

–85

–90

Figure 9. Highest Digital Spur vs. f

= 200MSPS,

DATA

= 310MSPS,

DATA

0 50 100 150 200 250 300

DATA

OUT

(MHz)

OUT

over f

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

–70

SPUR LEVEL (dBc)

–75

–80

–85

–90

Figure 10. Highest Digital Spur vs. f

= 150MSPS,

DATA

= 250MSPS, 2

DATA

0 50 100 150 200 250 300 350 400 450 500

DATA

OUT

DATA

–

(MHz)

OUT

over f

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

–70

SPUR LEVEL (dBc)

–75

–80

–85

–90

Figure 11. Highest Digital Spur vs. f

= 125MSPS,

DATA

0 50 100 150 200 250 300 350 400 450 500

DATA

OUT

(MHz)

OUT

, 8× Interpolation, f

OUT

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

, 2× Interpolation,

, 4× Interpolation,

= 125 MSPS,

DATA

08910-009

08910-010

08910-011

Rev. A | Page 15 of 72

Page 16

AD9148 Data Sheet

–

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

Full-Scale Current = 20 mA, 4× Interpolation, f

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

–10

–20

–30

–40

–50

–60

POWER LEVEL (dBm)

–70

–80

–90

Figure 14. 4× Interpolation, f

0dBFS, SE COND HARMONIC –6dBFS, SECOND HARMONIC –12dBFS, SECOND HARMONIC –18dBFS, SECOND HARMONIC

0 50 100 150 200 250 300

Figure 12. Second Harmonic vs. f

10mA, SECOND HARMONIC 10mA, THIRD HARMONI C

20mA, SECOND HARMONI C 20mA, THIRD HARMONI C 30mA, SECOND HARMONI C 30mA, THIRD HARMONI C

0 50 100 150 200 250 300

Figure 13. Second Harmonic vs. f

Digital Scale = 0 dBFS, 4× Interpolation, f

0 100 200 300 400 500 600

(MHz)

OUT

(MHz)

OUT

over Full-Scale Current,

OUT

FREQUENCY ( MHz)

= 150 MSPS, f

DATA

over Digital Scale,

DATA

= 150 MSPS

DATA

OUT

= 150 MSPS

= 131 MHz

08910-012

08910-013

08910-014

–35

–40

–45

–50

–55

–60

–65

–70

HARMONIC LEVE L (dBc)

–75

–80

–85

–90

Full-Scale Current = 20 mA, 4× Interpolation, f

–10

–20

–30

–40

–50

–60

POWER LEVEL (dBm)

–70

–80

–90

Figure 16. 2× Interpolation, f

0dBFS, T HIRD HARMONIC –6dBFS, T HIRD HARMONIC –12dBFS, T HIRD HARMONIC –18dBFS, T HIRD HARMONIC

0 50 100 150 200 250 300

Figure 15. Third Harmonic vs. f

0 100 200 300 400 500 600

(MHz)

OUT

over Digital Scale,

OUT

FREQUENCY ( MHz)

= 310 MSPS, f

DATA

OUT

–10

–20

–30

–40

–50

–60

POWER LEVEL (dBm)

–70

–80

–90

0 100 200 30 0 400 500 600 700 800 900 1000

Figure 17. 8× Interpolation, f

FREQUENCY ( MHz)

= 125 MSPS, f

DATA

OUT

= 150 MSPS

= 131 MHz

08910-015

08910-016

08910-017

Rev. A | Page 16 of 72

Page 17

Data Sheet AD9148

–

–35

–40

–45

IMD (dBc)

–50

–55

–60

–65

–70

–75

–80

–85

–90

–95

–100

= 200MSPS

DATA

= 310MSPS

DATA

0 50 100 150 200 250 300 350

Figure 18. IMD vs. f

OUT

over f

(MHz)

, 2× Interpolation,

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

= 125MSPS

DATA

–70

IMD (dBc)

–75

–80

–85

–90

–95

–100

0 50 100 150 200 250 300 350 400 450 500

Figure 19. IMD vs. f

(MHz)

OUT

, 8× Interpolation, f

OUT

= 125 MSPS,

DATA

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

IMD (dBc)

–40

–45

–50

–55

–60

–65

–70

–75

–80

–85

–90

–95

–100

0dBFS –6dBFS –12dBFS –18dBFS

0 50 100 150 200 250 300

Figure 20. IMD vs. f

= 150 MSPS, Full-Scale Current = 20 mA

DATA

(MHz)

OUT

over Digital Scale, 4× Interpolation,

OUT

08910-018

08910-019

08910-020

–35

–40

–45

IMD (dBc)

–50

–55

–60

–65

–70

–75

–80

–85

–90

–95

–100

= 150MSPS

DATA

= 250MSPS

DATA

0 50 100 150 200 250 300 350 400 450 500

Figure 21. IMD vs. f

OUT

over f

(MHz)

DATA

, 4× Interpolation,

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

–35

–40

–45

–50

–55

–60

–65

–70

IMD (dBc)

–75

–80

–85

–90

–95

–100

4× Interpolation, f

–35

–40

–45

–50

–55

–60

–65

–70

IMD (dBc)

–75

–80

–85

–90

–95

–100

10mA 20mA 30mA

0 50 100 150 200 250 300

Figure 22. IMD vs. f

PLL OFF PLL ON

0 50 100 150 200 250 300

Figure 23. IMD vs. f

(MHz)

OUT

over Full-Scale Current,

OUT

= 150 MSPS, Digital Scale = 0 dBFS

DATA

(MHz)

OUT

, PLL On and Off,

OUT

Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

08910-021

08910-022

08910-023

Rev. A | Page 17 of 72

Page 18

AD9148 Data Sheet

–

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

Figure 24. Single-Tone NSD Performance vs. f

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

Figure 25. Single-Tone NSD Performance vs. f

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

1×, 200MSPS 2×, 200MSPS 4×, 200MSPS 8×, 100MSPS

0 50 100 150

= 200 MSPS, Full-Scale Current = 20 mA

4× f

DATA

2×, 200MSPS 4×, 200MSPS 8×, 100MSPS

0 50 100 150

= 200 MSPS, Full-Scale Current = 20 mA, PLL On

4× f

DATA

0dB –6dB –12dB –18dB

0 50 100 150 200 25 0 300 350 400

OUT

200

(MHz)

200

(MHz)

250 300

OUT

250 300

OUT

Figure 26. Single-Tone NSD Performance vs. f

= 200 MSPS, Full-Scale Current = 20 mA

4× f

DATA

350

400

, Digital Scale = 0 dBFS,

350

400

, Digital Scale = 0 dBFS,

over Digital Scale,

OUT

08910-024

08910-025

08910-026

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

Figure 27. Eight-Tone NSD Performance vs. f

1×, 200MSPS 2×, 200MSPS 4×, 200MSPS 8×, 100MSPS

0 50 100 150

OUT

200

(MHz)

250 300

, Digital Scale = 0 dBFS,

OUT

Full-Scale Current = 20 mA

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

Figure 28. Single-Tone NSD Performance vs. f

2×, 200MSPS 4×, 200MSPS 8×, 100MSPS

0 50 100 150

OUT

200

(MHz)

250 300

OUT

Full-Scale Current = 20 mA, PLL On

144

–146

–148

–150

–152

–154

–156

NSD (dBm/Hz)

–158

–160

–162

–164

–166

Figure 29. Eight-Tone NSD Performance vs. f

0dB –6dB –12dB –18dB

0 50 100 150 200 25 0 300 350 400

= 200 MSPS, Full-Scale Current = 20 mA

4× f

DATA

OUT

(MHz)

OUT

350

400

350

400

, Digital Scale = 0 dBFS,

over Digital Scale,

08910-027

08910-028

08910-029

Rev. A | Page 18 of 72

Page 19

Data Sheet AD9148

–

–55

–60

–65

–70

–75

ACLR (dBc)

–80

–85

–90

–95

Figure 30. One-Carrier W-CDMA ACLR vs. f

–55

–60

–65

–70

–75

ACLR (dBc)

–80

–85

–90

–95

Figure 31. One-Carrier W-CDMA ACLR vs. f

–55

–60

–65

–70

–75

ACLR (dBc)

–80

–85

–90

–95

Figure 32. One-Carrier W-CDMA ACLR vs. f

0dB, PLL O N 0dB, PLL O FF –3dB, PLL O FF –6dB, PLL O FF

0 50 100 150 200 250 300 350

4× Interpolation, f

0dB, PLL O N 0dB, PLL O FF –3dB, PLL O FF –6dB, PLL O FF

0 50 100 150 200 250 300 350

4× Interpolation, f

0dB, PLL ON 0dB, PLL OF F –3dB, PLL O FF –6dB, PLL O FF

0 50 100 150 200 250 300 350

4× Interpolation, f

OUT

(MHz)

DATA

(MHz)

DATA

(MHz)

DATA

, Adjacent Channel,

OUT

= 184.32 MHz

, Alternate Channel,

OUT

= 184.32 MHz

, Second Alternate Channel,

OUT

= 184.32 MHz

CENTER 150.00MHz #RES BW 30kHz

RMS RESULTS

ARRIER POWER 5.000MHz 3.840MHz –78.88 –92.35 –77. 98 –91.45

13.47dBm/ 10. 00MHz 3.840MHz –82.12 –95.59 –82.65 –96.12

08910-030

.84000MHz 15.00MHz 3.840MHz –82.18 –95.65 –82.28 –95.75

VBW 300kHz

FREQ

OFFSET

SWEEP 112. 5ms (601 PTS)

REF BW

Figure 33. One-Carrier W-CDMA ACLR, f

4× Interpolation, f

CENTER 150.00MHz #RES BW 30kHz

RMS RESULTS

ARRIER POWER 5.000MHz 3.840MHz –74.50 –87.27 –73. 79 –86.56

12.77dBm/ 10. 00MHz 3.840MHz –82.72 –95.49 –82.99 –95.76

08910-031

.84000MHz 15.00MHz 3.840MHz –82.97 –95.74 –83.54 –96.31

VBW 300kHz

FREQ

OFFSET

= 184.32 MHz, PLL Off

DATA

SWEEP 112. 5ms (601 PTS)

REF BW

Figure 34. One-Carrier W-CDMA ACLR, f

4× Interpolation, f

START 1.0MHz

08910-032

#RES BW 30kHz SWEEP 1.6 85s (601 PTS)

Figure 35. One-Carrier W-CDMA, f

= 184.32 MHz, PLL On

DATA

VBW 30kHz

OUT

= 150 MHz, f

LOWER

dBc dBm

LOWER

dBc dBm

SPAN 34.68MHz

UPPER

dBc dBm

= 150 MHz,

OUT

SPAN 34.68MHz

UPPER

dBc dBm

= 150 MHz,

OUT

STOP 368. 6MHz

= 737.28 MSPS,

DAC

08910-033

08910-034

08910-035

4× Interpolation, −3 dBFS

Rev. A | Page 19 of 72

Page 20

AD9148 Data Sheet

–

CENTER 150.00MHz #RES BW 30kHz

TOTAL CARRI ER POWER –13.30 dBm/15 .3600MHz

REF CARRIER PO WER –19. 14dBm/3.84000M Hz

RCC FILTER: ON FILTER ALPHA 0.22

1 –19.14dBm

2 –19.29dBm 3 –19.24dBm 4 –19.61dBm

VBW 300kHz

FREQ

OFFSET

INTEG BW

5.000MHz 3.840MHz –72.59 –91.81 –72. 99 –92.22

10.00MHz 3.840MHz –73.58 –92.81 –74. 45 –93.67

15.00MHz 3.840MHz –75.18 –94.40 –75. 28 –94.51

Figure 36. Four-Carrier W-CDMA, f

4× Interpolation, −3 dBFS, PLL Off

CENTER 150.00MHz #RES BW 30kHz

TOTAL CARRI ER POWER –13.28 dBm/15 .3600MHz

REF CARRIER PO WER –19. 07dBm/3.84000M Hz

RCC FILTER: ON FILTER ALPHA 0.22

1 –19.07dBm

2 –19.42dBm 3 –19.28dBm 4 –19.45dBm

VBW 300kHz

FREQ

OFFSET

INTEG BW

5.000MHz 3.840MHz –64.50 –64.39 –83.56

10.00MHz 3.840MHz –65.12 –65.20 –84.37

15.00MHz 3.840MHz –65.40

Figure 37. Four-Carrier W-CDMA, f

4× Interpolation, −3 dBFS, PLL On

SWEEP 193. 2ms (601 PTS)

= 150 MHz, f

OUT

SWEEP 193. 2ms (601 PTS)

= 150 MHz, f

OUT

SPAN 59.58MHz

LOWER

dBc dBm

SPAN 59.58MHz

LOWER

dBc dBm

–83.67 –84.29 –84.57 –65.35 –84.52

dBc dBm

DAC

dBc dBm

DAC

UPPER

8910-036

= 737.28 MSPS,

UPPER

8910-037

= 737.28 MSPS,

START 1.0MHz #RES BW 30kHz SWEEP 1.6 85s (601 PTS)

Figure 38. Four-Carrier W-CDMA, f

VBW 30kHz

= 150 MHz, f

OUT

STOP 368. 6MHz

= 737.28 MSPS,

DAC

4× Interpolation, −3 dBFS

–82

–84

–86

–88

–90

–92

–94

–96

–98

CROSSTALK (dB)

–100

–102

–104

–106

–108

–110

0 50 100 150 200 250 300

(MHz)

OUT

Figure 39. Crosstalk (DAC Set 1 to DAC Set 2), 4× Interpolation,

= 150 MSPS, Digital Scale = 0 dBFS, Full-Scale Current = 20 mA

DATA

08910-038

08910-039

Rev. A | Page 20 of 72

Page 21

Data Sheet AD9148

TERMINOLOGY

Integral Nonlinearity (INL)

INL is defined as the maximum deviation of the actual analog output from the ideal output, determined by a straight line drawn from zero scale to full scale.

Differential Nonlinearity (DNL)

DNL is the measure of the variation in analog value, normalized to full scale, associated with a 1 LSB change in digital input code.

Monotonicity

A DAC is monotonic if the output either increases or remains constant as the digital input increases.

Offset Error

The deviation of the output current from the ideal of zero is called offset error. For IOUTx_P, 0 mA output is expected when the inputs are all 0s. For IOUTx_N, 0 mA output is expected when all inputs are set to 1.

Gain Error

The difference between the actual and ideal output span. The actual span is determined by the difference between the output when all inputs are set to 1 and the output when all inputs are set to 0.

Output Compliance Range

The range of allowable voltage at the output of a current-output DAC. Operation beyond the maximum compliance limits can cause either output stage saturation or breakdown, resulting in nonlinear performance.

Temp er at u re D ri ft

Temperature drift is specified as the maximum change from the ambient (25°C) value to the value at either T

MIN

or T

MAX

. For offset and gain drift, the drift is reported in ppm of full-scale range (FSR) per degrees Celsius. For reference drift, the drift is reported in ppm per degrees Celsius.

Power Supply Rejection (PSR)

The maximum change in the full-scale output as the supplies are varied from minimum to maximum specified voltages.

Settling Time

The time required for the output to reach and remain within a specified error band around its final value, measured from the start of the output transition.

In-Band Spurious Free Dynamic Range (SFDR)

The difference, in decibels, between the peak amplitude of the output signal and the peak spurious signal between dc and the frequency equal to half the input data rate.

Out-of-Band Spurious Free Dynamic Range (SFDR)

The difference, in decibels, between the peak amplitude of the output signal and the peak spurious signal within the band that starts at the frequency of the input data rate and ends at the Nyquist frequency of the DAC output sample rate. Normally, energy in this band is rejected by the interpolation filters. This specification, therefore, defines how well the interpolation filters work and the effect of other parasitic coupling paths on the DAC output.

Total Harmonic Distortion (THD)

THD is the ratio of the rms sum of the first six harmonic components to the rms value of the measured fundamental. It is expressed as a percentage or in decibels.

Signal-to-Noise Ratio (SNR)

SNR is the ratio of the rms value of the measured output signal to the rms sum of all other spectral components below the Nyquist frequency, excluding the first six harmonics and dc. The value for SNR is expressed in decibels.

Interpolation Filter

An interpolation filter up-samples the input digital data by a multiple of f

(interpolation rate) and then filters out the

DATA

undesired spectral images created by the up-sampling process.

Adjacent Channel Leakage Ratio (ACLR)

The ratio in dBc between the measured power within a channel relative to its adjacent channel.

Complex Image Rejection

In a traditional two-part upconversion, two images are created around the second IF frequency. These images have the effect of wasting transmitter power and system bandwidth. By placing the real part of a second complex modulator in series with the first complex modulator, either the upper or lower frequency image near the second IF can be rejected.

Rev. A | Page 21 of 72

Page 22

AD9148 Data Sheet

SERIAL PERIPHERAL INTERFACE

SDO

SDIO

SPI

PORT

08910-040

CLK

Figure 40. SPI Por

The serial port is a flexible, synchronous serial communications port allowing easy interface to many industry-standard microcontrollers and microprocessors. The serial I/O is compatible with most synchronous transfer formats, including both the

Motorola SPI and Intel

® SSR protocols. The interface allows

read/write access to all registers that configure the AD9148. Single- or multiple-byte transfers are supported, as well as MSBfirst or LSB-first transfer formats. The serial interface ports can be configured as a single pin I/O (SDIO) or two unidirectional pins for input/output (SDIO/SDO).

GENERAL OPERATION OF THE SERIAL INTERFACE

There are two phases to a communication cycle with the AD9148. Phase 1 is the instruction cycle (the writing of an instruction byte into the device), coincident with the first eight SCLK rising edges. The instruction byte provides the serial port controller with information regarding the data transfer cycle, Phase 2 of the communication cycle. The Phase 1 instruction byte defines whether the upcoming data transfer is a read or a write, and the starting register address for the first byte of the data transfer. The first eight SCLK rising edges of each communication cycle are used to write the instruction byte into the device.

A logic high on the port timing to the initial state of the instruction cycle. From this state, the next eight rising SCLK edges represent the instruction bits of the current I/O operation, regardless of the state of the internal registers or the other signal levels at the inputs to the SPI port. If the SPI port is in an instruction cycle or a data transfer cycle, none of the present data is written.

The remaining SCLK edges are for Phase 2 of the communication cycle. Phase 2 is the actual data transfer between the device and the system controller. Phase 2 of the communication cycle is a transfer of one or more data bytes. Registers change immediately upon writing to the last bit of each transfer byte.

pin followed by a logic low resets the SPI

DATA FORMAT

The instruction byte contains the information shown in Tab le 1 1.

Table 11. SPI Instruction Byte

I7 (MSB) I6 I5 I4 I3 I2 I1 I0 (LSB)

R/W

A6 A5 A4 A3 A2 A1 A0

R/W, Bit 7 of the instruction byte, determines whether a read or a write data transfer occurs after the instruction byte write. Logic high indicates a read operation, and Logic 0 indicates a write operation.

A6 through A0, Bit 6 through Bit 0 of the instruction byte, determine the register that is accessed during the data transfer portion of the communication cycle. For multibyte transfers, this address is the starting byte address. The remaining register addresses are generated by the device based on the LSB-first bit (Register 0x00, Bit 6).

SPI PIN DESCRIPTIONS

Serial Clock (SCLK)

The serial clock pin synchronizes data to and from the device and runs the internal state machines. The maximum frequency of SCLK is 40 MHz. All data input is registered on the rising edge of SCLK. All data is driven out on the falling edge of SCLK.

Chip Select (CS)

Active low input starts and gates a communication cycle. It allows more than one device to be used on the same serial communications lines. The SDO and SDIO pins go to a high impedance state when this input is high. Chip select should stay low during the entire communication cycle.

Serial Data I/O (SDIO)

Data is always written into the device on this pin. However, this pin can be used as a bidirectional data line. The configuration of this pin is controlled by Register 0x00, Bit 7. The default is Logic 0, configuring the SDIO pin as unidirectional.

Serial Data Output (SDO)

Data is read from this pin for protocols that use separate lines for transmitting and receiving data. In the case where the device operates in a single bidirectional I/O mode, this pin does not output data and is set to a high impedance state.

Rev. A | Page 22 of 72

Page 23

Data Sheet AD9148

SPI OPTIONS

The serial port can support both MSB-first and LSB-first data formats. This functionality is controlled by the LSB first bit (Register 0x00, Bit 6). The default is MSB first (LSB first = 0).

When LSB first = 0 (MSB first), the instruction and data bit must be written from MSB to LSB. Multibyte data transfers in MSBfirst format start with an instruction byte that includes the register address of the most significant data byte. Subsequent data bytes should follow from the high address to the low address. In MSBfirst mode, the serial port internal byte address generator decrements for each data byte of the multibyte communication cycle.

When LSB first = 1 (LSB first), the instruction and data bit must be written from LSB to MSB. Multibyte data transfers in LSBfirst format start with an instruction byte that includes the register address of the least significant data byte followed by multiple data bytes. The serial port internal byte address generator increments for each byte of the multibyte communication cycle.

The serial port controller data address decrements from the data address written toward 0x00 for multibyte I/O operations if the MSB-first mode is active. The serial port controller address increments from the data address written toward 0x1F for multibyte I/O operations if the LSB-first mode is active.

INSTRUCTI ON CYCLE DATA TRANS FER CYCLE

SCLK

SDIO

SDO

R/W A6 A5 A4 A3 A2 A1 A0 D7 D6ND5

D7 D6ND5

Figure 41. Serial Register Interface Timing MSB First

D00D10D20D3

INSTRUCTION CY CLE DATA TRANSF ER CYCLE

SCLK

SDIO

SDO

A0 A1 A2 A3 A4 A5 A6 R/W D00D10D2

D00D10D2

D7ND6ND5ND4

08910-042

Figure 42. Serial Register Interface Timing LSB First

CLK

SDIO

DCSB

PWH

SCLK

PWL

INSTRUCTIO N BIT 6INSTRUCTION BIT 7

08910-043

Figure 43. Timing Diagram for SPI Register Write

SCLK

SDIO

SDO

DATA BIT n – 1DATA BIT n

08910-044

Figure 44. Timing Diagram for SPI Register Read

08910-041

Rev. A | Page 23 of 72

Page 24

AD9148 Data Sheet

SPI REGISTER MAP

Table 12. Register Map

Addr

0x00 Comm SDIO

0x01 Power

0x03 Data format Binary

0x04 Interrupt

0x05 Interrupt

0x06 Event Flag 0 PLL lock

0x07 Event Flag 1 AED compare

0x08 Clock

0x0A PLL Control 0 PLL enable PLL

0x0C PLL Control 1 PLL Loop Bandwidth[2:0] 0 1 0 0 1 0xF1 0x0D PLL Control 2 N2[1:0] PLL cross

0x0E PLL Status 0 PLL Control Voltage[3:0] 0x0F PLL Status 1 VCO Band Readback[5:0] 0x10 Sync Control 0 Syn c

0x11 Sync Control 1 Sync Phase Request[5:0] 0x00 0x12 Sync Status 0 Sync lost Sync

0x14 Data receiver

0x15 Data receiver

0x17 FIFO Status/

0x18 FIFO Status

0x19 FIFO Status/

0x1A FIFO Status

0x1C HB1 control Enable

0x1D HB2 control HB2[2:0] Bypass

0x1E HB3 control Bypass

control

Enable 0

Enable 1

receiver control

control

status

Control Port A

Port A

Control Port B

Port B

direction Power-

Down DAC Set 1

format Enable PLL

lock lost

Enable AED

lost

CLK duty correction

enable

One DCI 0x00

LVDS rcvr frame high

FIFO Warning 1

pre mod

digital gain and phase adjustment

LSB/ MSB first

PowerDown DAC Set 2

Q first enable

Enable PLL lock

PLL lock Sync lock

REFCLK duty correction

manual enable

FIFO rate/ data rate toggle

locked

LVDS rcvr frame low

FIFO Warning 2

Bypass

−1

sinc

HB3[2:0] Bypass

Software reset

Powerdown data receiver

Dual-port mode

Enable sync lock lost

lost

CLK cross correction

Rising

LVDS rcvr DCI high

FIFO reset aligned

HB1[1:0] Bypass

DAC SPI select

0x00

Bus swap Byte mode Byte

Enable sync lock

compare pass

Sync lock FIFO SPI

pass

REFCLK cross correction

control enable

LVDS rcvr DCI low

FIFO SPI align ack

FIFO Level[7:0]

FIFO SPI align ack

FIFO Level[7:0]

0x00

swap

Enable

Enable AED compare fail

AED compare fail

0 1 1 1 0x37

Manual VCO Band[5:0] 0x40

edge sync

LVDS rcvr Port B high

FIFO SPI align requesting

FIFO SPI aligned

Enable SED compare fail

aligned SED

compare fail

N0[1:0] N1[1:0] 0xD9

LVDS rcvr Port B low

0x20

Enable FIFO Warning 1

0x00

FIFO Warning 1

Sync Averaging[2:0] 0x08

LVDS rcvr Port A high

FIFO Phase Offset[2:0] 0x00

Enable FIFO Warning 2

FIFO Warning 2

LVDS rcvr Port A low

HB1

HB2

HB3

0x00

0x40

0x00

0x81

Rev. A | Page 24 of 72

Page 25

Data Sheet AD9148

Addr

0x1F Chip ID Chip ID[7:0] 0x20 0x201 Coeff I Byte 0 0 Coeff_1i[3:0] Coeff_0i[2:0] 0x00

0x211

0x221

0x231

0x241

0x251

0x261

0x271

0x281

0x291

0x2A1

0x2B1

0x2C1

0x2D1

0x2E1

0x2F1

0x301

0x311

0x321

0x331

0x341

0x351

0x361

0x371

0x381

0x391

0x3A1

0x3B1

0x3C1

0x3D1

0x3E1

0x3F1 0x40 SED control/

0x411

0x421

0x431

0x441

0x501

0x511

Name

Coeff I Byte 1 Coeff_3i[2:0] Coeff_2i[4:0] 0xC0

Coeff I Byte 2 Coeff_4i[2:0] 0 Coeff_3i[6:3] 0xEF

Coeff I Byte 3 0 Coeff_4i[9:3] 0x7F

Coeff Q Byte 0 0 Coeff_1q[3:0] Coeff_0q[2:0] 0x69

Coeff Q Byte 1 Coeff_3q[2:0] Coeff_2q[4:0] 0xE6

Coeff Q Byte 2 Coeff_4q[2:0] 0 Coeff_3q[6:3] 0x0D

Coeff Q Byte 3 0 Coeff_4q[9:3] 0x00

I phase adj LSB Phase Word I[7:0] 0x00

I phase adj MSB

Q phase adj LSB

Q phase adj MSB

I DC offset LSB DC Offset I[7:0] 0x00

I DC offset MSB

Q DC offset LSB

Q DC offset MSB

IDAC FSC adj IDAC FSC Adj[7:0] 0xF9

IDAC control IDAC sleep IDAC FSC Adj[9:8] 0x01

AUX IDAC data AUX IDAC Data[7:0] 0x00

AUX IDAC control

QDAC FSC adj QDAC FSC Adj[7:0] 0xF9

QDAC control QDAC sleep QDAC FSC Adj[9:8] 0x01

AUX QDAC data

AUX QDAC control

SED_S0_L SED Compare Pattern Sample0[7:0] 0xB6

SED_S0_H SED Compare Pattern Sample0[15:8] 0x7A

SED_S1_L SED Compare Pattern Sample1[7:0] 0x45

SED_S1_H SED Compare Pattern Sample1[15:8] 0xEA

SED3_S2_L SED Compare Pattern Sample2[7:0] 0x16

SED3_S2_H SED Compare Pattern Sample2[15:8] 0x1A

SED4_S3_L SED Compare Pattern Sample3[7:0] 0xC6

SED4_S3_H SED Compare Pattern Sample3[15:8] 0xAA

status

SED_R_L SED Status Rising Edge Samples[7:0]

SED_R_H SED Status Rising Edge Samples[15:8]

SED_F_L SED Status Falling Edge Samples[7:0]

SED_F_H SED Status Falling Edge Samples[15:8]

I gain control I Gain[7:0] 0x40

Q gain control Q Gain[7:0] 0x40

Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0 Default

Phase Word I[9:8] 0x00

Phase Word Q[7:0] 0x00

Phase Word Q[9:8] 0x00

DC Offset I[15:8] 0x00

DC Offset Q[7:0] 0x00

DC Offset Q[15:8] 0x00

AUX IDAC sign

AUX QDAC sign

SED compare enable

AUX IDAC current direction

AUX QDAC current direction

Port B error detected

AUX IDAC powerdown

AUX QDAC powerdown

Port A error detected

AUX IDAC Data[9:8] 0x00

AUX QDAC Data[7:0] 0x00

AUX QDAC Data[9:8] 0x00

Auto-

clear enable

Port B compare failed

Port A compare failed

Compare passed

0x00

Rev. A | Page 25 of 72

Page 26

AD9148 Data Sheet

Addr

0x54 FTW (LSB) FTW[7:0] 0x00 0x55 FTW FTW [15:8] 0x00 0x56 FTW FTW[23:16] 0x00 0x57 FTW (MSB) FTW[31:24] 0x00

0x58 Phase offset

0x59 Phase offset

0x5A DDS/mod

0x5C Die Temp

0x5D Die Temp

0x5E Die temp LSB D ie Temp[ 7:0] 0x5F Die temp MSB Die Temp[15:8] 0x72 DCI delay DCI Delay[1:0] 0x00

SPI REGISTER DESCRIPTIONS

Name

(MSB)

(LSB)

control

Control 0

Control 1

Register 0x20 to Register 0x3F and Register 0x41 to Register 0x51 configure DAC 1 (I) and DAC 2 (Q) data paths with DAC SPI select = 0 (Register 0x00[4]). Register 0x20

to Register 0x3F and Register 0x41 to Register 0x51 configure DAC 3 (I) and DAC 4 (Q) data paths with DAC SPI select = 1 (Register 0x00[4]).

Bit 7 Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1 Bit 0 Default

NCO Phase Offset[15:8] 0x00

NCO Phase Offset[7:0] 0x00

Bypass DDS/MOD

Latch

0 0 0 0 1 0 1 0 0x20

Frame NCO

reset ack

Frame NCO reset request

FTW update ack

FTW update request

Sideband

temp data

select

Temp sensor power-down

0x80

0x01

Table 13. Register Descriptions

Addr

(Hex) Bit Name Function Default

Comm 00 7 SDIO SDIO operation. 0

0 = SDIO operates as an input only.

1 = SDIO operates as bidirectional input/output.

6 LSB/MSB first SPI communication LSB first (default is MSB first). 0

0 = MSB first.

1 = LSB first.

5 Software Reset Software reset. 0

Reset is asserted when this bit transitions from 0 to 1.

4 DAC SPI select

Selects which DAC data path Register 0x20 to Register 0x3F and Register 0x41 to Register 0x51 configure.

0 = DAC 1 (I path) and DAC 2 (Q path) are configured. 0

1 = DAC 3 (I path) and DAC 4 (Q path) are configured.

Power Control 01 7

Power-Down

Power down DAC 1 and power down DAC 2. 0

DAC Set 1

Power-Down

Power down DAC 3 and power down DAC 4. 0

DAC Set 2

Power-down data receiver

Power down the input data receiver. 0

Rev. A | Page 26 of 72

Page 27

Data Sheet AD9148

Addr

(Hex) Bit Name Function Default

Data Format 03 7 Binary format

6 Q first enable Indicates I/Q data pairing on data input; I first (0), Q first (1). 0

5 Dual-port mode Number of input data ports used. 1

Single port (0), dual port (1).

4 Bus swap 0 = normal data input bus pin out (MSB to LSB). 0

1 = inverted data input bus pin out (LSB to MSB).

3 Byte mode 0 = data input bus is 16-bit wide on each port. 0

1 = data input bus is two 8-bit wide buses on Port A.

2 Byte swap 0 = normal data input bus pin out (MSB to LSB). 0

1 = inverted data input bus pin out (LSB to MSB).

Interrupt Enable 0 04 7 Enable PLL lock lost Enables interrupt for PLL lock lost. 0

6 Enable PLL lock Enables interrupt for PLL lock. 0

4 Enable sync lock Enables interrupt for sync lock. 0

Enable sync lock lost

Enable FIFO SPI aligned

Input data is in twos complement format (0) or unsigned binary format (1).

Enables interrupt for sync lock lost. 0

Enables interrupt for FIFO SPI aligned. 0

Interrupt Enable 1 05 4

Enable FIFO Warning 1

Enable FIFO Warning 2

Enable AED compare pass

Enable AED compare fail

Enable SED compare fail

Enables interrupt for FIFO Warning 1. 0

Enables interrupt for FIFO Warning 2. 0

Enables interrupt for AED compare pass. 0

Enables interrupt for AED compare fail. 0

Enables interrupt for SED compare fail. 0

Rev. A | Page 27 of 72

Page 28

AD9148 Data Sheet

Addr

(Hex) Bit Name Function Default

Event Flag 0 (All bits are high when interrupt is active. Clear interrupt by writing respective bit high.)

4 Sync lock

2 FIFO SPI aligned

1 FIFO Warning 1

0 FIFO Warning 2

Event Flag 1(All bits are high when interrupt is active. Clear interrupt by writing respective bit high).

06 7 PLL lock lost

6 PLL lock

5 Sync lock lost

07 4 AED compare pass

3 AED compare fail

1 = indicates that the PLL that was previously locked has unlocked from the reference signal.

1 = indicates that the PLL has locked to the reference clock input.

1 = indicates that the sync logic that was previously locked has lost alignment.

1 = indicates that the sync logic achieved sync alignment. This is indicated when no phase changes are requested for at least a few full averaging cycles.

1 = indicates that a FIFO reset originating from a serial portbased request has successfully completed.

1 = indicates that the difference between the FIFO read and write pointers is 1.

1 = indicates that the difference between the FIFO read and write pointers is 2.

1 = indicates that the SED logic detected a valid input data pattern comparison against the preprogrammed expected values.

1 = indicates that the SED logic detected an invalid input data pattern comparison against the preprogrammed expected values. This automatically clears when eight valid I/Q data pairs are received.

2 SED compare fail

Clock Receiver Control 08 7

3:0 0111 Always set these bits to 0111 0111

PLL Control 0 0A 7 PLL enable Enables PLL clock multiplier. 0

5:0 Manual VCO band VCO band used in manual mode. 0

PLL Control 1 0C 7:5 PLL loop bandwidth Selects PLL loop filter bandwidth. 110

000 = narrowest bandwidth.

…

111 = widest bandwidth.

CLK duty correction

REFCLK duty correction

CLK cross correction

REFCLK cross correction

PLL manual enable

1 = indicates that the SED logic detected an invalid input data pattern comparison against the preprogrammed expected values.

Enables duty-cycle correction on CLK input. 0

Enables duty-cycle correction on REFCLK input. 0

Enables differential crossing correction on CLK input. 1

Enables differential crossing correction on REFCLK input. 1

Enables PLL band selection mode (0 = auto, and 1 = manual). 1

4:0 01001 Set these bits to 01001 for optimal PLL operation. 10001

Rev. A | Page 28 of 72

Page 29

Data Sheet AD9148

Addr

(Hex) Bit Name Function Default

PLL Control 2 0D 7:6 N2 REFCLK-to-PLL controller clock rate (f

). 11

PC_CLK

00 = 2.

01 = 4.

10 = 8.

11 = 16.

PLL cross

must always be less than 50 MHz.

PC_CLK

Enables PLL cross-point control.

control enable

3:2 N0 VCO-to-DACCLK divider. 001

00 = 1.

01 = 2.

10 = 4.

11 = 4.

1:0 N1 DACCLK-to-REFCLK divider. 01

00 = 2.

01 = 4.

10 = 8.

11 = 16.

PLL Status 0 0E 3:0 PLL control voltage PLL VCO control voltage readback value.

Readonly

PLL Status 1 0F 5:0 VCO band readback VCO band value.

Readonly

Sync Control 0 10 7 Sync enable Enables synchronization logic. 0

FIFO rate/data

Operates synchronization at the FIFO reset rate (0)/data rate (1). 0

rate toggle

3 Rising edge sync

Rising edge of CLK samples sync input (1), falling edge of

CLK samples sync input (0).

2:0 Sync averaging Average sync input of number of samples. 000

000 = 1.

001 = 2.

010 = 4.

011 = 8.

100 = 16.

101 = 32.

110 = 64.

111 = 128.

Rev. A | Page 29 of 72

Page 30

AD9148 Data Sheet

Addr

Sync Control 1 11 5:0 Sync phase request Offset of internal divided by 64 clock phase after sync. 000000

000000 = 0 DAC clocks.

…

111111 = 63 DAC clocks.

(Hex) Bit Name Function Default

Sync Status 0 12 7 Sync Lost Synchronization lost.

6 Sync locked Synchronization found.

Data Receiver Control 14 6 One DCI 0 = two DCIs used, DCIA_x and DCIB_x. 0

1 = one DCI used, DCIA_x.

Data Receiver Status 15 7

LVDS receiver frame high

LVDS receiver frame low

LVDS receiver DCI high

LVDS receiver DCI low

LVDS receiver Port B high

LVDS receiver Port B low

LVDS receiver Port A high

Frame input LVDS level > 1.7 V.

Frame input LVDS level < 0.7 V.

DCI input LVDS level > 1.7 V.

DCI input LVDS level < 0.7 V.

Port B input LVDS level > 1.7 V.

Port B input LVDS level < 0.7 V.

Port A input LVDS level > 1.7 V.

Readonly

FIFO Status/ Control Port A

6 FIFO Warning 2 FIFO read and write pointers within ±2

5 FIFO reset aligned FIFO read and write pointers aligned after chip reset.

2:0 FIFO phase offset

000 = 0 offset from optimal phase.

…

111 = 7 offset from optimal phase.

The optimal value is 0.

17 7 FIFO Warning 1 FIFO read and write pointers within ±1.

LVDS receiver Port A low

FIFO SPI align acknowledge

FIFO SPI align requesting

Port A input LVDS level < 0.7 V.

FIFO read and write pointers aligned after SPI driven FIFO reset.

Request FIFO read and write pointers alignment via SPI. 0

FIFO read and write pointer phase offset from optimal phase following FIFO reset.

Rev. A | Page 30 of 72

Readonly

000

Page 31

Data Sheet AD9148

Addr

(Hex) Bit Name Function Default

FIFO Status Port A 18 7:0 FIFO Level Thermometer encoded measure of the FIFO level.

Readonly

FIFO Status/ Control Port B

6 FIFO Warning 2 FIFO read and write pointers within ±2.

19 7 FIFO Warning 1 FIFO read and write pointers within ±1.

Readonly

5 FIFO reset aligned FIFO read and write pointers aligned after chip reset.

Readonly

FIFO SPI align acknowledge

FIFO SPI align

FIFO read and write pointers aligned after SPI driven FIFO reset.

Request FIFO read and write pointers alignment via SPI. 0

Readonly

requesting

2:0 FIFO phase offset

FIFO read and write pointer phase offset from optimal

000

phase following FIFO reset.

000 = 0 offset from optimal phase.

…

111 = 7 offset from optimal phase.

The optimal value is 0.

FIFO Status Port B 1A 7:0 FIFO level Thermometer encoded measure of the FIFO level.

Readonly

HB1 Control 1C 7 Enable pre mod

Enable fS/2 modulation stage that precedes Stage 1 interpolation filter.

6 Bypass sinc-1 Sinc-1 filter bypass. 1

2:1 HB1[1:0]

Modulation mode for first stage interpolation filter (f

HB1

= 2 × f

IN1

00 = input signal modulated by dc. Filter pass band is from −0.2 to +0.2 of f

HB1

01 = input signal modulated by dc. Filter pass band is

HB1

/2. Filter pass band is

HB1

/2. Filter pass band is

HB1

from 0.05 to 0.45 of f

10 = input signal modulated by f from 0.3 to 0.7 of f

11 = input signal modulated by f from 0.55 to 0.95 of f

0 Bypass HB1 First stage interpolation filter bypass. 0

Rev. A | Page 31 of 72

Page 32

AD9148 Data Sheet

Addr

(Hex) Bit Name Function Default

HB2 Control 1D HB2[2:0]

Modulation mode for second stage interpolation filter (f

HB2

= 2 × f

IN2

000 = input signal modulated by dc. Filter pass band is from −0.1 to +0.1 of f

HB2

001 = input signal modulated by dc. Filter pass band is

HB2

/4. Filter pass band is

HB2

/4. Filter pass band is

HB2

/2. Filter pass band is

HB2

/2. Filter pass band is

HB2

/4. Filter pass band is

HB2

from 0.025 to 0.225 of f

010 = input signal modulated by f from 0.15 to 0.35 of f

011 = input signal modulated by f from 0.275 to 0.475 of f

100 = input signal modulated by f from 0.4 to 0.6 of f

101 = input signal modulated by f from 0.525 to 0.725 of f

110 = input signal modulated by 3 f from 0.65 to 0.85 of fHB2.

111 = input signal modulated by 3 f from 0.775 to 0.975 of f

HB2

/4. Filter pass band is

HB2

0 Bypass HB2 Second stage interpolation filter bypass. 0

HB3 Control 1E 7

Bypass digital gain and phase adjustment

1 = bypass digital gain and phase compensation. 1

000

3:1 HB3[2:0]

Modulation mode for third stage interpolation filter

= 2 × f

HB3

000 = input signal modulated by dc. Filter pass band is from −0.1 to +0.1 of f

001 = input signal modulated by dc. Filter pass band is from 0.025 to 0.225 of f

010 = input signal modulated by f from 0.15 to 0.35 of f

011 = input signal modulated by f from 0.275 to 0.475 of f

100 = input signal modulated by f from 0.4 to 0.6 of f

101: Input signal modulated by f from 0.525 to 0.725 of f

110 = input signal modulated by 3 f from 0.65 to 0.85 of f

111 = input signal modulated by 3 f from 0.775 to 0.975 of f

IN3

HB3

/4. Filter pass band is

HB3

/4. Filter pass band is

HB3

/2. Filter pass band is

HB3

/2. Filter pass band is

HB3

/4. Filter pass band is

HB3

/4. Filter pass band is

HB3

000

0 Bypass HB3 Third stage interpolation filter bypass. 1

Chip ID 1F 7:0 Chip ID Chip ID readback. 20

Rev. A | Page 32 of 72

Page 33

Data Sheet AD9148

Addr

Coeff I Byte 0 20 7 0 Set this bit to 0. 0

6:3 Coeff_1i[3:0]

2:0 Coeff_0i

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Coeff I Byte 1 21 7:5 Coeff_3i[2:0]

4:0 Coeff_2i

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Coeff I Byte 2 22 7:5 Coeff_4i[2:0]

(Hex) Bit Name Function Default

I-Path DAC Sinc

-1

Filter Coefficient 2 in twos complement

format.

I-Path DAC Sinc

-1

Filter Coefficient 1 in twos complement

format.

I-Path DAC Sinc

-1

Filter Coefficient 4 (LSB) in twos

complement format.

I-Path DAC Sinc

-1

Filter Coefficient 3 in twos complement

format.

I-Path DAC Sinc

-1

Filter Coefficient 5 (LSB) in twos

complement format.

4 0 Set this bit to 0. 0

3:0 Coeff_3i[6:3]

Set I-Path DAC Sinc

-1

Filter Coefficient 4 (MSB) in twos

complement format.

DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Coeff I Byte 3 23 7 0 Set this bit to 0. 0

6:0 Coeff_4i[9:3]

I-Path DAC Sinc

-1

Filter Coefficient 5 (MSB) in twos

complement format.

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Coeff Q Byte 0 24 7 0 Set this bit to 0. 0

6:3 Coeff_1q[3:0]

Q-Path DAC Sinc

-1

Filter Coefficient 2 in twos complement

format.

2:0 Coeff_0q

Q-Path DAC Sinc

-1

Filter Coefficient 1 in twos complement

format.

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

Coeff Q Byte 1 25 7:5 Coeff_3q[2:0]

Q-Path DAC Sinc

-1

Filter Coefficient 4 (LSB) in twos

complement format.

4:0 Coeff_2q

Q-Path DAC Sinc

-1

Filter Coefficient 3 in twos complement

format.

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

Rev. A | Page 33 of 72

Page 34

AD9148 Data Sheet

Addr

Coeff Q Byte 2 26 7:5 Coeff_4q[2:0]

4 0 Set this bit to 0. 0

3:0 Coeff_3q[6:3]

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

Coeff Q Byte 3 27 7 0 Set this bit to 0. 0

6:0 Coeff_4q[9:3]

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

I Phase Adj LSB 28 7:0 Phase Word I[7:0] See Register 0x29. 0

(Hex) Bit Name Function Default

Q-Path DAC Sinc

-1

Filter Coefficient 5 (LSB) in twos

complement format.

Q-Path DAC Sinc

-1

Filter Coefficient 4 (MSB) in twos

complement format.

Q-Path DAC Sinc

-1

Filter Coefficient 5 (MSB) in twos

complement format.

I Phase Adj MSB 29 1:0 Phase Word I[9:8]

Phase Word I[9:0] is used to insert a phase offset between the I and Q data paths. The adjustment range is ±1.75°.

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Q Phase Adj LSB 2A 7:0 Phase Word Q[7:0] See Register 0x2B. 0

Q Phase Adj MSB 2B 1:0 Phase Word Q[9:8]

Phase Word Q[9:0] is used to insert a phase offset between the I and Q data paths. The adjustment range is ±1.75°.

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

I DC Offset LSB 2C 7:0 DC Offset I[7:0] See Register 0x2D. 0

I DC Offset MSB 2D 7:0 DC Offset I[15:8]

DC Offset I[15:0] is a value added directly to the samples written to the IDAC. The LSB bit weight is 20. The adjustment range is ±10 mA.

Set DAC SPI select = 0 to configure DAC 1 path. 0

Set DAC SPI select = 1 to configure DAC 3 path.

Q DC Offset LSB 2E 7:0 DC Offset Q[7:0] See Register 0x2F. 0

Q DC Offset MSB 2F 7:0 DC Offset Q[15:8]

DC Offset Q[15:0] is a value added directly to the samples written to the QDAC. The LSB bit weight is 2

. The adjustment

range is ±10 mA.

Set DAC SPI select = 0 to configure DAC 2 path. 0

Set DAC SPI select = 1 to configure DAC 4 path. 0

Rev. A | Page 34 of 72

Page 35

Data Sheet AD9148

Addr

(Hex) Bit Name Function Default

IDAC FSC Adj 30 7:0 IDAC FSC Adj

0x000 = 8.64 mA.

...

0x200 = 20.14 mA.

…

0x3FF = 31.66 mA.

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

IDAC Control 31 7 IDAC sleep I DAC sleep mode (fast wake-up mode). 0

1:0 IDAC FSC Adj[9:8] IDAC full-scale current adjustment (MSB part) 01

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Aux IDAC Data 32 7:0 AUX IDAC Data

IDAC full-scale current adjustment (LSB part). IDAC FS Adj[9:0] sets the full-scale current of the IDAC. The full-scale current can be adjusted from 8.64 mA to 31.6 mA in step sizes of approximately 22.5 μA.

Auxiliary IDAC data (LSB part). AUX IDAC Data[9:0] sets the magnitude of the aux DAC current. The range is 0 mA to 2 mA, and the step size is 2 μA.

0x000 = 0.000 mA.

0x001 = 0x002 mA.

…

0x3FF = 2.046 mA.

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Aux IDAC Control 33 7 AUX IDAC sign Auxiliary IDAC output sign. 0

0 = positive, current is directed to the AUXx_P pin.

1 = negative, current is directed to the AUXx_N pin.

0 = source.

1 = sink.

1:0 AUX IDAC Data[9:8] Auxiliary IDAC data (MSB part). 00

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select =1 to configure DAC 3 path.

AUX IDAC current direction

AUX IDAC power-down

Auxiliary IDAC current direction. 0

Auxiliary IDAC power-down. 0

Rev. A | Page 35 of 72

Page 36

AD9148 Data Sheet

Addr

(Hex) Bit Name Function Default

QDAC FSC Adj 34 7:0 QDAC FSC Adj

0x000 = 8.64 mA

...

0x200 = 20.14mA

…

0x3FF = 31.66 mA

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

QDAC Control 35 7 QDAC sleep Q DAC sleep mode (fast wake-up mode). 0

1:0 QDAC FSC Adj[9:8] QDAC full-scale current adjustment (MSB part). 01

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

Aux QDAC Data 36 7:0 AUX QDAC Data

0x000 = 0.000 mA.

Q DAC full-scale current adjustment (LSB part). QDAC FS Adj[9:0] sets the full-scale current of the QDAC. The full-scale current can be adjusted from 8.64 mA to 31.6 mA in step sizes of approximately 22.5 μA.

Auxiliary QDAC data (LSB part). AUX QDAC Data[9:0] sets the magnitude of the AUX DAC current. The range is 0 mA to 2 mA, and the step size is 2 μA.

0x001 = 0x002 mA.

…

0x3FF = 2.046 mA.

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

Aux QDAC Control 37 7 AUX QDAC sign Auxiliary QDAC output sign. 0

0 = positive, current is directed to the AUXx_P pin.

1 = negative, current is directed to the AUXx_N pin.

0 = source.

1 = sink.

1:0

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

AUX QDAC current direction

AUX QDAC power-down

AUX QDAC Data[9:8]

Auxiliary QDAC current direction. 0

Auxiliary QDAC power-down. 0

Auxiliary QDAC data (MSB part). 00

Rev. A | Page 36 of 72

Page 37

Data Sheet AD9148

Addr

(Hex) Bit Name Function Default

SED_S0_L 38 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S0_H 39 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S1_L 3A 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S1_H 3B 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED Compare Pattern Sample0[7:0]

SED Compare Pattern Sample0[15:8]

SED Compare Pattern Sample1[7:0]

SED Compare Pattern Sample1[15:8]

Compare Pattern Sample0[15:0] is the word that is compared with Data Sample 0 captured at the input interface by the rising edge of DCI.

Compare Pattern Sample1[15:0] is the word that is compared with Data Sample 1 captured at the input interface by the falling edge of DCI.

SED_S2_L 3C 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S2_H 3D 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S3_L 3E 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED_S3_H 3F 7:0

Set DAC SPI select = 0 to configure Port A.

Set DAC SPI select = 1 to configure Port B.

SED Compare Pattern Sample2[7:0]

SED Compare Pattern Sample2[15:8]

SED Compare Pattern Sample3 [7:0]

SED Compare Pattern Sample3[15:8]

Compare Pattern Sample2[15:0] is the word that is compared with Data Sample 2 captured at the input interface by the rising edge of DCI.

Compare Pattern Sample3[15:0] is the word that is compared with Data Sample 3 captured at the input interface by the falling edge of DCI.

Rev. A | Page 37 of 72

Page 38

AD9148 Data Sheet

Addr

SED Control/Status 40 7 SED compare enable Enables the SED circuitry. 0

(Hex) Bit Name Function Default

3 Auto-clear enable Enables the auto reset after eight valid sample sets. 0

0 Compare passed Pass status determined for last sample set. 0

SED_R_L 41 7:0

Set DAC SPI select = 0 to read back errors on Port A.

Set DAC SPI select = 1 to read back errors on Port B.

SED_R_H 42 7:0

Set DAC SPI select = 0 to read back errors on Port A.

Set DAC SPI select = 1 to read back errors on Port B.

SED_F_L 43 7:0

Port B error detected

Port A error detected

Port B compare failed

Port A compare failed

SED Status Rising Edge Samples[7:0]

SED Status Rising Edge Samples[15:8]

SED Status Falling Edge Samples[7:0]

Status of last compare on Port B. 0

Status of last compare on Port A. 0

Fail status determined for last sample set on Port B. 0

Fail status determined for last sample set on Port A. 0

SED Status Rising Edge Samples[15:0] indicate which bits were received in error.

SED Status Falling Edge Samples[15:0] indicate which bits were received in error.

Readonly

Set DAC SPI select = 0 to read back errors on Port A.

Set DAC SPI select = 1 to read back errors on Port B.

SED_F_H 44 7:0

Set DAC SPI select = 0 to read back errors on Port A.

Set DAC SPI select = 1 to read back errors on Port B.

I Gain Control 50 7:0 IGain

Set DAC SPI select = 0 to configure DAC 1 path.

Set DAC SPI select = 1 to configure DAC 3 path.

Q Gain Control 51 7:0 QGain

Set DAC SPI select = 0 to configure DAC 2 path.

Set DAC SPI select = 1 to configure DAC 4 path.

SED Status Falling Edge Samples[15:8]

SED Status Falling Edge Samples[15:0] indicate which bits were received in error.

IGain[7:0] is a value that directly scales the samples written to the IDAC. The bit weighting is MSB = 21 and LSB = 2−6, which yields a multiplier range of 0 to 3.984375.

QGain[7:0] is a value that directly scales the samples written to the QDAC. The bit weighting is MSB = 2 which yields a multiplier range of 0 to 3.984375.

and LSB = 2−6,

Readonly

Rev. A | Page 38 of 72

Page 39

Data Sheet AD9148

Addr

FTW (LSB) 54 7:0 FTW[7:0] See Register 0x57. 0

FTW 55 7:0 FTW[15:8] See Register 0x57. 0

FTW 56 7:0 FTW [23:16] See Register 0x57. 0

(Hex) Bit Name Function Default

FTW (MSB) 57 7:0 FTW [31:24]

Phase Offset MSB 58 7:0

Phase Offset LSB 59 7:0

DDS/Mod Control 5A 7 Bypass DDS/MOD 1 = bypass NCO. 1

5 Frame NCO reset ack

3 FTW update ack

2 FTW update request

0 Sideband select 0 = The modulator output high-side image. 0

NCO Phase Offset[15:8]

NCO Phase Offset[7:0]

Frame NCO reset request

FTW[31:0] is the 32-bit frequency tuning word that determines the frequency of the complex carrier generated by the on-chip NCO. The frequency is not updated when the FTW registers are written. The values are only updated when Register 0x5A[2] transitions from 0 to 1.

See Register 0x59. 0

NCO Phase Offset[15:0] sets the phase of the complex carrier signal when the NCO is reset. The phase offset spans between 0º and 360º. Each bit represents an offset of 0.0055°. Value is in twos complement format.

1 = indicates that the NCO has been reset due to an extended FRAMEx_x pulse signal.

0→1 = The NCO is reset on the first extended FRAMEx_x pulse after this bit transitions from 0 to 1.

1 = indicates that the FTW has been updated with the SPI value.

0→1 = FTW is updated with the SPI value on a 0-to-1 transition of this bit.

1 = The modulator output low-side image. The image is spectrally inverted compared to the input data.

Die Temp Control 0 5C 1 Latch temp data

Die Temp Control 1 5D 7:0 00001010

Die Temp (LSBs) 5E 7:0 Die Temp[7:0] Die Temp[15:0] indicates the approximate die temperature.

Die Temp (MSBs) 5F 7:0 Die Temp[15:8] Die Temp[15:0] indicates the approximate die temperature.

DCI Delay 72 1:0 DCI delay Programmable delay added DCI. 00

00 = no added delay.

01 = 200 ps delay.

10 = 400 ps delay.

11 = 600 ps delay.

Tem p sensor power-down

0 → 1 = latches temp sensor data. This should be completed before the Die Temp[15:0] is readback.

1 = powers down the aux ADC that converts die temperature. 1

Set these bits to 00001010 for optimal temperature sensor operation.

100000

Readonly

Rev. A | Page 39 of 72

Page 40

AD9148 Data Sheet

INPUT DATA PORTS

The AD9148 can operate in three data input modes: dual-port mode, single-port mode, and byte mode. In dual-port mode, DAC 1 and DAC 2 receive data from Port A, and DAC 3 and DAC 4 receive data from Port B. In single-port mode, all four DACs receive data from Port A. In byte mode, all four DACs receive data from Port A, but the port is split into two 8-bit wide buses. In all modes, the data input timing is relative to a DCI signal provided with the data.

DUAL-PORT MODE

In dual-port mode, the DCI signal indicates to which DAC the data is intended. On the rising edge of DCI, data is latched into DAC 1 and DAC 3. On the falling edge of DCI, data is latched into DAC 2 and DAC 4. This pattern repeats continuously.

There is a SPI programmable option (Register 0x14[6]) to provide one DCI for both input ports or two DCIs, where each DCI is associated with one input port. Two DCIs are useful when the data for each port is coming from a different data source. These cases are illustrated in Figure 45 and Figure 46.

DCIA

A[15:0]

B[15:0]

DCIA

A[15:0]

DCIB

B[15:0]

DAC1 DAC2 DAC1 DAC2 DAC1 DAC2 DAC1 DAC2

DAC3 DAC4 DAC3 DAC4 DAC3 DAC4 DAC3 DAC4

Figure 45. Timing Diagram for Dual-Port Mode, One DCI

DAC1 DAC2 DAC1 DAC2 DAC1 DAC2 DAC1 DAC2

DAC3 DAC4 DAC3 DAC4 DAC3 DAC4 DAC3 DAC4

Figure 46. Timing Diagram for Dual-Port Mode, Two DCI

08910-045

08910-046

Each data sample, by default, is expected to be formatted as an MSB sent to Bit 15 and an LSB sent to Bit 0 for each port. The

AD9148 contains an option to swap the bus (Register 0x03[4]).

When this bus swap bit is set, the MSB should be sent to Bit 0, and the LSB should be sent to Bit 15 for each port.

SINGLE-PORT MODE

In single-port mode, a FRAME signal must be provided along with the DCI signal and the data. The FRAME signal indicates to which DAC the data is intended. When FRAME goes high, the first data-word goes to DAC 1, and the second data-word goes to DAC 2. When FRAME goes low, the first data-word goes to DAC 3, and the second data-word goes to DAC 4. This pattern repeats continuously as illustrated in Figure 47.

DCIA

FRAMEA

A[15:0]

Each data sample, by default, is expected to be formatted as an MSB sent to Bit 15 and an LSB sent to Bit 0. When the bus swap bit is set (Register 0x03[4]), the MSB should be sent to Bit 0, and the LSB should be sent to Bit 15 for each port.

The FRAME signal is sampled with the same internal signal as the data and has the same set-up and hold timing relative to DCI. If desired, only the first FRAME pulse needs to be generated. This initializes the internal clock phases inside the device, and data latches just as if the periodic FRAME signal were sent.

DAC1 DAC2 DAC3 DAC4 DAC1 DAC2 DAC3 DAC4

Figure 47. Timing Diagram for Single-Port Mode

08910-047

Rev. A | Page 40 of 72

Page 41

Data Sheet AD9148

Ω

BYTE MODE

In byte mode, a FRAME signal must be provided along with the DCI signal and the data. The most significant byte of the data should correspond with DCI being high, and the least significant byte of the data should correspond with DCI being low. The FRAME signal indicates to which DAC the data is intended. When FRAME is high, data on the top half of the port (A[15:8]) is sent to DAC 1 and data on the bottom half of the port (A[7:0]) is sent to DAC 3. When the FRAME is low, data on the top half of the port is sent to DAC 2 and data on the bottom half of the port is sent to DAC 4. This pattern repeats continuously as shown in Figure 48.

DCIA

FRAMEA

A[15:8]

A[7:0]

DAC1H DAC1L DAC2H DAC2L DAC1H DAC1L DAC2H DAC2 L

DAC3H DAC3L DAC4H DAC4L DAC3H DAC3L DAC4H DAC4L

Figure 48. Timing Diagram for Byte Mode

08910-048

The AD9148 also includes a byte swap feature. By default, the bytes should be formatted as an MSB sent to Bit 15 on Bus 1 and Bit 7 on Bus 2. When byte swap is enabled (Register 0x03[2]), an MSB should be sent to Bit 8 on Bus 1 and Bit 0 on Bus 2. This is described in Ta ble 14 .

Table 14. Byte Swap Formatting

Byte Swap Byte A[15:8] A[7:0]

0 MSB Data Set 1[15:8] Data Set 2[15:8] 0 LSB Data Set 1[7:0] Data Set 2[7:0] 1 MSB Data Set 1[8:15]

Data Set 2[8:15]

1 LSB Data Set 1[0:7] Data Set 2[0:7]

DATA INTERFACE OPTIONS

To enable optimization of the data interface, some additional options have been provided in the following registers:

• Data format (Register 0x03)

• Data receiver control (Register 0x14)

• Data receiver status (Register 0x15)

Depending on the data rate and DCI vs. data skew, the internal DCI can be inverted to meet the valid data timing window.

RECOMMENDED FRAME INPUT BIAS CIRCUITRY

Because the frame signal can be used as a reference clock in the byte mode or as a trigger to reset the FIFO, it is recommended that the frame input be tied to LVDS logic low when it is not used (that is, when it is not driven by an ASIC or FPGA). The external bias circuit shown in Figure 49 is recommended for this purpose. This bias circuit applies to both FRAMEA and FRAMEB ports.

150

51Ω

DVDD18

FRAMEP

FRAMEN

(1.8V)

Figure 49. External Bias Circuit

AD9148

100Ω

8910-145

Rev. A | Page 41 of 72

Page 42

AD9148 Data Sheet

FIFO OPERATION

DATA

PORT A

DCIA

FRAME

FIFO RATE/

DATA RATE

FRAMEB

DCIB

INPUT

LATCH

DATA

ASSEMBLER

WRITE PTR A

WRITE PTR B

WRITE PTR

RESET

LOGIC

RESET

32 BITS

REG 0

REG 1

REG 2

REG 3

REG 4

REG 5

REG 6

REG 7

READ

RESET

FIFO A

OFS[2:0]

FIFO B

OFS[2:0]

32 BITS

PTR

READ POINTER AREAD POINTER B

DATA

PATHS

÷INT

DAC1

AND

DAC2

DACCLK

SYNC

ONE

DCI

DATA

PORT B

INPUT

LATCH

DATA

ASSEMBLER

INTERFACE

MODE

Figure 50. Block Diagram of FIFO

The AD9148 contains two 32-bit wide, 8-word deep FIFOs (one per dual DAC) designed to relax the timing relationship between the data arriving at the DAC input ports and the internal DAC data rate clock. The FIFOs can also be used to provide an adjustable pipeline delay between the DCIx clocks and the DACCLK allowing realignment of data input in a multichip system. This significantly increases the timing budget of the interface.

Figure 50 shows the block diagram of the datapath through the FIFO. The data is latched into the device, is formatted, and is then written into the FIFO register determined by the FIFO write pointer. The value of the write pointer is incremented every time a new word is loaded into the FIFO. Meanwhile, data is read from the FIFO register determined by the read pointer and fed into the digital datapath. The value of the read pointer is updated every time data is read into the datapath from the FIFO. This happens at the data rate, that is the DACCLK rate divided by the interpolation ratio. The difference between the write and read pointers represents the FIFO pipeline delay and is

REG 0

REG 1

REG 2

REG 3

REG 4

REG 5

REG 6

REG 7

3232

DATA

PATHS

DAC3

AND

DAC4

08910-049

important to take into account when understanding the overall pipeline delay of the AD9148.

In single-port and byte interface modes, the incoming digital data is sampled at twice the data rate (DCIA). The data is then assembled based on the interface mode. At the output of the data assembler block, the data samples for DAC 1 and DAC 2 are written to FIFO A and the data samples for DAC 3 and DAC 4 are written to FIFO B at the data rate.

Valid data is transmitted through the FIFO as long as the FIFO does not overflow or become empty. An overflow or empty condition of the FIFO is the same as the write pointer and the read pointer being equal. When both pointers are equal, an attempt is made to read and write a single FIFO register simultaneously. This simultaneous register access leads to unreliable data transfer through the FIFO and must be avoided.

Rev. A | Page 42 of 72

Page 43

Data Sheet AD9148

Nominally, data is written to the FIFO at the same rate as data is read from the FIFO. This keeps the data level in the FIFO constant. If data is written to the FIFO faster than data is read, the data level in the FIFO increases. If the data is written to the device slower than data is read, the data level in the FIFO decreases. For a maximum timing margin, the FIFO level should be maintained near half full, which is the same as maintaining a difference of 4 between the write pointer and read pointer values.

SYNCHRONIZING AND RESETTING THE FIFO

To avoid any concurrent reads and writes to the same FIFO address and to assure a fixed pipeline delay, it is important to reset the state of the FIFO pointers to known states. The pipeline delay in the

AD9148 comes from two sources, FIFO delay and the delay

though the signal processing in the DAC.

To assure a fixed and predictable pipeline delay in the signal processing, the FIFO read operation is synchronized with the DACCLK and, more importantly, in case of interpolation, its divided down version so that the same edge of the slowest clock in the signal processing reads the same data in the FIFO. The synchronization is performed by resetting the FIFO read pointer to a known state relative to the slowest clock used in the signal processing. This synchronization is enabled by setting Bit 7 in Register 0x10 to 1, and it uses the REFCLK/SYNC signal for its reference.

To manage the FIFO pipeline delay, the FIFO write pointer must be synchronized with the read pointer to avoid concurrent access to the FIFO and to potentially compensate for any data input phase mismatch. This synchronization can be performed either at the data rate (see the Data Rate Synchronization section) or at the FIFO rate (see the FIFO Rate Synchronization section).

FIFO Synchronization Modes

To benefit from the advantages of the FIFO functionality in the different modes of operations, PLL on/off, standalone, or multichip synchronization, the FIFO can operate in the following ways:

• Synchronization at the data rate

• Synchronization at the FIFO rate (data rate/FIFO depth)

• No synchronization

As discussed in the Input Data Ports section, in single-port mode and byte mode, the FRAME input is used as a data select signal that indicates to which DAC the input data is intended to be written. When synchronization is needed, the FRAME signal is given another function, initializing the FIFO write pointer address. When the FRAME signal is asserted high for at least the time interval needed to load complete data to the four DACs (which correspond to one DCI period in dual-port mode and two DCI periods in single-port mode or byte mode), the FIFO write pointer is reset to a value dependent on the synchronization mode selected and the FIFO phase offset bits of the corresponding FIFO Status/Control Port x register, Register 0x17 or Register 0x19.

Rev. A | Page 43 of 72

Data Rate Synchronization

In this mode, the REFCLK/SYNC signal is used to reset the FIFO read pointer to 0. The edge of the CLK used to sample the SYNC signal is selected by Bit 3 of Register 0x10. If the PLL is used, REFCLK is used as a SYNC signal, and the FIFO read pointer is reset at the REFCLK rate divided by 64. The data rate synchronization is selected by setting Bit 6 of Register 0x10 to 0.

As previously mentioned, the FRAME signal is used to reset the FIFO write pointer. When the FRAME is asserted, the FIFO write pointer is reset to the address defined in Bits[2:0] of the corresponding FIFO Status/Control Port x register (Register 0x17 or Register 0x19) the next time the read pointer becomes 0 (see Figure 51).

The data rate synchronization, the write pointer of the FIFO, and the read pointer of the FIFO are synchronized at the SYNC rate and have a fixed phase offset.

SYNC

RDPTRA

RDPTRB

FRAMEA

WRPTRA

FRAMEB

WRPTRB

34567

345670123456

FIFO_A WRITE RESET

701234567012

FIFO_B WRITE RESET

012344567012

01234

RESET VALUE FOR REGISTER 0x17[2:0] = 0b100

RESET VALUE FOR REGISTER 0x19[2:0] = 0b100

Figure 51. Timing of the FRAME Input vs. Write Pointer Value in Data Rate

Synchronization

FIFO Rate Synchronization

In this mode, the REFCLK/SYNC signal is used to reset the FIFO read pointer to 0. The edge of the CLK_x used to sample the SYNC signal is selected by Bit 3 of Register 0x10. As previously mentioned, the FRAME signal is used to reset the FIFO write pointer. In the FIFO rate synchronization mode, the FIFO write pointer is reset immediately after the FRAME signal is asserted high for at least the time interval needed to load complete data to the four DACs, and the FIFO write pointer is reset to the address defined in Bits[2:0] of the corresponding FIFO Status/Control Port x register, Register 0x17 or Register 0x19 (see Figure 52

SYNC

FIFO_A AND FIFO_B READ RESET

RDPTRA

RDPTRB

FRAMEA

FRAMEB

01234

012345670123

FIFO_A WRITE

RESET

RPTRA

44567012345

RPTRB

2346 0 12 34567

RESET VALUE FOR REGISTER 0x17[2:0] = 0b100

FIFO_B WRITE

RESET

Figure 52. Timing of the FRAME Input vs. Write Pointer Value in

FIFO Rate Synchronization

56701

RESET VALUE FOR REGISTER 0x19[2:0] = 0b110

08910-050

08910-051

Page 44

AD9148 Data Sheet

No Synchronization

In this mode, Bit 7 in Register 0x10 is set to 0, the pipeline delay in the signal processing is not controlled, and the read pointer of the FIFO is never reset. However, to assure that the FIFO can operate safely and there is no concurrent access to FIFO from the write and read pointer to the same address, it is important to ensure that the phase offset between the two pointers is greater than 2. In consequence, the only FIFO reset that can be used safely is the data rate synchronization, Bit 6 of Register 0x10 set to 0, where the FIFO is reset with a fixed offset of 4 between the write and read pointers. Because there is no SYNC signal, the reset of the FIFO write pointer can only be done by a FRAME signal or an SPI command.

FIFO Reset Commands

Depending on the configuration of the system, the FIFO reset can be done manually or periodically for a multichip system. The AD9148 provides two ways to resetting the FIFO pointers: SPI interface or periodic reset using the FRAME signal.

The SPI also gives access to each FIFO phase offset in Bits [2:0] of the corresponding FIFO status/control registers, Register 0x17 and Register 0x19. The value in these three bits corresponds either to the offset between the write and read pointer in the data rate synchronization or to the absolute address of the FIFO write pointer in the FIFO rate synchronization.

SPI Command for Manual Reset

If a manual reset is acceptable, the FIFO pointer addresses can be reset using the SPI interface.

To initialize the FIFO data level through the SPI, Bit 3 of Register 0x17 (FIFO Port A) or Bit 3 of Register 0x19 (FIFO Port B) should be toggled from 0 to 1 and back. When the write to the register is complete, the corresponding FIFO data level is initialized.

The recommended procedure for a SPI FIFO data level initialization is

Request FIFO Port A or FIFO Port B level reset by setting

1. Bit 3 in Register 0x17 or Bit 3 in Register 0x19 to Logic 1. The FIFO phase offset, Bits [2:0] in Register 0x17 or Bits [2:0] in Register 0x19, should also be written at the same time to set the desired value of offset between the FIFO write and read pointers.

Verify that the part acknowledges the request by ensuring

2. that Bit 4 in Register 0x17 or Bit 4 in Register 0x19 is set to Logic 1.

Remove the request by resetting Bit 3, Register 0x17 or Bit 3,

The FIFO SPI aligned flag in the Event Flag 0 register, Bit 2

4. in Register 0x06, is set when the reset of the write pointer has been realized. Bit 4 in Register 0x17 or Bit 4 in Register 0x19 is reset to 0 to indicate which FIFO has generated this flag.

Note that the SPI writes to Register 0x17 or Register 0x19 should be done while maintaining a constant value in the FIFO phase offset bits.

FIFO Reset Using FRAME Signal

The FIFO pointers can also be reset using the FRAME signals. If only one DCI is used, only the FRAMEA signal is used for the FIFO reset. This mode is enabled by setting Bit 6 in Register 0x10.

As discussed in the FIFO Synchronization Modes section, the FRAME input is used to initialize the FIFO data level value. When the FRAME signal is asserted high for at least the time interval needed to load the complete data to the four DACs, the write pointer is reset depending on the mode of synchronization chosen.

• Data rate synchronization (default), Bit 6 of Register 0x10,

is set to 0. When read pointer reaches 0, write pointer reset to FIFO offset phase.

• FIFO rate synchronization, Bit 6 of Register 0x10, is set to 1.

On the rising edge of the FRAME signal, write pointer reset to FIFO start level.

MONITORING THE FIFO STATUS

The FIFO initialization and status can be read from Register 0x17. This register provides information about the FIFO initialization method and whether the initialization was successful. The MSB of Register 0x17 is a FIFO warning flag that can optionally trigger a

IRQ

device emptying (FIFO level is 1) or overflowing (FIFO level is 7). This is an indication that the data may soon be corrupted, and action should be taken.

The FIFO data level can be read from Register 0x18 at any time. The SPI reported FIFO data level is denoted as a 7-bit thermometer code of the write counter state relative to the absolute read counter being 0. The optimum FIFO data level of four is, therefore, reported as a value of 00001111 in the status register.

Note that, depending on the timing relationship between DCI and the main DACCLK, the FIFO level value can be off by a ±1 count. Therefore, it is important to keep the difference between the read and write points to at least 2.

. This flag is an indication that the FIFO is close to

Rev. A | Page 44 of 72

Page 45

Data Sheet AD9148

DEVICE SYNCHRONIZATION

SYNCHRONIZING MULTIPLE DEVICES

System demands may require that the outputs of multiple DACs be synchronized with each other or with a system clock. Systems that support transmit diversity or beam-forming, where multiple antennas are used to transmit a correlated signal, require multiple DAC outputs to be phase aligned with each other. Systems with a time-division multiplexing transmit chain may require one or more DACs to be synchronized with a system-level reference clock.

Multiple devices are considered synchronized to each other when the state of the clock generation state machines is identical for all parts and time aligned data is being read from the FIFOs of all parts simultaneously. Devices are considered synchronized to a system clock when there is a fixed and known relationship between the clock generation state machine and the data being read from the FIFO and a particular clock edge of the system clock. The

AD9148 has provisions for enabling multiple devices to be

synchronized to each other or to a system clock.

The AD9148 supports synchronization in two different modes, data rate mode and FIFO rate mode. The two modes are distinguished by the lowest rate clock that the synchronization logic attempts to synchronize. In data rate mode, the input data rate represents the lowest synchronized clock. In FIFO rate mode, the FIFO rate, which is the data rate divided by the FIFO depth of 8, represents the lowest rate clock. The advantage of the FIFO

LENGTH TRACES

YSTEM CLOC

FPGA

Figure 53. Typical Circuit Diagram for Synchronizing Devices with Clock Multiplication Enabled

LOW SKEW

CLOCK DRIVER

MATCHED

LENGTH TRACES

MATCHED

rate synchronization is increased setup and hold times of DCI relative to the CLK input. When in data rate synchronization mode, the elasticity of the FIFO is not used to absorb timing variations between the data source and DAC, resulting in tighter setup and hold time requirements.

The method chosen for providing the DAC sampling clock directly impacts the synchronization methods available. When the device clock multiplier is used, only data rate synchronization is available. When the DAC sampling clock is sourced directly, both data rate mode and FIFO rate mode synchronization are available.

SYNCHRONIZATION WITH CLOCK MULTIPLICATION

When using the clock multiplier to generate the DACCLK, the REFCLK/SYNC input signal acts as both the reference clock for the PLL-based clock multiplier and as the synchronization signal. To synchronize devices, the REFCLK/SYNC signal must be distributed with low skew to all of the devices to be synchronized. Skew between the REFCLK/SYNC signals of different devices show up directly as a timing mismatch at the DAC outputs.

The frequency of the REFCLK/SYNC signal is typically equal to the input data rate. The FRAME signal and DCI signals can be created in the FPGA along with the data. A circuit diagram of a typical configuration is shown in Figure 53.

REFCLK/SYNC

FRAME DCI

REFCLK/SYNC FRAME

DCI

OUT1

OUT2

08910-052

Rev. A | Page 45 of 72

Page 46

AD9148 Data Sheet

The following procedure outlines the steps required to synchronize multiple devices. The procedure assumes that the REFCLK/SYNC signal is applied to all of the devices and the PLL of each device is phase locked to it. Each individual device must follow this procedure.

The procedure for synchronization when using the PLL follows:

Configure for data rate, periodic synchronization by writing

1. 0xC0 to the sync control register (Register 0x10).

Read the sync status register (Register 0x12) and verify that

2. the sync locked bit (Bit 6) is set high indicating that the device achieved back-end synchronization and that the sync lost bit (Bit 7) is low. These levels indicate that the clocks are running with a constant and known phase relative to the sync signal.

Reset the FIFO by strobing the FRAME signal high for at

3. least the time interval needed to load complete data to the four DACs. Resetting the FIFO ensures that the correct data is being read from the FIFO. This completes the synchronization procedure, and at this stage, all devices should be synchronized.

SKEW

To maintain synchronization, the skew between REFCLK/SYNC signals of the devices must be less than t

nanoseconds. There

SKEW

is also a setup and hold time to be observed between the DCI and data of each device and the REFCLK/SYNC signal. When resetting the FIFO, the FRAME signal must be held high for at least the time interval needed to load complete data to the four DACs (one DCI period for dual-port mode and two DCI periods for single-port or byte mode). A timing diagram of the input signals is shown in Figure 54.

The example in Figure 54 shows a REFCLK/SYNC frequency equal to the data rate. Whereas this is the most common situation, it is not strictly required for proper synchronization. Any REFCLK/SYNC frequency that satisfies the following equations is acceptable:

SYNC

N = 1, 2, 3, or 4.

where

= f

DACCLK

/2N and f

SYNC

≤ f

DATA

For example, a configuration with 4× interpolation and clock frequencies of f

= 200 MHz, f

DATA

= 1600 MHz, f

VCO

= 100 MHz would be a viable solution.

SYNC

= 800 MHz, and

DACCLK

REFCLK(1)

REFCLK(2)

SU_DCItH_DCI

DCI(2)

FRAME(2)

Figure 54. Timing Diagram Required for Synchronizing Two Devices

08910-053

Rev. A | Page 46 of 72

Page 47

Data Sheet AD9148

SAMPLE RATE CLOCK

SYNC CLOCK

FPGA

LOW SKEW

CLOCK DRIVER

LOW SKEW

CLOCK DRIVER

Figure 55. Typical Circuit Diagram for Synchronizing Devices to a System Clock

MATCHED

LENGTH TRACES

MATCHED

LENGTH TRACES

SYNCHRONIZATION WITH DIRECT CLOCKING

When directly sourcing the DAC sample rate clock to CLK, a separate REFCLK/SYNC input signal is required for synchronization. To synchronize devices, the CLK signals and the REFCLK/SYNC signals must be distributed with low skew to all of the devices being synchronized. This configuration is shown below in Figure 55.

Data Rate Mode Synchronization

The following procedure outlines the steps required to synchronize multiple devices in data rate mode. The procedure assumes that the CLK and REFCLK/SYNC signals are applied to all of the devices. Each individual device must follow the procedure.

The procedure for data rate synchronization when directly sourcing the DAC sampling clock follows:

Configure for data rate, periodic synchronization by

writing 0xC0 to the sync control register (Register 0x10). Additional synchronization options are available (see the Additional Synchronization Features section).

Poll the sync locked bit (Bit 6, Register 0x12) to verify that

the device is back-end synchronized. A high level on this bit indicates that the clocks are running with a constant and known phase relative to the sync signal.

Reset the FIFO by strobing the FRAME signal for at least the

time interval needed to load complete data to the four DACs Resetting the FIFO ensures that the correct data is being read from the FIFO of each of the devices simultaneously. This completes the synchronization procedure, and at this stage, all devices should be synchronized.

To ensure that each of the DACs are updated with the correct data on the same DACCLK edge, two timing relationships must be met on each DAC. DCI (and data) must meet the setup and hold times with respect to the rising edge of CLK, and REFCLK/SYNC

Rev. A | Page 47 of 72

CLK

REFCLK/SYNC

FRAME

DCI

CLK

REFCLK/SYNC

FRAME

DCI

OUT1

OUT2

08910-054

must also meet the setup and hold time with respect to the rising edge of CLK. When resetting the FIFO, the FRAME signal must be held high for at least the time interval needed to load complete data to the four DACs (one DCI period for dualport mode and two DCI periods for single-port or byte mode). When these conditions are met, the outputs of the DACs will be updated within t

SKEW

+ t

nanoseconds of each other. A timing

OUTDLY

diagram that illustrates the timing requirements of the input signals is shown in Figure 56.

SKEW

CLK(1)

CLK(2)

H_DCI

SU_DCI

SYNC(2)

DCI(2)

FRAME(2)

Figure 56. Synchronization Signal Timing Requirements in Data Rate Mode,

2× Interpolation

SU_SYNC

H_SYNC

Figure 56 shows the synchronization signal timing with 2× interpolation, so that f

= ½ × f

DCI

. The REFCLK/SYNC input

CLK

is shown equal to the DCI rate. The maximum frequency at which the device can be resynchronized in data rate mode can be expressed as

DATA

f2=

SYNC

for any positive integer, N.

08910-055

Page 48

AD9148 Data Sheet

Generally, for values of N equal to or greater than 3, the FIFO rate synchronization mode is chosen.

FIFO Rate Mode Synchronization

The following procedure outlines the steps required to synchronize multiple devices in FIFO rate mode. The procedure assumes that the CLK and REFCLK/SYNC signals are applied to all of the devices. Each individual device must follow the procedure.

The procedure for FIFO rate synchronization when directly sourcing the DAC sampling clock follows:

Configure for FIFO rate, periodic synchronization by writing

1. 0x80 to the sync control register (Register 0x10). Additional synchronization options are available and are described in the Additional Synchronization Features section.

Poll the sync locked bit (Bit 6, Register 0x12) to verify that

2. the device is back-end synchronized. A high level on this bit indicates that the clocks are running with a constant and known phase relative to the sync signal.

Reset the FIFO by strobing the FRAME signal high for at

3. least the time interval needed to load complete data to the four DACs. Resetting the FIFO ensures that the correct data is being read from the FIFO of each of the devices simultaneously. This completes the synchronization procedure, and at this stage, all devices should be synchronized.

To ensure that each of the DACs is updated with the correct data on the same DACCLK edge, two timing relationships must be met on each DAC. DCI (and data) must meet the setup and hold times with respect to the rising edge of CLK, and REFCLK/ SYNC must also meet the setup and hold time with respect to the rising edge of CLK. When resetting the FIFO, the FRAME signal must be held high for at least the time interval needed to load complete data to the four DACs (one DCI period for dualport mode, and two DCI periods for single-port or byte mode). When these conditions are met, the outputs of the DACs will be updated within t

SKEW

+ t

nanoseconds of each other. A

OUTDLY

timing diagram that illustrates the timing requirements of the input signals is shown in Figure 57.

SKEW

CLK(1)

CLK(2)

SU_SYNC

2× Interpolation

H_SYNC

Rev. A | Page 48 of 72

SYNC(2)

DCI(2)

FRAME(2)

Figure 57. Synchronization Signal Timing Requirements in FIFO Rate Mode,

08910-056

Figure 57 shows the synchronization signal timing with 2× interpolation, so that f

DCI

= ½ × f

. The REFCLK/SYNC input

CLK

is shown equal to the FIFO rate. The maximum frequency at which the device can be resynchronized in FIFO rate mode can be expressed as

SYNC

DATA

28×

for any positive integer, N.

ADDITIONAL SYNCHRONIZATION FEATURES

The synchronization logic incorporates additional features that provide means for querying the status of the synchronization and for improving the robustness of the synchronization. For more information on these features, see the Sync Status Bits section and the Timing Optimization section.

Sync Status Bits

When the sync locked bit (Bit 6, Register 0x12) is set, it indicates that the synchronization logic has reached alignment. This is determined when the clock generation state machine phase is constant. This takes between (11 + Averaging) × 64 and (11 + Averaging) × 128 DACCLK cycles. This bit may optionally trigger

IRQ

, as described in the section. Interrupt Request Operation

When the sync lost bit (Bit 7, Register 0x12) is set, it indicates that a previously synchronized device has lost alignment. This bit is latched and remains set until cleared by overwriting the register. This bit may optionally trigger an

Timing Optimization

The REFCLK/SYNC signal is sampled by a version of the DACCLK. If sampling errors are detected, the opposite sampling edge can be selected to improve the sampling point. The sampling edge can be selected by setting Bit 3, Register 0x10 (1 = rising and 0 = falling).

The synchronization logic resynchronizes when a phase change between the REFCLK/SYNC signal and the state of the clock generation state machine exceeds a threshold. To mitigate the effects of jitter and prevent erroneous resynchronizations, the relative phase can be averaged. The amount of averaging is set by the sync averaging bits (Bits[2:0], Register 0x10) and can be set from 1 to 128. The higher the number of averages, the more slowly the device recognizes and resynchronizes to a legitimate phase correction. Generally, the averaging should be made as large as possible while still meeting the allotted resynchronization time interval.

Additional information on synchronization can be found in the

AN-1093 Application Note, Synchronization of Multiple AD9122

TxDAC+ Converters.

Table 15. Synchronization Setup and Hold Times

Parameter Min Max Unit

−t

SKEW

−100 ps

SU_SYNC

+400 ps

H_SYNC

DACCLK

IRQ

section. Interrupt Request Operation

/2 +t

as described in the

/2 ps

DACCLK

Page 49

Data Sheet AD9148

INTERFACE TIMING

The timing diagram for the digital interface port is shown in Figure 59. The sampling point of the data bus nominally occurs 250 ps after each edge of the DCI signal and has an uncertainty of

ps when the DCI delay is set to 00b (Register 0x72[1:0]),

± 250

DACCLK/

REFCLK

as illustrated by the sampling interval. The data and FRAME signals must be valid throughout this sampling interval. The data and FRAME signals may change at any time between sampling intervals.

The setup (t

) and hold (tH) times with respect to the edges are

shown in Figure 59. The minimum setup and hold times are shown in Tab l e 1 6 .

Table 16. Data Port Setup and Hold Times

DCI Delay (Register 0x72, Bits[1:0])

Minimum Setup

(ns)

Time, t

Minimum Hold Time, tH (ns)

00 −0.02 0.52 01 −0.16 0.78 10 −0.28 1.03 11 −0.36 1.16

The data interface timing can be verified by using the SED circuitry. See the Interface Timing Validation section for details.

In data rate mode with synchronization enabled, a second timing constraint between DCI and DACCLK must be met in addition to the DCI-to-data timing shown in Tab le 17 . In data rate mode, only one FIFO slot is being used. The DCI to DACCLK timing restriction is required to prevent data being written to and read from the FIFO slot at the same time. The required timing between DCI and DACCLK is shown in Figure 58.

Figure 58. Timing Diagram for Input Data P ort (Data Rate Mode with Sync On)

Table 17. DCI to DACCLK Setup and Hold Times vs. DCI Delay Value

DCI Delay (Register 0x72,Bits[1:0])

00 −0.06 0.85 01 −0.22 1.14 10 −0.36 11 −0.45 1.59

DCI

DATA

SAMPLING

INTERVAL

SDCI

HDCI

Minimum Setup Time, t

SDCI

(ns)

Minimum Hold Time, t

HDCI

(ns)

1.43

08910-057

DAT

DCI

DATA

SAMPLING

INTERVAL

Figure 59. Timing Diagram for Input Data Ports

SAMPLING

INTERVAL

08910-058

Rev. A | Page 49 of 72

Page 50

AD9148 Data Sheet

yjy

()(

)

×+=×+

DIGITAL DATA PATH

The block diagram in Figure 60 shows the functionality of the complex digital data path. The digital processing includes a premodulation block, a programmable complex filter, three half-band interpolation filters with built-in coarse modulation, a quadrature modulator with a fine resolution NCO as well as phase, gain, and offset adjustment blocks.

PREMOD

PROG

–1

SINC FILTER

Figure 60. Block Diagram of Digital Data Path

HB2

HB1 HB3

DIGITAL

PHASE/GAIN/

OFFSET ADJ

There are two complex digital data paths that feed the four DACs. Each digital data path accepts I and Q data streams and processes them as a quadrature data stream, resulting in two quadrature data streams. All of the signal processing blocks can be used when the input data stream is represented as complex data.

The data path can be used to process an input data stream representing four independent real data streams as well; however, the functionality is somewhat restricted. The premodulation block can be used, as well as any of the nonshifted interpolation filter modes.

PREMODULATION

The half-band interpolation filters have selectable pass bands that allow the center frequencies to be moved in increments of ½ of their input data rate. The premodulation block provides a digital upconversion of the incoming waveform by ½ of the incoming data rate, f

Functionally, the premodulation

DATA.

multiplies the incoming data samples alternatively by +1 and −1. This can be used to frequency shift baseband input data to the center of the interpolation filters pass band.

PROGRAMMABLE INVERSE SINC FILTER

The AD9148 provides a programmable inverse sinc filter to compensate the DAC roll-off over frequency. Because this filter is implemented before the interpolation filter, its coefficients must be changed depending on the interpolation rate and DAC output center frequency.

08910-059

Filter Structure

The programmable inverse sinc filter is a nine-tap complex FIR filter using complex conjugate coefficients. The z-transfer function is

()

×+

xjx

×+=

HjH

×+×+×+×+

−−−−

×+×+×+×+=

zczczczcc

−−−−

zczczczc

where:

and xQ are the in-phase (real) and quadrature (imaginary)

filter input, respectively.

and yQ are the in-phase (real) and quadrature (imaginary)

filter output, respectively.

and HQ are the in-phase (real) and quadrature (imaginary)

filter coefficients, respectively.

, c1, c2, c3, and c4 are the complex filter coefficient, and

their

complex conjugate.

The filter coefficients must be calculated and programmed into the AD9148 registers to perform the operation desired.

Filter Implementation

To perform the complex filtering of the complex input, the filter is divided in four filters working in parallel, two sets of H two sets of H

(see Figure 61).

xjxHjHyjy

×+⋅

Figure 61. Complex Filter Implementation

()

QQQ

–

and

xHxHjxHxH

×+×⋅+×−×=

08910-060

The coefficients for the filter are stored in SPI Register 0x20 to Register 0x27 in twos-complement format. They have variable length, three bits to 10 bits.

Rev. A | Page 50 of 72

Page 51

Data Sheet AD9148

Table 18. Programmable Inverse Sinc Filter Coefficient Widths and Ranges

Coefficient Width Minimum Maximum

c0 in-phase (real) 3 100b 011b

−4 3 c

quadrature (imaginary) 3 0100b 011b

−4 c

in-phase (real) 4 1000b 0111b

−8 c

quadrature (imaginary) 4 1000b 0111b

−8 c

in-phase (real) 5 10000b 01111b

−16 c

quadrature (imaginary) 5 10000b 01111b

−16 c

in-phase (real) 7 1000000b 0111111b

−64 c

quadrature (imaginary) 7 1000000b 0111111b

−64 c

in-phase (real) 10 1000000000b 0111111111b

−1024 c

quadrature (imaginary) 10 1000000000b 0111111111b

−1024 1023

The real and imaginary filters are implemented using the structure described in Figure 62 and Figure 63.

INPUT

–1

0REALc1REALc2REALc3REALc4REAL

–1

5REAL

OUTPUT

Figure 62. Real Filter implementation

INPUT

–1

0IMG

1IMGc2IMGc3IMGc4IMG

–

–1

–

–1

5IMG

OUTPUT

Figure 63. Imaginary Filter implementation

The AD9148 evaluation tools provide software that allows for the processing of the filter coefficients based on the DAC sampling frequency, the amount of interpolation used (combination of HB1, HB2, and HB3), and the desired center frequency. This center frequency is limited to

[−0.4 ×

+ 0.5 × signalBW, 0.4 × f

DAC

− 0.5 × signalBW]

DAC

The bandwidth of the inverse sinc filter equals the maximum allowable signal bandwidth of the interpolation filters (0.8 × f

DATA

08910-061

08910-062

When there is no interpolation used, the real filter coefficients can be fixed at (no imaginary coefficients)

= 2 ; C8 = 2

= −4 ; C7 = −4

= 10 ; C6 = 10

= −35 ; C5 = −35

= 401

INTERPOLATION FILTERS

The transmit path contains three interpolation filters. Each of the three interpolation filters provides a 2× increase in output data rate. The filters can be cascaded to provide 2×, 4×, or 8× interpolation ratios. Each of the half-band filter stages offers a different combination of bandwidths and operating modes.

The bandwidth of the three half-band filters with respect to the data rate at the filter input is as follows:

• Bandwidth of HB1 = 0.8 × f

• Bandwidth of HB2 = 0.5 × f

• Bandwidth of HB3 = 0.4 × f

The usable bandwidth is defined as the frequency over which the filters have a pass-band ripple of less than ±0.001 dB and an image rejection of greater than +85 dB. As is discussed in the Half-Band Filter 1 (HB1) section, the image rejection usually sets the usable bandwidth of the filter, not the pass-band flatness.

The half-band filters operate in several modes, providing programmable pass-band center frequencies as well as signal modulation. The HB1 filter has four modes of operation, and the HB2 and HB3 filters each have eight modes of operation.

1023

IN1

IN2

IN3

Rev. A | Page 51 of 72

Page 52

AD9148 Data Sheet

Half-Band Filter 1 (HB1)

HB1 has four modes of operation, as shown in Figure 64. The shape of the filter response is identical in each of the four modes. The four modes are distinguished by two factors: the filter center frequency and whether the filter modulates the input signal.

MAGNITUDE (dB)

–100

–20

–40

–60

–80

01.81. 61.41.21.00.80.60.40.2

MODE 0

MODE 1 MODE 3

Figure 64. HB1 Filter Modes

MODE 2

2.0

(×

)

IN1

08910-063

As is shown in Figure 64, the center frequency in each mode is offset by ½ of the input data rate (f

) of the filter. Mode 0 and

IN1

Mode 1 do not modulate the input signal. Mode 2 and Mode 3 modulate the input signal by f

. When operating in Mode 0

IN1

and Mode 2, the I and Q paths operate independently and no mixing of the data between channels occurs. When operating in Mode 1 and Mode 3, mixing of the data between the I and Q paths occurs; therefore, the data input into the filter is assumed complex. Tab l e 19 summarizes the HB1 modes.

Table 19. 2× Interpolation Filter Modes (Register 0x1C to Register 0x1E)

Interpolation Fac tor

Filter Modes

Pre-Mod HB1 HB2 HB3

CENTER

DAC

)

2 0 0 Off Off 0 2 1 1 Off Off f 2 0 2 Off Off f 2 1 3 Off Off −f

DAC

/4 /2

0.02

–0.02

–0.04

MAGNITUDE (dB)

–0.06

–0.08

–0.10

000.360.320.280. 240.200.160.120.080.04

(×

)

IN1

.40

08910-064

Figure 65. Pass-Band Detail of HB1

Table 20. HB1 Pass-Band and Stop-Band Performance by Bandwidth

Bandwidth (% of f

IN1

)

Pass-Band Flatness (dB)

Stop-Band Rejection (dB)

80 0.001 85

80.4 0.0012 80

81.2 0.0033

82.0 0.0076

83.6 0.0271

70 60 50

85.6 0.1096 40

Half-Band Filter 2 (HB2)

HB2 has eight modes of operation, as shown in Figure 66 and Figure 67. The shape of the filter response is identical in each of the eight modes. The eight modes are distinguished by two factors, the filter center frequency and whether the input signal is modulated by the filter.

–20

–40

MODE 0

MODE 2 MODE 6

MODE 4

Figure 65 shows the pass-band filter response for HB1. In most applications, the usable bandwidth of the filter is limited by the image suppression provided by the stop-band rejection and not by the pass-band flatness. Ta b le 2 0 shows the pass-band flatness and stop-band rejection the HB1 filter supports at different bandwidths.

Rev. A | Page 52 of 72

MAGNITUDE (dB)

–60

–80

–100

021.81. 61. 41. 21.00.80.60.40.2

(× f

)

IN2

Figure 66. HB2, Even Filter Modes

08910-065

Page 53

Data Sheet AD9148

–20

–40

–60

MAGNITUDE (dB)

–80

–100

021.81. 61. 41. 21.00.80.60.40.2

MODE 1

Figure 67. HB2, Odd Filter Modes

MODE 3

(× f

)

IN2

MODE 5

MODE 7

08910-066

As shown in Figure 66 and Figure 67, the center frequency in each mode is offset by ¼ of the input data rate (f

) of the filter.

IN2

Mode 0 through Mode 3 do not modulate the input signal. Mode 4 through Mode 7 modulate the input signal by f

. When operating

IN2

in Mode 0 and Mode 4, the I and Q paths operate independently, and no mixing of the data between channels occurs. When operating in the other six modes, mixing of the data between the I and Q paths occurs; therefore, the data input to the filter is assumed complex. Ta b le 2 1 summarizes the HB2 modes.

Table 21. 4× Interpolation Filter Modes (Register 0x1C to Register 0x1E)

Interpolation Fac tor

Filter Modes

Pre-Mod HB1 HB2 HB3

CENTER

DAC

)

4 0 0 0 Off 0 4 1 1 1 Off f 4 0 2 2 Off f 4 1 3 3 Off 3f 4 0 0 4 Off f 4 1 1 5 Off −3f 4 0 2 6 Off −f 4 1 3 7 Off −f

DAC

/8 /4

DAC

/8 /4 /8

Figure 68 shows the pass-band filter response for HB2. In most applications, the usable bandwidth of the filter is limited by the image suppression provided by the stop-band rejection and not by the pass-band flatness. Tab l e 2 2 shows the pass-band flatness and stop-band rejection the HB2 filter supports at different bandwidths.

0.02

–0.02

–0.04

MAGNITUDE (dB)

–0.06

–0.08

–0.10

000.280. 240.200.160.120.080. 04

Figure 68. Pass-Band Detail of HB2

(×

)

IN2

.32

08910-067

Table 22. HB2 Pass-Band and Stop-Band Performance by Bandwidth

Bandwidth (% of f

IN2

)

Pass-Band Flatness (dB)

Stop-Band Rejection (dB)

50 0.001 85

50.8 0.0012 80

52.8 0.0028

56.0 0.0089 60 0.0287

70 60 50

64.8 0.1877 40

Half-Band Filter 3 (HB3)

HB3 has eight modes of operation that function the same as HB2. The primary difference between HB2 and HB3 are the filter bandwidths. Table 2 3 summarizes the filter modes for HB3.

Table 23. 8× Interpolation Filter Modes (Register 0x1C to Register 0x1E)

Interpolation Fac tor

Filter Modes

Pre-Mod HB1 HB2 HB3

CENTER

DAC

)

8 0 0 0 0 0 8 0 2 2 1 f 8 0 0 4 2 8 0 2 6 3 8 0 0 0 4 8 0 2 2 5 8 0 0 4 6 8 0 2 6 7 −f

DAC

3f f

DAC

−3f

−f

DAC

/8 /4

DAC

/8 /4 /8

Rev. A | Page 53 of 72

Page 54

AD9148 Data Sheet

Figure 69 shows the pass-band filter response for HB3. In most applications, the usable bandwidth of the filter is limited by the image suppression provided by the stop-band rejection and not by the pass-band flatness. Tab l e 2 4 shows the pass-band flatness and stop-band rejection the HB3 filter supports at different bandwidths.

0.02

–0.02

–0.04

MAGNITUDE (dB)

–0.06

–0.08

–0.10

00.240.200.160.120.080.04

(×

)

IN3

0.28

08910-068

Figure 69. Pass-Band Detail of HB3

Table 24. HB3 Pass-Band and Stop-Band Performance by Bandwidth

Bandwidth (% of f

IN3

)

Pass-Band Flatness (dB)

Stop-Band Rejection (dB)

40 0.001 85

40.8 0.0014 80

42.4 0.002 70

45.6 0.0093 60

49.8 0.03 50

55.6 0.1 40

The maximum bandwidth can be achieved if the signal carrier frequency is placed directly at the center of one of the filter pass bands. In this case, the entire quadrature bandwidth of the interpolation filter (0.8 × f

) is available. The available signal

DATA

bandwidth decreases as the carrier frequency of the signal moves away from the center frequency of the filter. The worst-case carrier frequency is one that falls directly between the center frequency of two adjacent filters. Figure 70 shows how the signal bandwidth changes as a function of placement in the spectrum and interpolation rate.

×2 MODE ×4 MODE

0.4

)

0.3

DAC

0.2

0.1

COMPLEX BW (×

DC–1/2 1/2–3/8 3/8–1/4 1/4–1/8 1/8

CARRIER FREQUENCY

×8 MODE

0.15

0.075

0.0375

(×

DAC

)

Figure 70. Complex Signal Bandwidth as a Function of Output Frequency

FINE MODULATION

The fine modulation makes use of a numerically controlled oscillator, a phase shifter, and a complex modulator to provide a means for modulating the signal by a programmable carrier signal. A block diagram of the fine modulator is shown in Figure 71. The fine modulator allows the signal to be placed anywhere in the output spectrum with very fine frequency resolution.

I DATA

Q DATA

The quadrature modulator is used to mix the carrier signal generated by the NCO with the I and Q signal. The NCO produces a quadrature carrier signal to translate the input signal to a new center frequency. A complex carrier signal is a pair of sinusoidal waveforms of the same frequency, offset 90° from each other. The frequency of the complex carrier signal is set via the FTW[31:0] value in Register 0x54 through Register 0x57.

The NCO operating frequency, f frequency of the complex carrier signal can be set from dc up to f

The generated quadrature carrier signal is mixed with the I and Q data. The quadrature products are then summed into the I and Q data paths, as shown in Figure 71.

INTERPOLATIO N

FTW[31:0]

NCO PHASE OFFSET

WORD [15:0]

SPECTRAL INVERSION

INTERPO LATIO N

COSINE

NCO

SINE

–

–1

Figure 71. Fine Modulator Block Diagram

, is at the DAC rate. The

NCO

/2. The frequency tuning word (FTW) is calculated as

DAC

FTW

CENTER

DAC

2×=

OUT_I

OUT_Q

08910-069

08910-070

Rev. A | Page 54 of 72

Page 55

Data Sheet AD9148

When using the fine modulator, the maximum signal bandwidth of

0.8 × f

is always achieved.

DATA

Updating the Frequency Tuning Word

The frequency tuning word registers are not updated immediately upon writing as the other configuration registers do. After loading the FTW registers with the desired values, Bit 2 of Register 0x5A must transition from 0 to 1 for the new FTW to take effect.

Phase Offset Adjustment

A 16-bit phase offset may be added to the output of the phase accumulator via the serial port. This static phase adjustment results in an output signal that is offset by a constant angle relative to the nominal signal. This allows the user to phase align the NCO output with some external signal, if necessary. This can be especially useful when NCOs of multiple AD9148s are programmed for synchronization. The phase offset allows for the adjustment of the output timing between the devices. The static phase adjustment is sourced from the NCO Phase Offset[15:0] value located in Register 0x58 and Register 0x59.

QUADRATURE PHASE CORRECTION

The purpose of the quadrature phase correction block is to enable compensation of the phase imbalance of the analog quadrature modulator following the DAC. If the quadrature modulator has a phase imbalance, the unwanted sideband appears with significant energy. Tuning the quadrature phase adjust value can optimize image rejection in single sideband radios.

Ordinarily, the I and Q channels have an angle of precisely 90° between them. The quadrature phase adjustment is used to change the angle between the I and Q channels. When I Phase Adj, Bits[9:0] (Register 0x28 and Register 0x29), are set to 1000000000b, the I DAC output moves approximately 1.75° away from the Q DAC output, creating an angle of 91.75° between the channels. When I Phase Adj, Bits[9:0] (Register 0x28 and Register 0x29), are set to 0111111111b, the I DAC output moves approximately 1.75° toward the Q DAC output, creating an angle of 88.25° between the channels.

Q Phase Adj, Bits[9:0] (Register 2A and Register 2B), work in a similar fashion. When Q Phase Adj, Bits[9:0] (Register 2A and Register 2B), are set to 1000000000b, the Q DAC output moves approximately 1.75° away from the I DAC output, creating an angle of 91.75° between the channels. When Q Phase Adj[9:0] is set to 0111111111b, the Q DAC output moves approximately

1.75° toward the I DAC output, creating an angle of 88.25° between the channels.

Based on these two endpoints, the combined resolution of the phase compensation register is approximately 3.5°/1024 or

0.00342° per code. When both I Phase Adj, Bits[9:0] (Register 0x28 and Register 0x29), and Q Phase Adj, Bits[9:0] (Register 2A and Register 2B), are used, the full phase adjustment range is ±3.5°.

DC OFFSET CORRECTION

The dc value of the I data path and the Q data path can be independently controlled by adjusting I DC Offset, Bits[15:0], and Q DC Offset, Bits[15:0], values in Register 0x2C through Register 0x2F. These values are added directly to the data path values. Care should be taken not to overrange the transmitted values.

Figure 72 shows how the DAC offset current varies as a function of I DC Offset, Bits[15:0], and Q DC Offset, Bits[15:0], values. With the digital inputs fixed at midscale (0x000, twos complement data format), Figure 72 shows the nominal I

OUTxP

and I

OUTxN

currents as the DC offset value is swept from 0 to 65,535. Because I the sum of I

(mA)

OUTxP

0x0000 0x4000 0x8000 0xC000 0xFFFF

and I

OUTxP

Figure 72. DAC Output Currents vs. DC Offset Value

are complementary current outputs,

OUTxN

and I

is always 20 mA.

OUTxN

DAC OFFSET VALUE

(mA)

OUTxN

DIGITAL GAIN CONTROL

The last block in each datapath is an 8-bit scalar (Register 0x50 and Register 0x51) that can be used for digital gain control. The IGain Control, Bits[7:0] (Register 0x50), and QGain control, Bits[7:0] (Register 0x51), values directly scale the samples written to the IDAC and QDAC, respectively. The bit weighting is MSB = 2

-6

and LSB = 2

, which yields a multipler range of 0 to 3.984375.

The scale factor for each data path is calculated as

rScaleFacto =

IGain

]0:7[ QGain

]0:7[

Take care not to overrange the DAC when using a scale factor greater than 1.

08910-131

Rev. A | Page 55 of 72

Page 56

AD9148 Data Sheet

CLOCK GENERATION

DAC INPUT CLOCK CONFIGURATIONS

The AD9148 DAC sample clock (DACCLK) can be sourced directly or by clock multiplying. Clock multiplying employs the on-chip, phased-locked loop (PLL) that accepts a reference clock (REFCLK_x) operating at a submultiple of the desired DACCLK rate, most commonly the data input frequency. The PLL then multiplies the reference clock up to the desired DACCLK frequency, which can then be used to generate all the internal clocks required by the DAC. The clock multiplier provides a high quality clock that meets the performance requirements of most applications. Using the on-chip clock multiplier removes the burden of generating and distributing the high speed DACCLK.

The second mode bypasses the clock multiplier circuitry and allows DACCLK to be sourced directly through the CLK_x pins. This mode enables the user to source a very high quality clock directly to the DAC core. Sourcing the DACCLK directly through the CLK_x pins may be necessary in demanding applications that require the lowest possible DAC output noise, particularly at higher output frequencies.

DAC DAC

DRIVING THE CLK_x AND REFCLK_x INPUTS

The REFCLK_x and CLK_x differential inputs share similar clock receiver input circuitry. Figure 1 shows a simplified circuit diagram of the input, along with a recommended drive circuit. The on-chip clock receiver has a differential input impedance of about 10 kΩ. It is self-biased to a common-mode voltage of about

1.25 V. The recommended circuit for driving the input is a pair of ac coupling capacitors and a differential 100 Ω termination.

The minimum input drive level to either of the clock inputs is 100 mV ppd. The optimal performance is achieved when the clock input signal is between 500 mV ppd and 1.6 V ppd. Whether using the on-chip clock multiplier or sourcing the DACCLK directly, it is necessary that the input clock signal to the device have low jitter and fast edge rates to optimize the DAC noise performance.

DIRECT CLOCKING

When a high quality, sample rate clock is connected to the AD9148, it provides the lowest noise spectral density at the DAC outputs. To select the differential CLK inputs as the source for the DAC sampling clock, set the PLL enable bit to 0 (Register 0x0A, Bit 7). Setting this bit to 0 powers down the internal PLL clock multiplier and selects the input from the CLK_x pins as the source for the internal DACCLK.

The device also has duty-cycle correction circuitry and differential input level correction circuitry. Enabling these circuits may provide improved performance in some cases. The control bits for these functions can be found in Register 0x08.

LVPECL

DRIVER

100Ω

CLK_P/ REFCLK_P

5kΩ

CLK_N/ REFCLK_N

1.25V

LVDS

DRIVER

1000pF

200Ω

1000pF

Figure 73. Clock Receiver Circuitry and Recommended Drive Circuitry using LVPECL (Left) and LVDS (Right)

1000pF

100Ω

CLK_P/ REFCLK_P

5kΩ

CLK_N/ REFCLK_N

1.25V

08910-071

Rev. A | Page 56 of 72

Page 57

Data Sheet AD9148

REFCLK_P/REFCLK_N

(PIN B9 AND PIN A9)

0x0D[7:6]

÷N2

PC_CLK

CLK_P/CLK_N

(PIN B6 AND PIN A6)

0x06[7:6]

PLL LOCK

PLL LOCK LOST

PHASE

DETECTION

PLL ENABLE

Figure 74. PLL Clock Multiplication Circuit

LOOP

FILTER

÷N1

0x0D[1:0]N10x0D[3:2]

0x0A[7]

÷N0

ADC

VCO

0x0E[3:0] PLL CONTROL VOLTAGE

DACCLK

08910-072

Table 25. PLL Settings

Address PLL SPI Control Register Bit Optimal Setting

PLL Loop Bandwidth 0x0C [7:5] 110 PLL Control 1 Register 0x0C [4:0] 01001 PLL Cross Control Enable 0x0D [4] 1

CLOCK MULTIPLICATION

The on-chip PLL clock multiplier circuit can be used to generate the DAC sample rate clock from a lower frequency reference clock. When the PLL clock multiplier is enabled (Register 0x0A[7] = 1), the clock multiplication circuit generates the DAC sample clock from the lower rate REFCLK input. The functional diagram of the clock multiplier is shown in Figure 74.

The clock multiplication circuit operates such that the VCO outputs a frequency, f frequency multiplied by N0 × N1.

= f

VCO

REFCLK

The DAC sample clock frequency, f

= f

DACCLK

REFCLK

The output frequency of the VCO must be chosen to keep f the optimal operating range of 1.0 GHz to 2.1 GHz. The frequency

, equal to the REFCLK input signal

VCO

× (N0 × N1)

, is equal to

DACCL K

× N1

VCO

Therefore, it is required that the optimal PLL band select value be determined for each individual device.

PLL BAND

1000 220020001800160014001200

Figure 75. PLL Lock Range Overtemperature for a Typical Device

VCO FREQUENCY (MHz)

08910-073

of the reference clock and the values of N1 and N0 must be chosen so that the desired DACCLK frequency can be synthesized and the VCO output frequency is in the correct range.

PLL Bias Settings

There are four bias settings for the PLL circuitry that should be programmed to their nominal values. The PLL values shown in Tabl e 25 are the recommended settings for these parameters.

Configuring the VCO Tuning Band

The PLL VCO has a valid operating range from approximately

1.0 GHz to 2.1 GHz covered in 63 overlapping frequency bands. For any desired VCO output frequency, there may be several valid PLL band select values. The frequency bands of a typical device are shown in Figure 75. Device-to-device variations and

Automatic VCO Band Select

The device has an automatic VCO band select feature on chip; using this feature is a simple and reliable method for configuring the VCO frequency band. To use the automatic VCO band select feature, enable the PLL by writing 0xC0 to Register 0x0A and enable the auto band select mode by writing 0x80 to Register 0x0A. When this value is written, the device executes an automated routine that determines the optimal VCO band setting for the device. The setting selected by the device ensures that the PLL remains locked over the full −40°C to +85°C operating temperature range of the device without further adjustment. (The PLL remains locked over the full temperature range even if the temperature during initialization is at one of the temperature extremes.)

operating temperature affect the actual band frequency range.

Rev. A | Page 57 of 72

Page 58

AD9148 Data Sheet

Manual VCO Band Select

The device also has a manual band select mode that allows the user to select the VCO tuning band. When in manual mode (enabled by setting Bit 6, Register 0x0A to 1), the VCO band is set directly with the value written to the manual VCO band bit enabled (Bits[5:0], Register 0x0A). To properly select the VCO band, complete the following sequence:

Put the device in manual band select mode.

Sweep the VCO band over a range of bands that results in

2. the PLL being locked.

Verify that the PLL is locked and read the VCO control

3. voltage for each band.

Select the band that results in the control voltage being

4. closest to the center of the range (that is, 1000). See Tab le 2 6 for more details.

The resulting VCO band should be the optimal setting for the device. This band should be written to the manual VCO band register value.

If desired, an indication of where the VCO is within the operating frequency band can be determined by querying the VCO control voltage. Tabl e 26 shows how to interpret the VCO control voltage value.

Table 26. VCO Control Voltage Range Indications

VCO Control Voltage Indication

1111 Move to a higher VCO band. 1110 1101

1100 1011 1010 1001

1000 0111 0110 0101

0100 0011 0010 0001 Move to a lower VCO band. 0000

VCO is operating in the higher end of frequency band.

VCO is operating with an optimal region of the frequency band.

VCO is operating in the lower end of frequency band.

Rev. A | Page 58 of 72

Page 59

Data Sheet AD9148

ANALOG OUTPUTS

TRANSMIT DAC OPERATION

Figure 77 shows a simplified block diagram of one pair of the transmit path DACs. The DAC core consists of a current source array, switch core, digital control logic, and full-scale output current

⎞ ⎟ ⎠

OUTFS

⎞ ⎟

⎠

) is nominally

control. The DAC full-scale output current (I 20 mA. The output currents from the IOUTx_P and IOUTx_N pins are complementary, meaning that the sum of the two currents always equals the full-scale current of the DAC. The digital input code to the DAC determines the effective differential current delivered to the load.

The DAC has a 1.2 V band gap reference with an output impedance of 5 k. The reference output voltage appears on the VREF pin. When using the internal reference, the VREF pin should be decoupled to AVSS with a 0.1 µF capacitor. The internal reference should only be used for external circuits that draw dc currents of 2 µA or less. For dynamic loads or static loads greater than 2 µA, the VREF pin should be buffered. If desired, an external reference (between 1.10 V to 1.30 V) can be applied to the VREF pin.

A 10 k external resistor, R RESET

pin to AVSS. This resistor, along with the reference

, must be connected from the

SET

control amplifier, sets up the correct internal bias currents for the DAC. Because the full-scale current is inversely proportional to this resistor, the tolerance of R

is reflected in the full-scale

SET

output amplitude.

The full-scale current can be calculated by

OUTFS

REF

SET

⎛

⎜ ⎝

×+×= DAC gain

⎜

⎝

where DAC gain is set individually for the I and Q DACs in Register 0x30, Register 0x31, Register 0x34, and Register 0x35, respectively.

For nominal values of VREF (1.2 V), R gain (512), the full-scale current of the DAC is typically 20.16 mA. The DAC full-scale current can be adjusted from 8.66 mA to

31.66 mA by setting the DAC gain parameter setting as shown in Figure 76.

(mA)

OUTFS

200 400 600 800

DAC GAIN CODE

Figure 76. DAC Full-Scale Current vs. DAC Gain Code

Transmit DAC Transfer Function

The output currents from the IOUTx_P and IOUTx_N pins are complementary, meaning that the sum of the two currents always equals the full-scale current of the DAC. The digital input code to the DAC determines the effective differential current delivered to the load. IOUTx_P provides the maximum output current when all bits are high. The output currents vs. DACCODE for the DAC outputs are expressed as

DACCODE

= I

⎡

POUT

⎢

⎣

– I

OUTFS

I ×

OUT_N

where DACCODE = 0 to 2

⎤

OUT_P

OUTFS

⎥

⎦

(2)

− 1.

(10 k), and DAC

SET

1000

08910-074

(1)

0.1µ

RSET

1.2V

5kΩ

VREF

I120

10kΩ

Figure 77. Simplified Block Diagram of the DAC Core

I DAC GAIN

CURRENT

SCALING

Q DAC GAIN

I DAC

Q DAC

IOUT1_P/ IOUT3_P

IOUT1_N/IOUT3_N

IOUT2_N/IOUT4_N

IOUT2_P/ IOUT4_P

08910-075

Rev. A | Page 59 of 72

Page 60

AD9148 Data Sheet

Transmit DAC Output Configurations

The optimum noise and distortion performance of the AD9148 is realized when it is configured for differential operation. The common-mode error sources of the DAC outputs are reduced significantly by the common-mode rejection of a transformer or differential amplifier. These common-mode error sources include even-order distortion products and noise. The enhancement in distortion performance becomes more significant as the frequency content of the reconstructed waveform increases and/or its amplitude increases. This is due to the first-order cancellation of various dynamic common-mode distortion mechanisms, digital feedthrough, and noise.

IOUT1_P/ IOUT3_P

IOUT1_N/IOUT3_N

IOUT2_N/IOUT4_N

IOUT2_P/ IOUT4_P

Figure 78. Basic Transmit DAC Output Circuit

OUTI

–

OUTQ

–

8910-076

Figure 78 shows the most basic DAC output circuitry. A pair of resistors, R output currents to a differential voltage output, V

, are used to convert each of the complementary

. Because

OUT

the current outputs of the DAC are high impedance, the differential driving point impedance of the DAC outputs, R 2 × R

. Figure 79 illustrates the output voltage waveforms.

PEAK

, is equal to

OUT

Transmit DAC Linear Output Signal Swing

The DAC outputs have a linear output compliance voltage range of ±1 V that must be adhered to achieve optimum performance. The linear output signal swing is dependent on the full-scale output current, I

, and the common-mode level of the output.

OUTFS

AUXILIARY DAC OPERATION

The AD9148 has four 10-bit auxiliary DACs (AUX1, AUX2, AUX3, and AUX4). The full-scale output current on these DACs is derived from the 1.2 V band gap reference and external resistor. The gain scale from the reference amplifier current, I DAC reference current is 16.67 with the auxiliary DAC gain set to full-scale. This gives a full-scale current of approximately 2 mA for each auxiliary DAC.

The magnitude of the AUX1 DAC current is controlled via Bits[1:0], Register 0x33 (MSBs) and Bits[7:0], Register 0x32 (LSBs) when DAC SPI select = 0 (Bit 4, Register 0x00). The magnitude of the AUX2 DAC current is controlled via Bits[1:0], Register 0x37 (MSBs) and Bits[7:0], Register 0x36 (LSBs) when DAC SPI select = 0 (Bit 4, Register 0x00). Likewise, the magnitudes of AUX3 DAC current and AUX4 DAC current are controlled via Register 0x33 to Register 0x32 and Register 0x37 to Register 0x36, respectively when DAC SPI Select = 1 (Register 0x00, Bit[4]).

The auxiliary DAC structure is shown in Figure 80. There are two output signals on each auxiliary DAC. One signal is P, and the other is N. The auxiliary DAC outputs are not differential. Only one side of the auxiliary DAC (P or N) is active at one time. The inactive side goes into a high impedance state (100 k). Control of the P side and N side for the auxiliary DACs is via Bit 7, Register 0x33 and Bit 7, Register 0x37 (DAC SPI select is 0 to control AUX1 and AUX2, and DAC SPI select is 1 to control AUX3 and AUX4).

0mA TO 2mA

(SOURCE)

AUXDAC[9:0]

0mA TO 2mA (SINK)

, to the auxiliary

REF

TIME

Figure 79. Voltage Output Waveforms

The common-mode signal voltage, VCM, is calculated by

08910-077

AUXDAC

DIRECTION

(SOURCE/SINK)

AUXDAC

SIGN

(P/N)

AUX_P

AUX_N

08910-078

Figure 80. Auxiliary DAC Structure

V ×=

The peak output voltage, V

PEAK

= IFS × RO

, is calculated by

PEAK

With this circuit configuration, the single-ended peak voltage is the same as the peak differential output voltage.

Rev. A | Page 60 of 72

Page 61

Data Sheet AD9148

[

]

In addition, the P or N output can act as a current source or a current sink. When sourcing current, the output compliance voltage is 0 V to 1.6 V. When sinking current, the output compliance voltage is 0.8 V to 1.6 V. The auxiliary DAC current direction is programmable via Bit 6, Register 0x33 and Bit 6, Register 0x37 (DAC SPI select is 0 to control AUX1 and AUX2, and DAC SPI select is 1 to control AUX3 and AUX4). The choice of sinking or sourcing should be made at circuit design time. There is no advantage to switching between sourcing and sinking current after the circuit is in place.

These auxiliary DACs can be used for local oscillator (LO) cancellation when the DAC output is followed by a quadrature modulator. More information and example application circuits are given in the Interfacing to Modulators section.

INTERFACING TO MODULATORS

The AD9148 interfaces to the ADL537x family with a minimal number of components. An example of the recommended interface circuitry is shown in Figure 81.

AD9148

IOUT1_P

IOUT1_N

IOUT2_N

IOUT2_P

RBIP

50Ω

RBIN

50Ω

RBQN

50Ω

RBQP

50Ω

RLI

100Ω

RLQ

100Ω

Figure 81. Typical Interface Circuitry Between the AD9148 and ADL537x

Family of Modulators

The baseband inputs of the ADL537x family require a dc bias of 500 mV. The nominal midscale output current on each output of the DAC is 10 mA (1/2 the full-scale current). Therefore, a single 50 Ω resistor to ground from each of the DAC outputs results in the desired 500 mV dc common-mode bias for the inputs to the ADL537x. The signal level can be reduced by the addition of the load resistor in parallel with the modulator inputs (RLI, RLQ). The peak-to-peak voltage swing of the transmitted signal is

××

×=

FSSIGNAL

[]

+×

ADL537x

IBBP

IBBN

QBBN

QBBP

08910-079

Baseband Filter Implementation

Most applications require a baseband anti-imaging filter between the DAC and modulator to filter out Nyquist images and broadband DAC noise. The filter can be inserted between the I-to-V resistors at the DAC output and the signal level setting resistor across the modulator input. Doing this establishes the input and output impedances for the filter.

Figure 83 shows a fifth-order low-pass filter. A common-mode choke is used between the I-to-V resistors and the remainder of the filter. This removes the common-mode signal produced by the DAC and prevents the common-mode signal from being converted to a differential signal, which would appear as unwanted spurious signals in the output spectrum. The common-mode choke or balun may not be needed if the layout between the DAC and IQ modulator is optimized and balanced. Splitting the second filter capacitor into two and grounding the center point creates a common-mode low-pass filter, providing additional common-mode rejection of high frequency signals. A purely differential filter passes common-mode signals.

Driving the ADL5375-15 with the AD9148

The ADL5375-15 requires a 1500 mV dc bias and therefore requires a slightly more complex interface than most other Analog Devices, Inc., modulators. It is necessary to level shift the DAC output from a 500 mV dc bias to the 1500 mV dc bias that the ADL5375-15 requires. Level shifting can be achieved with a purely passive network, as shown in Figure 82. In this network, the dc bias of the DAC remains at 500 mV, while the input to the ADL5375-15 is 1500 mV. Note that this passive level shifting network introduces approximately 2 dB of loss in the ac signal.

AD9148

IOUT1_P

IOUT1_N

IOUT2_N

IOUT2_P

RBIP

45.3Ω

RBIN

45.3Ω

RBQN

45.3Ω

RBQP

45.3Ω

RSIN

1kΩ

RSIP

1kΩ

RSQN

1kΩ

RSQP

1kΩ

RLIP

3480Ω

RLIN

3480Ω

RLQN

3480Ω

RLQP

3480Ω

Figure 82. Passive Level Shifting Network for Biasing

the ADL5375-15 from the AD9148

ADL5375-15

IBBP

IBBN

QBBN

QBBP

08910-081

IDAC

QDAC

50Ω

MABACT0043

50Ω

(OPTIONAL)

33nH

2pF

33nH

22pF

56nH

3pF

100Ω6p F

ADL537x

08910-080

Figure 83. DAC Modulator Interface with Fifth-Order, Low Pass Filter

Rev. A | Page 61 of 72

Page 62

AD9148 Data Sheet

Reducing LO Leakage and Unwanted Sidebands

Analog Devices modulators can introduce unwanted signals at the LO frequency due to dc offset voltages in the I and Q baseband inputs as well as feedthrough paths from the LO input to the output. The LO feedthrough can be nulled by applying the correct dc offset voltages at the DAC output. This can be done either by using the auxiliary DACs (Register 0x32, Register 0x33, Register 0x36, and Register 0x37) or by using the digital dc offset adjustments (Register 0x2C to Register 0x2F). Using the auxiliary DACs has the advantage that none of the main DAC dynamic range is used for performing the dc offset adjustment. The disadvantage is that the common-mode level of the output signal changes as a function of the auxiliary DAC current. The opposite is true when the digital offset adjustment is used.

Good sideband suppression requires both gain and phase matching of the I and Q signals. The phase adjust (Register 0x28 to Register 0x2B) and gain control (Register 0x50 and Register 0x51) registers can be used to calibrate I and Q transmit paths to optimize the sideband suppression. As an alternative to the digital gain scaling, the DAC full-scale output current (Register 0x30, Register 0x31, Register 0x34, and Register 0x35) can also be adjusted to calibrate the I and Q transmit paths; however, changing the DAC full-scale output current affects the common-mode voltage level.

For more information on correcting imperfections in IQ modulators to improve RF signal fidelity, refer to the AN-1039

Application Note.

Rev. A | Page 62 of 72

Page 63

Data Sheet AD9148

DEVICE POWER DISSIPATION

The AD9148 has four supply rails: AVDD33, IOVDD, DVDD18, and CVDD18.

The AVDD33 supply powers the DAC core circuitry. The power dissipation of the AVDD33 supply rail is independent of the digital operating mode and sample rate. The current drawn from the AVDD33 supply rail is typically 98 mA (320 mW) when the full-scale current of the four main DACs (DAC 1, DAC 2, DAC 3, and DAC 4) is set to the nominal value of 20 mA. Changing the full-scale current directly impacts the supply current drawn from the AVDD33 rail. For example, if the full-scale current of the four main DACs is changed to 10 mA, the AVDD33 supply current drops by 40 mA to 58 mA.

The IOVDD voltage supplies the serial port I/O pins (SCLK,

RESET

240

pin, and

260

280

300

8910-082

SDIO, SDO, CSB, TCK, TDI, TDO, TMS), the

IRQ

the

pin. The voltage applied to the IOVDD pin can range from 1.8 V to 3.3 V. The current drawn by the IOVDD supply pin is typically 1 mA.

The DVDD18 supply powers all of the digital signal processing blocks of the device. The power consumption from this supply is a function of which digital blocks are enabled and the frequency at which the device is operating.

The CVDD18 supply powers the clock receiver and clock distribution circuitry. The power consumption from this supply varies directly with the operating frequency of the device. CVDD18 also powers the PLL. The power dissipation of the PLL is typically 80 mW.

Figure 84 to Figure 89 detail the power dissipation of the AD9148 under a variety of operating conditions. All of the graphs are taken with data being supplied to all four DACs. The power consumption of the device does not vary significantly with changes in the coarse modulation mode selected or analog output frequency. Graphs of the total power dissipation are shown along with the power dissipation of the DVDD18 and CVDD18 supplies.

2.75

2.50

2.25

2.00

1.75

TION (W)

1.50

1.25

1.00

0.75

POWE R DISSIP

0.50

0.25

Figure 84. Total Power Dissipation vs. f

1× 2× 4× 8×

204060

100

Inverse Sinc Filter Disabled

120

140

160

180

200

(MSPS)

DATA

with Coarse Modulation, PLL, and

DATA

220

3.25

3.00

2.75

2.50

2.25

2.00

TION (W)

1.75

1.50

1.25

1.00

POWER DISSIP

0.75

0.50

0.25

1× 2× 4× 8×

204060

100

120

140

DATA

Figure 85. Total Power Dissipation vs. f

Inverse Sinc Filter Disabled

2.00

1× 2× 4× 8×

204060

100

120

140

DATA

1.75

1.50

1.25

TION (W)

1.00

0.75

POWER DISSIP

0.50

0.25

Figure 86. DVDD18 Power Dissipation vs. f

and Inverse Sinc Filter Disabled

2.50

1× 2× 4× 8×

204060

100

120

140

DATA

2.25

2.00

1.75

1.50

TION (W)

1.25

1.00

0.75

POWER DISSIP

0.50

0.25

Figure 87. DVDD18 Power Dissipation vs. f

and Inverse Sinc Filter Disabled

160

180

200

220

240

260

240

260

280

(MSPS)

with Fine Modulation, PLL, and

DATA

160

180

200

(MSPS)

DATA

160

(MSPS)

DATA

220

with Coarse Modulation, PLL,

180

200

220

with Fine Modulation, PLL,

300

8910-083

8910-084

8910-085

Rev. A | Page 63 of 72

Page 64

AD9148 Data Sheet

0.35

0.30

0.25

0.20

0.15

POWER (W)

0.10

0.05

0 100 200 300 400 500 600 700 800 900 1000

Figure 88. CVDD18 Power Dissipation vs. f

DAC

(MSPS)

, PLL Disabled

DAC

08910-086

0.25

0.23

0.20

0.18

0.15

0.13

0.10

POWER (W)

0.08

0.05

0.03

204060

100

120

Figure 89. DVDD18 Power Dissipation vs. f

DATA

140

(MSPS)

160

180

200

220

240

Due to Inverse Sinc Filter

DATA

260

280

300

08910-087

Rev. A | Page 64 of 72

Page 65

Data Sheet AD9148

TEMPERATURE SENSOR

The AD9148 has a diode-based temperature sensor for measuring the temperature of the die. The temperature reading is accessed by Register 0x5E and Register 0x5F. The temperature of the die can be calculated as

where T

DieTemp

DIE

is the die temperature in degrees Celsius. The

DIE

130

)700,13]0:15[( −

temperature accuracy is ±5°C typical over the +85°C to

−35°C range. A typical plot of the AD9148 die temperature vs. die temperature code readback is shown in Figure 90.

27,500

25,000

22,500

20,000

17,500

15,000

DIE CODE READBACK

12,500

MEASURED DIE TEMPERATURE CALCULATED DIE TE MPERATURE +5°C –5°C

Estimates of the ambient temperature can be made if the power dissipation of the device is known. For example, if the device power dissipation is 800 mW and the measured die temperature is 50°C, then the ambient temperature can be calculated as

= T

– PD × TJA = 50 – 0.8 × 18 = 35.6°C

DIE

where:

is the ambient temperature in degrees Celsius.

is the thermal resistance from junction to ambient of the

AD9148 as shown in Tabl e 7.

To use the temperature sensor, it must be enabled by setting Bit 0, Register 0x5C to 0. Before the temperature sensor data can be read back, it must be latched by toggling Bit 1, Register 0x5C from 0 to 1. In addition, to get accurate readings, the die temperature control register (Register 0x5D) should be set to 0x02.

10,000

7,500

–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90 100

TEMPERATURE (°C)

Figure 90. Die Temperature vs. Die Temperature Code Readback

08910-189

Rev. A | Page 65 of 72

Page 66

AD9148 Data Sheet

INTERRUPT REQUEST OPERATION

The AD9148 provides an interrupt request output signal (Pin H4, IRQ

) that can be used to notify an external host processor of significant device events. Upon assertion of the interrupt, the device should be queried to determine the precise event that

IRQ

occurred. The Pull the

IRQ

pin is an open-drain, active low output.

pin high external to the device. This pin may be tied to the interrupt pins of other devices with open-drain outputs to wired-OR these pins together.

Ten different event flags provide visibility into the device. These 10 flags are located in the two event flag registers (Register 0x06 and Register 0x07). The behavior of each of the event flags is independently selected in the interrupt enable registers (Register 0x04 and Register 0x05). When the flag interrupt enable is active, the event flag latches and triggers an external interrupt. When the flag interrupt is disabled, the event flag

IRQ

simply monitors the source signal, and the external remains inactive.

Figure 91 shows the event flag signals propagate to the

IRQ

-related circuitry. shows how the

Figure 91

IRQ

output. The interupt_enable signal represents one bit from the interrupt enable register. The event_flag signal represents one bit from the event flag register. The event_flag_source signal represents one of the device signals that can be monitored such as the PLL_locked signal from the PLL phase detector or the FIFO Warning 1 signal from the FIFO controller.

When an interrupt enable bit is set high, the corresponding event flag bit reflects a positively tripped (that is, latched on the rising edge of the event_flag_source version of the event_flag_source signal.

IRQ

This signal also asserts the external

. When an interrupt enable bit is set low, the event flag bit reflects the current status of the event_flag_source signal, and the event flag has no effect on the

IRQ

external

The latched version of an event flag (the interupt_source signal) can be cleared in two ways. The recommended way is by writing 1 to the corresponding event flag bit. A hardware or software reset also clears the interupt_source.

INTERRUPT SERVICE ROUTINE

Interrupt request management starts by selecting the set of event flags that require host intervention or monitoring. Those events that require host action should be enabled so that the host is notified when they occur. For events requiring host intervention,

IRQ

upon interrupt request:

Read the status of the event flag bits that are being

•

Set the interupt enable bit low so that the unlatched

•

Perform any actions that may be required to quiet the

•

Read the event flag to verify that the actions taken have

•

Clear the interrupt by writing 1 to the event flag bit.

•

Set the interrupt enable bits of the events to be monitored.

Noted that some of the event_flag_source signals are latched signals. These are cleared by writing to the corresponding event flag bit. Details of each of the event flags can be found in Table 12 .

activation, run the following routine to clear an

monitored.

event_flag_source can be monitored directly.

event_source_flag. In many cases, no specific actions may be required.

quieted the event_flag_source.

INTERRUPT_ENABL E

EVENT_FL AG_SOURCE

RITE_1_TO _EVENT_FLAG

DEVICE_RESET

INTERRUPT SOURCE

OTHER

INTERRUPT

SOURCES

Figure 91. Simplified Schematic of

IRQ

Circuitry

EVENT_FLAG

IRQ

08910-088

Rev. A | Page 66 of 72

Page 67

Data Sheet AD9148

INTERFACE TIMING VALIDATION

The AD9148 provides on-chip sample error detection (SED) circuitry that simplifies verification of the input data interface. The SED compares the input data samples captured at the digital input pins with a set of comparison values. The comparison values are loaded into registers through the SPI port. Differences between the captured values and the comparison values are detected and stored. Options are available for customizing SED test sequencing and error handling.

SED OPERATION

The SED circuitry operates on two data sets, one for each data port, each made up of four 16-bit input words, denoted as S0, S1, S2, and S3. To properly align the input samples, the first data-word (that is, S0) is indicated by asserting FRAME for at least one complete input sample.

Figure 92 shows the input timing of the interface for each port. The FRAME signal can be issued once at the start of the data transmission, or it can be asserted repeatedly at intervals coinciding with the S0 and S1 data-words.

FRAMEA/ FRAMEB

A[15:0]/ B[15:0]

Figure 92. Timing Diagram of Extended FRAME Signal Required to Align

Input Data for SED

The SED has five flag bits (Register 0x40, Bit 0, Bit 1, Bit 2, Bit 5 and Bit 6) that indicate the results of the input sample comparisons. The sample error detected bit (Bit 5, Register 0x40 for Port A and Bit 6, Register 0x40 for Port B) is set when an error is detected and remains set until cleared. The SED also provides registers that indicate which input data bits experienced errors (Register 0x41 through Register 0x44). These bits are latched and indicate the accumulated errors detected until cleared.

The autoclear mode has two effects: it activates the compare fail bits and the compare pass bit (Register 0x40, Bit 2, Bit 1, and Bit 0) and changes the behavior of Register 0x41 through Register 0x44. The compare pass bit sets if the last comparison indicated that the sample was error free. The compare fail bit sets if an error is detected. The compare fail bit is cleared automatically by the reception of eight consecutive error-free comparisons. When autoclear mode is enabled (Bit 3, Register 0x40), Register 0x41 through Register 0x44 accumulate errors as previously described but reset to all 0s after eight consecutive error-free sample comparisons are made.

The sample error, compare pass, and compare fail flags can be

IRQ

configured to trigger an

when active, if desired. This is done by enabling the appropriate bits in the event flag register (Register 0x07).

S3S1S0 S2 S0 S1

08910-089

Rev. A | Page 67 of 72

SED EXAMPLE

Normal Operation

The following example illustrates the SED configuration for continuously monitoring the input data and assertion of an when a single error is detected.

Write to the following registers to enable the SED and load

1. the comparison values:

Register 0x40  0x80 Register 0x00[4]  0 (to configure Port A SED) Register 0x38  S0[7:0] Register 0x39  S0[15:8] Register 0x3A  S1[7:0] Register 0x3B  S1[15:8] Register 0x3C  S2[7:0] Register 0x3D  S2[15:8] Register 0x3E  S3[7:0] Register 0x3F  S3[15:8] Register 0x00[4]  1 (to configure Port B SED) Register 0x38  S0[7:0] Register 0x39  S0[15:8] Register 0x3A  S1[7:0] Register 0x3B  S1[15:8] Register 0x3C  S2[7:0] Register 0x3D  S2[15:8] Register 0x3E  S3[7:0]

Register 0x3F  S3[15:8] Comparison values can be chosen arbitrarily; however, choosing values that require frequent bit toggling provides the most robust test.

Enable the SED error detect flag to assert the

IRQ

Begin transmitting the input data pattern.

IRQ

is asserted, read Register 0x40 and Register 0x41 through

If Register 0x44 with Bit 4, Register 0x00 = 0 for Port A and with Bit 4, Register 0x00 = 1 for Port B, to verify that a SED error was detected, and determine which input bits were in error. The bits in Register 0x41 through Register 0x44 are latched; therefore, the bits indicate any errors that occurred on those bits throughout the test and not just the errors that caused the error detected flag to be set.

Note that the FRAME signal is not required during normal operation when the device is configured for dual-port mode. To enable the alignment of the S0 sample as previously described requires the use of both the FRAMEA and FRAMEB signals.

The timing diagrams for single-port and byte modes are the same as during normal operation and are shown in Figure 47 and Figure 48, respectively. For single-port and byte mode, only FRAMEA and the IRQs for Port A should be used. The FRAMEA rising edge should always be aligned with the first sample of the data transmission. There should not be another rising edge until four complete words of data are received. This means four data samples for dualport mode and eight data samples for single-port and byte modes.

IRQ

pin.

Page 68

AD9148 Data Sheet

EXAMPLE START-UP ROUTINE

To ensure reliable start-up of the AD9148, certain sequences should be followed. An example start-up routine using the following device configuration is used for this example.

•

Interpolation = 4×, using HB1 = ’00’ and HB2 = ’000’

•

Input data = baseband data Dual port mode with 1 DCI

•

PLL = enabled

•

Fine NCO = enabled

•

Inverse sinc filter = disabled Synchronization = enabled

•

= 122.88 MSPS

DATA

= 140 MHz

OUT

= 122.88 MHz

REFCLK

DERIVED PLL SETTINGS

The following PLL settings can be derived from the device configuration:

•

N1 = f

•

N0 = f

DACCLK

= 4 × f

VCO

= f

× Interpolation = 491.52 MHz

DATA

= 1966.08 MHz (1 GHz < f

DACCLK

DACCLK/fREFCLK

VCO/fDACCLK

= 4

< 2 GHz)

VCO

DERIVED NCO SETTINGS

The following NCO settings can be derived from the device configuration:

•

= 140 MHz

OUT

DACCLK

= f

× Interpolation = 491.52 MHz

DATA

FTW = 140/(491.52) × 2

= 0x48, EAAAAA

START-UP SEQUENCE

The power clock and register write sequencing for reliable device start-up follows:

Power up the device (no specific power supply sequence is

•

required)

•

Apply a stable REFCLK input signal.

•

Apply a stable DCI input signal. Issue a hardware reset (optional)

•

Configure device registers with the following write

sequence:

0x0C  0xC9 0x0D  0xD9 0x0A  0xC0 0x0A  0x80 0x10  0x48 0x14  0x40 0x17  0x08 0x17  0x00 0x19  0x08 0x19  0x00 0x1C  0x40 0x1D  0x00 0x1E  0x01 0x54  0xAA 0x55  0xAA 0x56  0xEA 0x57  0x48 0x5A  0x01 0x5A  0x00

DEVICE VERIFICATION SEQUENCE

The following device polling can be conducted to verify that the device is working properly:

Read 0x06, Expect Bit 7 = 0, Bit 6 = 1, Bit 5 = 0, Bit 4 = 1,

•

Bit 2 = 1

•

Read 0x12, Expect Bit 6 = 1

•

Read 0x18, Expect 0x0F (0x07 is also normal)

•

Read 0x1A, Expect 0x0F (0x07 is also normal)

Rev. A | Page 68 of 72

Page 69

Data Sheet AD9148

OUTLINE DIMENSIONS

A1 BALL CORNER

1.30 MAX

12.10

12.00 SQ

11.90

TOP VIEW

DETAIL A

10.40

BSC SQ

0.80 REF

0.80 BSC

910 811121314 7 564231

BOTTOM VIEW

DETAIL A

0.96

0.70

G H J

L M N P

0.35 NOM

0.30 MIN

A1 BALL CORNER

SEATING

PLANE

COMPLIANT TO JEDEC STANDARDS MO-205-AE WITH EXCEPTION TO PACKAGE HEIGHT.

0.50

0.45

0.40

BALL DIAMETER

COPLANARITY

0.12

06-15-2010-A

Figure 93. 196-Ball Chip Scale Package, Ball Grid Array [CSP_BGA]

BC-196-7

Dimensions shown in millimeters

Rev. A | Page 69 of 72

Page 70

AD9148 Data Sheet

12.10

A1 BALL

PAD CORNER

12.00 SQ

11.90

TOP VIEW

11.20

REF SQ

8.20 SQ

10.40

BSC SQ

0.80

BSC

0.80 REF

91011121314 8 7 564231

BOTTOM VIEW

A1 BALL PAD CORNER

A B C D E F G H

J K L M N P

1.50

1.32

1.17

DETAIL A

0.75 REF

0.24 REF

SEATING

PLANE

COMPLIANT TO JEDEC STANDARDS MO-192.

DETAIL A

BALL DIAMETER

0.53

0.48

0.43

1.09

0.99

0.89

0.38

0.33

0.28

COPLANARITY

0.12

Figure 94. 196-Ball Ball Grid Array, Thermally Enhanced [BGA_ED]

BP-196-1

Dimensions shown in millimeters

ORDERING GUIDE

Model1 Temperature Range Package Description Package Option

AD9148BBCZ −40°C to +85°C 196-Ball Chip Scale Package Ball Grid Array [CSP_BGA] BC-196-7 AD9148BBCZRL −40°C to +85°C 196-Ball Chip Scale Package Ball Grid Array [CSP_BGA] BC-196-7 AD9148BBPZ −40°C to +85°C 196-Ball Ball Grid Array, Thermally Enhanced [BGA_ED] AD9148BBPZRL −40°C to +85°C 196-Ball Ball Grid Array, Thermally Enhanced [BGA_ED] AD9148-EBZ DAC Only Evaluation Board [BGA_ED] AD9148-M5372-EBZ AD9148 + ADL5372 Evaluation Board [BGA_ED] AD9148-M5375-EBZ AD9148 + ADL5375-0.5 Evaluation Board [BGA_ED] BP-196-1

Z = RoHS Compliant Part.

BP-196-1 BP-196-1 BP-196-1

BP-196-1

03-02-2010-A

Rev. A | Page 70 of 72

Page 71

Data Sheet AD9148

NOTES

Rev. A | Page 71 of 72

Page 72

AD9148 Data Sheet

NOTES

Rev. A | Page 72 of 72

Datasheet AD9148 Datasheet (ANALOG DEVICES)

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES

APPLICATIONS

TYPICAL SIGNAL CHAIN

GENERAL DESCRIPTION

PRODUCT HIGHLIGHTS

TABLE OF CONTENTS

REVISION HISTORY

FUNCTIONAL BLOCK DIAGRAM

SPECIFICATIONS

DC SPECIFICATIONS

INPUT/OUTPUT SIGNAL SPECIFICATIONS

DIGITAL INPUT DATA TIMING SPECIFICATIONS

AC SPECIFICATIONS

ABSOLUTE MAXIMUM RATINGS

THERMAL RESISTANCE

MAXIMUM SAFE POWER DISSIPATION

ESD CAUTION

PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS

TYPICAL PERFORMANCE CHARACTERISTICS

TERMINOLOGY

SERIAL PERIPHERAL INTERFACE

GENERAL OPERATION OF THE SERIAL INTERFACE

DATA FORMAT

SPI PIN DESCRIPTIONS

Serial Data I/O (SDIO)

Serial Data Output (SDO)

SPI OPTIONS

SPI REGISTER MAP

SPI REGISTER DESCRIPTIONS

INPUT DATA PORTS

DUAL-PORT MODE

SINGLE-PORT MODE

BYTE MODE

DATA INTERFACE OPTIONS

RECOMMENDED FRAME INPUT BIAS CIRCUITRY

FIFO OPERATION

SYNCHRONIZING AND RESETTING THE FIFO

FIFO Synchronization Modes

Data Rate Synchronization

FIFO Rate Synchronization

No Synchronization

FIFO Reset Commands

SPI Command for Manual Reset

FIFO Reset Using FRAME Signal

MONITORING THE FIFO STATUS

DEVICE SYNCHRONIZATION

SYNCHRONIZING MULTIPLE DEVICES

SYNCHRONIZATION WITH CLOCK MULTIPLICATION

SYNCHRONIZATION WITH DIRECT CLOCKING

Data Rate Mode Synchronization

FIFO Rate Mode Synchronization

ADDITIONAL SYNCHRONIZATION FEATURES

Sync Status Bits

Timing Optimization

INTERFACE TIMING

DIGITAL DATA PATH

PREMODULATION

PROGRAMMABLE INVERSE SINC FILTER

Filter Structure

Filter Implementation

INTERPOLATION FILTERS

Half-Band Filter 1 (HB1)

Half-Band Filter 2 (HB2)

Half-Band Filter 3 (HB3)

FINE MODULATION

Updating the Frequency Tuning Word

Phase Offset Adjustment

QUADRATURE PHASE CORRECTION

DC OFFSET CORRECTION

DIGITAL GAIN CONTROL

CLOCK GENERATION

DAC INPUT CLOCK CONFIGURATIONS

DRIVING THE CLK_x AND REFCLK_x INPUTS

DIRECT CLOCKING

CLOCK MULTIPLICATION

PLL Bias Settings