APPLICATIONS
Multimedia Audio
LCD Driver
ASIC Input or Output Amplifier
8-Lead SO
(R Suffix)
Headphone Driver
GENERAL DESCRIPTION
The AD8531, AD8532 and AD8534 are single, dual and quad
rail-to-rail input and output single-supply amplifiers featuring
250 mA output drive current. This high output current makes
these amplifiers excellent for driving either resistive or capacitive
loads. AC performance is very good with 3 MHz bandwidth,
8-Lead TSSOP
(RU Suffix)
5 V/µs slew rate and low distortion. All are guaranteed to oper-
ate from a +3 volt single supply as well as a +5 volt supply.
The very low input bias currents enable the AD853x to be used
for integrators and diode amplification and other applications
OUT A
–IN A
+IN A
V–
requiring low input bias current. Supply current is only 750 µA
per amplifier at 5 volts, allowing low current applications to
control high current loads.
14-Lead Epoxy DIP
(N Suffix)
Applications include audio amplification for computers, sound
ports, sound cards and set-top boxes. The AD853x family is
very stable and capable of driving heavy capacitive loads, such as
those found in LCDs.
The ability to swing rail-to-rail at the inputs and outputs enables
designers to buffer CMOS DACs, ASICs or other wide output
swing devices in single-supply systems.
The AD8531, AD8532 and AD8534 are specified over the
extended industrial (–40°C to +85°C) temperature range. The
AD8531 is available in SO-8 and SOT-23-5 packages. The
AD8532 is available in 8-lead plastic DIP, SO-8, 8-lead MSOP
and 8-lead TSSOP surface-mount packages. The AD8534 is
available in 14-lead plastic DIP, narrow SO-14 and 14-lead
TSSOP surface-mount packages. All TSSOP and SOT versions
are available in tape and reel only.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
N, R, RM, RT, RU Package . . . . . . . . . . –65°C to +150°C
Lead Temperature Range (Soldering, 60 sec) . . . . . . . +300°C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; the functional operation of
the device at these or any other conditions above those indicated in the operational
sections of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
For supplies less than +6 volts, the differential input voltage is equal to ±V
AD8531AR–40°C to +85°C8-Lead SOICSO-8
AD8531ART*–40°C to +85°C5-Lead SOT-23RT-5A7A
AD8532AR–40°C to +85°C8-Lead SOICSO-8
AD8532ARM*–40°C to +85°C8-Lead MSOPRM-8ARA
AD8532AN–40°C to +85°C8-Lead Plastic DIPN-8
AD8532ARU*–40°C to +85°C8-Lead TSSOPRU-8
AD8534AR–40°C to +85°C14-Lead SOICSO-14
AD8534AN–40°C to +85°C14-Lead Plastic DIPN-14
AD8534ARU*–40°C to +85°C14-Lead TSSOPRU-14
*Available in reels only.
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD8531/AD8532/AD8534 feature proprietary ESD protection circuitry, perm anent
damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper
ESD precautions are recommended to avoid performance degradation or loss of functionality.
2.5
–V
OL
2
+V
OH
1.5
OUT
6V
1
0.5
0
020406080100 120 140 160 180 200
R
LOAD –
V
Figure 1. Output Voltage vs. Load. VS = ±2.5 V, RL Is Connected to GND (0 V)
Figure 14. Closed-Loop Output
Impedance vs. Frequency
5
4
3
2
OUTPUT SWING – Volts p-p
PHASE SHIFT – Degrees
1
0
1k10k10M100k1M
FREQUENCY – Hz
Figure 12. Closed-Loop Output
Voltage Swing vs. Frequency
100
90
100mV/div
10
0%
MARKER 41mV/ Hz
VS = +5V
A
T
FREQUENCY = 1kHz
Figure 15. Voltage Noise Density
vs. Frequency
VS = +2.7V
T
A
R
L
V
IN
= 1000
V
= +258C
A
= +258C
= 2kV
= 2.5V p-p
Figure 13. Closed-Loop Output
Voltage Swing vs. Frequency
Figure 16. Voltage Noise Density
vs. Frequency
1
0.1
CURRENT NOISE DENSITY – pA/ Hz
0.01
10100100k
Figure 17. Current Noise Density
vs. Frequency
1k
FREQUENCY – Hz
VS = +5V
T
= +258C
A
10k
110
VS = +5V
T
= +258C
100
A
90
80
70
60
50
COMMON-MODE REJECTION – dB
40
1k10k10M100k1M
FREQUENCY – Hz
Figure 18. Common-Mode Rejection vs. Frequency
–6–
Figure 19. Power Supply Rejection
vs. Frequency
REV. B
Page 7
AD8531/AD8532/AD8534
SMALL SIGNAL OVERSHOOT – %
50
40
0
30
20
10
60
CAPACITANCE – pF
10100100001000
–OS
+OS
VS = +5V
T
A
= +258C
R
L
= 2kV
TEMPERATURE – 8C
SUPPLY CURRENT/AMPLIFIER – mA
0.9
0.65
0.5
–40
–200 20406080
0.85
0.7
0.6
0.55
0.8
0.75
VS = 5V
VS = 3V
500 ns/DIV
20mV/DIV
VS = 62.5V
V
IN
= 650mV
A
V
= 1
R
L
= 2kV
C
L
= 300pF
T
A
= +258C
0V
140
VS = +5V
120
T
= +258C
A
100
80
60
PSRR+
40
20
0
–20
POWER SUPPLY REJECTION – dB
–40
–60
PSRR–
10k
1k
FREQUENCY – Hz
100k1M10M100
Figure 20. Power Supply Rejection
vs. Frequency
50
VS = +5V
T
= +258C
A
R
= 600V
L
40
30
20
10
SMALL SIGNAL OVERSHOOT – %
0
10100100001000
–OS
+OS
CAPACITANCE – pF
Figure 23. Small Signal Overshoot
vs. Load Capacitance
50
VS = +2.7V
= +258C
T
A
40
= 2kV
R
L
30
20
10
SMALL SIGNAL OVERSHOOT – %
0
10100100001000
–OS
+OS
CAPACITANCE – pF
Figure 21. Small Signal Overshoot
vs. Load Capacitance
50
VS = +2.7V
T
= +258C
A
= 600V
R
40
L
30
20
–OS
10
SMALL SIGNAL OVERSHOOT – %
0
10100100001000
+OS
CAPACITANCE – pF
Figure 24. Small Signal Overshoot
vs. Load Capacitance
Figure 22. Small Signal Overshoot
vs. Load Capacitance
Figure 25. Supply Current per
Amplifier vs. Temperature
0.80
TA = +258C
0.70
0.60
0.50
0.40
0.30
0.20
0.10
SUPPLY CURRENT/AMPLIFIER – mA
0.00
0.75 1.00
Figure 26. Supply Current per
Amplifier vs. Supply Voltage
REV. B
1.502.002.503.00
SUPPLY VOLTAGE – 6Volts
0V
20mV/DIV
VS = 61.35V
VIN = 650mV
= 1
A
V
= 2kV
R
L
C
= 300pF
L
= +258C
T
A
500 ns/DIV
Figure 27. Small Signal Transient
Response
–7–
Figure 28. Small Signal Transient
Response
Page 8
AD8531/AD8532/AD8534
50mA
100mA100mA
20mA
V
B2
M5
M8
M12
M15
M16
M11
OUT
M3
M4
M2
M1
IN–
IN+
V
B3
M6
M7M10
20mA
M13
50mA
V+
V–
M9
M14
VS = 62.5V
= 1
A
100
90
10
0%
500mV
Figure 29. Large Signal Transient
Response
V
R
L
T
A
500ns
= 2kV
= +258C
100
90
10
0%
500mV
Figure 30. Large Signal Transient
Response
APPLICATIONS
THEORY OF OPERATION
The AD8531/AD8532/AD8534 is an all-CMOS, high output
current drive, rail-to-rail input/output operational amplifier.
This is the latest entry in Analog Devices’ expanding family of
single-supply devices for the multimedia and telecom marketplaces. Its high output current drive and stability with heavy
capacitive loads makes the AD8531/AD8532/AD8534 an excellent choice as a drive amplifier for LCD panels.
Figure 32 illustrates a simplified equivalent circuit for the AD8531/
AD8532/AD8534. Like many rail-to-rail input amplifier configurations, it is comprised of two differential pairs, one n-channel
(M1–M2) and one p-channel (M3–M4). These differential pairs
are biased by 50 µA current sources, each with a compliance
limit of approximately 0.5 V from either supply voltage rail. The
differential input voltage is then converted into a pair of differential output currents. These differential output currents are
then combined in a compound folded-cascade second gain
stage (M5–M9). The outputs of the second gain stage at M8
and M9 provide the gate voltage drive to the rail-to-rail output
stage. Additional signal current recombination for the output
stage is achieved through the use of transistors M11–M14.
In order to achieve rail-to-rail output swings, the AD8531/
AD8532/AD8534 design employs a complementary commonsource output stage (M15–M16). However, the output voltage
swing is directly dependent on the load current, as the difference
between the output voltage and the supply is determined by the
AD8531/AD8532/AD8534’s output transistors on-channel
resistance (see Figures 8 and 9). The output stage also exhibits
voltage gain by virtue of the use of common-source amplifiers;
as a result, the voltage gain of the output stage (thus, the openloop gain of the device) exhibits a strong dependence to the total
load resistance at the output of the AD8531/AD8532/AD8534.
As a result of the design of the output stage for maximum load
current capability, the AD8531/AD8532/AD8534 does not have
any internal short-circuit protection circuitry. Direct connection of
the AD8531/AD8532/AD8534’s output to the positive supply
in single-supply applications will destroy the device. In those
applications where some protection is needed, but not at the
expense of reduced output voltage headroom, a low value resistor in series with the output, as shown in Figure 33, can be
used. The resistor, connected within the feedback loop of the
amplifier, will have very little effect on the performance of the
amplifier other than limiting the maximum available output voltage swing. For single +5 V supply applications, resistors less than
20 Ω are not recommended.
+5V
R
V
IN
AD8532
20V
X
V
OUT
Figure 33. Output Short-Circuit Protection
–8–
REV. B
Page 9
AD8531/AD8532/AD8534
Power Dissipation
Although the AD8531/AD8532/AD8534 is capable of providing
load currents to 250 mA, the usable output load current drive
capability will be limited to the maximum power dissipation
allowed by the device package used. In any application, the
absolute maximum junction temperature for the AD8531/
AD8532/AD8534 is 150°C, and should never be exceeded for
the device could suffer premature failure. Accurately measuring
power dissipation of an integrated circuit is not always a
straightforward exercise, so Figure 34 has been provided as a
design aid for either setting a safe output current drive level or
in selecting a heatsink for the three package options available on
the AD8531/AD8532/AD8534.
1.5
PDIP
u
= 1038C/W
JA
1
SOIC
u
= 1588C/W
JA
TSSOP
u
= 2408C/W
JA
0.5
POWER DISSIPATION – Watts
0
010025
TEMPERATURE – 8C
5075
TJ MAX = 1508C
FREE AIR
NO HEATSINK
85
Figure 34. Maximum Power Dissipation vs. Ambient
Temperature
These thermal resistance curves were determined using the
AD8531/AD8532/AD8534 thermal resistance data for each
package and a maximum junction temperature of 150°C. The
following formula can be used to calculate the internal junction
temperature of the AD8531/AD8532/AD8534 for any application:
TJ = P
DISS
×θ
JA
+ T
A
where TJ = junction temperature;
= power dissipation;
P
DISS
θ
= package thermal resistance,
JA
junction-to-case; and
= Ambient temperature of the circuit.
T
A
To calculate the power dissipated by the AD8531/AD8532/
AD8534, the following equation can be used:
P
where I
= I
DISS
= is output load current;
LOAD
V
= is supply voltage; and
S
V
= is output voltage.
OUT
LOAD
× (VS–V
OUT
)
The quantity within the parentheses is the maximum voltage
developed across either output transistor. As an additional
design aid in calculating available load current from the
AD8531/AD8532/AD8534, Figure 1 illustrates the AD8531/
AD8532/AD8534 output voltage as a function of load resistance.
Power Calculations for Varying or Unknown Loads
Often, calculating power dissipated by an integrated circuit to
determine if the device is being operated in a safe range is not
as simple as it might seem. In many cases power cannot be
directly measured. This may be the result of irregular output
waveforms or varying loads; indirect methods of measuring
power are required.
There are two methods to calculate power dissipated by an
integrated circuit. The first can be done by measuring the package temperature and the board temperature. The other is to
directly measure the circuit’s supply current.
Calculating Power by Measuring Ambient and Case
Temperature
Given the two equations for calculating junction temperature:
= T
T
+ P θ
J
A
JA
where TJ is junction temperature, and TA is ambient tempera-
ture. θ
where T
is the junction to ambient thermal resistance.
JA
= TC + P
T
J
is case temperature and θJA and θ
C
θ
JC
are given in the
JC
data sheet.
The two equations can be solved for P (power):
T
+ P θ
A
P = (TA – TC )/ (
= TC + P
JA
θ
JC
θ
–
θ
)
JC
JA
Once power has been determined it is necessary to go back and
calculate the junction temperature to assure that it has not been
exceeded.
The temperature measurements should be directly on the package and on a spot on the board that is near the package but
definitely not touching it. Measuring the package could be difficult. A very small bimetallic junction glued to the package could
be used or it could be done using an infrared sensing
device if the spot size is small enough.
Calculating Power by Measuring Supply Current
Power can be calculated directly knowing the supply voltage and
current. However, supply current may have a dc component
with a pulse into a capacitive load. This could make rms current
very difficult to calculate. It can be overcome by lifting the supply pin and inserting an rms current meter into the circuit. For
this to work you must be sure all of the current is being delivered by the supply pin you are measuring. This is usually a good
method in a single supply system; however, if the system uses
dual supplies, both supplies may need to be monitored.
Input Overvoltage Protection
As with any semiconductor device, whenever the condition
exists for the input to exceed either supply voltage, the device’s
input overvoltage characteristic must be considered. When an
overvoltage occurs, the amplifier could be damaged depending
on the magnitude of the applied voltage and the magnitude of
the fault current. Although not shown here, when the input
voltage exceeds either supply by more than 0.6 V, pn-junctions
internal to the AD8531/AD8532/AD8534 energize allowing
current to flow from the input to the supplies. As illustrated in
the simplified equivalent input circuit (Figure 32), the AD8531/
AD8532/AD8534 does not have any internal current limiting
resistors, so fault currents can quickly rise to damaging levels.
REV. B
–9–
Page 10
AD8531/AD8532/AD8534
This input current is not inherently damaging to the device as
long as it is limited to 5 mA or less. For the AD8531/AD8532/
AD8534, once the input voltage exceeds the supply by more
than 0.6 V the input current quickly exceeds 5 mA. If this
condition continues to exist, an external series resistor should
be added. The size of the resistor is calculated by dividing the
maximum overvoltage by 5 mA. For example, if the input
voltage could reach 10 V, the external resistor should be (10 V/
5 mA) = 2 kΩ. This resistance should be placed in series with
either or both inputs if they are exposed to an overvoltage condition. For more information on general overvoltage characteristics of amplifiers refer to the 1993 Seminar Applications Guide,
available from the Analog Devices Literature Center.
Output Phase Reversal
Some operational amplifiers designed for single-supply operation exhibit an output voltage phase reversal when their inputs
are driven beyond their useful common-mode range. The
AD8531/AD8532/AD8534 is free from reasonable input voltage
range restrictions provided that input voltages no greater than
the supply voltage rails are applied. Although the device’s output will not change phase, large currents can flow through
internal junctions to the supply rails, as was pointed out in the
previous section. Without limit, these fault currents can easily
destroy the amplifier. The technique recommended in the
input overvoltage protection section should therefore be applied
in those applications where the possibility of input voltages
exceeding the supply voltages exists.
Capacitive Load Drive
The AD8531/AD8532/AD8534 exhibits excellent capacitive
load driving capabilities. It can drive up to 10 nF directly as
shown in Figures 21 through 24. However, even though the
device is stable, a capacitive load does not come without a
penalty in bandwidth. As shown in Figure 35, the bandwidth is
reduced to under 1 MHz for loads greater than 10 nF. A “snubber” network on the output won’t increase the bandwidth, but
it does significantly reduce the amount of overshoot for a given
capacitive load. A snubber consists of a series R-C network
, CS), as shown in Figure 36, connected from the output of
(R
S
the device to ground. This network operates in parallel with the
load capacitor, C
, to provide phase lag compensation. The actual
L
value of the resistor and capacitor is best determined empirically.
4
3.5
3
2.5
2
VS = 62.5V
= 1kV
R
L
T
= +258C
A
+5V
C
L
47nF
V
OUT
V
100mV p-p
AD8532
IN
R
5V
S
C
1mF
S
Figure 36. Snubber Network Compensates for Capacitive
Loads
The first step is to determine the value of the resistor, RS. A
good starting value is 100 Ω. This value is reduced until the
small-signal transient response is optimized. Next, C
is deter-
S
mined—10 µF is a good starting point. This value is reduced to
the smallest value for acceptable performance (typically, 1 µF).
For the case of a 47 nF load capacitor on the AD8531/AD8532/
AD8534, the optimal snubber network is a 5 Ω in series with
1 µF. The benefit is immediately apparent as seen in the scope
photo in Figure 37. The top trace was taken with a 47 nF load
and the bottom trace with the 5 Ω—1 µF snubber network in
place. The amount of overshoot and ringing is dramatically reduced. Table I below illustrates a few sample snubber networks
for large load capacitors:
Table I. Snubber Networks for Large Capacitive Loads
Load CapacitanceSnubber Network
(CL)(R
, CS)
S
0.47 nF300 Ω, 0.1 µF
4.7 nF30 Ω, 1 µF
47 nF5 Ω, 1 µF
50mV
ONLY
100
90
10
0%
50mV
10ms
47nF LOAD
SNUBBER
IN CIRCUIT
Figure 37. Overshoot and Ringing Is Reduced by Adding
a Snubber Network in Parallel with the 47 nF Load
1.5
BANDWIDTH – MHz
1
0.5
0
0.011000.1
CAPACITIVE LOAD – nF
110
Figure 35. Unity-Gain Bandwidth vs. Capacitive Load
–10–
REV. B
Page 11
AD8531/AD8532/AD8534
R
L
600V
C1
22mF
A2
7
6
5
3
1
2
A1
+5V
R1
10kV
R2
10kV
R11
10kV
R7
10kV
6
7
5
A1
+12V
+5V
R8
100kV
R9
100kV
C2
1mF
R12
10kV
R14
50V
A2
1
2
3
R3
10kV
R6
10kV
R13
10kV
C3
47mF
V
O1
V
O2
C4
47mF
A1, A2 = 1/2 AD8532
GAIN =
R3
R2
SET: R7, R10, R11 = R2
SET: R6, R12, R13 = R3
V
IN
R10
10kV
R5
50V
A High Output Current, Buffered Reference/Regulator
Many applications require stable voltage outputs relatively close
in potential to an unregulated input source. This “low dropout” type of reference/regulator is readily implemented with a
rail-to-rail output op amp, and is particularly useful when using
a higher current device such as the AD8531/AD8532/AD8534.
A typical example is the 3.3 V or 4.5 V reference voltage developed from a 5 V system source. Generating these voltages
requires a three terminal reference, such as the REF196 (3.3 V)
or the REF194 (4.5 V), both which feature low power, with
sourcing outputs of 30 mA or less. Figure 38 shows how such a
reference can be outfitted with an AD8531/AD8532/AD8534
buffer for higher currents and/or voltage levels, plus sink and
source load capability.
+V
S
+5V
0.1mF
V
C
ON/OFF
CONTROL
INPUT CMOS HI
(OR OPEN) = ON
LO = OFF
V
S
COMMON
C1
0.1mF
R1
10kV
1%
C3
3
U1
REF196
2
6
V
=
OUT2
3.3V
4
(SeeText)
C4
1mF
R3
R4
3.3kV
U2
AD8531
R2
10kV 1%
C2
0.1mF
V
=
OUT1
3.3V @ 100mA
C5
100mF/16V
TANTALUM
R5
0.2V
V
OUT
COMMON
To scale V
resistor R3 (shown dotted) is added, causing, the new V
to another (higher) output level, the optional
OUT2
OUT1
to
become:
V
OUT1=VOUT 2
× 1+
R2
R3
The circuit can either be used as shown, as a 5 V to 3.3 V
reference/regulator, or with ON/OFF control. By driving Pin 3
of U1 with a logic control signal as noted, the output is switched
ON/OFF. Note that when ON/OFF control is used, resistor R4
must be used with U1 to speed ON-OFF switching.
A Single-Supply, Balanced Line Driver
The circuit in Figure 39 is a unique line driver circuit topology
used in professional audio applications and has been modified
for automotive and multimedia audio applications. On a single
+5 V supply, the line driver exhibits less than 0.7% distortion
into a 600 Ω load from 20 Hz to 15 kHz (not shown) with an
input signal level of 4 V p-p. In fact, the output drive capability
of the AD8531/AD8532/AD8534 maintains this level for loads
as small as 32 Ω. For input signals less than 1 V p-p, the THD
is less than 0.1%, regardless of load. The design is a transformerless, balanced transmission system where output common-mode
rejection of noise is of paramount importance. As with the
transformer-based system, either output can be shorted to
ground for unbalanced line driver applications without changing
the circuit gain of 1. Other circuit gains can be set according to
the equation in the diagram. This allows the design to be easily
configured for inverting, noninverting or differential operation.
Figure 38. A High Output Current Reference/Regulator
The low dropout performance of this circuit is provided by
stage U2, an AD8531 connected as a follower/buffer for the basic
reference voltage produced by U1. The low voltage saturation
characteristic of the AD8531/AD8532/AD8534 allows up to
100 mA of load current in the illustrated use, as a 5 V to 3.3 V
converter with good dc accuracy. In fact, the dc output voltage
change for a 100 mA load current delta measured less than
1 mV. This corresponds to an equivalent output impedance of
< 0.01 Ω. In this application, the stable 3.3 V from U1 is ap-
plied to U2 through a noise filter, R1–C1. U2 replicates the U1
voltage within a few millivolts, but at a higher current output at
, with the ability to both sink and source output current(s)
V
OUT1
—unlike most IC references. R2 and C2 in the feedback path of
U2 provide additional noise filtering.
Transient performance of the reference/regulator for a 100 mA
step change in load current is also quite good and is largely
determined by the R5–C5 output network. With values as
shown, the transient is about 20 mV peak and settles to within
2 mV in less than 10 µs for either polarity. Although room exists
for optimizing the transient response, any changes to the R5–C5
network should be verified by experiment to preclude the possibility of excessive ringing with some capacitor types.
REV. B
–11–
Figure 39. A Single-Supply, Balanced Line Driver for
Multimedia and Automotive Applications
Page 12
AD8531/AD8532/AD8534
V
IN
3
2
1
U1A
AD8532
+V
S
4
R1
31.6kV
C1
0.01mF
C2
0.01mF
R2
31.6kV
R5
31.6kV
R6
31.6kV
R4
49.9V
HI
LO
500Hz
AND UP
DC –
500Hz
6
5
7
C3
0.01mF
U1B
AD8532
C4
0.02mF
R7
15.8kV
R3
49.9V
270mF
270mF
100kV
+V
S
10mF
100kV
100kV
C
IN
10mF
R
IN
100kV
0.1mF100mF/25V
+V
S
TO U1
+5V
COM
+
100kV
+
A Single-Supply Headphone Amplifier
Because of its speed and large output drive, the AD8531/AD8532/
AD8534 makes an excellent headphone driver, as illustrated in
Figure 40. Its low supply operation and rail-to-rail inputs and
outputs give a maximum signal swing on a single +5 V supply.
To ensure maximum signal swing available to drive the headphone, the amplifier inputs are biased to V+/2, which in this
case is 2.5 V. The 100 kΩ resistor to the positive supply is
equally split into two 50 kΩ resistors, with their common point
bypassed by 10 µF to prevent power supply noise from contami-
nating the audio signal.
The audio signal is then ac-coupled to each input through a
10 µF capacitor. A large value is needed to ensure that the
20 Hz audio information is not blocked. If the input already has
the proper dc bias, the ac coupling and biasing resistors are not
required. A 270 µF capacitor is used at the output to couple the
amplifier to the headphone. This value is much larger than that
used for the input because of the low impedance of the head-
phones, which can range from 32 Ω to 600 Ω. An additional
16 Ω resistor is used in series with the output capacitor to pro-
tect the op amp’s output stage by limiting capacitor discharge
current. When driving a 48 Ω load, the circuit exhibits less than
0.3% THD+N at output drive levels of 4 V p-p.
+V + 5V
50kV
+V + 5V
1mF/0.1mF
In this two-way example, the LO signal is a dc-500 Hz
LP woofer output, and the HI signal is the HP (>500 Hz)
tweeter output. U1B forms an LP section at 500 Hz, while
U1A provides a HP section, covering frequencies ≥500 Hz.
16V
16V
270mF
270mF
50kV
50kV
+V
100kV
100kV
10mF
10mF
50kV
LEFT
INPUT
10mF
50kV
50kV
RIGHT
INPUT
10mF
Figure 40. A Single-Supply, Stereo Headphone Driver
A Single-Supply, Two-Way Loudspeaker Crossover Network
Active filters are useful in loudspeaker crossover networks for
reasons of small size, relative freedom from parasitic effects, the
ease of controlling low/high channel drive and the controlled driver
1/2
AD8532
1/2
AD8532
damping provided by a dedicated amplifier. Both Sallen-Key
(SK) and multiple-feedback (MFB) filter architectures are useful in implementing active crossover networks. The circuit
shown in Figure 41 is a single-supply, two-way active crossover
which combines the advantages of both filter topologies. This
active crossover exhibits less than 0.4% THD+N at output levels
of 1.4 V rms using general purpose unity-gain HP/LP stages.
LEFT
HEADPHONE
RIGHT
HEADPHONE
Figure 41. A Single-Supply, Two-Way Active Crossover
The crossover example frequency of 500 Hz can be shifted
lower or higher by frequency scaling of either resistors or
capacitors. In configuring the circuit for other frequencies,
complementary LP/HP action must be maintained between
sections, and component values within the sections must be in
the same ratio. Table II provides a design aid to adaptation,
with suggested standard component values for other frequencies.
Table II. RC Component Selection for Various
Crossover Frequencies
For Sallen-Key stage U1A: R1 = R2, and C1 = C2, etc.
2
For Multiple Feedback stage U1B: R6 = R5, R7 = R5/2, and
C4 = 2C3.
For additional information on the active filters and active crossover networks, please consult the data sheet for the OP279, a
dual rail-to-rail high-output current operational amplifier.
–12–
REV. B
Page 13
AD8531/AD8532/AD8534
Direct Access Arrangement for Telephone Line Interface
Figure 42 illustrates a +5 V only transmit/receive telephone line
interface for 600 Ω transmission systems. It allows full duplex
transmission of signals on a transformer coupled 600 Ω line in a
differential manner. Amplifier A1 provides gain that can be
adjusted to meet the modem output drive requirements. Both
A1 and A2 are configured to apply the largest possible signal on
a single supply to the transformer. Because of the high output
current drive and low dropout voltage of the AD8531/AD8532/
AD8534s, the largest signal available on a single +5 V supply is
approximately 4.5 V p-p into a 600 Ω transmission system.
Amplifier A3 is configured as a difference amplifier for two
reasons: (1) It prevents the transmit signal from interfering with
the receive signal and (2) it extracts the receive signal from the
transmission line for amplification by A4. A4’s gain can be
adjusted in the same manner as A1’s to meet the modem’s input
signal requirements. Standard resistor values permit the use of
SIP (Single In-line Package) format resistor arrays. Couple this
with the AD8531/AD8532/AD8534’s 8-lead SOIC or TSSOP
footprint and this circuit offers a compact, cost-sensitive solution.
P1
Tx GAIN
TO TELEPHONE
LINE
1:1
Z
O
600V
T1
MIDCOM
671-8005
A1, A2 = 1/2 AD8532
A3, A4 = 1/2 AD8532
6.2V
6.2V
R11
10kV
360V
R9
10kV
R12
10kV
ADJUST
R3
R5
10kV
R6
10kV
R10
10kV
2
A3
3
2kV
1
7
1
9.09kV
A1
A2
R13
10kV
R2
2
3
6
5
R14
14.3kV
6
5
R1
10kV
A4
C1
0.1mF
+5V DC
10mF
P2
Rx GAIN
ADJUST
2kV
7
C2
0.1mF
TRANSMIT
TxA
R7
10kV
R8
10kV
RECEIVE
RxA
Figure 42. A Single-Supply Direct Access Arrangement
for Modems
REV. B
–13–
Page 14
AD8531/AD8532/AD8534
* AD8531/AD8532/AD8534 SPICE Macro-model 3/96, REV. B
* 5-Volt Version ARG / ADSC
*
* Copyright 1996 by Analog Devices
*
* Refer to “README.DOC” file for License Statement. Use of this model
* indicates your acceptance of the terms and provisions in the License
* Statement.
*
* Node assignments
*noninverting input
*|inverting input
*||positive supply
*|||negative supply
*||||output
*|||||
.SUBCKT AD8531/AD8532/AD8534_5 12995040
*
* INPUT STAGE
*
M1 32650NIXL=6UW=25U
M2 47650NIXL=6UW=25U
M3 8255PIXL=6UW=25U
M4 9755PIXL=6UW=25U
EOS71POLY(1)25985E-3 0.451
IIN11985P
IIN22985P
IOS210.5P
I199550U
I265050U
R19934.833K
R29944.833K
R38504.833K
R49504.833K
D3599DX
D4506DX
*
* GAIN STAGE
*
EREF980POLY(2)99050000.5
+0.5
G19821POLY(2)43980
+145U+145U
RG219818.078E6
CC214014P
D12122DX
D22321DX
V199221.37
V223501.37
*