Datasheet AD8200R, AD8200CHIPS Datasheet (Analog Devices)

High Common-Mode Voltage, Single Supply

–IN

GND

A1

A2

+IN

NC

+V

S

OUT

AD8200

5V

OUTPUT

INDUCTIVE
LOAD

4 TERM

SHUNT

CLAMP

DIODE

BATTERY 14V

POWER
DEVICE

COMMON NC = NO CONNECT

a

FEATURES
High Common-Mode Voltage Range –2 V to +24 V at a

5 V Supply Voltage

Operating Temperature Range

Die: –40C to +150C

8-Lead SOIC: –40C to +125C
Supply Voltage Range: 4.7 V to 12 V
Low-Pass Filter (One Pole or Two Pole)

EXCELLENT AC AND DC PERFORMANCE
15 V/C Max Offset Drift
20 ppm/C Max Gain Drift
80 dB CMRR Min DC to 10 kHz

PLATFORMS
Transmission Control
Diesel Injection Control
Engine Management
Semi-Active Suspension Control
Vehicle Dynamics Control

+IN

–IN

FUNCTIONAL BLOCK DIAGRAM

NC A1 A2

200k200k

NC = NO CONNECT

Difference Amplifier

AD8200

SOIC (R) Package

DIE Form

+V

S

AD8200

10k

10k

GND

G = X10

+IN

A1

–IN

100k

G = X2

+IN

A2

–IN

OUT

GENERAL DESCRIPTION

The AD8200 is a single-supply difference amplifier for amplifying
and low-pass filtering small differential voltages in the presence
of a large common-mode voltage. The input CMV range extends
from –2 V to +24 V at a typical supply voltage of 5 V.

The AD8200 is offered in die and packaged form. Both package
options are specified over wide temperature ranges, making the
AD8200 well suited for use in many automotive platforms. The
SOIC package is specified over a temperature range of –40°C to
+125°C. The die is specified from –40°C to +150°C.

BATTERY

INDUCTIVE

CLAMP

DIODE

14V

4 TERM

SHUNT

POWER
DEVICE

COMMON NC = NO CONNECT

LOAD

+IN

AD8200

–IN

NC

GND

5V

+V

OUT

S

A1

OUTPUT

A2

Figure 1. High-Line Current Sensor

REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

Automotive platforms demand precision components for better
system control. The AD8200 provides excellent ac and dc per­formance that keeps errors to a minimum in the user’s system.
Typical offset and gain drift in the SOIC package are 6 µV/°C
and 10 ppm/°C, respectively. The device also delivers a mini­mum CMRR of 80 dB from dc to 10 kHz.

The AD8200 features an externally accessible 100 kΩ resistor at
the output of the preamp A1, which can be used for low-pass
filter applications, and for establishing gains other than 20.

Figure 2. Low-Line Current Sensor

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2000

AD8200–SPECIFICATIONS

SINGLE SUPPLY

(TA = 25C, VS = 5 V, VCM = 0 V, RL = 10 k, Pin 5 to ground, unless otherwise noted.)

AD8200 SOIC AD8200 DIE

Parameter Condition Min Typ Max Min Typ Max Unit

SYSTEM GAIN

Initial 20 20
Error V

≥ 0.1 V dc –1 +1 –1 +1 %

O

vs. Temperature 10 20 25 30 ppm/°C

OFFSET VOLTAGE

Offset Voltage (RTI) VCM = 0.15 V –1 +1 –1 +1 mV
vs. Temperature 6 15 12 25 µV/°C

INPUT

Input Impedance

Differential 320 400 480 320 400 480 kΩ
Common-Mode 160 200 240 160 200 240 kΩ

CMV Continuous –2 +24 –2 +24 V

Common-Mode Rejection

1

VCM = 10 V
f = 1 kHz 80 80 dB
f = 10 kHz

2

80 80 dB

PREAMPLIFIER

Gain 10 10
Gain Error –1 +1 –1 +1 %
Output Voltage Range 0.02 4.8 0.02 4.8 V
Output Resistance 97 100 103 97 100 103 kΩ

OUTPUT BUFFER

Gain 2 2
Gain Error –1 +1 –1 +1 %
Output Voltage Range 0.02 4.8 0.02 4.8 V
Output Resistance 2 2 Ω

DYNAMIC RESPONSE

3 dB Bandwidth 30 50 30 50 kHz
Slew Rate 0.22 0.22 V/µs

NOISE

0.1 Hz to 10 Hz 10 10 µV p-p
Spectral Density, 1 kHz, RTI 300 300 nV/√Hz

POWER SUPPLY

Operating Range 4.7 12 4.7 12 V
Quiescent Current vs. Temp V

= 0.1 V dc 0.25 1 0.25 1 mA

O

PSRR VS = 4.7 V to 12 V 75 80 75 80 dB

TEMPERATURE RANGE

For Specified Performance –40 +125 –40 +150 °C

NOTES

1

Source Imbalance < 2 Ω.

2

The AD8200 preamplifier exceeds 80 dB CMRR at 10 kHz. However, since the signal is available only by way of a 100 k Ω resistor, even the small amounts of pin­to-pin capacitance between Pins 1, 8 and 3, 4 may couple an input common-mode signal larger than the greatly attenuated preamplifier output. The effect of pin-to­pin coupling may be neglected in all applications using filter capacitors at Node 3.

Specifications subject to change without notice.

–2–

REV. 0

AD8200

ABSOLUTE MAXIMUM RATINGS*

PIN CONFIGURATION

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12.5 V

Transient Input Voltage (300 ms) . . . . . . . . . . . . . . . . . . 44 V

Continuous Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . 35 V

Reversed Supply Voltage Protection . . . . . . . . . . . . . . . 0.3 V

Operating Temperature . . . . . . . . . . . (Die) – 40°C to +150°C

. . . . . . . . . (SOIC) –40°C to +125°C

Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C

Output Short Circuit Duration . . . . . . . . . . . . . . . . Indefinite

1

–IN

2

GND

3

A1

(Not to Scale)

A2

4

NC = NO CONNECT

AD8200

TOP VIEW

Lead Temperature Range (Soldering 60 sec) . . . . . . . . 300°C

*Stresses beyond those listed under Absolute Maximum Ratings may cause

permanent damage to the device. This is a stress rating only; the functional
operation of the device at these or any other conditions above those indicated in
the operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD8200R –40°C to +125°C Plastic SOIC SO-8
AD8200CHIPS –40°C to +150°C DIE Form

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD8200 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

8

+IN

7

NC

6

+V

S

5

OUT

WARNING!

ESD SENSITIVE DEVICE

+IN

–IN 1

METALLIZATION PHOTOGRAPH

+V

S

6

8

2

GND

3

A1

OUT5

A24

REV. 0

–3–

AD8200–Typical Performance Characteristics

(TA = 25C, VS = 5 V, VCM = 0 V, RL = 10 k unless otherwise
noted.)

30

25

20

15

10

5

POSITIVE COMMON-MODE RANGE – Volts

0

253

SUPPLY VOLTAGE – Volts

+V

CM

–V

CM

4

TPC 1. Input Common-Mode Range vs. Supply

0

OUTPUT VOLTAGE – mV

–10

–15

–20

–25

–30

–35

–5

RL = 10k TO GND

2

SUPPLY VOLTAGE – Volts

RL =

4

TPC 2. Output Voltage – VS vs. Supply

0

–2

–4

–6

–8

–10

NEGATIVE COMMON-MODE RANGE – Volts

–12

GAIN – dB

–10

–15

–20

30

25

20

15

10

5

0

–5

1k

10k 100k 1M

FREQUENCY – Hz

TPC 4. Gain vs. Frequency

100

95

90

85

80

75

CMRR – dB

70

65

60

55

50

53

10

100 100k

1k 10k 1M

FREQUENCY – Hz

TPC 5. Common-Mode Rejection vs. Frequency

5

4

3

2

OUTPUT VOLTAGE – Volts

1

0

10

100 1k 10k
LOAD RESISTANCE – 

TPC 3. Output Voltage Swing vs. Load Resistance

–4–

100

90

80

70

60

50

PSRR – dB

40

30

20

10

0

10

100 10k1k 100k

FREQUENCY – Hz

TPC 6. Power Supply Rejection vs. Frequency

REV. 0

AD8200

TEK RUN: 2.5MS/s HI RES

V

, RL = 10k

OUT

1

V

IN

2

CH1 500mV 50mV M 20s CH1 1.5VCH2

T

TPC 7. Pulse Response

THEORY OF OPERATION

The AD8200 consists of a preamp and buffer arranged as shown
in Figure 3. Like-named resistors have equal values.

The preamp incorporates a dynamic bridge (subtractor) circuit.
Identical networks (within the shaded areas), consisting of R
R

, RC, and RG, attenuate input signals applied to Pins 1 and 8.

B

,

A

Note that when equal amplitude signals are asserted at inputs 1
and 8, and the output of A1 is equal to the common potential
(i.e., zero), the two attenuators form a balanced-bridge network.
When the bridge is balanced, the differential input voltage at A1
and thus its output, will be zero.

Any common-mode voltage applied to both inputs will keep the
bridge balanced and the A1 output at zero. Because the resistor
networks are carefully matched, the common-mode signal rejec­tion approaches this ideal state.

However, if the signals applied to the inputs differ, the result is a
difference at the input to A1. A1 responds by adjusting its output
to drive R

, by way of RG, to adjust the voltage at its inverting

B

input until it matches the voltage at its noninverting input.

By attenuating voltages at Pins 1 and 8, the amplifier inputs are
held within the power supply range, even if Pin 1 and Pin 8 input
levels exceed the supply, or fall below Common (Ground.) The
input network also attenuates normal (differential) mode volt­ages. R

and RG form an attenuator that scales A1 feedback,

C

forcing large output signals to balance relatively small differen­tial inputs. The resistor ratios establish the preamp gain at ten.

Because the differential input signal is attenuated, and then
amplified to yield an overall gain of ten, the amplifier A1 oper­ates at a higher noise gain, multiplying deficiencies such as input
offset voltage and noise with respect to Pins 1 and 8.

+IN

R

R

R

R

G

–IN

R

A

A

A1

R

R

CM

R

B

B

R

R

G

C

C

CM

A3

100k

(TRIMMED)

AD8200

A2

R

F

R

F

TEK RUN: 2.5MS/s AVERAGE

1

V

, RL = 10k

OUT

MAGNIFIED V

V

3

2

CH1 1V CH 2 10mV M 20s CH1 1.36V
CH3 100mV

IN

OUT

TPC 8. Settling Time

To minimize these errors while extending the common-mode
range, a dedicated feedback loop is employed to reduce the
range of common-mode voltage applied to A1, for a given over­all range at the inputs. By offsetting the range of voltage applied
to the compensator, the input common-mode range is also offset
to include voltages more negative than the power supply. Ampli­fier A3 detects the common-mode signal applied to A1 and
adjusts the voltage on the matched R

resistors to reduce the

CM

common-mode voltage range at the A1 inputs. By adjusting the
common voltage of these resistors, the common-mode input
range is extended while, at the same time, the normal mode
signal attenuation is reduced, leading to better performance
referred to input.

The output of the dynamic bridge taken from A1 is connected
to Pin 3 by way of a 100 kΩ series resistor, provided for low-
pass filtering and gain adjustment. The resistors in the input
networks of the preamp and the buffer feedback resistors are
ratio-trimmed for high accuracy.

The output of the preamp drives a gain-of-two buffer-amplifier
A2, implemented with carefully matched feedback resistors R

.

F

The two-stage system architecture of the AD8200 enables the
user to incorporate a low-pass filter prior to the output buffer.
By separating the gain into two stages, a full-scale rail-to-rail
signal from the preamp can be filtered at Pin 3, and a half-scale
signal resulting from filtering can be restored to full scale by the
output buffer amp. The source resistance seen by the inverting
input of A2 is approximately 100 kΩ, to minimize the effects of
A2’s input bias current. However, this current is quite small and
errors resulting from applications that mismatch the resistance
are correspondingly small.

APPLICATIONS

The AD8200 difference amplifier is intended for applications
where it is required to extract a small differential signal in the
presence of large common-mode voltages. The input resistance
is nominally 200 kΩ, and the device can tolerate common-mode
voltages higher than the supply voltage and lower than ground.

The open collector output stage will source current to within
20 mV of ground.

REV. 0

COM

Figure 3. Simplified Schematic

–5–

AD8200

CURRENT SENSING
High-Line, High-Current Sensing

Basic automotive applications making use of the large common­mode range are shown in Figures 1 and 2. The capability of the
device to operate as an amplifier in primary battery supply cir­cuits is shown in Figure 1, Figure 2 illustrates the ability of the
device to withstand voltages below system ground.

Low Current Sensing

The AD8200 can also be used in low current sensing applica­tions, such as a 4–20 mA current loop shown in Figure 4. In
such applications, the relatively large shunt resistor can degrade
the common-mode rejection. Adding a resistor of equal value in
the low-impedance side of the input corrects for this error.

+IN

–IN

NC

AD8200

GND

5V

OUT

+V

S

A1

NC = NO CONNECT

OUTPUT

A2

10

1%

+

10

1%

Figure 4. 4–20 mA Current Loop Receiver

GAIN ADJUSTMENT

The default gain of the preamplifier and buffer are ×10 and ×2
respectively, resulting in a composite gain of ×20. With the
addition of external resistor(s) or trimmer(s), the gain may be
lowered, raised, or finely calibrated.

Gains Less than 20

See Figure 5. Since the preamplifier has an output resistance of
100 kΩ, an external resistor connected from Pins 3 and 4 to
GND will decrease the gain by a factor R

+V

S

NC+IN

S

V

DIFF

2

V

V

CM

DIFF

2

AD8200

100k

10k10k

EXT

OUT+V

A2A1GND–IN

/(100 kΩ + R

OUT

20R

R

EXT

= 100k

EXT

+ 100k

20 – GAIN

GAIN =

R

EXT

EXT

GAIN

).

Gains Greater than 20

Connecting a resistor from the output of the buffer amplifier to
its noninverting input, as shown in Figure 6, will increase the

/(R

gain. The gain is now multiplied by the factor R
100 kΩ); for example, it is doubled for R

EXT

= 200 kΩ. Overall

EXT

EXT

–

gains as high as 50 are achievable in this way. Note that the
accuracy of the gain becomes critically dependent on resistor
value at high gains. Also, the effective input offset voltage at
Pins 1 and 8 (about six times the actual offset of A1) limits the
part’s use in very high-gain, dc-coupled applications.

+V

S

+V

NC+IN

V

DIFF

2

AD8200

V

V

CM

DIFF

2

NC = NO CONNECT

100k

GND

OUT

S

10k10k

R

EXT

A2A1–IN

GAIN =

R

EXT

OUT

R

EXT

= 100k

20R

– 100k

GAIN – 20

EXT

GAIN

Figure 6. Adjusting for Gains Greater than 20

GAIN TRIM

Figure 7 shows a method for incremental gain trimming using
a trimpot and external resistor R

EXT

.

The following approximation is useful for small gain ranges:

∆G ≈ (10 MΩ ÷ R

Thus, the adjustment range would be ±2% for R
±10% for R

V

EXT

CM

= 1 MΩ, etc.

V

DIFF

2

V

DIFF

2

NC = NO CONNECT

+IN

–IN

NC

AD8200

GND

) %

EXT

= 5 MΩ;

EXT

5V

OUT

+V

S

A1

A2

R

EXT

OUT

GAIN TRIM
20k MIN

R

EXT

NC = NO CONNECT

Figure 5. Adjusting for Gains Less than 20

The overall bandwidth is unaffected by changes in gain using
this method, although there may be a small offset voltage due to
the imbalance in source resistances at the input to the buffer. In
many cases this can be ignored, but if desired, can be nulled by
inserting a resistor equal to 100 kΩ minus the parallel sum of

and 100 kΩ, in series with Pin 4. For example, with R

R

EXT

EXT

= 100 kΩ (yielding a composite gain of ×10), the optional offset
nulling resistor is 50 kΩ (see Figure 11.)

–6–

Figure 7. Incremental Gain Trim

REV. 0

AD8200

40LOG (f2/f1)

f

1

ATTENUATION

f

2

f

2

2

/f

1

FREQUENCY

A 1-POLE FILTER, CORNER f1, AND
A 2-POLE FILTER, CORNER f

2

, HAVE

THE SAME ATTENUATION –40LOG (f

2/f1

)

AT FREQUENCY f

2

2

/f

1

20dB/DECADE

40dB/DECADE

Internal Signal Overload Considerations

When configuring gain for values other than 20, the maximum
input voltage with respect to the supply voltage and ground
must be considered, since either the preamplifier or the output
buffer will reach its full-scale output (approximately V

– 0.2 V)

S

with large differential input voltages. The input of the AD8200
is limited to (V

– 0.2) ÷ 10, for overall gains ≤10, since the

S

preamplifier, with its fixed gain of ×10, reaches its full-scale
output before the output buffer. For gains greater than 10, the
swing at the buffer output reaches its full-scale first and limits
the AD8200 input to (VS – 0.2) ÷ G, where G is the overall gain.

LOW-PASS FILTERING

In many transducer applications it is necessary to filter the sig­nal to remove spurious high-frequency components, including
noise, or to extract the mean value of a fluctuating signal with a
peak-to-average ratio (PAR) greater than unity. For example, a
full-wave rectified sinusoid has a PAR of 1.57, a raised cosine
has a PAR of 2, and a half-wave sinusoid has a PAR of 3.14.
Signals having large spikes may have PARs of 10 or more.

When implementing a filter, the PAR should be considered so
the output of the AD8200 preamplifier (A1) does not clip before
A2, since this nonlinearity would be averaged and appear as an
error at the output. To avoid this error, both amplifiers should
be made to clip at the same time. This condition is achieved
when the PAR, is no greater than the gain of the second ampli­fier (2 for the default configuration). For example, if a PAR of 5
is expected, the gain of A2 should be increased to 5.

Low-pass filters can be implemented in several ways using the
features provided by the AD8200. In the simplest case, a single­pole filter (20 dB/decade) is formed when the output of A1 is
connected to the input of A2 via the internal 100 kΩ resistor by
strapping Pins 3 and 4, and a capacitor added from this node to
ground, as shown in Figure 8. If a resistor is added across the
capacitor to lower the gain, the corner frequency will increase; it
should be calculated using the parallel sum of the resistor and
100 kΩ.

5V

OUT

5V

V

DIFF

2

V

V

CM

DIFF

2

NC = NO CONNECT

+IN

AD8200

–IN

NC

GND

OUT

+V

S

A2

A1

255k

C

F

= 1Hz – F

C

OUT

C

Figure 9. 2-Pole Low-Pass Filter

A 2-pole filter (with a roll-off of 40 dB/decade) can be imple­mented using the connections shown in Figure 9. This is a
Sallen-Key form based on a ×2 amplifier. It is useful to remem­ber that a 2-pole filter with a corner frequency f
filter with a corner at f
frequency (f
(f

). This is illustrated in Figure 10. Using the standard resis-

2/f1

2

/f1). The attenuation at that frequency is 40 Log

2

have the same attenuation at the

1

and a 1-pole

2

tor value shown, and equal capacitors (Figure 9), the corner
frequency is conveniently scaled at 1 Hz-µF (0.05 µF for a 20 Hz
corner). A maximally flat response occurs when the resistor is
lowered to 196 kΩ and the scaling is then 1.145 Hz-µF. The
output offset is raised by about 5 mV (equivalent to 250 ␮V at
the input pins).

V

V

CM

DIFF

2

V

DIFF

2

NC = NO CONNECT

+IN

–IN

NC

+V

AD8200

GND

A1

S

OUT

A2

F

C IN FARADS

C

1

=

C

2C10

5

Figure 8. A Single-Pole, Low-Pass Filter Using the Internal

Ω

Resistor

100 k

If the gain is raised using a resistor, as shown in Figure 8, the
corner frequency is lowered by the same factor as the gain is
raised. Thus, using a resistor of 200 kΩ (for which the gain
would be doubled) the corner frequency is now 0.796 Hz-µF,
(0.039 µF for a 20 Hz corner frequency.)

REV. 0

Figure 10. Comparative Responses of 1- and 2-Pole Low­Pass Filters

–7–

AD8200

HIGH LINE CURRENT SENSING WITH LPF AND GAIN
ADJUSTMENT

Figure 11 is another refinement of Figure 1, including gain
adjustment and low-pass filtering.

BATTERY

CLAMP

DIODE

14V

NC = NO CONNECT

4 TERM

SHUNT

POWER

DEVICE

INDUCTIVE
LOAD

COMMON

+IN

–IN

5V

+V

NC

OUT

S

AD8200

A2

GND

A1

V

OS/IB

NULL

C

5% CALIBRATION RANGE

= 0.796Hz – F

F

C

(0.22F FOR f = 3.6 Hz)

OUTPUT
4V/AMP

191k

20k

Figure 11. High-Line Current Sensor Interface. Gain = ×40,
Single-Pole, Low-Pass Filter

A power device that is either ‘ON’ or ‘OFF’ controls the current
in the load. The average current is proportional to the duty cycle
of the input pulse, and is sensed by a small value resistor. The
average differential voltage across the shunt is typically 100 mV,
although its peak value will be higher by an amount that depends
on the inductance of the load and the control frequency. The
common-mode voltage, on the other hand, extends from roughly
1 V above ground, when the switch is ‘ON,’ to about 1.5 V
above the battery voltage, when the device is ‘OFF,’ and the
clamp diode conducts. If the maximum battery voltage spikes
up to 20 V, the common-mode voltage at the input can be as
high as 21.5 V.

To produce a full-scale output of 4 V, a gain ×40 is used, adjust­able by ±5% to absorb the tolerance in the shunt. There is
sufficient headroom to allow 10% overrange (to 4.4 V). The
roughly triangular voltage across the sense resistor is averaged
by a single-pole, low-pass filter, here set with a corner frequency
= 3.6 Hz, which provides about 30 dB of attenuation at 100 Hz.
A higher rate of attenuation can be obtained using a two-pole
filter having f

= 20 Hz, as shown in Figure 12. Although this

C

circuit uses two separate capacitors, the total capacitance is less
than half that needed for the single-pole filter.

DRIVING CHARGE REDISTRIBUTION A/D
CONVERTERS

When driving CMOS ADCs, such as those embedded in popular
microcontrollers, the charge injection (⌬Q) can cause a signifi­cant deflection in the output voltage of the AD8200. Though
generally of short duration, this deflection may persist until after
the sample period of the ADC has expired, due to the relatively
high open-loop output impedance of the AD8200. Including an
R-C network in the output can significantly reduce the effect.
The capacitor helps to absorb the transient charge, effectively
lowering the high-frequency output impedance of the AD8200.
For these applications, the output signal should be taken from the
midpoint of the R

LAG–CLAG

combination as shown in Figure 13.

Since the perturbations from the analog-to-digital converter are
small, the output impedance of the AD8200 will appear to be
low. The transient response will, therefore, have a time constant
governed by the product of the two LAG components, C
R

. For the values shown in Figure 13, this time constant is

LAG

LAG

×

programmed at approximately 10 µs. Therefore, if samples are
taken at several tens of microseconds or more, there will be
negligible charge “stack-up.”

5V

+IN

–IN

AD8200

A2

10k

10k

R

1k

LAG

C

LAG

0.01F

PROCESSOR

A/D

Figure 13. Recommended Circuit for Driving CMOS A/D

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

C02054–4.5–10/00 (rev. 0)

BATTERY

CLAMP

DIODE

14V

NC = NO CONNECT

4 TERM

SHUNT

POWER

DEVICE

INDUCTIVE
LOAD

COMMON

+IN

–IN

5V

+V

NC

AD8200

A1

GND

OUT

S

A2

127k

C

= 1Hz – F

F

C

(0.05F FOR f

C

Figure 12. Illustration of 2-Pole Low-Pass Filtering

432k

50k

= 20Hz)

C

OUTPUT

–8–

0.1574 (4.00)

0.1497 (3.80)

PIN 1

0.0098 (0.25)

0.0040 (0.10)

SEATING

0.1968 (5.00)

0.1890 (4.80)

85

0.0500 (1.27)
BSC

0.0192 (0.49)

0.0138 (0.35)

PLANE

8-Lead SOIC Package

(SO-8)

0.2440 (6.20)

0.2284 (5.80)

41

0.0688 (1.75)

0.0532 (1.35)

0.0098 (0.25)

0.0075 (0.19)

0.0196 (0.50)

0.0099 (0.25)

8

0.0500 (1.27)

0

0.0160 (0.41)

PRINTED IN U.S.A.

 45

REV. 0