Datasheet AD811AR-16-REEL7, AD811AR-16-REEL, AD811AR-16, AD811AN, AD811ACHIPS Datasheet (Analog Devices)

High Performance

1
2
3
4

8
7
6
5

AD811

32

12019

18
17
16
15
14

9

10 11

12 13

4
5
6
7
8

AD811

NC
NC

+V

S

NC
OUTPUT

–V

S

NC

NC
NC

NC

–IN

+IN

NC

NC

NC

NC

NC

NC
–IN
+IN

–V

S

NC

OUTPUT
NC

+V

S

NC = NO CONNECT

NC = NO CONNECT

NCNCNC

a

FEATURES
High Speed

140 MHz Bandwidth (3 dB, G = +1)
120 MHz Bandwidth (3 dB, G = +2)
35 MHz Bandwidth (0.1 dB, G = +2)
2500 V/␮s Slew Rate
25 ns Settling Time to 0.1% (For a 2 V Step)
65 ns Settling Time to 0.01% (For a 10 V Step)

Excellent Video Performance (R

0.01% Differential Gain, 0.01ⴗ Differential Phase

Voltage Noise of 1.9 nV√Hz

Low Distortion: THD = –74 dB @ 10 MHz
Excellent DC Precision

3 mV max Input Offset Voltage

Flexible Operation

Specified for ⴞ5 V and ⴞ15 V Operation
ⴞ2.3 V Output Swing into a 75 ⍀ Load (V

APPLICATIONS
Video Crosspoint Switchers, Multimedia Broadcast

Systems
HDTV Compatible Systems
Video Line Drivers, Distribution Amplifiers
ADC/DAC Buffers
DC Restoration Circuits
Medical—Ultrasound, PET, Gamma and Counter

Applications

PRODUCT DESCRIPTION

The AD811 is a wideband current-feedback operational ampli­fier, optimized for broadcast quality video systems. The –3 dB
bandwidth of 120 MHz at a gain of +2 and differential gain and

phase of 0.01% and 0.01° (R

= 150 Ω) make the AD811 an

L

excellent choice for all video systems. The AD811 is designed to
meet a stringent 0.1 dB gain flatness specification to a band­width of 35 MHz (G = +2) in addition to the low differential
gain and phase errors. This performance is achieved whether

driving one or two back terminated 75 Ω cables, with a low

power supply current of 16.5 mA. Furthermore, the AD811 is

specified over a power supply range of ±4.5 V to ±18 V.

0.10

0.09

0.08

%

0.07

0.06

0.05

0.04

0.03

DIFFERENTIAL GAIN –

0.02

0.01

REV. D

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

GAIN

6

5

PHASE

SUPPLY VOLTAGE –

=150 ⍀)

L

RF = 649V

= 3.58MHz

F

C

100 IRE
MODULATED RAMP
R

= 150V

L

6

Volts

= ⴞ5 V)

S

0.20

0.18

0.16

0.14

0.12

0.10

0.08

0.06

0.04

DIFFERENTIAL PHASE – Degrees

0.02

15

1413121110987

Video Op Amp

AD811

CONNECTION DIAGRAMS

8-Lead Plastic (N-8)

Cerdip (Q-8)

SOIC (SO-8) Packages

16-Lead SOIC (R-16) Package 20-Lead SOIC (R-20) Package

1

NC
NC

2
3

–IN
NC

4

+IN

5

NC

6
7

–V

S

AD811

8

NC

NC = NO CONNECT

The AD811 is also excellent for pulsed applications where tran­sient response is critical. It can achieve a maximum slew rate of

greater than 2500 V/µs with a settling time of less than 25 ns to

0.1% on a 2 volt step and 65 ns to 0.01% on a 10 volt step.

The AD811 is ideal as an ADC or DAC buffer in data acquisi­tion systems due to its low distortion up to 10 MHz and its wide
unity gain bandwidth. Because the AD811 is a current feedback
amplifier, this bandwidth can be maintained over a wide range
of gains. The AD811 also offers low voltage and current noise of

1.9 nV/√Hz and 20 pA/√Hz, respectively, and excellent dc accu-

racy for wide dynamic range applications.

12

G = +2
R

9

R

6

3

GAIN – dB

0

–3

–6

1M

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 1999

20-Lead LCC (E-20A) Package

1

16
15
14
13
12
11
10

9

= 150V

L

= R

G

NC
NC
+V

S

NC
OUTPUT

NC
NC
NC

FB

NC

2

NC

3

NC

4

–IN

NC

5

+IN

6
7

NC

–V

8

S

9

NC

10

NC

NC = NO CONNECT

VS = 615V

VS = 65V

10M 100M

FREQUENCY – Hz

AD811

20
19
18

17

16
15

14
13
12
11

NC
NC
NC
+V

S

NC
OUTPUT
NC
NC
NC

NC

AD811–SPECIFICATIONS

(@ TA = +25ⴗC and VS = ⴞ15 V dc, R

Model Conditions V

S

Min Typ Max Min Typ Max Units

= 150 Ω unless otherwise noted)

LOAD

AD811J/A

1

AD811S

2

DYNAMIC PERFORMANCE

Small Signal Bandwidth (No Peaking)

–3 dB

G = +1 R
G = +2 R
G = +2 R
G = +10 R

0.1 dB Flat
G = +2 R

Full Power Bandwidth

3

Slew Rate V

Settling Time to 0.1% 10 V Step, A
Settling Time to 0.01% 65 65 ns
Settling Time to 0.1% 2 V Step, A
Rise Time, Fall Time R
Differential Gain f = 3.58 MHz ±15 V 0.01 0.01 %

= 562 Ω±15 V 140 140 MHz

FB

= 649 Ω±15 V 120 120 MHz

FB

= 562 Ω±5 V 80 80 MHz

FB

= 511 Ω±15 V 100 100 MHz

FB

= 562 Ω±5 V 25 25 MHz

FB

= 649 Ω±15 V 35 35 MHz

R

FB

V

= 20 V p-p ±15 V 40 40 MHz

OUT

= 4 V p-p ±5 V 400 400 V/µs

OUT

= 20 V p-p ±15 V 2500 2500 V/µs

V

OUT

= 649, A

FB

= –1 ±15 V 50 50 ns

V

= –1 ±5 V 25 25 ns

V

= +2 ±15 V 3.5 3.5 ns

V

Differential Phase f = 3.58 MHz ±15 V 0.01 0.01 Degree
THD @ f
Third Order Intercept

= 10 MHz V

C

4

= 2 V p-p, A

OUT

@ f

= 10 MHz ±5 V 36 36 dBm

C

= +2 ± 15 V –74 –74 dBc

V

±15 V 43 43 dBm

INPUT OFFSET VOLTAGE ±5 V, ±15 V 0.5 3 0.5 3 mV

to T

T

MIN

Offset Voltage Drift 55µV/°C

MAX

55mV

INPUT BIAS CURRENT

–Input ±5 V, ±15 V 2 5 2 5 µA

to T

T

MIN

+Input ±5 V, ±15 V 2 10 2 10 µA

T

TRANSRESISTANCE T

MAX

to T

MIN

MAX

to T

MIN

MAX

V

= ±10 V

OUT

= ∞±15 V 0.75 1.5 0.75 1.5 MΩ

R

L

= 200 Ω±15 V 0.5 0.75 0.5 0.75 MΩ

R

L

= ±2.5 V

V

OUT

R

= 150 Ω±5 V 0.25 0.4 0.125 0.4 MΩ

L

15 30 µA
20 25 µA

COMMON-MODE REJECTION

(vs. Common Mode)

V

OS

to T

T

MIN

MAX

to T

T

MIN

Input Current (vs. Common Mode) T

MAX

POWER SUPPLY REJECTION V

V

OS

+Input Current T
–Input Current T

V

= ±2.5 ±5 V 56 60 50 60 dB

CM

V

= ±10 V ±15 V 60 66 56 66 dB

CM

to T

MIN

MAX

= ±4.5 V to ±18 V

S

T

to T

MIN

MAX

to T

MIN

MAX

to T

MIN

MAX

60 70 60 70 dB

13 1 3µA/V

0.3 2 0.3 2 µA/V

0.4 2 0.4 2 µA/V
INPUT VOLTAGE NOISE f = 1 kHz 1.9 1.9 nV/√Hz
INPUT CURRENT NOISE f = 1 kHz 20 20 pA/√Hz

OUTPUT CHARACTERISTICS

Voltage Swing, Useful Operating Range

5

±5 V ±2.9 ±2.9 V
±15 V ±12 ±12 V

Output Current T
Short-Circuit Current 150 150 mA

= +25°C 100 100 mA

J

Output Resistance (Open Loop @ 5 MHz) 9 9 Ω

INPUT CHARACTERISTICS

+Input Resistance 1.5 1.5 MΩ
–Input Resistance 14 14 Ω
Input Capacitance +Input 7.5 7.5 pF
Common-Mode Voltage Range ±5 V ±3 ±3V

±15 V ±13 ±13 V

POWER SUPPLY

Operating Range ±4.5 ±18 ±4.5 ±18 V
Quiescent Current ±5 V 14.5 16.0 14.5 16.0 mA

±15 V 16.5 18.0 16.5 18.0 mA

TRANSISTOR COUNT # of Transistors 40 40

NOTES

1

The AD811JR is specified with ±5 V power supplies only, with operation up to ±12 volts.

2

See Analog Devices’ military data sheet for 883B tested specifications.

3

FPBW = slew rate/(2 π V

4

Output power level, tested at a closed loop gain of two.

5

Useful operating range is defined as the output voltage at which linearity begins to degrade.

Specifications subject to change without notice.

PEAK

).

–2–

REV. D

AD811

WARNING!

ESD SENSITIVE DEVICE

ABSOLUTE MAXIMUM RATINGS

1

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±18 V

AD811JR Grade Only . . . . . . . . . . . . . . . . . . . . . . . . .±12 V

Internal Power Dissipation

2

. . . . . . . . Observe Derating Curves

Output Short Circuit Duration . . . . . Observe Derating Curves

Common-Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . ±V

S

Differential Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . .±6 V

Storage Temperature Range (Q, E) . . . . . . . . –65°C to +150°C

Storage Temperature Range (N, R) . . . . . . . . –65°C to +125°C

Operating Temperature Range

AD811J . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .0°C to +70°C

AD811A . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C

AD811S . . . . . . . . . . . . . . . . . . . . . . . . . . . –55°C to +125°C

Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300°C

NOTES

1

Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.

2

8-Lead Plastic Package: θJA = 90°C/W

8-Lead Cerdip Package: θJA = 110°C/W
8-Lead SOIC Package: θJA = 155°C/W
16-Lead SOIC Package: θJA = 85°C/W
20-Lead SOIC Package: θJA = 80°C/W
20-Lead LCC Package: θJA = 70°C/W

ORDERING GUIDE

Temperature Package

Model Range Option*

AD811AN –40°C to +85°C N-8
AD811AR-16 –40°C to +85°C R-16
AD811AR-20 –40°C to +85°C R-20
AD811JR 0°C to +70°C SO-8
AD811SQ/883B –55°C to +125°C Q-8
5962-9313101MPA –55°C to +125°C Q-8
AD811SE/883B –55°C to +125°C E-20A
5962-9313101M2A –55°C to +125°C E-20A
AD811JR-REEL 0°C to +70°C SO-8
AD811JR-REEL7 0°C to +70°C SO-8
AD811AR-16-REEL –40°C to +85°C R-16
AD811AR-16-REEL7 –40°C to +85°C R-16
AD811AR-20-REEL –40°C to +85°C R-20
AD811ACHIPS –40°C to +85°CDie
AD811SCHIPS –55°C to +125°CDie

*E = Ceramic Leadless Chip Carrier; N = Plastic DIP; Q = Cerdip; SO (R) =

Small Outline IC (SOIC).

MAXIMUM POWER DISSIPATION

The maximum power that can be safely dissipated by the
AD811 is limited by the associated rise in junction temperature.
For the plastic packages, the maximum safe junction tempera-

ture is +145°C. For the cerdip and LCC packages, the maxi­mum junction temperature is +175°C. If these maximums are

exceeded momentarily, proper circuit operation will be restored
as soon as the die temperature is reduced. Leaving the device in
the “overheated” condition for an extended period can result in
device burnout. To ensure proper operation, it is important to
observe the derating curves in Figures 17 and 18.

While the AD811 is internally short circuit protected, this may
not be sufficient to guarantee that the maximum junction tem­perature is not exceeded under all conditions. One important
example is when the amplifier is driving a reverse terminated

75 Ω cable and the cable’s far end is shorted to a power supply.
With power supplies of ±12 volts (or less) at an ambient tem­perature of +25°C or less, if the cable is shorted to a supply rail,

then the amplifier will not be destroyed, even if this condition
persists for an extended period.

ESD SUSCEPTIBILITY

ESD (electrostatic discharge) sensitive device. Electrostatic
charges as high as 4000 volts, which readily accumulate on the
human body and on test equipment, can discharge without
detection. Although the AD811 features proprietary ESD pro­tection circuitry, permanent damage may still occur on these
devices if they are subjected to high energy electrostatic dis­charges. Therefore, proper ESD precautions are recommended
to avoid any performance degradation or loss of functionality.

METALIZATION PHOTOGRAPH

Contact Factory for Latest Dimensions.

Dimensions Shown in Inches and (mm).

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD811 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

REV. D

–3–

AD811–Typical Performance Characteristics

20

TA = +258C

15

10

5

COMMON-MODE VOLTAGE RANGE – 6Volts

0

020

5

SUPPLY VOLTAGE – 6Volts

10

15

Figure 1. Input Common-Mode Voltage Range vs. Supply

35

30

VS = 615V

25

20

15

10

OUTPUT VOLTAGE – Volts p–p

5

VS = 65V

20

TA = +258C

15

10

5

MAGNITUDE OF THE OUTPUT VOLTAGE – 6 Volts

0

020

NO LOAD

= 150V

R

L

5

SUPPLY VOLTAGE – 6 Volts

10

15

Figure 4. Output Voltage Swing vs. Supply

21

18

15

12

9

6

QUIESCENT SUPPLY CURRENT – mA

VS = 615V

VS = 65V

0

10

100
LOAD RESISTANCE – V

1k

10k

Figure 2. Output Voltage Swing vs. Resistive Load

10

NONINVERTING INPUT

65 TO 615V

0

INVERTING

–10

–20

INPUT BIAS CURRENT – mA

–30

–40

–60 140

JUNCTION TEMPERATURE – 8C

INPUT

VS = 615V

VS = 65V

100 120806040200–20

Figure 3. Input Bias Current vs. Junction Temperature

3
–60

–40

200–20

JUNCTION TEMPERATURE – 8C

60

40 100

80

120

140

Figure 5. Quiescent Supply Current vs. Junction
Temperature

10

8

6

4
2

0

–2
–4

INPUT OFFSET VOLTAGE – mV

–6

–8

–10

–40

–60

JUNCTION TEMPERATURE – 8C

VS = 65V

VS = 615V

120100806040200–20

140

Figure 6. Input Offset Voltage vs. Junction Temperature

REV. D–4–

AD811

200

0

1.6k

120

40

600

80

400

160

1.4k1.2k1.0k

800

VALUE OF FEEDBACK RESISTOR (RFB) – V

–3dB BANDWIDTH – MHz

PEAKING – dB

8

6

4

2

BANDWIDTH

PEAKING

VO = 1V p–p
V

S

= 615V

R

L

= 150V

GAIN = +2

10

0

250

200

VS = 615V

150

100

SHORT CIRCUIT CURRENT – mA

50

–60

–40

VS = 65V

JUNCTION TEMPERATURE – 8C

100 120806040200–20

140

Figure 7. Short Circuit Current vs. Junction Temperature

10

VS = 65V

1

2.0

1.5

1.0

0.5

TRANSRESISTANCE – MV

0

–60 140

–40

JUNCTION TEMPERATURE – 8C

VS = 65V
R

= 150V

L

= 62.5V

V

OUT

VS = 615V

= 200V

R

L

V

= 610V

OUT

100 120806040200–20

Figure 10. Transresistance vs. Junction Temperature

100

NONINVERTING CURRENT VS = 65 TO 15V

10

INVERTING CURRENT VS = 65 TO 15V

100

10

0.1

CLOSED-LOOP OUTPUT RESISTANCE – V

0.01
10k 100M

100k 10M1M

FREQUENCY – Hz

VS = 615V

GAIN = +2
R

= 649V

FB

Figure 8. Closed-Loop Output Resistance vs. Frequency

10

RISE TIME

8

6

OVERSHOOT

4

RISETIME – ns

2

0
400

Figure 9. Rise Time and Overshoot vs. Value of
Feedback Resistor, R

600

800

VALUE OF FEEDBACK RESISTOR (RFB) – V

FB

VS = 615V
V
R
GAIN = +2

= 1V p–p

O

= 150V

L

1.4k1.2k1.0k

60

40

20

OVERSHOOT – %

0

1.6k

NOISE VOLTAGE – nV/ Hz

1

10

100 100k10k1k

VOLTAGE NOISE VS = 615V

VOLTAGE NOISE VS = 65V

FREQUENCY – Hz

NOISE CURRENT – pA/ Hz

1

Figure 11. Input Noise vs. Frequency

Figure 12. 3 dB Bandwidth and Peaking vs. Value of R

FB

REV. D

–5–

AD811

110

100

90

80

70

CMRR – dB

60

50

40

30

1k

V

IN

649V

649V

150V

150V

FREQUENCY – Hz

V

OUT

VS = 65V

1M100k10k

VS = 615V

10M

Figure 13. Common-Mode Rejection vs. Frequency

80

70

60

50

40

PSRR – dB

30

20

10

5

1k

VS = 615V

VS = 65V

CURVES ARE FOR WORST
CASE CONDITION WHERE
ONE SUPPLY IS VARIED
WHILE THE OTHER IS
HELD CONSTANT.

FREQUENCY – Hz

RF = 649V
A

= +2

V

1M100k10k

10M

Figure 14. Power Supply Rejection vs. Frequency

25

20

15

10

OUTPUT VOLTAGE – Volts p–p

5

GAIN = +10
OUTPUT LEVEL FOR 3% THD

0
100k 1M 100M10M

VS = 615V

VS = 65V

FREQUENCY – Hz

Figure 16. Large Signal Frequency Response

–50

V

= 2V p–p

OUT

R

= 100V

L

GAIN = +2

–70

–90

2ND HARMONIC

3RD HARMONIC

–110

HARMONIC DISTORTION – dBc

2ND HARMONIC

–130

1k 10M

10k

65V SUPPLIES

615V SUPPLIES

3RD HARMONIC

100k 1M

FREQUENCY – Hz

Figure 17. Harmonic Distortion vs. Frequency

2.5

16-LEAD SOIC

2.0

1.5

8-LEAD MINI-DIP

1.0

TOTAL POWER DISSIPATION – Watts

0.5
–50 80

–40

8-LEAD SOIC

AMBIENT TEMPERATURE – 8C

TJ MAX = +1458C

20-LEAD SOIC

60 70503020100–10–20–30 40

90

Figure 15. Maximum Power Dissipation vs. Temperature
for Plastic Packages

3.4

3.2

3.0

2.8

2.6

2.4

2.2

2.0

1.8

1.6

1.4

8-LEAD CERDIP

1.2

1.0

0.8

TOTAL POWER DISSIPATION – Watts

0.6

0.4
–40

–60

20-LEAD LCC

AMBIENT TEMPERATURE – 8C

TJ MAX = +1758C

100 120806040200–20

Figure 18. Maximum Power Dissipation vs. Temperature
for Hermetic Packages

–6–

140

REV. D

Typical Characteristics, Noninverting Connection–

R

FB

+V

S

V

TO

OUT

0.1mF

R

G

0.1mF

AD811

+

–V

S

V

IN

HP8130
PULSE
GENERATOR

50V

Figure 19. Noninverting Amplifier Connection

TEKTRONIX
P6201 FET
PROBE

R

L

AD811

9

G = +1
R

= 150V

6

L

R

=

G

3

0

–3

GAIN – dB

–6

–9

–12

1M

FREQUENCY – Hz

Figure 22. Closed-Loop Gain vs. Frequency, Gain = +1

26

VS = 615V
R

= 750V

FB

VS = 65V
R

= 619V

FB

10M 100M

1V

100

90

V

IN

10

V

OUT

0%

10ns

1V

Figure 20. Small Signal Pulse Response, Gain = +1

100mV

100

90

V

IN

10

V

OUT

0%

10ns

23

20

17

GAIN – dB

14

11

8

1M

G = +10
R = 150V

L

10M 100M

FREQUENCY – Hz

V = 615V

S

R = 511V

FB

V = 65V

S

R = 442V

FB

Figure 23. Closed-Loop Gain vs. Frequency, Gain = +10

1V

100

V

90

IN

10

V

OUT

0%

20ns

1V

Figure 21. Small Signal Pulse Response, Gain = +10

REV. D

10V

Figure 24. Large Signal Pulse Response, Gain = +10

–7–

AD811

–Typical Characteristics, Inverting Connection

R

FB

+V

S

V

IN

HP8130
PULSE
GENERATOR

0.1mF

R

G

AD811

0.1mF

–V

S

V

TO

OUT

TEKTRONIX
P6201 FET
PROBE

R

L

Figure 25. Inverting Amplifier Connection

1V

100

90

V

IN

10

V

OUT

0%

10ns

1V

Figure 26. Small Signal Pulse Response, Gain = –1

6

3

0

–3

GAIN – dB

–6

–9

–12

1M

G = –1
R

= 150V

L

10M 100M

FREQUENCY – Hz

VS = 615V
R

V = 65V

S

R = 562V

FB

FB

= 590V

Figure 28. Closed-Loop Gain vs. Frequency, Gain = –1

26

23

20

17

GAIN – dB

14

11

8

1M

G = –10
R

= 150V

L

10M 100M

FREQUENCY – Hz

VS = 65V
R

= 442V

FB

VS = 615V
R

= 511V

FB

Figure 29. Closed-Loop Gain vs. Frequency, Gain = –10

100mV

100

90

V

IN

10

V

OUT

0%

10ns

1V

Figure 27. Small Signal Pulse Response, Gain = –10

1V

100

90

V

IN

10

V

OUT

0%

10V

Figure 30. Large Signal Pulse Response, Gain = –10

–8–

20ns

REV. D

AD811

APPLICATIONS
General Design Considerations

The AD811 is a current feedback amplifier optimized for use in
high performance video and data acquisition applications. Since
it uses a current feedback architecture, its closed-loop –3 dB
bandwidth is dependent on the magnitude of the feedback resis­tor. The desired closed-loop gain and bandwidth are obtained
by varying the feedback resistor (R
and varying the gain resistor (R

) to tune the bandwidth,

FB

) to get the correct gain. Table I

G

contains recommended resistor values for a variety of useful
closed-loop gains and supply voltages.

Table I. –3 dB Bandwidth vs. Closed-Loop Gain and
Resistance Values

VS = ⴞ15 V
Closed-Loop –3 dB BW
Gain R

FB

R

G

(MHz)

+1 750 Ω 140
+2 649 Ω 649 Ω 120
+10 511 Ω 56.2 Ω 100
–1 590 Ω 590 Ω 115
–10 511 Ω 51.1 Ω 95

VS = ⴞ5 V
Closed-Loop –3 dB BW
Gain R

FB

R

G

(MHz)

+1 619 Ω 80
+2 562 Ω 562 Ω 80
+10 442 Ω 48.7 Ω 65
–1 562 Ω 562 Ω 75
–10 442 Ω 44.2 Ω 65

VS = ⴞ10 V
Closed-Loop –3 dB BW
Gain R

FB

R

G

(MHz)

+1 649 Ω 105
+2 590 Ω 590 Ω 105
+10 499 Ω 49.9 Ω 80
–1 590 Ω 590 Ω 105
–10 499 Ω 49.9 Ω 80

Figures 11 and 12 illustrate the relationship between the feed­back resistor and the frequency and time domain response char­acteristics for a closed-loop gain of +2. (The response at other
gains will be similar.)

The 3 dB bandwidth is somewhat dependent on the power supply
voltage. As the supply voltage is decreased for example, the
magnitude of internal junction capacitances is increased, causing
a reduction in closed-loop bandwidth. To compensate for this,
smaller values of feedback resistor are used at lower supply
voltages.

Achieving the Flattest Gain Response at High Frequency

Achieving and maintaining gain flatness of better than 0.1 dB at
frequencies above 10 MHz requires careful consideration of
several issues.

Choice of Feedback and Gain Resistors

Because of the above-mentioned relationship between the 3 dB
bandwidth and the feedback resistor, the fine scale gain flatness
will, to some extent, vary with feedback resistor tolerance. It is,
therefore, recommended that resistors with a 1% tolerance be
used if it is desired to maintain flatness over a wide range of
production lots. In addition, resistors of different construction
have different associated parasitic capacitance and inductance.
Metal-film resistors were used for the bulk of the characteriza­tion for this data sheet. It is possible that values other than those
indicated will be optimal for other resistor types.

Printed Circuit Board Layout Considerations

As to be expected for a wideband amplifier, PC board parasitics
can affect the overall closed loop performance. Of concern are
stray capacitances at the output and the inverting input nodes. If
a ground plane is to be used on the same side of the board as
the signal traces, a space (3/16" is plenty) should be left around
the signal lines to minimize coupling. Additionally, signal lines
connecting the feedback and gain resistors should be short
enough so that their associated inductance does not cause
high frequency gain errors. Line lengths less than 1/4" are
recommended.

Quality of Coaxial Cable

Optimum flatness when driving a coax cable is possible only
when the driven cable is terminated at each end with a resistor
matching its characteristic impedance. If the coax was ideal,
then the resulting flatness would not be affected by the length of
the cable. While outstanding results can be achieved using inex­pensive cables, it should be noted that some variation in flatness
due to varying cable lengths may be experienced.

Power Supply Bypassing

Adequate power supply bypassing can be critical when optimiz­ing the performance of a high frequency circuit. Inductance in
the power supply leads can form resonant circuits that produce
peaking in the amplifier’s response. In addition, if large current
transients must be delivered to the load, then bypass capacitors

(typically greater than 1 µF) will be required to provide the best

settling time and lowest distortion. Although the recommended

0.1 µF power supply bypass capacitors will be sufficient in many

applications, more elaborate bypassing (such as using two paral­leled capacitors) may be required in some cases.

REV. D

–9–

AD811

p

Driving Capacitive Loads

The feedback and gain resistor values in Table I will result in
very flat closed-loop responses in applications where the load
capacitances are below 10 pF. Capacitances greater than this
will result in increased peaking and overshoot, although not
necessarily in a sustained oscillation.

There are at least two very effective ways to compensate for this
effect. One way is to increase the magnitude of the feedback
resistor, which lowers the 3 dB frequency. The other method is
to include a small resistor in series with the output of the ampli­fier to isolate it from the load capacitance. The results of these

two techniques are illustrated in Figure 32. Using a 1.5 kΩ

feedback resistor, the output ripple is less than 0.5 dB when driv­ing 100 pF. The main disadvantage of this method is that it
sacrifices a little bit of gain flatness for increased capacitive load
drive capability. With the second method, using a series resistor,
the loss of flatness does not occur.

R

FB

+V

S

0.1mF

R

G

RS (OPTIONAL)

C

L

0.1mF

AD811

–V

S

V

IN

R

T

V

OUT

R

L

Figure 31. Recommended Connection for Driving a Large
Capacitive Load

12

100

90

80

70

– V

60

S

50

40

VALUE OF R

30

20

10

0

10

LOAD CAPACITANCE –

G = +2
V = 615V

S

R VALUE SPECIFIED

S

IS FOR FLATTEST
FREQUENCY RESPONSE

100

F

1000

Figure 33. Recommended Value of Series Resistor vs. the
Amount of Capacitive Load

Figure 33 shows recommended resistor values for different load
capacitances. Refer again to Figure 32 for an example of the
results of this method. Note that it may be necessary to adjust
the gain setting resistor, R
results due to the divider formed by the series resistor, R

, to correct for the attenuation which

G

, and

S

the load resistance.

Applications which require driving a large load capacitance at a
high slew rate are often limited by the output current available
from the driving amplifier. For example, an amplifier limited to
25 mA output current cannot drive a 500 pF load at a slew rate

greater than 50 V/µs. However, because of the AD811’s 100 mA
output current, a slew rate of 200 V/µs is achievable when driv-

ing this same 500 pF capacitor (see Figure 34).

2V

100

V

90

IN

100ns

R

= 1.5kV

9

6

3

GAIN – dB

0

–3

–6

1M

G = +2

= 615V

V

S

= 10kV

R

L

= 100pF

C

L

FB

= 0

R

S

= 649V

R

FB

= 30V

R

S

10M 100M

FREQUENCY – Hz

Figure 32. Performance Comparison of Two Methods for
Driving a Capacitive Load

10

V

OUT

0%

5V

Figure 34. Output Waveform of an AD811 Driving a
500 pF Load. Gain = +2, R

= 10 k

R

S

Ω

= 649 Ω, RS = 15 Ω,

FB

–10–

REV. D

Operation as a Video Line Driver

The AD811 has been designed to offer outstanding perfor­mance at closed-loop gains of one or greater, while driving
multiple reverse-terminated video loads. The lowest differential

gain and phase errors will be obtained when using ±15 volt
power supplies. With ±12 volt supplies, there will be an insig-

nificant increase in these errors and a slight improvement in

gain flatness. Due to power dissipation considerations, ±12 volt

supplies are recommended for optimum video performance.
Excellent performance can be achieved at much lower supplies
as well.

The closed-loop gain vs. frequency at different supply voltages
is shown in Figure 36. Figure 37 is an oscilloscope photograph

of an AD811 line driver’s pulse response with ±15 volt supplies.

The differential gain and phase error vs. supply are plotted in
Figures 38 and 39, respectively.

Another important consideration when driving multiple cables
is the high frequency isolation between the outputs of the
cables. Due to its low output impedance, the AD811 achieves
better than 40 dB of output to output isolation at 5 MHz driv-

ing back terminated 75 Ω cables.

75V CABLE

75V

75V

75V CABLE

75V

75V

V

#1

OUT

V

#2

OUT

V

IN

75V CABLE

75V

0.1mF

649V649V

+V

AD811

–V

S

0.1mF

S

Figure 35. A Video Line Driver Operating at a Gain of +2

AD811

1V

100

V

90

IN

10

V

OUT

0%

1V

Figure 37. Small Signal Pulse Response, Gain = +2,

= ±15 V

V

S

0.10

0.09

0.08

0.07

0.06

a.

0.05

0.04

DIFFERENTIAL GAIN – %

0.03

0.02

0.01

DRIVING A SINGLE, BACK TERMINATED,
75V COAX CABLE

b.

DRIVING TWO PARALLEL,
BACK TERMINATED, COAX CABLES

b

a

6

5

SUPPLY VOLTAGE – 6Volts

Figure 38. Differential Gain Error vs. Supply Voltage for
the Video Line Driver of Figure 35

10ns

RF= 649V
F

= 3.58MHz

C

100 IRE
MODULATED
RAMP

15

1413121110987

12

9

6

3

GAIN – dB

0

–3

–6

1

G = +2

= 150V

R

L

= R

R

G

FB

10 100

FREQUENCY – MHz

VS = 615V
R

VS = 65V

= 562V

R

FB

= 649V

FB

Figure 36. Closed-Loop Gain vs. Frequency, Gain = +2

REV. D

0.20

0.18

0.16

0.14

a.

0.12

0.10

0.08

0.06

DIFFERENTIAL PHASE – Degrees

0.04

0.02

6

5

DRIVING A SINGLE, BACK TERMINATED,
75V COAX CABLE

b.

DRIVING TWO PARALLEL,
BACK TERMINATED, COAX CABLES

a

SUPPLY VOLTAGE – 6Volts

b

RF = 649V

= 3.58MHz

F

C

100 IRE
MODULATED
RAMP

Figure 39. Differential Phase Error vs. Supply Voltage for
the Video Line Driver of Figure 35

–11–

15

1413121110987

AD811

An 80 MHz Voltage-Controlled Amplifier Circuit

The voltage-controlled amplifier (VCA) circuit of Figure 40
shows the AD811 being used with the AD834, a 500 MHz,
4-quadrant multiplier. The AD834 multiplies the signal input
by the dc control voltage, V

. The AD834 outputs are in the

G

form of differential currents from a pair of open collectors,
ensuring that the full bandwidth of the multiplier (which ex­ceeds 500 MHz) is available for certain applications. Here,
the AD811 op amp provides a buffered, single-ended ground­referenced output. Using feedback resistors R8 and R9 of

511 Ω, the overall gain ranges from –70 dB, for V

+12 dB, (a numerical gain of four), when V

= 0 dB to

G

= +1 V. The over-

G

all transfer function of the VCA is:

V

= 4 (X1 – X2)(Y1 – Y2)

OUT

which reduces to V

= 4 VG VIN using the labeling conven-

OUT

tions shown in Figure 40. The circuit’s –3 dB bandwidth of
80 MHz, is maintained essentially constant—independent of

gain. The response can be maintained flat to within ±0.1 dB

from dc to 40 MHz at full gain with the addition of an optional
capacitor of about 0.3 pF across the feedback resistor R8. The

circuit produces a full-scale output of ±4 V for a ±1 V input,
and can drive a reverse-terminated load of 50 Ω or 75 Ω to ±2 V.

The gain can be increased to 20 dB (×10) by raising R8 and R9
to 1.27 kΩ, with a corresponding decrease in –3 dB bandwidth

to about 25 MHz. The maximum output voltage under these

conditions will be increased to ±9 V using ±12 V supplies.

The gain-control input voltage, VG, may be a positive or nega­tive ground-referenced voltage, or fully differential, depending
on the user’s choice of connections at Pins 7 and 8. A positive
value of V
ing the sign of V

results in an overall noninverting response. Revers-

G

simply causes the sign of the overall response

G

to invert. In fact, although this circuit has been classified as a
voltage-controlled amplifier, it is also quite useful as a general­purpose four-quadrant multiplier, with good load-driving capa­bilities and fully-symmetrical responses from X- and Y-inputs.

The AD811 and AD834 can both be operated from power

supply voltages of ±5 V. While it is not necessary to power them

from the same supplies, the common-mode voltage at W1 and
W2 must be biased within the common-mode range of the
AD811’s input stage. To achieve the lowest differential gain and
phase errors, it is recommended that the AD811 be operated

from power supply voltages of ±10 volts or greater. This VCA
circuit is designed to operate from a ±12 volt dual power

supply.

FB

+12V

C1

0.1mF

+

V

G

–

V

IN

R1 100V

R2 100V

8765
X2

X1 +V

AD834

Y1 Y2 W2
1234

249V

W1

S

U1

–V

S

R3

*R8 = R9 = 511V FOR X4 GAIN

R4

182V

R5

182V

= 1.27kV FOR X10 GAIN

294V

294V

R6

R7

R8*

R9*

U3

AD811

C2

0.1mF

FB

V

OUT

R

L

–12V

Figure 40. An 80 MHz Voltage-Controlled Amplifier

–12–

REV. D

AD811

A Video Keyer Circuit

By using two AD834 multipliers, an AD811, and a 1 V dc
source, a special form of a two-input VCA circuit called a
video keyer can be assembled. “Keying” is the term used in
reference to blending two or more video sources under the
control of a third signal or signals to create such special effects
as dissolves and overlays. The circuit shown in Figure 41 is a
two-input keyer, with video inputs V
input V

. The transfer function (with V

G

and VB, and a control

A

at the load) is

OUT

given by:

V

= G VA + (1–G) V

OUT

B

where G is a dimensionless variable (actually, just the gain of
the “A” signal path) that ranges from 0 when V
when V

= +1 V. Thus, V

G

varies continuously between V

OUT

= 0, to 1

G

A

and VB as G varies from 0 to 1.

Circuit operation is straightforward. Consider first the signal
path through U1, which handles video input V
clearly zero when V
ensures that it is unity when V

= 0 and the scaling we have chosen

G

= +1 V; this takes care of the

G

first term of the transfer function. On the other hand, the V

. Its gain is

A

G

input to U2 is taken to the inverting input X2 while X1 is
biased at an accurate +1 V. Thus, when V
to video input V
whereas when V

is already at its full-scale value of unity,

B

= +1 V, the differential input X1–X2 is zero.

G

= 0, the response

G

This generates the second term.

The bias currents required at the output of the multipliers are
provided by R8 and R9. A dc-level-shifting network comprising
R10/R12 and R11/R13 ensures that the input nodes of the
AD811 are positioned at a voltage within its common-mode
range. At high frequencies C1 and C2 bypass R10 and R11
respectively. R14 is included to lower the HF loop gain, and is
needed because the voltage-to-current conversion in the
AD834s, via the Y2 inputs, results in an effective value of the

feedback resistance of 250 Ω; this is only about half the value

required for optimum flatness in the AD811’s response. (Note
that this resistance is unaffected by G: when G = 1, all the
feedback is via U1, while when G = 0 it is all via U2). R14
reduces the fractional amount of output current from the multi­pliers into the current-summing inverting input of the AD811,
by sharing it with R8. This resistor can be used to adjust the
bandwidth and damping factor to best suit the application.

To generate the 1 V dc needed for the “1–G” term an AD589

reference supplies 1.225 V ± 25 mV to a voltage divider consist-

ing of resistors R2 through R4. Potentiometer R3 should be
adjusted to provide exactly +1 V at the X1 input.

In this case, we have shown an arrangement using dual supplies

of ±5 V for both the AD834 and the AD811. Also, the overall

gain in this case is arranged to be unity at the load, when it is

driven from a reverse-terminated 75 Ω line. This means that the

“dual VCA” has to operate at a maximum gain of 2, rather

113V

V

G

(0 TO +1V dc)

V

A

(61V FS)

R5

1.87kV

174V

100V

1.02kV

V

B

(61V FS)

+5V

R7

45.3V

R6

226V

876 5

+5V

R1

U4

AD589

R2

R3

R4

X2 X1 +VSW1

U1

AD834

Y1 Y2 W2–V

1

876 5

X2

Y1 Y2 W2–V

1

2

X1 +V

U1

AD834

2

S

34

–5V

W1

S

S

34

–5V

C1

0.1mF

R8

29.4V

R9

29.4V

R10

2.49kV

+5V

C2

0.1mF

R11

2.49kV

R14

SEE TEXT

R12

6.98kV

R13

6.98kV

–5V

SETUP FOR DRIVING
REVERSE-TERMINATED LOAD

Z

200V

200V

C3

0.1mF

C4

0.1mF

O

LOAD

GND

LOAD

GND

TO PIN 6

AD811

TO Y2

FB

AD811

FB

+5V

U3

–5V

V

OUT

Z

INSET

O

V

OUT

REV. D

Figure 41. A Practical Video Keyer Circuit

–13–

AD811

than 4 as in the VCA circuit of Figure 40. However, this cannot
be achieved by lowering the feedback resistor, since below a

critical value (not much less than 500 Ω) the AD811’s peaking

may be unacceptable. This is because the dominant pole in the
open-loop ac response of a current-feedback amplifier is con­trolled by this feedback resistor. It would be possible to operate
at a gain of X4 and then attenuate the signal at the output.
Instead, we have chosen to attenuate the signals by 6 dB at the
input to the AD811; this is the function of R8 through R11.

Figure 42 is a plot of the ac response of the feedback keyer,

when driving a reverse terminated 50 Ω cable. Output noise and

adjacent channel feedthrough, with either channel fully off and
the other fully on, is about –50 dB to 10 MHz. The feedthrough
at 100 MHz is limited primarily by board layout. For V

= +1 V,

G

the –3 dB bandwidth is 15 MHz when using a 137 Ω resistor for
R14 and 70 MHz with R14 = 49.9 Ω. For further information

regarding the design and operation of the VCA and video keyer
circuits, refer to the application note “Video VCA’s and Keyers
Using the AD834 & AD811” by Brunner, Clarke, and Gilbert,
available FREE from Analog Devices.

R14 = 49.9

0

–10

–20

–30
–40

–50

–60

CLOSED-LOOP GAIN – dB

–70

–80

10k

ADJACENT

FEEDTHROUGH

100k

GAIN

CHANNEL

FREQUENCY – Hz

V

R14 = 137

10M1M

V

100M

Figure 42. A Plot of the AC Response of the Video Keyer

–14–

REV. D

OUTLINE DIMENSIONS

PIN 1

0.299 (7.60)

0.291 (7.40)

0.419 (10.65)

0.404 (10.26)

1

16

9

8

0.018 (0.46)

0.014 (0.36)

0.050 (1.27)
BSC

0.107 (2.72)

0.089 (2.26)

0.413 (10.50)

0.398 (10.10)

0.010 (0.25)

0.004 (0.10)

0.015 (0.38)

0.007 (1.18)

0.045 (1.15)

0.020 (0.50)

0.364 (9.246)

0.344 (8.738)

Dimensions shown in inches and (mm).

AD811

PIN 1

0.165 6 0.01

(4.19 6 0.25)

0.125 (3.18)

0.200.(5.08)

0.200 (5.08)

0.125 (3.18)

8- Lead Plastic DIP (N) Package

0.39 (9.91)
MAX

8

MIN

0.018 6 0.003
(0.46 6 0.08)

5

0.25

0.033
(0.84)

NOM

(6.35)

0.035 6 0.01
(0.89 6 0.25)

14

0.10 (2.54)
BSC

0.31

(7.87)

0.18 6 0.03
(4.57 6 0.75)

SEATING
PLANE

8-Lead Cerdip (Q) Package

0.005 (0.13)

PIN 1

MAX

0.023 (0.58)

0.014 (0.36)

MIN

0.055 (1.4)
MAX

85

1

4

0.100 (2.54) BSC

0.405 (10.29) MAX

0.070 (1.78)

0.030 (0.76)

0.310 (7.87)

0.220 (5.59)

0.060 (1.52)

0.015 (0.38)

0.150
(3.81)
MIN

SEATING
PLANE

0.30 (7.62)

0.320 (8.13)

0.290 (7.37)

15°

0°

158

REF

08

0.011 6 0.003
(0.28 6 0.08)

0.015 (0.38)

0.008 (0.20)

0.082 ± 0.018

(2.085 ± 0.455)

20-Lead LCC (E-20A) Package

0.350 ± 0.008 SQ

0.050

(1.27)

(8.89 ± 0.20) SQ

NO. 1 PIN

INDEX

0.040 x 45°
(1.02 x 45°)

REF 3 PLCS

(0.635 ± 0.075)

0.020 x 45°
(0.51 x 45°)

16-Lead SOIC (R-16) Package

0.025 ± 0.003

C1592b–0–8/99

REF

20-Lead Wide Body SOIC (R-20) Package

8-Lead SOIC (SO-8) Package

0.1 968 (5.00)

0.1 890 (4.80)

0.1574 (4.00)

85

0.1497 (3.80)

PIN 1

0.0500 (1.27)

0.0098 (0.25)

0.0040 (0.10)
SEATING

PLANE

41

BSC

0.0192 (0.49)

0.0138 (0.35)

0.2440 (6.20)

0.2284 (5.80)

0.0688 (1.75)

0.0532 (1.35)

0.0098 (0.25)

0.0075 (0.19)

0.0196 (0.50)

0.0099 (0.25)

88

0.0500 (1.27)

08

0.0160 (0.41)

3 458

0.015 (0.38)

0.007 (0.18)

1

0.50 (1.27)
BSC

0.512 (13.00)

0.496 (12.60)

0.019 (0.48)

0.014 (0.36)

1120

0.300 (7.60)

0.292 (7.40)

10

0.011 (0.28)

0.004 (0.10)

0.050 (1.27)

0.016 (0.40)

0.419 (10.65)

0.394 (10.00)

0.104 (2.64)

0.093 (2.36)

PRINTED IN U.S.A.

REV. D

–15–