Datasheet AD607ARS Datasheet (Analog Devices)

Low Power Mixer

a

FEATURES
Complete Receiver-on-a-Chip: Monoceiver

–15 dBm 1 dB Compression Point
–8 dBm Input Third Order Intercept
500 MHz RF and LO Bandwidths

Linear IF Amplifier

Linear-in-dB Gain Control
Manual Gain Control

Quadrature Demodulator

On-Board Phase-Locked Quadrature Oscillator
Demodulates IFs from 400 kHz to 12 MHz
Can Also Demodulate AM, CW, SSB

Low Power

25 mW at 3 V
CMOS Compatible Power-Down

Interfaces to AD7013 and AD7015 Baseband Converters

APPLICATIONS
GSM, CDMA, TDMA, and TETRA Receivers
Satellite Terminals
Battery-Powered Communications Receivers

®

Mixer

3 V Receiver IF Subsystem

AD607

PIN CONFIGURATION

20-Lead SSOP

(RS Suffix)

FDIN

COM1

PRUP

LOIP

RFLO

RFHI

GREF

MXOP

VMID

IFHI

1

2

3

4

5

AD607

TOP VIEW

6

(Not to Scale)

7

8

9

10

20

VPS1

19

FLTR

18

IOUT

17

QOUT

16

VPS2

15

DMIP

14

IFOP

13

COM2

12

GAIN

11

IFLO

GENERAL DESCRIPTION

The AD607 is a 3 V low power receiver IF subsystem for opera­tion at input frequencies as high as 500 MHz and IFs from
400 kHz to 12 MHz. It consists of a mixer, IF amplifiers, I and
Q demodulators, a phase-locked quadrature oscillator, and a
biasing system with external power-down.

The AD607’s low noise, high intercept mixer is a doubly­balanced Gilbert cell type. It has a nominal –15 dBm input
referred 1 dB compression point and a –8 dBm input referred
third-order intercept. The mixer section of the AD607 also
includes a local oscillator (LO) preamplifier, which lowers the
required LO drive to –16 dBm.

In MGC operation, the AD607 accepts an external gain-control
voltage input from an external AGC detector or a DAC.

Monoceiver is a registered trademark of Analog Devices, Inc.

The I and Q demodulators provide in-phase and quadrature
baseband outputs to interface with Analog Devices’ AD7013
(IS54, TETRA, MSAT) and AD7015 (GSM) baseband con­verters. A quadrature VCO phase-locked to the IF drives the I
and Q demodulators. The I and Q demodulators can also
demodulate AM; when the AD607’s quadrature VCO is phase
locked to the received signal, the in-phase demodulator becomes
a synchronous product detector for AM. The VCO can also be
phase-locked to an external beat-frequency oscillator (BFO),
and the demodulator serves as a product detector for CW or
SSB reception. Finally, the AD607 can be used to demodulate
BPSK using an external Costas Loop for carrier recovery.

REV. B

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2000

AD607–SPECIFICATIONS

(@ TA = 25ⴗC, Supply = 3.0 V, IF = 10.7 MHz, unless otherwise noted)

Model AD607ARS

Conditions Min Typ Max Unit

DYNAMIC PERFORMANCE

MIXER

Maximum RF and LO Frequency Range For Conversion Gain > 20 dB 500 MHz
Maximum Mixer Input Voltage For Linear Operation; Between RFHI and RFLO ± 54 mV
Input 1 dB Compression Point RF Input Terminated in 50 Ω –15 dBm
Input Third-Order Intercept RF Input Terminated in 50 Ω –5 dBm
Noise Figure Matched Input, Max Gain, f = 83 MHz, IF = 10.7 MHz 14 dB

Matched Input, Max Gain, f = 144 MHz, IF = 10.7 MHz 12 dB
Maximum Output Voltage at MXOP Z
Mixer Output Bandwidth at MXOP –3 dB, Z

= 165 Ω, at Input Compression ± 1.3 V

IF

= 165 Ω 45 MHz

IF

LO Drive Level Mixer LO Input Terminated in 50 Ω –16 dBm
LO Input Impedance LOIP to VMID 1 kΩ
Isolation, RF to IF RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz 30 dB
Isolation, LO to IF RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz 20 dB
Isolation, LO to RF RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz 40 dB
Isolation, IF to RF RF = 240 MHz, IF = 10.7 MHz, LO = 229.3 MHz 70 dB

IF AMPLIFIERS

Noise Figure Max Gain, f = 10.7 MHz 17 dB
Input 1 dB Compression Point IF = 10.7 MHz –15 dBm
Output Third-Order Intercept IF = 10.7 MHz 18 dBm
Maximum IF Output Voltage at IFOP Z

= 600 Ω±560 mV

IF

Output Resistance at IFOP From IFOP to VMID 15 Ω
Bandwidth –3 dB at IFOP, Max Gain 45 MHz

GAIN CONTROL (See Figures 43 and 44)

Gain Control Range Mixer + IF Section, GREF to 1.5 V 90 dB
Gain Scaling GREF to 1.5 V 20 mV/dB

GREF to General Reference Voltage V

R

75/V

R

dB/V
Gain Scaling Accuracy GREF to 1.5 V, 80 dB Span ± 1dB
Bias Current at GAIN 5 µA
Bias Current at GREF 1 µA
Input Resistance at GAIN, GREF 1MΩ

I AND Q DEMODULATORS

Required DC Bias at DMIP VPOS/2 V dc
Input Resistance at DMIP From DMIP to VMID 50 kΩ
Input Bias Current at DMIP 2 µA
Maximum Input Voltage IF > 3 MHz ±150 mV

IF ≤ 3 MHz ±75 mV
Amplitude Balance IF = 10.7 MHz, Outputs at 600 mV p-p, F = 100 kHz ± 0.2 dB
Quadrature Error IF = 10.7 MHz, Outputs at 600 mV p-p, F = 100 kHz –1.2 Degrees
Phase Noise in Degrees IF = 10.7 MHz, F = 10 kHz –100 dBc/Hz
Demodulation Gain Sine Wave Input, Baseband Output 18 dB
Maximum Output Voltage R
Output Offset Voltage Measured from I

≥ 20 kΩ±1.23 V

L

OUT

, Q

to VMID –150 10 +150 mV

OUT

Output Bandwidth Sine Wave Input, Baseband Output 1.5 MHz

PLL

Required DC Bias at FDIN VPOS/2 V dc
Input Resistance at FDIN From FDIN to VMID 50 kΩ
Input Bias Current at FDIN 200 nA
Frequency Range 0.4 to 12 MHz
Required Input Drive Level Sine Wave Input at Pin 1 400 mV
Acquisition Time to ± 3° IF = 10.7 MHz 16.5 µs

POWER-DOWN INTERFACE

Logical Threshold For Power Up on Logical High 2 V dc
Input Current for Logical High 75 µA
Turn-On Response Time To PLL Locked 16.5 µs
Standby Current 550 µA

POWER SUPPLY

Supply Range 2.7 5.5 V
Supply Current Midgain, IF = 10.7 MHz 8.5 mA

OPERATING TEMPERATURE

T

MIN

to T

MAX

Operation to 2.7 V Minimum Supply Voltage –25 +85 °C

Operation to 4.5 V Minimum Supply Voltage –40 +85 °C

Specifications subject to change without notice.

–2–

REV. B

AD607

WARNING!

ESD SENSITIVE DEVICE

ABSOLUTE MAXIMUM RATINGS

Supply Voltage VPS1, VPS2 to COM1, COM2 . . . . . . . 5.5 V

Internal Power Dissipation

2

. . . . . . . . . . . . . . . . . . . . 600 mW

1

2.7 V to 5.5 V Operating Temperature Range

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –25°C to +85°C

4.5 V to 5.5 V Operating Temperature Range

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . – 40°C to +85°C

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300°C

NOTES

1

Stresses above those listed under Absolute Maximum Rating may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.

2

Thermal Characteristics: 20-lead SSOP Package: θJA = 126°C/W.

Model Range Description Option

AD607ARS –25°C to +85°C 20-Lead Plastic RS-20

ORDERING GUIDE

Temperature Package Package

for 2.7 V to 5.5 V SSOP
Operation; –40°C
to +85°C for 4.5 V
to 5.5 V Operation

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD607 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high-energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

REV. B

–3–

AD607

PIN FUNCTION DESCRIPTIONS

Pin Mnemonic Reads Function

1 FDIN Frequency Detector Input PLL input for I/Q demodulator quadrature oscillator, ±400 mV

drive required from external oscillator. Must be biased at VP/2.
2 COM1 Common #1 Supply common for RF front end and main bias.
3 PRUP Power-Up Input 3 V/5 V CMOS compatible power-up control; logical high =

powered-up; max input level = VPS1 = VPS2.
4 LOIP Local Oscillator Input LO input, ac coupled ± 54 mV LO input required (–16 dBm for

50 Ω input termination).
5 RFLO RF “Low” Input Usually connected to ac ground.
6 RFHI RF “High” Input AC coupled, ±56 mV, max RF input for linear operation.
7 GREF Gain Reference Input High impedance input, typically 1.5 V, sets gain scaling.
8 MXOP Mixer Output High impedance, single-sided current output, ±1.3 V max voltage

output (±6 mA max current output).
9 VMID Midsupply Bias Voltage Output of the midsupply bias generator (VMID = VPOS/2).
10 IFHI IF “High” Input AC coupled IF input, ± 56 mV max input for linear operation.
11 IFLO IF “Low” Voltage Reference node for IF input; auto-offset null.
12 GAIN Gain Control Input High impedance input, 0 V–2 V using 3 V supply, max gain at

V = 0.
13 COM2 Common #2 Supply common for IF stages and demodulator.
14 IFOP IF Output Low impedance, single-sided voltage output, 5 dBm (±560 mV)

max.
15 DMIP Demodulator Input Signal input to I and Q demodulators ±150 mV max input at IF

> 3 MHz for linear operation; ±75 mV max input at IF < 3 MHz

for linear operation. Must be biased at V
16 VPS2 VPOS Supply #2 Supply to high-level IF, PLL, and demodulators.
17 QOUT Quadrature Output Low impedance Q baseband output; ± 1.23 V full scale in 20 kΩ

min load; ac coupled.
18 IOUT In-Phase Output Low impedance I baseband output; ± 1.23 V full scale in 20 kΩ

min load; ac coupled.
19 FLTR PLL Loop Filter Series RC PLL Loop filter, connected to ground.
20 VPS1 VPOS Supply #1 Supply to mixer, low level IF, PLL, and gain control.

/2.

P

PIN CONNECTION

20-Lead SSOP (RS-20)

FDIN

COM1

PRUP

LOIP

RFLO

RFHI

GREF

MXOP

VMID

IFHI

1

2

3

4

5

AD607

TOP VIEW

6

(Not to Scale)

7

8

9

10

20

VPS1

19

FLTR

18

IOUT

17

QOUT

16

VPS2

15

DMIP

14

IFOP

13

COM2

12

GAIN

11

IFLO

–4–

REV. B

HP8656B

IEEE

RF_OUT

SYNTHESIZER

HP8656B

IEEE

RF_OUT

SYNTHESIZER

HP8656B

IEEE

RF_OUT

SYNTHESIZER

HP6633A

IEEE

VPOS

VNEG

SPOS

SNEG

DCPS

HP34401A

CPIB

HI

LO

I

DMM

DP8200

IEEE

VPOS

VNEG
SPOS

SNEG

V

REF

HP8764B

0

0

1

1

S0

S1

V

50⍀

50⍀

MXOP

RFHI

LOIP

L

R

X

IFOPIFHI

PLL

IOUT

QOUT

DMIP

FDIN

BIAS

VPOS

PRUP

GAIN

HP8764B

0

0

1

1

S0

S1

V

50⍀

50⍀

HP8594E

RF_IN

IEEE

SPEC

AN

HP8765B

0

1

C

S0

S1V

R5

1k⍀

CHARACTERIZATION

BOARD

HP8765B

0

1C

S0 S1V

P6205

X10

OUT

FET PROBE

TEK1105

IN1 OUT1

IN2 OUT2

PROBE

SUPPLY

Typical Performance Characteristics–AD607

REV. B

HP8720C

IEEE_488

NETWORK AN

HP346B

28V

NOISE SOURCE

HP8656B

IEEE

SYNTHESIZER

HP6633A

IEEE

DCPS

DP8200

IEEE

V

REF

PORT_1

PORT_2

NOISE

RF_OUT

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

HP8765B

0

1C

S0 S1

V

Figure 1. Mixer/Amplifier Test Set

CHARACTERIZATION

BOARD

RFHI

LOIP

DMIP

FDIN

VPOS

PRUP

GAIN

Figure 2. Mixer Noise Figure Test Set

X

R

L

PLL

BIAS

MXOP

IFOPIFHI

IOUT

QOUT

C

–5–

HP8765B

S1 V

50⍀

0

1

S0

RF_IN 28V_OUT

NOISE FIGURE METER

HP8970A

AD607

CHARACTERIZATION

BOARD

HP8656B

IEEE

DCFM

IEEE

DUAL SYNTHESIZER

IEEE

IEEE

RF_OUT

SYNTHESIZER

HP3326A

OUTPUT_1

OUTPUT_2

HP6633A

DCPS

DP8200

V

REF

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

HP346B

28V

NOISE SOURCE

HP6633A

IEEE

DCPS

DP8200

IEEE

V

REF

50⍀

50⍀

NOISE

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

HP8764B

RFHI

LOIP

DMIP

FDIN

VPOS

PRUP

GAIN

R

PLL

L

BIAS

MXOP

X

IFOPIFHI

IOUT

QOUT

X10

FET

P6205

OUT

PROBE

TEK1103

IN1 OUT1

IN2 OUT2

PROBE SUPPLY

HP8970A

RF_IN 28V_OUT

NOISE FIGURE METER

Figure 3. IF Amp Noise Figure Test Set

CHARACTERIZATION

BOARD

0

1
0

1

RFHI

LOIP

S0

S1

V

DMIP

FDIN

VPOS

PRUP

GAIN

MXOP

X

R

L

IFOPIFHI

OUT

OUT

IN1

IN2

1103

PROBE

SUPPLY

OUT1

OUT2

HP8765B

0

1C

S0

S1V

HP8765B

C

0

1

S0S1

V

HP8694E

RF_IN

CH1

CH2

CH3

CH4

TRIG IEEE_488

OSCILLOSCOPE

SPEC AN

HP54120

DIGITAL

IEEE

PLL

BIAS

IOUT

QOUT

P6205

X10

FET PROBE

P6205

X10

FET PROBE

Figure 4. PLL/Demodulator Test Set

–6–

REV. B

IEEE

IEEE

GPIB

HP6633A

DCPS

DP8200

V

REF

HP34401A

DMM

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

HI

LO

I

499k

R1

⍀

Figure 5. GAIN Pin Bias Test Set

CHARACTERIZATION

BOARD

R

LOIP

IFHI

DMIP

FDIN

VPOS

PRUP

GAIN

L

PLL

BIAS

CHARACTERIZATION

BOARD

AD607

MXOPRFHI

X

IFOP

IOUT

QOUT

HP3325B

IEEE

SYNTHESIZER

HP6633A

IEEE

DCPS

HP6633A

IEEE

DCPS

HP34401A

GPIB

DMM

IEEE

IEEE

GPIB

HP6633A

DCPS

DP8200

V

HP34401A

DMM

RF_OUT

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

REF

HI

LO

I

R1

499k⍀

Figure 6. Demodulator Bias Test Set

CHARACTERIZATION

RFHI

LOIP

IFHI

DMIP

FDIN

R1

HI

LO

I

10k⍀

VPOS

PRUP

GAIN

BOARD

R

L

PLL

BIAS

R

PLL

L

BIAS

X

RF_IN

MXOP

IFOP

IOUT

QOUT

HP8594E

IEEE

SPEC AN

RFHI

LOIP

IFHI

DMIP

FDIN

VPOS

PRUP

GAIN

MXOP

X

IFOP

IOUT

QOUT

REV. B

Figure 7. Power-Up Threshold Test Set

–7–

AD607

CHARACTERIZATION

BOARD

FL6082A

MOD_OUT

HP6633A

DCPS

DP8200

V

REF

HP8112

PULSE_OUT

RF_OUT

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

IEEE

IEEE

IEEE

IEEE

PULSE GENERATOR

RFHI

LOIP

IFHI

MXOP

R

X

L

IFOP

50⍀

DMIP

FDIN

VPOS

PRUP

GAIN

PLL

IOUT

QOUT

BIAS

Figure 8. Power-Up Test Set

CHARACTERIZATION

BOARD

X10

FET PROBE

X10

FET PROBE

P6205

P6205

OUT

OUT

IN1

IN2

1103

OUT1

OUT2

PROBE SUPPLY

HP54120

CH1

CH2

CH3

CH4

TRIG

DIGITAL

OSCILLOSCOPE

NOTE: MUST BE 3 RESISTOR POWER DIVIDER

IEEE_488

HP8656B

SYNTHESIZER

HP6633A

IEEE

DCPS

RF_OUTIEEE

VPOS

VNEG

SPOS

SNEG

MXOPRFHI

R

X

PLL

L

BIAS

IFOP

IOUT

QOUT

R1
1k⍀

P6205

X10

FET PROBE

OUT

IN1 OUT1

IN2 OUT2

PROBE SUPPLY

LOIP

IFHI

DMIP

FDIN

VPOS

PRUP

GAIN

Figure 9. IF Output Impedance Test Set

1103

RF_IN

HP8594E

IEEE

SPEC AN

–8–

REV. B

IEEE

IEEE

IEEE

FL6082A

MOD_OUT

HP6633A

DCPS

DP8200

V

REF

RF_OUT

VPOS

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

CHARACTERIZATION

BOARD

MXOPRFHI

R

X

PLL

L

BIAS

IFOP

IOUT

QOUT

P6205

X10

FET PROBE

P6205

X10

FET PROBE

OUT

OUT

PROBE SUPPLY

20⍀

dB

LOIP

IFHI

DMIP

FDIN

VPOS

PRUP

GAIN

Figure 10. PLL Settling Time Test Set

AD607

1103

OUT1

IN1

OUT2

IN2

HP54120

CH1

CH2

CH3

CH4

TRIG IEEE_488

DIGITAL

OSCILLOSCOPE

HP3325B

IEEE RF_OUT

SYNTHESIZER

HP3326

OUTPUT_1

DCFM

IEEE

OUTPUT_2

DUAL SYNTHESIZER

HP6633A

VPOS

DCPS

DP8200

V

REF

VNEG

SPOS

SNEG

VPOS

VNEG

SPOS

SNEG

IEEE

IEEE

CHARACTERIZATION

BOARD

R

PLL

L

BIAS

MXOP

X

IFOP

IOUT

QOUT

P6205

X10

FET PROBE

P6205

X10

FET PROBE

OUT

IN1

OUT

IN2

PROBE SUPPLY

1103

OUT1

OUT2

RFHI

LOIP

IFHI

DMIP

FDIN

VPOS

PRUP

GAIN

Figure 11. Quadrature Accuracy Test Set

HP8765B

0

1C

S0

S1

V

RF_IN

HP8694E

SPEC AN

IEEE

REV. B

–9–

AD607

VPOS

GND

FDIN

PRUP

LOIP

RFHI

MXOP

*

IFHI

NOTE: CONNECTIONS MARKED * ARE DC COUPLED.

R14

54.9⍀

R8

51.1⍀

R6

51.1⍀

0.1␮F

C15

0.1␮F

C11

10nF

R7

51.1⍀

R13

301⍀

332⍀

C10
1nF

R5

C9
1nF

0.1␮F
C13

R9

51.1⍀

Figure 12. Characterization Board

4.99k⍀
R10

0⍀
R12

C16
1nF

C7

1nF

0.1␮F

R1

1k⍀

0.1␮F
C2

R2
316⍀

C6

0.1␮F

C8

0.1␮F

C1

C3

10nF

C5
1nF

IOUT
*

QOUT
*

IFOP
*

GAIN
*

DMIP
*

1

FDIN

2

COM1

3

PRUP

4

LOIP

AD607

5

RFLO

6

RFHI

GREF

7

MXOP

8

9

VMID

10

IFHI

VPS1

FLTR

IOUT

QOUT

VPS2

DMIP

IFOP

COM2

GAIN

IFLO

20

19

18

17

16

15

14

13

12

11

–10–

REV. B

20

p

30

20

1000.1

25

10

15

0

5

INTERMEDIATE FREQUENCY – MHz

–5

–10

110

CONVERSION GAIN – dB

V

GAIN

= 0.3V

V

GAIN

= 0.6V

V

GAIN

= 1.8V

V

GAIN

= 1.2V

V

GAIN

= 2.4V

19

18

17

16

15

14

SSB NF – dB

13

12

11

VPOS = 5V, IF = 10MHz

10

50 25070 90 110 130 150 170 190 210 230

VPOS = 5V, IF = 20MHz

VPOS = 3V, IF = 20MHz

VPOS = 3V, IF = 10MHz

RF FREQUENCY – MHz

Figure 13. Mixer Noise Figure vs. Frequency

AD607

Figure 16. Mixer Conversion Gain vs. IF, T = 25°C,
VPOS = 3 V, VREF = 1.5 V

4500

4000

3500

3000

2500

2000

1500

RESISTANCE – ⍀

1000

500

0

50 100 150 200 300 350 400 450

C SHUNT COMPONENT

R SHUNT COMPONENT

FREQUENCY – MHz

Figure 14. Mixer Input Impedance vs. Frequency,
VPOS = 3 V, V GAIN = 0.8 V

4.0

3.5

3.0

F

2.5

2.0

1.5

CAPACITANCE –

1.0

0.5

0

5002500

GAIN – dB

–10

–20

80

70

60

50

40

30

20

10

0

CUBIC FIT OF IF_GAIN (TEMP)

IF AMP GAIN

CUBIC FIT OF CONV_GAIN (TEMP)

MIXER CG

–30 –10 10 20 30 40 50 60 80 90 100 110 120–40 –20 0

TEMPERATURE – ⴗC

70

130–50

Figure 17. Mixer Conversion Gain and IF Amplifier Gain
vs. Temperature, VPOS = 3 V, VGAIN = 0.3 V, VREF = 1.5 V,
IF = 10.7 MHz, RF = 250 MHz

30

25

V

= 0.54V

GAIN

20

15

10

V

= 1.62V

GAIN

5

0

–5

CONVERSION GAIN – dB

–10

–15

–20

50 100 150 200 250 350 400 450 500 550

Figure 15. Mixer Conversion Gain vs. Frequency,

°

C, VPOS = 2.7 V, VREF = 1.35 V, IF = 10.7 MHz

T = 25

RADIO FREQUENCY – MHz

REV. B

V

= 0.00V

GAIN

V

= 1.08V

GAIN

V

= 2.16V

GAIN

6003000

Figure 18. Mixer Conversion Gain and IF Amplifier Gain
vs. Supply Voltage, T = 25
IF = 10.7 MHz, RF = 250 MHz

–11–

GAIN – dB

80

70

60

50

40

30

20

10

IF AMP GAIN

CUBIC FIT OF IF_GAIN (V

CUBIC FIT OF CONV_GAIN (V

MIXER CG

2.8 3.2 3.6 3.8 4 4.2 4.4 4.6 5 5.2 5.4 5.6 5.82.6 3 3.4

SUPPLY – Volts

°

)

POS

)

POS

4.8

62.4

C, VGAIN = 0.3 V, VREF = 1.5 V,

AD607

, f(fm)

80

V

= 0.3V

GAIN

V

= 0.6V

GAIN

V

= 1.2V

GAIN

V

= 1.8V

GAIN

V

= 2.4V

GAIN

110

INTERMEDIATE FREQUENCY – MHz

1000.1

IF AMPLIFIER GAIN – dB

70

60

50

40

30

20

10

0

–10

Figure 19. IF Amplifier Gain vs. Frequency,
T = 25

ERROR – dB

°

C, VPOS = 3 V, VREF = 1.5 V

10

8

6

4

2

0

–2

–4

–6

–8

–10

0.2 0.4 0.6 0.8 1.2 1.4 1.6 1.8 2.2 2.4 2.6 2.8

12

GAIN VOLTAGE – Volts

IF AMP

MIXER

Figure 20. Gain Error vs. Gain Control Voltage,
Representative Part

–90.00

–100.00

–110.00

–120.00

–130.00

PHASE NOISE – dBc

–140.00

–150.00

1.00E+03 1.00E+05

CARRIER FREQUENCY OFFSET

1.00E+04 1.00E+06

1.00E+071.00E+02

– Hz

Figure 22. PLL Phase Noise L (F) vs. Frequency,
VPOS = 3 V, C3 = 0.1

2.5

2

FLTR PIN VOLTAGE – Volts

30

1.5

Figure 23. PLL Loop Voltage at FLTR (K

µ

F, IF = 10.7 MHz

110

PLL FREQUENCY – MHz

) vs. Frequency

VCO

1000.1

996.200␮s 1.00870ms 1.02120ms

TIMEBASE

= 2.5␮s/DIV

MEMORY 1

= 100.0mV/DIV

TIMEBASE

= 2.50␮s/DIV

MEMORY 2

= 20.00mV/DIV

TIMEBASE

= 2.50␮s/DIV

DELTA T

= 16.5199␮s

START

= 1.00048ms

TRIGGER ON EXTERNAL AT POS. EDGE AT 134.0mV

DELAY

OFFSET

DELAY

OFFSET

DELAY

STOP

= 1.00870ms

= 127.3mV

= 1.00870ms

= 155.2mV

= 1.00870ms

= 1.01700ms

Figure 21. PLL Acquisition Time

8

7

6

5

4

COUNT

3

2

1

0

QUADRATURE ANGLE – Degrees

9186 87 88 89 90 92 93

Figure 24. Demodulator Quadrature Angle, Histogram,

°

C, VPOS = 3 V, IF = 10.7 MHz

T = 25

–12–

9485

95

REV. B

AD607

30

25

20

15

COUNT

10

5

0

–2

–101 2

IQ GAIN BALANCE – dB

Figure 25. Demodulator Gain Balance, Histogram,

°

C, VPOS = 3 V, IF = 10.7 MHz

T = 25

20

19

18

IGAIN – dB

17

16

15

14

13

12

11

10

0

I_GAIN_CORR

QUADRATIC FIT OF I_GAIN_CORR (IFF)

0.2 0.4 0.6 0.8

BASEBAND FREQUENCY – MHz

1.0 1.2 1.4 1.6 1.8 2.0

Figure 26. Demodulator Gain vs. Frequency

IGAIN – dB

20

19

18

17

16

15

14

13

12

11

10

2.5

CUBIC FIT OF I_GAIN_CORR (TEMP)

3.5 4 4.5

3

I_GAIN_CORR

5 5.5 6

SUPPLY – Volts

Figure 28. Demodulator Gain vs. Supply Voltage

40

35

30

25

20

COUNT

15

10

0

17.2517.4 17.6 17.8 18 18.2 18.4

17

DEMODULATOR GAIN – dB

18.6 18.8

Figure 29. Demodulator Gain Histogram,
T = 25

°

C, VPOS = 3 V, IF = 10.7 MHz

20

I_GAIN_CORR

0 1020304050

TEMPERATURE – ⴗC

60 70 80 90 100 110 120 130

IGAIN – dB

19

18

17

16

15

14

13

12

11

10

–50

CUBIC FIT OF I_GAIN_CORR (TEMP)

–40 –30 –20 –10

Figure 27. Demodulator Gain vs. Temperature

REV. B

–13–

AD607

40.2127ms 40.2377ms 40.2627ms

TIMEBASE

MEMORY 1

TIMEBASE

MEMORY 2

TIMEBASE

DELTA T

START

Figure 30. Power-Up Response Time to PLL Stable

15

10

SUPPLY CURRENT – mA

5

0

Figure 31. Power Supply Current vs. Gain Control Voltage,
GREF = 1.5 V

= 500␮s/DIV

= 100.0mV/DIV

= 5.00␮s/DIV

= 60.00mV/DIV

= 5.00␮s/DIV

= 15.7990␮s

= 40.2327ms

TRIGGER ON EXTERNAL AT POS. EDGE AT 40.0mV

0.5 1.5 2
GAIN VOLTAGE – Volts

DELAY

= 40.2377ms

OFFSET

= 154.0mV

DELAY

= 40.2377ms

OFFSET

= 209.0mV

DELAY

= 40.2377ms

STOP

= 40.2485ms

1

2.5

PRODUCT OVERVIEW

The AD607 provides most of the active circuitry required to
realize a complete low power, single-conversion superhetero­dyne receiver, or most of a double-conversion receiver, at input
frequencies up to 500 MHz, and with an IF of from 400 kHz to
12 MHz. The internal I/Q demodulators, and their associated
phase locked-loop, which can provide carrier recovery from the
IF, support a wide variety of modulation modes, including n­PSK, n-QAM, and AM. A single positive supply voltage of 3 V
is required (2.7 V minimum, 5.5 V maximum) at a typical sup­ply current of 8.5 mA at midgain. In the following discussion,

will be used to denote the power supply voltage, which will

V

P

be assumed to be 3 V.

Figure 32 shows the main sections of the AD607. It consists of a
variable-gain UHF mixer and linear four-stage IF strip, which
together provide a voltage controlled gain range of more than
90 dB; followed by dual demodulators, each comprising a multi­plier followed by a two-pole, 2 MHz low-pass filter; and driven
by a phase-locked loop providing the inphase and quadrature
clocks. A biasing system with CMOS compatible power-down
completes the AD607.

Mixer

The UHF mixer is an improved Gilbert cell design, and can
operate from low frequencies (it is internally dc-coupled) up to
an RF input of 500 MHz. The dynamic range at the input of the
mixer is determined, at the upper end, by the maximum input
signal level of ±56 mV between RFHI and RFLO up to which
the mixer remains linear, and, at the lower end, by the noise
level. It is customary to define the linearity of a mixer in terms
of the 1 dB gain-compression point and third-order intercept,
which for the AD607 are –15 dBm and –8 dBm, respectively, in
a 50 Ω system.

RFHI

RFLO

VPS1

VPS2

PRUP

LOIP

MIDPOINT

BIAS

GENERATOR

BIAS

GENERATOR

MXOP

VMID

COM1 COM2

VMID

PTAT

VOLTAGE

BPF

IFHI

IFLO

Figure 32. Functional Block Diagram

–14–

IFOP

BPF OR

LPF

DMIP

AD607

VQFO

IOUT

FDIN

FLTR

QOUT

GAIN

GREF

REV. B

AD607

The mixer’s RF input port is differential, that is, pin RFLO is
functionally identical to RFHI, and these nodes are internally
biased; we will generally assume that RFLO is decoupled to ac
ground. The RF port can be modeled as a parallel RC circuit as
shown in Figure 33.

AD607

C1

C1, C2, L1: OPTIONAL MATCHING CIRCUIT
C3: COUPLES RFLO TO AC GROUND

C2

RFHI

R

C

IN

L1

C3

RFLO

IN

Figure 33. Mixer Port Modeled as a Parallel RC Network;
an Optional Matching Network Is also Shown

The local oscillator (LO) input is internally biased at VP/2 via a
nominal 1000 Ω resistor internally connected from pin LOIP to
VMID. The LO interface includes a preamplifier which mini­mizes the drive requirements, thus simplifying the oscillator
design and reducing LO leakage from the RF port. Internally,
this single-sided input is actually differential; the noninverting
input is referenced to pin VMID. The LO requires a single­sided drive of ± 50 mV, or –16 dBm in a 50 Ω system.

The mixer’s output passes through both a low-pass filter and a
buffer, which provides an internal differential to single-ended
signal conversion with a bandwidth of approximately 45 MHz.
Its output at pin MXOP is in the form of a single-ended current.
This approach eliminates the 6 dB voltage loss of the usual
series termination by replacing it with shunt terminations at the
both the input and the output of the filter. The nominal conver­sion gain is specified for operation into a total IF bandpass filter
(BPF) load of 165 Ω, that is, a 330 Ω filter, doubly-terminated
as shown in Figure 33. Note that these loads are connected to
bias point VMID, which is always at the midpoint of the supply
(that is, V

P

/2).

The conversion gain is measured between the mixer input and
the input of this filter, and varies between 1.5 dB and 26.5 dB
for a 165 Ω load impedance. Using filters of higher impedance,
the conversion gain can always be maintained at its specified
value or made even higher; for filters of lower impedance, of say
Z

, the conversion gain will be lowered by 10 log10(165/ZO).

O

Thus, the use of a 50 Ω filter will result in a conversion gain that
is 5.2 dB lower. Figure 34 shows filter matching networks and
Table I lists resistor values.

R3

100nF

1nF

10

IFHI

IFLO

11

MXOP

VMID

R2

8

9

BPF

R1

100nF

Table I. AD607 Filter Termination Resistor Values for
Common IFs

Filter Filter Termination Resistor

IF Impedance Values1 for 24 dB of Mixer Gain

R1 R2 R3

450 kHz 1500 Ω 174 Ω 1330 Ω 1500 Ω
455 kHz 1500 Ω 174 Ω 1330 Ω 1500 Ω

6.5 MHz 1000 Ω 215 Ω 787 Ω 1000 Ω

10.7 MHz 330 Ω 330 Ω 0 Ω 330 Ω

NOTES

1

Resistor values were calculated such that R1+ R2 = Z
R1储 (R2 + Z

FILTER

) = 165 Ω.

FILTER

and

The maximum permissible signal level at MXOP is determined
by both voltage and current limitations. Using a 3 V supply and
VMID at 1.5 V, the maximum swing is about ±1.3 V. To attain
a voltage swing of ±1 V in the standard IF filter load of 165 Ω
load requires a peak drive current of about ±6 mA, which is well
within the linear capability of the mixer. However, these upper
limits for voltage and current should not be confused with issues
related to the mixer gain, already discussed. In an operational
system, the AGC voltage will determine the mixer gain, and
hence the signal level at the IF input pin IFHI; it will always be
less than ± 56 mV (–15 dBm into 50 Ω), which is the limit of
the IF amplifier’s linear range.

IF Amplifier

Most of the gain in the AD607 arises in the IF amplifier strip,
which comprises four stages. The first three are fully differential
and each has a gain span of 25 dB for the nominal AGC voltage
range. Thus, in conjunction with the mixer’s variable gain, the
total gain exceeds 90 dB. The final IF stage has a fixed gain of
20 dB, and it also provides differential to single-ended conversion.

The IF input is differential, at IFHI (noninverting relative to the
output IFOP) and IFLO (inverting). Figure 35 shows a simpli­fied schematic of the IF interface. The offset voltage of this
stage would cause a large dc output error at high gain, so it is
nulled by a low-pass feedback path from the IF output, also
shown in Figure 25. Unlike the mixer output, the signal at IFOP
is a low-impedance single-sided voltage, centered at V

/2 by the

P

dc feedback loop. It may be loaded by a resistance as low as
50 Ω, which will normally be connected to VMID.

IFHI

IFLO

OFFSET FEEDBACK

LOOP

10k⍀

10k⍀

AD607

VMID

IFOP

Figure 35. Simplified Schematic of the IF Interface

Figure 34. Suggested IF Filter Matching Network. The
Values of R1 and R2 Are Selected to Keep the Impedance
at Pin MXOP at 165

Ω

REV. B

–15–

AD607

The IF’s small-signal bandwidth is approximately 45 MHz from
IFHI and IFLO through IFOP. The peak output at IFOP is
± 560 mV at V

= 3 V and ± 400 mV at the minimum VP of

P

2.7 V. This allows some headroom at the demodulator inputs
(pin DMIP), which accept a maximum input of ±150 mV for
IFs > 3 MHz and ±75 mV for IFs ≤ 3 MHz (at IFs ≤ 3 MHz,
the drive to the demodulators must be reduced to avoid saturat­ing the output amplifiers with higher order mixing products that
are no longer removed by the onboard low-pass filters).

Since there is no band-limiting in the IF strip, the output­referred noise can be quite high; in a typical application and
at a gain of 75 dB it is about 100 mV rms, making post-IF filtering
desirable. IFOP may be also used as an IF output for driving
an A/D converter, external demodulator, or external AGC
detector. Figure 36 shows methods of matching the optional
second IF filter.

BPF

VPOS

2R

T

2R

T

AD607

IFOP

DMIP

R

T

a. Biasing DMIP from Power Supply (Assumes BPF AC
Coupled Internally)

Table II lists gain control voltages and scale factors for power
supply voltages from 2.7 V to 5.5 V.

Alternatively, pin GREF can be tied to an external voltage
reference, V

, provided, for example, by an AD1582 (2.5 V)

R

or AD1580 (1.21 V) voltage reference, to provide supply­independent gain scaling of V

/75 (volts per dB). When using

R

the Analog Devices’ AD7013 and AD7015 baseband converters,
the external reference may also be provided by the reference
output of the baseband converter (Figure 37). For example, the
AD7015 baseband converter provides a V

of 1.23 V; when

R

connected to GREF the gain scaling is 16.4 mV/dB (60 dB/V).
An auxiliary DAC in the AD7015 can be used to generate the
MGC voltage. Since it uses the same reference voltage, the
numerical input to this DAC provides an accurate RSSI value
in digital form, no longer requiring the reference voltage to have
high absolute accuracy.

C

AD7013 OR

AD7015

IADC

QADC

IADC

QADC

REFOUT
BYPASS

AUX DAC

(AD7015)
(AD7013)

AD607

IOUT

QOUT

VMID

GREF

GAIN

R

R

C

10nF

1nF

AD607

IFOP

DMIP

VMID

R

T

BPF

R

T

C

BYPASS

b. Biasing DMIP from VMID (Assumes BPF AC Coupled
Internally)

Figure 36. Input and Output Matching of the Optional
Second IF Filter

Gain Scaling and RSSI

The AD607’s overall gain, expressed in decibels, is linear-in-dB
with respect to the AGC voltage V
all sections is maximum when V
sively up to V
V

– 0.8 V). The gain of all stages changes in parallel. The AD607

P

= 2.2 V (for VP = 3 V; in general, up to a limit

G

at pin GAIN. The gain of

G

is zero, and reduces progres-

G

features temperature-compensation of the gain scaling. The gain
control scaling is proportional to the reference voltage applied to
the pin GREF. When this pin is tied to the midpoint of the
supply (VMID), the scale is nominally 20 mV/dB (50 dB/V) for

= 3 V. Under these conditions, the lower 80 dB of gain range

V

P

(mixer plus IF) corresponds to a control voltage of 0.4 V ≤

≤ 2.0 V. The final centering of this 1.6 V range depends on

V

G

the insertion losses of the IF filters used. More generally, the gain
scaling using these connections is V

/150 (volts per dB), so

P

becomes 33.3 mV/dB (30 dB/V) using a 5 V supply, with a
proportional change in the AGC range, to 0.33 V ≤ V

≤ 3 V,

G

Figure 37. Interfacing the AD607 to the AD7013 or AD7015
Baseband Converters

I/Q Demodulators

Both demodulators (I and Q) receive their inputs at pin DMIP.
Internally, this single-sided input is actually differential; the
noninverting input is referenced to pin VMID. Each demodula­tor comprises a full-wave synchronous detector followed by a
2 MHz, two-pole low-pass filter, producing single-sided outputs
at pins IOUT and QOT. Using the I and Q demodulators for
IFs above 12 MHz is precluded by the 400 kHz to 12 MHz
response of the PLL used in the demodulator section. Pin DMIP
requires an external bias source at V

/2; Figure 38 shows

P

suggested methods.

Outputs IOUT and QOUT are centered at V

/2 and can swing

P

up to ±1.23 V even at the low supply voltage of 2.7 V. They can
therefore directly drive the RX ADCs in the AD7015 baseband
converter, which require an amplitude of 1.23 V to fully load
them when driven by a single-sided signal. The conversion gain of
the I and Q demodulators is 18 dB (X8), requiring a maxi­mum input amplitude at DMIP of ±150 mV for IFs > 3 MHz.

–16–

REV. B

AD607

BPF

VPOS

2R

T

2R

T

AD607

IFOP

DMIP

R

T

a. Biasing DMIP from Power Supply (Assumes BPF
AC-Coupled Internally)

AD607

IFOP

DMIP

VMID

R

T

BPF

R

T

C

BYPASS

b. Biasing DMIP from VMID (Assumes BPF
AC-Coupled Internally)

Figure 38. Suggested Methods for Biasing Pin DMIP

/2

at V

P

For IFs < 3 MHz, the on-chip low-pass filters (2 MHz cutoff)
do not attenuate the IF or feedthrough products; thus, the
maximum input voltage at DMIP must be limited to ±75 mV
to allow sufficient headroom at the I and Q outputs for not only
the desired baseband signal but also the unattenuated higher­order demodulation products. These products can be removed
by an external low-pass filter. In the case of IS54 applications
using a 455 kHz IF and the AD7013 baseband converter, a simple
one-pole RC filter with its corner above the modulation band­width is sufficient to attenuate undesired outputs.

Phase-Locked Loop

The demodulators are driven by quadrature signals that are
provided by a variable frequency quadrature oscillator (VFQO),
phase locked to a reference signal applied to pin FDIN. When
this signal is at the IF, inphase and quadrature baseband outputs

are generated at IOUT and QOUT, respectively. The quadra­ture accuracy of this VFQO is typically –1.2° at 10.7 MHz. The
PLL uses a sequential-phase detector that comprises low power
emitter-coupled logic and a charge pump (Figure 39).

IU~
40␮A

V

F

SEQUENTIAL

PHASE

DETECTOR

R

REFERENCE CARRIER

(FDIN AFTER LIMITING)

F

U

D

~

I

D

40␮A

VARIABLE-

FREQUENCY

QUADRATURE

OSCILLATOR

C

R

I-CLOCK

90ⴗ

Q-CLOCK

(ECL OUTPUTS)

Figure 39. Simplified Schematic of the PLL and
Quadrature VCO

The reference signal may be provided from an external source,
in the form of a high-level clock, typically a low level signal
(± 400 mV) since there is an input amplifier between FDIN and
the loop’s phase detector. For example, the IF output itself can
be used by connecting DMIP to FDIN, which will then provide
automatic carrier recover for synchronous AM detection and
take advantage of any post-IF filtering. Pin FDIN must be
biased at V

/2; Figure 41 shows suggested methods.

P

The VFQO operates from 400 kHz to 12 MHz and is controlled
by the voltage between VPOS and FLTR. In normal operation,
a series RC network, forming the PLL loop filter, is connected
from FLTR to ground. The use of an integral sample-hold
system ensures that the frequency-control voltage on pin FLTR
remains held during power-down, so reacquisition of the carrier
typically occurs in 16.5 µs.

In practice, the probability of a phase mismatch at power-up is
high, so the worst-case linear settling period to full lock needs
to be considered in making filter choices. This is typically 16.5 µs
at an IF of 10.7 MHz for a ±100 mV signal at DMIP and FDIN.

REV. B

Table II. AD607 Gain and Manual Gain Control Voltage vs. Power Supply Voltage

Power Supply GREF Gain Control
Voltage (= VMID) Scale Factor Scale Factor Voltage Input Range
(V) (V) (dB/V) (mV/dB) (V)

2.7 1.35 55.56 18.00 0.360–1.800

3.0 1.5 50.00 20.00 0.400–2.000

3.5 1.75 42.86 23.33 0.467–2.333

4.0 2.0 37.50 26.67 0.533–2.667

4.5 2.25 33.33 30.00 0.600–3.000

5.0 2.5 30.00 33.33 0.667–3.333

5.5 2.75 27.27 36.67 0.733–3.667

NOTE
Maximum gain occurs for gain control voltage = 0 V.

–17–

AD607

Bias System

The AD607 operates from a single supply, VP, usually of 3 V, at
a typical supply current of 8.5 mA at midgain and T = 27°C,
corresponding to a power consumption of 25 mW. Any voltage
from 2.7 V to 5.5 V may be used.

The bias system includes a fast-acting active-high CMOS­compatible power-up switch, allowing the part to idle at 550 µA
when disabled. Biasing is proportional-to-absolute-temperature
(PTAT) to ensure stable gain with temperature.

An independent regulator generates a voltage at the midpoint
of the supply (V

/2) which appears at the VMID pin, at a low

P

impedance. This voltage does not shut down, ensuring that the
major signal interfaces (e.g., mixer-to-IF and IF-to-demodulators)
remain biased at all times, thus minimizing transient disturbances
at power-up and allowing the use of substantial decoupling
capacitors on this node. The quiescent consumption of this
regulator is included in the idling current.

EXTERNAL
FREQUENCY
REFERENCE

VPOS

50k⍀

50k⍀

AD607

FDIN

a. Biasing FDIN from Supply when Using
External Frequency Reference

AD607

EXTERNAL
FREQUENCY
REFERENCE

50k⍀

C

BYPASS

FDIN

VMID

USING THE AD607

In this section, we will focus on a few areas of special impor­tance and include a few general application tips. As is true of
any wideband high gain component, great care is needed in PC
board layout. The location of the particular grounding points
must be considered with due regard to possibility of unwanted
signal coupling, particularly from IFOP to RFHI or IFHI or both.

The high sensitivity of the AD607 leads to the possibility that
unwanted local EM signals may have an effect on the perfor­mance. During system development, carefully-shielded test
assemblies should be used. The best solution is to use a fully­enclosed box enclosing all components, with the minimum
number of needed signal connectors (RF, LO, I and Q outputs)
in miniature coax form.

The I and Q output leads can include small series resistors
(about 100 Ω) inside the shielded box without significant loss
of performance, provided the external loading during testing
is light (that is, a resistive load of more than 20 kΩ and capaci­tances of a few picofarads). These help to keep unwanted RF
emanations out of the interior.

The power supply should be connected via a through-hole
capacitor with a ferrite bead on both inside and outside leads.
Close to the IC pins, two capacitors of different value should be
used to decouple the main supply (V

) and the midpoint supply

P

pin, VMID. Guidance on these matters is also generally included
in applications schematics.

Gain Distribution

As in all receivers, the most critical decisions in effectively using
the AD607 relate to the partitioning of gain between the various
subsections (Mixer, IF Amplifier, Demodulators) and the place­ment of filters, so as to achieve the highest overall signal-to-noise
ratio and lowest intermodulation distortion.

Figure 41 shows the main RF/IF signal path at maximum and
minimum signal levels.

b. Biasing FDIN from VMID when Using
External Frequency Reference

Figure 40. Suggested Methods for Biasing Pin FDIN

/2

at V

P

ⴞ54mV

MAX INPUT

RFHI

CONSTANT

(ⴞ50mV)

LOIP

–16dBm

ⴞ1.3V

MAX OUTPUT

MXOP IFHI

IMPEDANCE)

ⴞ54mV

MAX INPUT

IF BPF IF BPF

330⍀330⍀

(TYPICAL

Figure 41. Signal Levels for Minimum and Maximum Gain

ⴞ560mV

MAX OUTPUT

(VMID)

(LOCATION OF OPTIONAL

SECOND IF FILTER)

ⴞ154mV

MAX INPUT

DMIPIFOP

–18–

I

Q

ⴞ1.23V

MAX OUTPUT

IOUT

QOUT

REV. B

AD607

As noted earlier, the gain in dB is reduced linearly with the
voltage V
and IF strip gains vary with V

on the GAIN pin. Figure 42 shows how the mixer

G

when GREF is connected to

G

VMID (1.5 V) and a supply voltage of 3 V is used. Figure 43
shows how these vary when GREF is connected to a 1.23 V
reference.

90dB

80dB

70dB

60dB

50dB

40dB

30dB

20dB

10dB

0dB

01V2V

(67.5dB)

IF GAIN

(21.5dB)

0.4V 1.8V

NORMAL OPERATING RANGE

MIXER GAIN

V

(7.5dB)

(1.5dB)

2.2V

g

Figure 42. Gain Distribution for GREF = 1.5 V

90dB

80dB

70dB

60dB

50dB

40dB

30dB

20dB

10dB

0dB

01V2V

(67.5dB)

IF GAIN

(21.5dB)

MIXER GAIN

0.328V 1.64V

NORMAL OPERATING RANGE

(7.5dB)

(1.5dB)

V

g

Figure 43. Gain Distribution for GREF = 1.23 V

Using the AD607 with a Fast PRUP Control Signal

If the AD607 is used in a system in which the PRUP signal (Pin

3) is applied with a rise time less than 35 µs, anomalous behav-
ior occasionally occurs. The problem is intermittent, so it will
not occur every time the part is powered up under these condi­tions. It does not occur for any other normal operating conditions
when the PRUP signal has a rise time slower than 35 µs. Symp-
toms of operation with too fast a PRUP signal include low gain,
oscillations at the I or Q outputs of the device or no valid data
occurring at the output of the AD607. The problem causes no
permanent damage to the AD607, so it will often operate nor­mally when reset.

Fortunately, there is a very simple solution to the fast PRUP
problem. If the PRUP signal (Pin 3) is slowed down so that
the rise time of the signal edge is greater than 35 µs, the
anomalous behavior will not occur. This can be realized by a
simple RC circuit connected to the PRUP pin, where R = 4.7 kΩ
and C = 1.5 nF. This circuit is shown in Figure 44.

FROM PRUP

CONTROL SIGNAL

4.7k⍀

1.5nF

AD607

PRUP

Figure 44. Proper Configuration of AD607 PRUP Signal

All designs incorporating the AD607 should include this circuitry.

Note that connecting the PRUP pin to the supply voltage will
not eliminate the problem since the supply voltage may have a
rise time faster than 35 µs. With this configuration, the 4.7 kΩ
series R and 1.5 nF shunt C should be placed between the
supply and the PRUP pin as shown in Figure 44.

AD607 EVALUATION BOARD

The AD607 evaluation board (Figures 45 and 46) consists of an
AD607, ground plane, I/O connectors, and a 10.7 MHz band­pass filter. The RF and LO ports are terminated in 50 Ω to
provide a broadband match to external signal generators to
allow a choice of RF and LO input frequencies. The IF filter is
at 10.7 MHz and has 330 Ω input and output terminations; the
board is laid out to allow the user to substitute other filters for
other IFs.

The board provides SMA connectors for the RF and LO port
inputs, the demodulated I and Q outputs, the manual gain con­trol (MGC) input, the PLL input, and the power-up input. In
addition, the IF output is also available at an SMA connector;
this may be connected to the PLL input for carrier recovery to
realize synchronous AM and FM detection via the I and Q
demodulators, respectively. Table III lists the AD607 Evalua­tion Board’s I/O Connectors and their functions.

REV. B

–19–

AD607

FDIN

VPOS

GND

FDIN

PRUP

LO

RF

C18

SHORT

R14

C17

51.1⍀
10nF

MOD FOR LARGE MAGNITUDE

AC-COUPLED INPUT

C13 0

C14 0

VMID

R8

51.1⍀

50k⍀

R12
OPEN

C15

0.1␮F

R15

C11

10nF

R7

51.1⍀

R6

51.1⍀

C10
1nF

332⍀

332⍀

R13

50k⍀

JUMPER

C12

0.1␮F

R12

4.7k⍀

1.5nF

C9

1nF

R5

R3

C17

FDIN

R10

4.99k⍀

C16

JUMPER

R4
OPEN

C7

1nF

AD607 EVALUATION BOARD

VPOS

R11

OPEN

1nF

(AS RECEIVED)

Figure 45. Evaluation Board

FDIN
COM1

PRUP

LOIP

RFLO

RFHI

GREF

MXOP

VMID
IFHI

AD607

FDIN

C1

R19

0.1␮F

R1

1k⍀

C2 0.1␮F

R2
316⍀

C6

0.1␮F

C8

0.1␮F

C19

ANYTHING

C20

SHORT

VPS1

FLTR

IOUT

QOUT

VPS2

DMIP

IFOP

COM2

GAIN

IFLO

RSOURCE

MOD FOR DC-COUPLED INPUT

C3 10nF

47pF

C5
1nF

C4

VMID

OPEN

R16
OPEN

R17

I

Q

IF

GAIN

R18

OPEN

VPOS

FDIN

Figure 46a. Evaluation Board Layout, Topside

–20–

REV. B

AD607

Figure 46b. Evaluation Board Layout, Bottom Side

Table III. AD607 Evaluation Board Input and Output Connections

Reference Connector Approximate
Designation Type Description Coupling Signal Level Comments

J1 SMA Frequency DC ± 400 mV This pin needs to be biased at VMID

Detector Input and ac coupled when driven by an

external signal generator.

J2 SMA Power Up DC CMOS Logic Tied to Positive Supply by Jumper J10.

Level Input

J3 SMA LO Input AC –16 dBm Input is terminated in 50 Ω.

(± 50 mV)

J4 SMA RF Input AC –15 dBm max Input is terminated in 50 Ω.

(± 54 mV)

J5 SMA MGC Input DC 0.4 V to 2.0 V Jumper is set for Manual Gain Control

(3 V Supply) Input; See Table I for Control Voltage
(GREF = VMID) Values.

J6 SMA IF Output AC NA This signal level depends on the

AD607’s gain setting.

J7 SMA Q Output AC NA This signal level depends on the

AD607’s gain setting.

J8 SMA I Output AC NA This signal level depends on the

AD607’s gain setting.

J9 Jumper Ties GREF NA NA Sets gain-control Scale Factor (SF);

to VMID SF = 75/VMID in dB/V, where

VMID = VPOS/2.

J10 Jumper Ties Power-Up NA NA Remove to test Power-Up/-Down.

to Positive
Supply

T1 Terminal Pin Power Supply DC DC 2.7 V to 5.5 V

Positive Input Draws 8.5 mA at midgain connection.
(VPS1, VPS2)

T2 Terminal Pin Power Supply DC 0 V

Return (GND)

REV. B

–21–

AD607

In operation (Figure 47), the AD607 evaluation board draws
about 8.5 mA at midgain (59 dB). Use high impedance probes
to monitor signals from the demodulated I and Q outputs and
the IF output. The MGC voltage should be set such that the
signal level at DMIP does not exceed ±150 mV; signal levels

HP 6632A

PROGRAMMABLE

POWER SUPPLY

2.7V–6V

FLUKE 6082A

SYNTHESIZED

SIGNAL GENERATOR

240MHz

HP 8656A

SYNTHESIZED

SIGNAL GENERATOR

240.02MHz

IEEE CONTROLLER

MCL

ZFSC–2–1

COMBINER

HP 9920

HP9121

DISK DRIVE

RF

LO

HP 8656A

SYNTHESIZED

SIGNAL GENERATOR

229.3MHz

above this will overload the I and Q demodulators. The inser­tion loss between IFOP and DMIP is typically 3 dB if a simple
low-pass filter (R8 and C2) is used and higher if a reverse­terminated bandpass filter is used.

HP 3326

SYNTHESIZED

SIGNAL GENERATOR

10.710MHz

VPOS FDIN

AD607

EVALUATION

BOARD

I OUTPUT

Q OUTPUT

MGC

DATA PRECISION

DVC8200

PROGRAMMABLE

VOLTAGE SOURCE

TEKTRONIX

11402A

OSCILLOSCOPE

WITH 11A32

PLUGIN

Figure 47. Evaluation Board Test Setup

IEEE–488 BUS

–22–

REV. B

OUTLINE DIMENSIONS

20 11

101

0.295 (7.50)

0.271 (6.90)

0.311 (7.9)

0.301 (7.64)

0.212 (5.38)

0.205 (5.21)

PIN 1

SEATING

PLANE

0.008 (0.203)

0.002 (0.050)

0.07 (1.78)

0.066 (1.67)

0.0256
(0.65)

BSC

0.078 (1.98)

0.068 (1.73)

0.009 (0.229)

0.005 (0.127)

0.037 (0.94)

0.022 (0.559)

8°
0°

Dimensions shown in inches and (mm).

20-Lead Plastic SSOP (RS-20)

AD607

C00543a–0–10/00 (rev. B)

REV. B

PRINTED IN U.S.A.

–23–