Datasheet AD600JR-REEL, AD600JR, AD600JN, AD600AR-REEL7, AD600AR-REEL Datasheet (Analog Devices)

Dual, Low Noise, Wideband

a

FEATURES
Two Channels with Independent Gain Control

“Linear in dB” Gain Response

Two Gain Ranges:

AD600: 0 dB to 40 dB
AD602: –10 dB to +30 dB

Accurate Absolute Gain: 0.3 dB
Low Input Noise: 1.4 nV/√Hz
Low Distortion: –60 dBc THD at 1 V Output
High Bandwidth: DC to 35 MHz (–3 dB)
Stable Group Delay: 2 ns
Low Power: 125 mW (Max) per Amplifier
Signal Gating Function for Each Amplifier
Drives High-Speed A/D Converters
MIL-STD-883-Compliant and DESC Versions Available

APPLICATIONS
Ultrasound and Sonar Time-Gain Control
High-Performance Audio and RF AGC Systems
Signal Measurement

PRODUCT DESCRIPTION

The AD600 and AD602 dual channel, low noise variable gain
amplifiers are optimized for use in ultrasound imaging systems,
but are applicable to any application requiring very precise gain,
low noise and distortion, and wide bandwidth. Each indepen­dent channel provides a gain of 0 dB to +40 dB in the AD600
and –10 dB to +30 dB in the AD602. The lower gain of the
AD602 results in an improved signal-to-noise ratio at the out­put. However, both products have the same 1.4 nV/√Hz input
noise spectral density. The decibel gain is directly proportional
to the control voltage, is accurately calibrated, and is supply­and temperature-stable.

To achieve the difficult performance objectives, a proprietary
circuit form—the X-AMP
nel of the X-AMP comprises a variable attenuator of 0 dB to
–42.14 dB followed by a high speed fixed gain amplifier. In this
way, the amplifier never has to cope with large inputs, and can
benefit from the use of negative feedback to precisely define the
gain and dynamics. The attenuator is realized as a seven-stage
R-2R ladder network having an input resistance of 100 Ω, laser­trimmed to ± 2%. The attenuation between tap points is 6.02 dB;
the gain-control circuit provides continuous interpolation between
these taps. The resulting control function is linear in dB.

X-AMP is a registered trademark of Analog Devices, Inc.
*Patented.

®

—has been developed. Each chan-

Variable Gain Amplifiers

AD600/AD602*

FUNCTIONAL BLOCK DIAGRAM

GAT1

SCALING

REFERENCE

C1HI

V

G

C1LO

GAIN CONTROL

INTERFACE

A1HI

A1LO

0dB

500

–12.04dB

–6.02dB

R – 2R LADDER NETWORK

The gain-control interfaces are fully differential, providing an
input resistance of ~15 MΩ and a scale factor of 32 dB/V (that
is, 31.25 mV/dB) defined by an internal voltage reference. The
response time of this interface is less than 1 µs. Each channel
also has an independent gating facility that optionally blocks
signal transmission and sets the dc output level to within a few
millivolts of the output ground. The gating control input is TTL
and CMOS compatible.

The maximum gain of the AD600 is 41.07 dB, and that of the
AD602 is 31.07 dB; the –3 dB bandwidth of both models is
nominally 35 MHz, essentially independent of the gain. The
signal-to-noise ratio (SNR) for a 1 V rms output and a 1 MHz
noise bandwidth is typically 76 dB for the AD600 and 86 dB for
the AD602. The amplitude response is flat within ±0.5 dB from
100 kHz to 10 MHz; over this frequency range the group delay
varies by less than ±2 ns at all gain settings.

Each amplifier channel can drive 100 Ω load impedances with
low distortion. For example, the peak specified output is ±2.5 V
minimum into a 500 Ω load, or ±1 V into a 100 Ω load. For a
200 Ω load in shunt with 5 pF, the total harmonic distortion for
a ±1 V sinusoidal output at 10 MHz is typically –60 dBc.

The AD600J and AD602J are specified for operation from 0°C
to 70°C, and are available in both 16-lead plastic DIP (N) and
16-lead SOIC (R). The AD600A and AD602A are specified for
operation from –40°C to +85°C and are available in both 16-lead
cerdip (Q) and 16-lead SOIC (R).

The AD600S and AD602S are specified for operation from
–55°C to +125°C and are available in a 16-lead cerdip (Q)
package and are MIL-STD-883 compliant. The AD600S and
AD602S are also available under DESC SMD 5962-94572.

PRECISION PASSIVE
INPUT ATTENUATOR

–22.08dB

–18.06dB

–30.1dB

–36.12dB

–42.14dB

62.5

GATING

INTERFACE

RF2

2.24k(AD600)
694(AD602)

RF1
20

FIXED-GAIN

AMPLIFIER

41.07dB(AD600)

31.07(AD602)

A1OP

A1CM

REV. B

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] World Wide Web Site: http://www.analog.com
Fax: [phone] © Analog Devices, Inc., 2001

AD600/AD602–SPECIFICATIONS

(Each amplifier section, at TA = 25C, VS = 5 V, –625 mV ≤ VG ≤

+625 mV, RL = 500 , and CL = 5 pF, unless otherwise noted. Specifications for AD600 and AD602 are identical unless otherwise noted.)

AD600J/AD602J AD600A/AD602A

Parameter Conditions Min Typ Max Min Typ Max Unit

INPUT CHARACTERISTICS

Input Resistance Pins 2 to 3; Pins 6 to 7 98 100 102 95 100 105 Ω
Input Capacitance 22pF
Input Noise Spectral Density
Noise Figure R

Common-Mode Rejection Ratio f = 100 kHz 30 30 dB

OUTPUT CHARACTERISTICS

–3 dB Bandwidth V
Slew Rate 275 275 V/µs
Peak Output

2

Output Impedance f ≤ 10 MHz 2 2 Ω
Output Short-Circuit Current 50 50 mA
Group Delay Change vs. Gain f = 3 MHz; Full Gain Range ±2 ±2ns
Group Delay Change vs. Frequency VG = 0 V, f = 1 MHz to 10 MHz ±2 ±2ns
Total Harmonic Distortion RL= 200 Ω, V

ACCURACY

AD600

Gain Error 0 dB to 3 dB Gain 0 +0.5 +1 –0.5 +0.5 +1.5 dB

Maximum Output Offset Voltage3VG = –625 mV to +625 mV 10 50 10 65 mV
Output Offset Variation VG = –625 mV to +625 mV 10 50 10 65 mV

AD602

Gain Error –10 dB to –7 dB Gain 0 +0.5 +1 –0.5 +0.5 +1.5 dB

Maximum Output Offset Voltage3VG = –625 mV to +625 mV 5 30 10 45 mV
Output Offset Variation VG = –625 mV to +625 mV 5 30 10 45 mV

GAIN CONTROL INTERFACE

Gain Scaling Factor 3 dB to 37 dB (AD600); –7 dB to +27 dB (AD602) 31.7 32 32.3 30.5 32 33.5 dB/V
Common-Mode Range –0.75 +2.5 –0.75 +2.5 V
Input Bias Current 0.35 1 0.35 1 µA
Input Offset Current 10 50 10 50 nA
Differential Input Resistance Pins 1 to 16; Pins 8 to 9 15 15 MΩ
Response Rate Full 40 dB Gain Change 40 40 dB/µs

SIGNAL GATING INTERFACE

Logic Input “LO” (Output ON) 0.8 0.8 V
Logic Input “HI” (Output OFF) 2.4 2.4 V
Response Time ON to OFF, OFF to ON 0.3 0.3 µs
Input Resistance Pins 4 to 3; Pins 5 to 6 30 30 kΩ
Output Gated OFF

Output Offset Voltage ±10 100 ± 10 400 mV
Output Noise Spectral Density 65 65 nV/√Hz
Signal Feedthrough @ 1 MHz

AD600 –80 –80 dB
AD602 –70 –70 dB

POWER SUPPLY

Specified Operating Range ±4.75 ±5.25 ±4.75 ±5.25 V
Quiescent Current 11 12.5 22 28 mA

NOTES

1

Typical open or short-circuited input; noise is lower when system is set to maximum gain and input is short-circuited. This figure includes the effects of both voltage
and current noise sources.

2

Using resistive loads of 500 Ω or greater, or with the addition of a 1 kΩ pull-down resistor when driving lower loads.

3

The dc gain of the main amplifier in the AD600 is X113; thus an input offset of only 100 µV becomes an 11.3 mV output offset. In the AD602, the amplifier’s gain is
X35.7; thus, an input offset of 100 µV becomes a 3.57 mV output offset.

Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All min
and max specifications guaranteed, although only those shown in boldface are tested on all production units.

Specifications subject to change without notice.

1

= 50 Ω, Maximum Gain 5.3 5.3 dB

S

1.4 1.4 nV/√Hz

RS = 200 Ω, Maximum Gain 2 2 dB

= 100 mV rms 35 35 MHz

OUT

RL ≥ 500 Ω±2.5 ± 3 ±2.5 ±3V

= ±1 V Peak, Rpd = 1 kΩ –60 –60 dBc

OUT

3 dB to 37 dB Gain –0.5 ± 0.2 +0.5 –1.0 ± 0.2 +1.0 dB
37 dB to 40 dB Gain –1 –0.5 0 –1.5 –0.5 +0.5 dB

–7 dB to +27 dB Gain –0.5 ±0.2 +0.5 –1.0 ± 0.2 +1.0 dB
27 dB to 30 dB Gain –1 –0.5 0 –1.5 –0.5 +0.5 dB

–2–

REV. B

AD600/AD602

ABSOLUTE MAXIMUM RATINGS

1

Supply Voltage ±VS . . . . . . . . . . . . . . . . . . . . . . . . . . . ± 7.5 V

Input Voltages

Pins 1, 8, 9, 16 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ± V

S

Pins 2, 3, 6, 7 . . . . . . . . . . . . . . . . . . . . . . ± 2 V Continuous

. . . . . . . . . . . . . . . . . . . . . . . . . ±V

Pins 4, 5 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±V

for 10 ms

S

S

Internal Power Dissipation2 . . . . . . . . . . . . . . . . . . . . 600 mW

Operating Temperature Range (J) . . . . . . . . . . . . 0°C to 70°C

Operating Temperature Range (A) . . . . . . . . –40°C to +85°C

Operating Temperature Range (S) . . . . . . . –55°C to +125°C

Storage Temperature Range . . . . . . . . . . . . –65°C to +150°C

Lead Temperature Range (Soldering 60 sec) . . . . . . . . . 300°C

NOTES

1

Stresses above those listed under Absolute Maximum Ratings may cause perma-

nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.

2

Thermal Characteristics:

16-Lead Plastic Package: θJA = 85°C/W
16-Lead SOIC Package: θJA = 100°C/W
16-Lead Cerdip Package: θJA = 120°C/W

ORDERING GUIDE

Model Range Range Option

Gain Temperature Package

1

AD600AQ 0 dB to 40 dB –40°C to +85°C Q-16
AD600AR 0 dB to 40 dB –40°C to +85°C R-16
AD600AR-REEL 0 dB to 40 dB –40°C to +85°C 13" Reel
AD600AR-REEL7 0 dB to 40 dB –40°C to +85°C 7" Reel
AD600JN 0 dB to 40 dB 0°C to 70°C N-16
AD600JR 0 dB to 40 dB 0°C to 70°C R-16
AD600JR-REEL 0 dB to 40 dB 0°C to 70°C 13" Reel
AD600JR-REEL7 0 dB to 40 dB 0°C to 70°C 7" Reel
AD600SQ/883B

2

0 dB to 40 dB –55°C to +125°C Q-16

AD602AQ –10 dB to +30 dB –40°C to +85°C Q-16
AD602AR –10 dB to +30 dB –40°C to +85°C R-16
AD602AR-REEL –10 dB to +30 dB –40°C to +85°C 13" Reel
AD602AR-REEL7 –10 dB to +30 dB –40°C to +85°C 7" Reel
AD602JN –10 dB to +30 dB 0°C to 70°C N-16
AD602JR –10 dB to +30 dB 0°C to 70°C R-16
AD602JR-REEL –10 dB to +30 dB 0°C to 70°C 13" Reel
AD602JR-REEL7 –10 dB to +30 dB 0°C to 70°C 7" Reel
AD602SQ/883B3–10 dB to +30 dB –55°C to +150°C

NOTES

1

N = Plastic DIP; Q = Cerdip; R = Small Outline IC (SOIC).

2

Refer to AD600/AD602 Military data sheet. Also available as 5962-9457201MEA.

3

Refer to AD600/AD602 Military data sheet. Also available as 5962-9457202MEA.

Q-16

PIN FUNCTION DESCRIPTIONS

Pin Mnemonic Description

1 C1LO CH1 Gain-Control Input “LO” (Positive

Voltage Reduces CH1 Gain).

2 A1HI CH1 Signal Input “HI” (Positive Voltage

Increases CH1 Output).

3 A1LO CH1 Signal Input “LO” (Usually Taken to

CH1 Input Ground)

4 GAT1 CH1 Gating Input (A Logic “HI” Shuts Off

CH1 Signal Path).

5 GAT2 CH2 Gating Input (A Logic “HI” Shuts Off

CH2 Signal Path).

6 A2LO CH2 Signal Input “LO” (Usually Taken to

CH2 Input Ground).

7 A2HI CH2 Signal Input “HI” (Positive Voltage

Increases CH2 Output).

8 C2LO CH2 Gain-Control Input “LO” (Positive

Voltage Reduces CH2 Gain).

9 C2HI CH2 Gain-Control Input “HI” (Positive

Voltage Increases CH2 Gain).

10 A2CM CH2 Common (Usually Taken to CH2

Output Ground).
11 A2OP CH2 Output.
12 VNEG Negative Supply for Both Amplifiers.
13 VPOS Positive Supply for Both Amplifiers.
14 A1OP CH1 Output.
15 A1CM CH1 Common (Usually Taken to CH1

Output Ground).
16 C1HI CH1 Gain-Control Input “HI” (Positive

Voltage Increases CH1 Gain).

CONNECTION DIAGRAM

16-Lead Plastic DIP (N) Package

16-Lead Plastic SOIC (R) Package

16-Lead Cerdip (Q) Package

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

+

7

8

AD600 / AD602

A1

REF

A2

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

11

A2OP

10

A2CM

9

C2HI

CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection. Although
the AD600/AD602 features proprietary ESD protection circuitry, permanent damage may occur on
devices subjected to high-energy electrostatic discharges. Therefore, proper ESD precautions are
recommended to avoid performance degradation or loss of functionality.

REV. B

–3–

WARNING!

ESD SENSITIVE DEVICE

AD600/AD602–Typical Performance Characteristics

0.45

0.35

20dB

17dB

0

–45

–90

100k 1M 100M10M

FREQUENCY – Hz

–0.05

–0.15

GAIN ERROR – dB

–0.25

–0.35

–0.45

0.25

0.15

0.05

–0.5–0.7

GAIN CONTROL VOLTAGE – Volts

0.50.30.1–0.1–0.3

0.7

10dB

7dB

0

–45

–90

100k 1M 100M10M

FREQUENCY – Hz

TPC 1. Gain Error vs. Gain Control
Voltage

10.0

9.8

9.6

9.4

9.2

9.0

8.8

8.6

GROUP DELAY – ns

8.4

8.2

8.0

GAIN CONTROL VOLTAGE – Volts

0.7–0.5–0.7 0.50.30.1–0.1–0.3

TPC 4. AD600 and AD602 Typical
Group Delay vs. V

102

101

100

99

98

97

96

95

INPUT IMPEDANCE – 

94

93

92

100k 1M 100M10M

FREQUENCY – Hz

C

GAIN = 40dB

GAIN = 20dB

GAIN = 0dB

TPC 2. AD600 Frequency and Phase
Response vs. Gain

VG = 0V
10dB/DIV
CENTER
FREQ 1MHz
10kHz/DIV

TPC 5. Third Order Intermodula­tion Distortion, V

= 500

R

L

–1

–2

OUTPUT OFFSET VOLTAGE – mV

–3

–4

Ω

6

5

4

3

2

1

0

–0.5

–0.7

GAIN CONTROL VOLTAGE – Volts

OUT

AD602

= 2 V p-p,

AD600

0.7

0.50.1 0.3–0.3 –0.1

TPC 3. AD602 Frequency and Phase
Response vs. Gain

–1.0

–1.2

–1.4

–1.6

–1.8

–2.0

–2.2

–2.4

–2.6

–2.8

–3.0

–3.2

NEGATIVE OUTPUT VOLTAGE LIMIT – Volts

–3.4

50

0

LOAD RESISTANCE – 

20001000500200100

TPC 6. Typical Output Voltage vs.

Load Resistance (Negative Output

Swing Limits First)

1µs

100

90

OUTPUTINPUT

10

0%

1V VOUT

1V VC

TPC 7. Input Impedance vs.
Frequency

TPC 8. Output Offset vs. Gain
Control Voltage (Control Channel
Feedthrough)

–4–

TPC 9. Gain Control Channel

Response Time. Top: Output Volt-

age, 2 V max, Bottom: Gain Con-

trol Voltage V

= ±625 mV

C

REV. B

AD600/AD602

50mV

100

90

OUTPUT

10

0%

INPUT

5V

100ns

TPC 10. Gating Feedthrough to
Output, Gating Off to On

500mV

100

90

OUTPUTINPUT

10

0%

1V

200ns

TPC 13. Input Stage Overload
Recovery Time

50mV

100

90

OUTPUTINPUT

10

0%

5V

100ns

TPC 11. Gating Feedthrough to
Output, Gating On to Off

1V

100

90

OUTPUTINPUT

10

0%

200mV

500ns

TPC 14. Output Stage Overload
Recovery Time

1V

100

90

OUTPUT

10

0%

INPUT

100mV

TPC 12. Transient Response,
Medium and High Gain

500mV

100

90

OUTPUTINPUT

10

0%

1V

TPC 15. Transient Response
Minimum Gain

500ns

500ns

10

AD600: G = 20dB

5

AD602: G = 10dB
BOTH: V

0

V
R

–5

T

–10

–15

–20

CMRR – dB

–25

–30

–35

–40

1k 10k 100k 1M 10M 100M

= 100mV RMS

CM

= 5V

S

= 500

L

= 25C

A

AD600

FREQUENCY – Hz

AD602

TPC 16. CMRR vs. Frequency

20

10

0

–10

–20

–30

–40

PSRR – dB

–50

–60

–70

–80

100k 1M 100M10M

AD600

AD602

FREQUENCY – Hz

AD600: G = 40dB
AD602: G = 30dB
BOTH: R

V
R

TPC 17. PSRR vs. Frequency

= 500

L

= 0V

IN

= 50

S

10

AD600: CH1 G = 40dB, V

0

–10

–20

–30

–40

–50

CROSSTALK – dB

–60

–70

–80

–90

100k 1M 100M10M

CH2 G = 20dB, V

AD602: CH1 G = 30dB, V

CH2 G = 0dB, V

BOTH: V

CROSSTALK = 20log

= 1V RMS1, RS = 50,

OUT

= 500

R

L

FREQUENCY – Hz

= 0

IN

= 100mV

IN

= 0

IN

= 316mV

IN

CH1 V

OUT

{}

CH2 V

IN

AD600

AD602

TPC 18. Crosstalk Between A1
and A2 vs. Frequency

REV. B

–5–

AD600/AD602

THEORY OF OPERATION

The AD600 and AD602 have the same general design and
features. They comprise two fixed gain amplifiers, each pre­ceded by a voltage-controlled attenuator of 0 dB to 42.14 dB
with independent control interfaces, each having a scaling factor
of 32 dB per volt. The gain of each amplifier in the AD600 is
laser trimmed to 41.07 dB (X113), thus providing a control
range of –1.07 dB to 41.07 dB (0 dB to 40 dB with overlap),
while the AD602 amplifiers have a gain of 31.07 dB (X35.8)
and provide an overall gain of –11.07 dB to 31.07 dB (–10 dB to
30 dB with overlap).

The advantage of this topology is that the amplifier can use
negative feedback to increase the accuracy of its gain; also, since
the amplifier never has to handle large signals at its input, the
distortion can be very low. A further feature of this approach is
that the small-signal gain and phase response, and thus the
pulse response, are essentially independent of gain.

The following discussion describes the AD600. Figure 1 is a
simplified schematic of one channel. The input attenuator is a
seven-section R-2R ladder network, using untrimmed resistors
of nominally R = 62.5 Ω, which results in a characteristic resis­tance of 125 Ω ± 20%. A shunt resistor is included at the input
and laser trimmed to establish a more exact input resistance of
100 Ω ± 2%, which ensures accurate operation (gain and HP
corner frequency) when used in conjunction with external resis­tors or capacitors.

GAT1

PRECISION PASSIVE
INPUT ATTENUATOR

–22.08dB

–18.06dB

–30.1dB

–36.12dB

–42.14dB

62.5

GATING

INTERFACE

RF2

2.24k(AD600)
694(AD602)

RF1
20

FIXED-GAIN

AMPLIFIER

41.07dB(AD600)

31.07(AD602)

A1OP

A1CM

C1HI

C1LO

A1HI

A1LO

SCALING

REFERENCE

V

G

GAIN CONTROL

INTERFACE

0dB

500

–12.04dB

–6.02dB

R – 2R LADDER NETWORK

Figure 1. Simplified Block Diagram of Single Channel of
the AD600 and AD602

The nominal maximum signal at input A1HI is 1 V rms (±1.4 V
peak) when using the recommended ± 5 V supplies, although
operation to ±2 V peak is permissible with some increase in HF
distortion and feedthrough. Each attenuator is provided with a
separate signal “LO” connection, for use in rejecting common­mode, the voltage between input and output grounds. Circuitry
is included to provide rejection of up to ±100 mV.

The signal applied at the input of the ladder network is attenu­ated by 6.02 dB by each section; thus, the attenuation to each of
the taps is progressively 0, 6.02, 12.04, 18.06, 24.08, 30.1, 36.12
and 42.14 dB. A unique circuit technique is employed to interpo­late between these tap points, indicated by the “slider” in Figure
1, providing continuous attenuation from 0 dB to 42.14 dB.

It will help, in understanding the AD600, to think in terms of a
mechanical means for moving this slider from left to right; in
fact, it is voltage controlled. The details of the control interface
are discussed later. Note that the gain is at all times exactly
determined, and a linear decibel relationship is automatically
guaranteed between the gain and the control parameter which
determines the position of the slider. In practice, the gain devi­ates from the ideal law, by about ±0.2 dB peak (see, for example,
Figure 6).

Note that the signal inputs are not fully differential: A1LO and
A1CM (for CH1) and A2LO and A2CM (for CH2) provide
separate access to the input and output grounds. This recog­nizes the practical fact that even when using a ground plane,
small differences will arise in the voltages at these nodes. It is
important that A1LO and A2LO be connected directly to the
input ground(s); significant impedance in these connections will
reduce the gain accuracy. A1CM and A2CM should be con­nected to the load ground(s).

Noise Performance

An important reason for using this approach is the superior
noise performance that can be achieved. The nominal resistance
seen at the inner tap points of the attenuator is 41.7 Ω (one third
of 125 Ω), which exhibits a Johnson noise spectral density (NSD)
of 0.84 nV/√Hz (that is, √4kTR) at 27°C, which is a large fraction
of the total input noise. The first stage of the amplifier contrib­utes a further 1.12 nV/√Hz, for a total input noise of 1.4 nV/√Hz.

The noise at the 0 dB tap depends on whether the input is
short-circuited or open-circuited: when shorted, the minimum
NSD of 1.12 nV/√Hz is achieved; when open, the resistance
of 100 Ω at the first tap generates 1.29 nV/√Hz, so the noise
increases to a total of 1.71 nV/√Hz. (This last calculation would
be important if the AD600 were preceded, for example, by a
900 Ω resistor to allow operation from inputs up to ±10 V rms.
However, in most cases the low impedance of the source will
limit the maximum noise resistance.)

It will be apparent from the foregoing that it is essential to use a
low resistance in the design of the ladder network to achieve low
noise. In some applications this may be inconvenient, requiring
the use of an external buffer or preamplifier. However, very few
amplifiers combine the needed low noise with low distortion at
maximum input levels, and the power consumption needed to
achieve this performance is fundamentally required to be quite
high (due to the need to maintain very low resistance values
while also coping with large inputs). On the other hand, there is
little value in providing a buffer with high input impedance,
since the usual reason for this—the minimization of loading of a
high resistance source—is not compatible with low noise.

Apart from the small variations just discussed, the signal-to­noise (S/N) ratio at the output is essentially independent of the
attenuator setting, since the maximum undistorted output is 1 V
rms and the NSD at the output of the AD600 is fixed at 113
times 1.4 nV/√Hz, or 158 nV/√Hz. Thus, in a 1 MHz bandwidth,
the output S/N ratio would be 76 dB. The input NSD of the
AD600 and AD602 are the same, but because of the 10 dB
lower gain in the AD602’s fixed amplifier, its output S/N ratio is
10 dB better, or 86 dB in a 1 MHz bandwidth.

–6–

REV. B

AD600/AD602

The Gain-Control Interface

The attenuation is controlled through a differential, high imped­ance (15 MΩ) input, with a scaling factor which is laser trimmed
to 32 dB per volt, that is, 31.25 mV/dB. Each of the two amplifiers
has its own control interface. An internal bandgap reference
ensures stability of the scaling with respect to supply and tempera­ture variations, and is the only circuitry common to both channels.

When the differential input voltage VG = 0 V, the attenuator
“slider” is centered, providing an attenuation of 21.07 dB, thus
resulting in an overall gain of 20 dB (= –21.07 dB + 41.07 dB).
When the control input is –625 mV, the gain is lowered by
20 dB (= 0.625 × 32), to 0 dB; when set to +625 mV, the gain
is increased by 20 dB, to 40 dB. When this interface is over­driven in either direction, the gain approaches either –1.07 dB
(= –42.14 dB + 41.07 dB) or 41.07 dB (= 0 + 41.07 dB),
respectively.

The gain of the AD600 can thus be calculated using the follow­ing simple expression:

Gain (dB) = 32 V

where V

is in volts. For the AD602, the expression is:

G

Gain (dB) = 32 V

Operation is specified for V

+ 20 (1)

G

+ 10 (2)

G

in the range from –625 mV dc

G

to +625 mV dc. The high impedance gain-control input ensures
minimal loading when driving many amplifiers in multiple- channel
applications. The differential input configuration provides flexibil­ity in choosing the appropriate signal levels and polarities for
various control schemes.

For example, the gain-control input can be fed differentially to
the inputs, or single-ended by simply grounding the unused
input. In another example, if the gain is to be controlled by a
DAC providing a positive only ground referenced output, the
“Gain Control LO” pin (either C1LO or C2LO) should be
biased to a fixed offset of +625 mV, to set the gain to 0 dB
when “Gain Control HI” (C1HI or C2HI) is at zero, and to
40 dB when at 1.25 V.

It is a simple matter to include a voltage divider to achieve other
scaling factors. When using an 8-bit DAC having a FS output of
+2.55 V (10 mV/bit) a divider ratio of 1.6 (generating 6.25 mV/
bit) would result in a gain setting resolution of 0.2 dB/ bit. Later,
we will discuss how the two sections of an AD600 or AD602
may be cascaded, when various options exist for gain control.

Signal-Gating Inputs

Each amplifier section of the AD600 and AD602 is equipped

with a signal gating function, controlled by a TTL or CMOS
logic input (GAT1 or GAT2). The ground references for these
inputs are the signal input grounds A1LO and A2LO, respec­tively. Operation of the channel is unaffected when this input is
LO or left open-circuited. Signal transmission is blocked when
this input is HI. The dc output level of the channel is set to
within a few millivolts of the output ground (A1CM or A2CM),
and simultaneously the noise level drops significantly. The
reduction in noise and spurious signal feedthrough is useful
in ultrasound beam-forming applications, where many amplifier
outputs are summed.

Common-Mode Rejection

A special circuit technique is used to provide rejection of volt­ages appearing between input grounds (A1LO and A2LO) and
output grounds (A1CM and A2CM). This is necessary because
of the “op amp” form of the amplifier, as shown in Figure 1.
The feedback voltage is developed across the resistor RF1 (which,
to achieve low noise, has a value of only 20 Ω). The voltage
developed across this resistor is referenced to the input common,
so the output voltage is also referred to that node.

To provide rejection of this common voltage, an auxiliary ampli­fier (not shown) is included, which senses the voltage difference
between input and output commons and cancels this error com­ponent. Thus, for zero differential signal input between A1HI
and A1LO, the output A1OP simply follows the voltage at A1CM.
Note that the range of voltage differences which can exist between
A1LO and A1CM (or A2LO and A2CM) is limited to about
±100 mV. Figure 50 (one of the typical performance curves at
the end of this data sheet) shows typical common-mode rejec­tion ratio versus frequency.

ACHIEVING 80 dB GAIN RANGE

The two amplifier sections of the X-AMP can be connected
in series to achieve higher gain. In this mode, the output of
A1 (A1OP and A1CM) drives the input of A2 via a high-pass
network (usually just a capacitor) that rejects the dc offset. The
nominal gain range is now –2 dB to +82 dB for the AD600 or
–22 dB to +62 dB for the AD602.

There are several options in connecting the gain-control inputs.
The choice depends on the desired signal-to-noise ratio (SNR)
and gain error (output ripple). The following examples feature
the AD600; the arguments generally apply to the AD602, with
appropriate changes to the gain values.

Sequential Mode (Maximum S/N Ratio)

In the sequential mode of operation, the SNR is maintained at

its highest level for as much of the gain control range possible,
as shown in Figure 2. Note here that the gain range is 0 dB to
80 dB. Figure 3 shows the general connections to accomplish
this. Both gain-control inputs, C1HI and C2HI, are driven in
parallel by a positive only, ground referenced source with a
range of 0 V to 2.5 V.

85

80

75

70

65

60

55

50

S/N RATIO – dB

45

40

35

30

–0.5

0.0

V

G

3.0

2.52.01.51.00.5

REV. B

Figure 2. S/N Ratio vs. Control Voltage Sequential Control
(1 MHz Bandwidth)

–7–

AD600/AD602

A1

–40.00dB

–40.00dB

C1HI C1LO

V

G1

VO1 = 0.592V

–0.51dB

C1HI C1LO

V

G1

VO1 = 0.592V

0dB

C1HI C1LO

V

G1

VO1 = 0.592V

–0.51dB

0dB

41.07dB

41.07dB

41.07dB

1.07dB

(a)

40.56dB

(b)

41.07dB

(c)

= 1.25V

V

C

V

C

INPUT

= 0V

V

C

INPUT

INPUT

= 2.5V

0dB

0dB

0dB

Figure 3. AD600 Gain Control Input Calculations for Sequential Control Operation

The gains are offset (Figure 4) such that A2’s gain is increased
only after A1’s gain has reached its maximum value. Note that
for a differential input of –700 mV or less, the gain of a single
amplifier (A1 or A2) will be at its minimum value of –1.07 dB;
for a differential input of +700 mV or more, the gain will be at
its maximum value of 41.07 dB. Control inputs beyond these
limits will not affect the gain and can be tolerated without dam­age or foldover in the response. See the Specifications Section of
this data sheet for more details on the allowable voltage range.
The gain is now

Gain (dB) = 32 V

where V

is the applied control voltage.

C

+41.07dB

20dB

C

A1 A2

40.56dB

*

+38.93dB

(3)

*

+1.07dB

GAIN

(dB)

0 0.625 1.25 1.875 2.5
020406080–2.14 82.14

*

Figure 4. Explanation of Offset Calibration for Sequential
Control

–0.56dB

–1.07dB

0.592 1.908

VC (V)

GAIN OFFSET OF 1.07dB, OR 33.44mV

A2

–41.07dB

–42.14dB

C1HI C1LO

V

G2

VO2 = 1.908V

–1.07dB

–41.63dB

C1HI C1LO

V

G2

V

O2

–2.14dB

C1HI C1LO

V

G2

V

O2

41.07dB

41.07dB

= 1.908V

38.93dB

41.07dB

= 1.908V

OUTPUT
0dB

OUTPUT
40dB

OUTPUT
80dB

When VC is set to zero, VG1 = –0.592 V and the gain of A1 is
+1.07 dB (recall that the gain of each amplifier section is 0 dB
for V

= 625 mV); meanwhile, VG2 = –1.908 V so the gain of

G

A2 is –1.07 dB. The overall gain is thus 0 dB (see Figure 3a).
When V
sets the gain of A1 to 40.56 dB, while V

= +1.25 V, VG1 = 1.25 V– 0.592 V = +0.658 V, which

C

= 1.25 V – 1.908 V =

G2

–0.658 V, which sets A2’s gain at –0.56 dB. The overall gain is
now 40 dB (see Figure 3b). When V

= +2.5 V, the gain of A1

C

is 41.07 dB and that of A2 is 38.93 dB, resulting in an overall
gain of 80 dB (see Figure 3c). This mode of operation is further
clarified by Figure 5, which is a plot of the separate gains of A1
and A2 and the overall gain versus the control voltage. Figure 6
is a plot of the gain error of the cascaded amplifiers versus the
control voltage.

Parallel Mode (Simplest Gain-Control Interface)

In this mode, the gain-control voltage is applied to both inputs
in parallel—C1HI and C2HI are connected to the control volt­age, and C1LO and C2LO are optionally connected to an offset
voltage of 0.625 V. The gain scaling is then doubled to 64 dB/V,
requiring only 1.25 V for an 80 dB change of gain. The ampli­tude of the gain ripple in this case is also doubled, as shown in
Figure 7, and the instantaneous signal-to-noise ratio at the
output of A2 decreases linearly as the gain is increased (Figure 8).

Low Ripple Mode (Minimum Gain Error)

As can be seen in Figures 6 and 7, the output ripple is periodic.
By offsetting the gains of Al and A2 by half the period of the
ripple, or 3 dB, the residual gain errors of the two amplifiers
can be made to cancel. Figure 9 shows the much lower gain rip
ple when configured in this manner. Figure 10 plots the S/N
ratio as a function of gain; it is very similar to that in the
“Parallel Mode.”

–8–

REV. B

OVERALL GAIN – dB

75

30

1.4

40

35

0.20.0

45

50

55

60

65

70

1.21.00.80.60.4

S/N RATIO – dB

V

C

80

35

1.4

45

40

0.20.0

50

55

60

65

70

75

1.21.00.80.60.4

V

C

S/N RATIO – dB

–10

AD600/AD602

90

80

70

60

50

40

30

20

10

0

–0.5

A1

0.0

COMBINED

V

C

A2

2.52.01.51.00.5

3.0

Figure 5. Plot of Separate and Overall Gains in Sequential

Figure 8. SNR for Cascaded Stages—Parallel Control

Control

5

4

3

2

1

0

–1

–2

–3

GAIN ERROR – dB

–4

–5

–6

–7

–8

–0.5

0.0
V

C

2.0 2.51.51.00.5

3.0

Figure 6. Gain Error for Cascaded Stages—Sequential
Control

5

4

3

2

1

0

–1

–2

REV. B

GAIN ERROR – dB

–3

–4

–5

0

–0.1

V

C

1.21.00.80.40.2 0.6

Figure 7. Gain Error for Cascaded Stages—Parallel
Control

–9–

1.2

1.0

0.8

0.6

0.4

0.2

0.0

–0.2

–0.4

GAIN ERROR – dB

–0.6

–0.8

–1.0

–1.2

0.0

0.1
V

C

1.3

1.21.11.00.90.70.60.50.40.30.2 0.8

Figure 9. Gain Error for Cascaded Stages—Low Ripple
Mode

Figure 10. ISNR vs. Control Voltage—Low Ripple Mode

AD600/AD602

APPLICATIONS

The full potential of any high performance amplifier can only be
realized by careful attention to details in its applications. The
following pages describe fully tested circuits in which many such
details have already been considered. However, as is always true
of high accuracy, high speed analog circuits, the schematic is
only part of the story; this is no less true for the AD600 and
AD602. Appropriate choices in the overall board layout and the
type and placement of power supply decoupling components are
very important. As explained previously, the input grounds
A1LO and A2LO must use the shortest possible connections.

The following circuits show examples of time-gain control for
ultrasound and for sonar, methods for increasing the output
drive, and AGC amplifiers for audio and RF/IF signal process­ing using both peak and rms detectors. These circuits also
illustrate methods of cascading X-AMPs for either maintaining
the optimal S/N ratio or maximizing the accuracy of the gain­control voltage for use in signal measurement. These AGC
circuits may be modified for use as voltage-controlled amplifiers
for use in sonar and ultrasound applications by removing the
detector and substituting a DAC or other voltage source for
supplying the control voltage.

Time-Gain Control (TGC) and Time-Variable Gain (TVG)

Ultrasound and sonar systems share a similar requirement: both
need to provide an exponential increase in gain in response to a
linear control voltage, that is, a gain control that is “linear in
dB.” Figure 11 shows the AD600/AD602 configured for a con­trol voltage ramp starting at –625 mV and ending at +625 mV
for a gain-control range of 40 dB. For simplicity, only the A1
connections are shown. The polarity of the gain-control voltage
may be reversed and the control voltage inputs C1HI and C1LO
reversed to achieve the same effect. The gain-control voltage
can be supplied by a voltage-output DAC such as the AD7242,
which contains two complete DACs, operates from ±5 V supplies,
has an internal reference of 3 V, and provides ±3 V of output
swing. As such it is well-suited for use with the AD600/AD602,
needing only a few resistors to scale the output voltage of the
DACs to the levels needed by the AD600/AD602.

CONTROL VOLTAGE,

V

+625mV

0dB 40dB

G

–625mV

A1
GAIN

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

VOLTAGE-OUTPUT

DAC

1

2

+

3

–

4

5

–

6

+

7

8

AD600 or AD602

A1

A2

REF

V

G

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

+5V

–5V

Figure 11. The Simplest Application of the X-AMP Is as a
TGC or TVG Amplifier in Ultrasound or Sonar. Only the A1
Connections Are Shown for Simplicity.

Increasing Output Drive

The AD600/AD602’s output stage has limited capability for
negative-load driving capability. For driving loads less than
500 Ω, the load drive may be increased by about 5 mA by con­necting a 1 kΩ pull-down resistor from the output to the negative
supply (Figure 12).

Driving Capacitive Loads

For driving capacitive loads of greater than 5 pF, insert a 10 Ω
resistor between the output and the load. This lowers the possi­bility of oscillation.

GAIN-CONTROL

VOLTAG E

C1LO

1

V

IN

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

2

+

–

–

+

A1

REF

A2

AD600

3

4

5

6

7

8

C1HI

16

A1CM

15

A1OP

14

VPOS

+5V

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

1k

–5V

ADDED

PULL-DOWN

RESISTOR

Figure 12. Adding a 1 kΩ Pull-Down Resistor Increases the
X-AMP’s Output Drive by About 5 mA. Only the A1 Con­nections Are Shown for Simplicity.

Realizing Other Gain Ranges

Larger gain ranges can be accommodated by cascading amplifi­ers. Combinations built by cascading two amplifiers include
–20 dB to +60 dB (using one AD602), –10 dB to +70 dB (1/2
of an AD602 followed by 1/2 of an AD600), and 0 dB to 80 dB
(one AD600). In multiple-channel applications, extra protection
against oscillations can be provided by using amplifier sections
from different packages.

An Ultralow Noise VCA

The two channels of the AD600 or AD602 may be operated in
parallel to achieve a 3 dB improvement in noise level, providing
1 nV/√Hz without any loss of gain accuracy or bandwidth.

In the simplest case, as shown in Figure 13, the signal inputs
A1HI and A2HI are tied directly together, the outputs A1OP
and A2OP are summed via R1 and R2 (100 Ω each), and the
control inputs C1HI/C2HI and C1LO/C2LO operate in parallel.
Using these connections, both the input and output resistances
are 50 Ω. Thus, when driven from a 50 Ω source and termi­nated in a 50 Ω load, the gain is reduced by 12 dB, so the gain
range becomes –12 dB to +28 dB for the AD600 and –22 dB to
+18 dB for the AD602. The peak input capability remains unaf­fected (1 V rms at the IC pins, or 2 V rms from an unloaded
50 Ω source). The loading on each output, with a 50 Ω load, is
effectively 200 Ω, because the load current is shared between
the two channels, so the overall amplifier still meets its specified
maximum output and distortion levels for a 200 Ω load. This
amplifier can deliver a maximum sine wave power of +10 dBm
to the load.

–10–

REV. B

AD600/AD602

GAIN-CONTROL

VOLTAG E

V

G

– +

C1LO

1

A1HI

2

+

A1LO

3

GAT1

V

IN

GAT2

A2LO

A2HI

C2LO

4

5

6

7

8

A1

–

REF

–

A2

+

AD600 or AD602

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

+5V

–5V

100

100

V

50

OUT

Figure 13. An Ultralow Noise VCA Using the AD600 or
AD602

A Low Noise, 6 dB Preamplifier

In some ultrasound applications, the user may wish to use a
high input impedance preamplifier to avoid the signal attenua­tion that would result from loading the transducer by the 100 Ω
input resistance of the X-AMP. High gain cannot be tolerated,
because the peak transducer signal is typically ±0.5 V, while the
peak input capability of the AD600 or AD602 is only slightly
more than ±1 V. A gain of two is a suitable choice. It can be shown
that if the preamplifier’s overall referred-to-input (RTI) noise is
to be the same as that due to the X-AMP alone (1.4 nV/√Hz),
then the input noise of a X2 preamplifier must be √(3/4) times
as large, that is, 1.2 nV/√Hz.

+5V

R1

49.9

R2

174

1F

R4

42.2

V

IN

R5

42.2

1F

R7

174

R8

49.9

–5V

+5V

–5V

R3
562

R6
562

1F

Q1
MRF904

INPUT
GROUND

Q2
MM4049

1F

0.1F

0.1F

100
R

OUTPUT

GROUND

OF X AMP

IN

Figure 14. A Low Noise Preamplifier for the AD600 and
AD602

An inexpensive circuit, using complementary transistor types
chosen for their low r

, is shown in Figure 14. The gain is

bb

determined by the ratio of the net collector load resistance to
the net emitter resistance, that is, it is an open-loop amplifier.
The gain will be X2 (6 dB) only into a 100 Ω load, assumed to
be provided by the input resistance of the X-AMP; R2 and R7
are in shunt with this load, and their value is important in defin­ing the gain. For small-signal inputs, both transistors contribute
an equal transconductance, which is rendered less sensitive to
signal level by the emitter resistors R4 and R5, which also play a
dominant role in setting the gain.

This is a Class AB amplifier. As VIN increases in a positive direc­tion, Q1 conducts more heavily and its r
that of Q2 increases. Conversely, more negative values of V

becomes lower while

e

IN

result in the re Of Q2 decreasing, while that of Q1 increases.
The design is chosen such that the net emitter resistance is
essentially independent of the instantaneous value of V

, result-

IN

ing in moderately low distortion. Low values of resistance and
moderately high bias currents are important in achieving the low
noise, wide bandwidth, and low distortion of this preamplifier.
Heavy decoupling prevents noise on the power supply lines from
being conveyed to the input of the X-AMP.

Table I. Measured Preamplifier Performance

Measurement Value Unit

Gain (f = 30 MHz) 6 dB
Bandwidth (–3 dB) 250 MHz
Input Signal for

1 dB Compression 1 V p-p

Distortion

= 200 mV p-p HD2 0.27 %

V

IN

HD3 0.14 %

= 500 mV p-p HD2 0.44 %

V

IN

HD3 0.58 %

System Input Noise 1.03 nV/√Hz

Spectral Density (NSD)
(Preamp plus X-AMP)

Input Resistance 1.4 kΩ
Input Capacitance 15 pF
Input Bias Current ±150 µA
Power Supply Voltage ±5V
Quiescent Current 15 mA

A Low Noise AGC Amplifier with 80 dB Gain Range

Figure 15 provides an example of the ease with which the AD600
can be connected as an AGC amplifier. A1 and A2 are cascaded,
with 6 dB of attenuation introduced by the 100 Ω resistor R1,
while a time constant of 5 ns is formed by C1 and the 50 Ω of
net resistance at the input of A2. This has the dual effect of (a)
lowering the overall gain range from {0 dB to 80 dB} to {6 dB to
74 dB} and (b) introducing a single-pole low-pass filter with a
–3 dB frequency of about 32 MHz. This ensures stability at the
maximum gain for a slight reduction in the overall bandwidth.
The capacitor C4 blocks the small dc offset voltage at the out­put of A1 (which might otherwise saturate A2 at its maximum
gain) and introduces a high pass corner at about 8 kHz, useful
in eliminating low frequency noise and spurious signals which
may be present at the input.

REV. B

–11–

AD600/AD602

RF

INPUT

5V

R3

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

46.4k

R4

3.74k

1

2

+

3

–

4

5

–

6

+

7

8

AD600

A1

REF

A2

V

G

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

C4

0.1F

+5V DEC

–5V DEC

R1

100

100pF

C1

1F

15pF

C2

C3

AD590

806

1%

5V

300A
(at 300K)

Q1
2N3904

+

R2

V

PTAT

–

RF
OUTPUT

Figure 15. This Accurate HF AGC Amplifier Uses Just Three Active Components

+5V

FB

+5V DEC

–5V DEC

POWER SUPPLY

DECOUPLING NETWORK

0.1F

0.1F

FB

–5V

A simple half-wave detector is used, based on Q1 and R2. The
average current into capacitor C2 is just the difference between
the current provided by the AD590 (300 µA at 300 K, 27°C)
and the collector current of Q1. In turn, the control voltage V

G

is the time integral of this error current. When VG (and thus the
gain) is stable, the rectified current in Q1 must, on average, exactly
balance the current in the AD590. If the output of A2 is too
small to do this, V

will ramp up, causing the gain to increase,

G

until Q1 conducts sufficiently. The operation of this control
system will now be described in detail.

First, consider the particular case where R2 is zero and the out­put voltage V

is a square wave at, say, 100 kHz, that is, well

OUT

above the corner frequency of the control loop. During the time

is negative, Q1 conducts; when V

V

OUT

is positive, it is cut

OUT

off. Since the average collector current is forced to be 300 µA, and
the square wave has a 50% duty-cycle, the current when conduct­ing must be 600 µA. With R2 omitted, the peak value of V

OUT

would be just the VBE of Q1 at 600 µA (typically about 700 mV)
or 2 V

peak-to-peak. This voltage, hence the amplitude at which

BE

the output stabilizes, has a strong negative temperature coefficient
(TC), typically –1.7 mV/°C. While this may not be troublesome
in some applications, the correct value of R2 will render the
output stable with temperature.

To understand this, first note that the current in the AD590 is
closely proportional to absolute temperature (PTAT). (In fact,
this IC is intended for use as a thermometer.) For the moment,
continue to assume that the signal is a square wave. When Q1 is
conducting, V

is the now the sum of VBE and a voltage which

OUT

is PTAT and which can be chosen to have an equal but opposite
TC to that of the base-to-emitter voltage. This is actually noth­ing more than the “bandgap voltage reference” principle in thinly
veiled disguise! When we choose R2 such that the sum of the
voltage across it and the V
age of about 1.2 V, V

of Q1 is close to the bandgap volt-

BE

will be stable over a wide range of

OUT

temperatures, provided, of course, that Q1 and the AD590
share the same thermal environment.

Since the average emitter current is 600 µA during each half-cycle
of the square wave, a resistor of 833 Ω would add a PTAT volt­age of 500 mV at 300 K, increasing by 1.66 mV/°C. In practice,

the optimum value of R2 will depend on the transistor used, and,
to a lesser extent, on the waveform for which the temperature
stability is to be optimized; for the devices shown and sine wave
signals, the recommended value is 806 Ω. This resistor also
serves to lower the peak current in Q1 and the 200 Hz LP
filter it forms with C2 helps to minimize distortion due to ripple

Note that the output amplitude under sine wave condi-

in V

G.

tions will be higher than for a square wave, since the average
value of the current for an ideal rectifier would be 0.637 times as
large, causing the output amplitude to be 1.88 (= 1.2/0.637) V, or

1.33 V rms. In practice, the somewhat nonideal rectifier results
in the sine wave output being regulated to about 1.275 V rms.

An offset of 375 mV is applied to the inverting gain-control
inputs C1LO and C2LO. Thus the nominal –625 mV to +625 mV
range for V

is translated upwards (at VG´) to –0.25 V for mini-

G

mum gain to +1 V for maximum gain. This prevents Q1 from
going into heavy saturation at low gains and leaves sufficient
“headroom” of 4 V for the AD590 to operate correctly at high
gains when using a 5 V supply.

In fact, the 6 dB interstage attenuator means that the overall
gain of this AGC system actually runs from –6 dB to +74 dB.
Thus, an input of 2 V rms would be required to produce a 1 V
rms output at the minimum gain, which exceeds the 1 V rms
maximum input specification of the AD600. The available gain
range is therefore 0 dB to 74 dB (or, X1 to X5000). Since the
gain scaling is 15.625 mV/dB (because of the cascaded stages)
the minimum value of V

´ is actually increased by 6 × 15.625 mV,

G

or about 94 mV, to –156 mV, so the risk of saturation in Q1
is reduced.

The emitter circuit of Q1 is somewhat inductive (due its finite f

t

and base resistance). Consequently, the effective value of R2
increases with frequency. This would result in an increase in the
stabilized output amplitude at high frequencies, but for the addi­tion of C3, determined experimentally to be 15 pF for the 2N3904
for maximum response flatness. Alternatively, a faster transistor
can be used here to reduce HF peaking. Figure 16 shows the ac
response at the stabilized output level of about 1.3 V rms. Fig­ure 17 demonstrates the output stabilization for sine wave
inputs of 1 mV to 1 V rms at frequencies of 100 kHz, 1 MHz
and 10 MHz.

–12–

REV. B

3dB

AGC OUTPUT CHANGE – dB

0.1

1

FREQUENCY – MHz

10

100

Figure 16. AC Response at the Stabilized Output Level
of 1.3 V RMS

+0.2

0

–0.2

–0.4

RELATIVE OUTPUT – dB

0.001 0.01 10.1
INPUT AMPLITUDE – V RMS

100kHz

1MHz

10MHz

Figure 17. Output Stabilization vs. RMS Input for
Sine Wave Inputs at 100 kHz, 1 MHz, and 10 MHz

While the “bandgap” principle used here sets the output ampli­tude to 1.2 V (for the square wave case), the stabilization point
can be set to any higher amplitude, up to the maximum output
of ± (V

– 2) V which the AD600 can support. It is only neces-

S

sary to split R2 into two components of appropriate ratio whose
parallel sum remains close to the zero-TC value of 806 Ω. This
is illustrated in Figure 18, which shows how the output can be
raised, without altering the temperature stability.

5V

R2A

300A
(at 300K)

Q1
2N3904

R2B

+

R2 = R2A R2B ≈ 806

V

PTAT

–

RF
OUTPUT

TO AD600 PIN 16

TO AD600 PIN 11

1F

15pF

AD590

C2

C3

Figure 18. Modification in Detector to Raise Output to
2 V RMS

AD600/AD602

A Wide Range, RMS-Linear dB Measurement System
(2 MHz AGC Amplifier with RMS Detector)

Monolithic rms-dc converters provide an inexpensive means to
measure the rms value of a signal of arbitrary waveform, and
they also may provide a low accuracy logarithmic (“decibel­scaled”) output. However, they have certain shortcomings. The
first of these is their restricted dynamic range, typically only
50 dB. More troublesome is that the bandwidth is roughly pro­portional to the signal level; for example, the AD636 provides a
3 dB bandwidth of 900 kHz for an input of 100 mV rms, but
has a bandwidth of only 100 kHz for a 10 mV rms input. Its
logarithmic output is unbuffered, uncalibrated and not stable
over temperature; considerable support circuitry, including at
least two adjustments and a special high TC resistor, is required
to provide a useful output.

All of these problems can be eliminated using an AD636 as
merely the detector element in an AGC loop, in which the differ­ence between the rms output of the amplifier and a fixed dc
reference are nulled in a loop integrator. The dynamic range
and the accuracy with which the signal can be determined are
now entirely dependent on the amplifier used in the AGC system.
Since the input to the rms-dc converter is forced to a constant
amplitude, close to its maximum input capability, the bandwidth is
no longer signal dependent. If the amplifier has an exactly expo­nential (“linear-dB”) gain-control law, its control voltage VG is
forced by the AGC loop to be have the general form:

V

V

OUT=VSCALE

log 10

Figure 19 shows a practical wide dynamic range rms-responding
measurement system using the AD600. Note that the signal
output of this system is available at A2OP, and the circuit can
be used as a wideband AGC amplifier with an rms-responding
detector. This circuit can handle inputs from 100 µV to 1 V rms
with a constant measurement bandwidth of 20 Hz to 2 MHz,
limited primarily by the AD636 rms converter. Its logarithmic
output is a loadable voltage, accurately calibrated to 100 mV/dB,
or 2 V per decade, which simplifies the interpretation of the
reading when using a DVM, and is arranged to be –4 V for an
input of 100 µV rms input, zero for 10 mV, and +4 V for a
1 V rms input. In terms of Equation 4, V
V

is 2 V.

SCALE

Note that the peak “log output” of ± 4 V requires the use of
±6 V supplies for the dual op amp U3 (AD712) although lower
supplies would suffice for the AD600 and AD636. If only ±5 V
supplies are available, it will be either necessary to use a reduced
value for V

(say 1 V, in which case the peak output would

SCALE

be only ±2 V) or restrict the dynamic range of the signal to
about 60 dB.

As in the previous case, the two amplifiers of the AD600 are
used in cascade. However, the 6 dB attenuator and low-pass
filter found in Figure 1 are replaced by a unity gain buffer
amplifier U3A, whose 4 MHz bandwidth eliminates the risk of
instability at the highest gains. The buffer also allows the use of
a high impedance coupling network (C1/R3) which introduces a
high-pass corner at about 12 Hz. An input attenuator of 10 dB
(X0.316) is now provided by R1 + R2 operating in conjunction
with the AD600’s input resistance of 100 Ω. The adjustment
provides exact calibration of the logarithmic intercept V
critical applications, but R1 and R2 may be replaced by a fixed

IN (RMS )

V

REF

is 10 mV and

REF

REF

(4)

in

REV. B

–13–

AD600/AD602

INPUT

1V RMS

MAX

(SINE WAVE)

R1

115

R2 200

U3A

1/2

AD712

15.625mV/dB

Figure 19. The Output of This Three-IC Circuit Is Proportional to the Decibel Value of the RMS Input

CAL
0dB

133k

V

G

V

RMS

C2
2F

–6V

DEC

AF/RF
OUTPUT

NC

NC

NC

C4

4.7F

1

2

3

4

5

6

7

NC = NO CONNECT

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

U2

AD636

COMM

VPOS

LDLO

V

RMS

14

13

12

11

10

9

8

1/2

AD712

NC

NC

NC

R7

56.2k

R6

3.16k

U3B

C3
1F

+100mV/dB

0V = 0dB(AT 10mV RMS)

+6V DEC

+316.2mV

+6V

FB

+6V

DEC

–6V

DEC

FB

–6V

POWER SUPPLY

DECOUPLING

NETWORK

V

OUT

0.1F

0.1F

C1

0.1F

C1LO

1

A1HI

2

+

A1LO

3

GAT1

4

GAT2

R3

R4

3.01k

A2LO

A2HI

C2LO

5

6

7

8

–

–

+

U1 AD600

R5

16.2k

A1

REF

A2

C1HI

16

A1CM

15

A1OP

14

VPOS

VNEG

A2OP

A2CM

C2HI

+6V
DEC

–6V
DEC

13

12

11

10

9

resistor of 215 Ω if very close calibration is not needed, since the
input resistance of the AD600 (and all other key parameters of it
and the AD636) are already laser trimmed for accurate operation.
This attenuator allows inputs as large as ±4 V to be accepted,
that is, signals with an rms value of 1 V combined with a crest
factor of up to 4.

The output of A2 is ac coupled via another 12 Hz high-pass
filter formed by C2 and the 6.7 kΩ input resistance of the
AD636. The averaging time constant for the rms-dc converter is
determined by C4. The unbuffered output of the AD636 (at Pin

8) is compared with a fixed voltage of 316 mV set by the posi­tive supply voltage of 6 V and resistors R6 and R7. (V

REF

is
proportional to this voltage, and systems requiring greater cali­bration accuracy should replace the supply dependent reference
with a more stable source.)

Any difference in these voltages is integrated by the op amp
U3B, with a time constant of 3 ms formed by the parallel sum of
R6/R7 and C3. Now, if the output of the AD600 is too high, V
rms will be greater than the “setpoint” of 316 mV, causing the
output of U3B—that is, V
grator is noninverting). A fraction of V

—to ramp up (note that the inte-

OUT

is connected to the

OUT

inverting gain-control inputs of the AD600, so causing the gain
to be reduced, as required, until V rms is exactly equal to 316 mV,
at which time the ac voltage at the output of A2 is forced to be
exactly 316 mV rms. This fraction is set by R4 and R5 such
that a 15.625 mV change in the control voltages of A1 and
A2—which would change the gain of the cascaded amplifiers by
1 dB—requires a change of 100 mV at V

. Notice here that

OUT

since A2 is forced to operate at an output level well below its
capacity, waveforms of high crest factor can be tolerated through­out the amplifier.

To check the operation, assume an input of 10 mV rms is
applied to the input, which results in a voltage of 3.16 mV rms
at the input to A1, due to the 10 dB loss in the attenuator. If the
system operates as claimed, V

(and hence VG) should be

OUT

zero. This being the case, the gain of both A1 and A2 will be
20 dB and the output of the AD600 will therefore be 100 times
(40 dB) greater than its input, which evaluates to 316 mV rms,

the input required at the AD636 to balance the loop. Finally,
note that unlike most AGC circuits, needing strong temperature
compensation for the internal “kT/q” scaling, these voltages,
and thus the output of this measurement system, are tempera­ture stable, arising directly from the fundamental and exact
exponential attenuation of the ladder networks in the AD600.

Typical results are presented for a sine wave input at 100 kHz.
Figure 20 shows that the output is held very close to the set­point of 316 mV rms over an input range in excess of 80 dB.

450

425

400

375

350

325

– mV

300

OUT

275

V

250

225

200

175

150

10V 100V 10V1V100mV10mV1mV

INPUT SIGNAL – V RMS

Figure 20. The RMS Output of A2 Is Held Close to the
“Setpoint” 316 mV for an Input Range of Over 80 dB

(This system can, of course, be used as an AGC amplifier, in
which the rms value of the input is leveled.) Figure 21 shows
the “decibel” output voltage. More revealing is Figure 22, which
shows that the deviation from the ideal output predicted by
Equation 1 over the input range 80 µV to 500 mV rms is within
±0.5 dB, and within ±1 dB for the 80 dB range from 80 µV to
800 mV. By suitable choice of the input attenuator R1 + R2,
this could be centered to cover any range from 25 mV to 250 mV
to, say, 1 mV to 10 V, with appropriate correction to the value
of V

REF

. (Note that V

is not affected by the changes in the

SCALE

–14–

REV. B

AD600/AD602

range.) The gain ripple of ±0.2 dB seen in this curve is the result of
the finite interpolation error of the X-AMP. Note that it occurs
with a periodicity of 12 dB—twice the separation between the
tap points (because of the two cascaded stages).

5

4

3

2

1

– Volts

0

OUT

V

–1

–2

–3

–4

–5

10V 100V 10V1V100mV10mV1mV

INPUT SIGNAL – V RMS

Figure 21. The dB Output of Figure 19’s Circuit Is Linear
Over an 80 dB Range

2.5

2.0

1.5

1.0

0.5

0

–0.5

–1.0

OUTPUT ERROR – dB

–1.5

–2.0

–2.5

10V 100V 10V1V100mV10mV1mV

INPUT SIGNAL – V RMS

Figure 22. Data from Figure 20 Presented as the Deviation
from the Ideal Output Given in Equation 4

This ripple can be canceled whenever the X-AMP stages are
cascaded by introducing a 3 dB offset between the two pairs
of control voltages. A simple means to achieve this is shown
in Figure 23: the voltages at C1HI and C2HI are “split” by
±46.875 mV, or ±1.5 dB. Alternatively, either one of these pins
can be individually offset by 3 dB and a 1.5 dB gain adjustment
made at the input attenuator (R1 + R2).

C1HI

U1

AD600

NC = NO CONNECT

16

15

14

13

12

11

10

–6V

DEC

9

A1CM

A1OP
VPOS

VNEG

A2OP

A2CM

C2HI

10k

+6V DEC

–6V DEC

–46.875mV

78.7 78.7

–6V DEC

C2
2F

+46.875mV

+6V
DEC

10k

3dB OFFSET

MODIFICATION

NC

NC

NC

1

2

3

4

5

6

7

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

U2

AD636

Figure 23. Reducing the Gain Error Ripple

The error curve shown in Figure 24 demonstrates that over the
central portion of the range the output voltage can be maintained
very close to the ideal value. The penalty for this modification is
the higher errors at the extremities of the range. The next two
applications show how three amplifier sections can be cascaded
to extend the nominal conversion range to 120 dB, with the
inclusion of simple LP filters of the type shown in Figure 15.
Very low errors can then be maintained over a 100 dB range.

2.5

2.0

1.5

1.0

0.5

0

–0.5

–1.0

OUTPUT ERROR – dB

–1.5

–2.0

–2.5

10V 100V 10V1V100mV10mV1mV

INPUT SIGNAL – V RMS

Figure 24. Using the 3 dB Offset Network, the Ripple
Is Reduced

100 dB to 120 dB RMS Responding Constant Bandwidth AGC
Systems with High Accuracy dB Outputs

The next two applications double as both AGC amplifiers and
measurement systems. In both, precise gain offsets are used to
achieve either (1) a very high gain linearity of ±0.1 dB over
the full 100 dB range, or (2) the optimal signal-to-noise ratio
at any gain.

REV. B

–15–

AD600/AD602

INPUT

1V RMS

MAX

(SINE WAVE)

R13

3.01k

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

+

7

8

U1 AD600

+5V

FB

+5V

DEC

–5V

DEC

FB

–5V

POWER SUPPLY

DECOUPLING

NETWORK

Q1

2N3906

11.3k

A1

A2

0.1F

0.1F

R14

301k

R12

REF

16

15

14

13

12

11

10

9

+5V DEC

C1HI

A1CM

A1OP

VPOS

VNEG

A2OP

A2CM

C2HI

–5V

10k

R15

19.6k

R16

6.65k

+5V
DEC

–5V
DEC

R6

C1

0.1F

R1

133k

C2

0.1F

R4

133k

–2dB
–62.5mV

R7

127

U3A

1/4

AD713

R5

1.58k

C3

220pF

0dB

R8

127R910k

R2

487

U3B

+2dB
+62.5mV

NC

–5V

DEC

NC

NC

R3

200

1/4

AD713

+5V

1

2

3

4

5

6

7

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

C5

22F

U4

AD636

COMM

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

VPOS

LDLO

V

RMS

1

2

+

3

–

4

5

–

6

+

7

8

U2 AD600

14

NC

13

12

NC

11

NC

10

9

8

A1

REF

A2

R10

3.16k

+316.2mV

NC = NO CONNECT

16

15

14

13

12

11

10

9

R11

46.4k

C6

4.7F

U3C

AD713

C1HI

A1CM

A1OP

VPOS

VNEG

A2OP

A2CM

C2HI

+5V DEC

1/4

C4

2F

+5V
DEC

–5V
DEC

V

OUT

V

LOG

Figure 25. RMS Responding AGC Circuit with 100 dB Dynamic Range

A 100 dB RMS/AGC System with Minimal Gain Error
(Parallel Gain with Offset)

Figure 25 shows an rms-responding AGC circuit, which can
equally well be used as an accurate measurement system. It
accepts inputs of 10 µV to 1 V rms (–100 dBV to 0 dBV) with
generous overrange. Figure 26 shows the logarithmic output, V

LOG

,
which is accurately scaled 1 V per decade, that is, 50 mV/dB, with
an intercept (V

= 0) at 3.16 mV rms (–50 dBV). Gain offsets

LOG

of ±2 dB have been introduced between the amplifiers, provided
by the ±62.5 mV introduced by R6–R9. These offsets cancel a
small gain ripple which arises in the X-AMP from its finite interpo­lation error, which has a period of 18 dB in the individual VCA
sections. The gain ripple of all three amplifier sections without
this offset (in which case the gain errors simply add) is shown in
Figure 27; it is still a remarkably low ±0.25 dB over the 108 dB
range from 6 µV to 1.5 V rms. However, with the gain offsets
connected, the gain linearity remains under ±0.1 dB over the
specified 100 dB range (Figure 28).

5

4

3

2

1

0

–1

–2

LOGARITHMIC OUTPUT – Volts

–3

–4

–5

1V10V 10V1V100mV10mV1mV100V

Figure 26. V

LOG

INPUT SIGNAL – V RMS

Plotted vs. VIN for Figure 25‘s Circuit

Showing 120 dB AGC Range

–16–

REV. B

AD600/AD602

2.0

1.5

1.0

0.5

0.1
0

–0.1

–0.5

GAIN ERROR – dB

–1.0

–1.5

–2.0

1V10V 10V1V100mV10mV1mV100V

INPUT SIGNAL – V RMS

Figure 27. Gain Error for Figure 25 Without the 2 dB
Offset Modification

2.0

1.5

1.0

0.5

0.1
0

–0.1

–0.5

GAIN ERROR – dB

–1.0

–1.5

–2.0

1V10V 10V1V100mV10mV1mV100V

INPUT SIGNAL – V RMS

Figure 28. Adding the 2 dB Offsets Improves the
Linearization

The maximum gain of this circuit is 120 dB. If no filtering were
used, the noise spectral density of the AD600 (1.4 nV/√Hz)
would amount to an input noise of 8.28 µV rms in the full band-
width (35 MHz). At a gain of one million, the output noise
would dominate. Consequently, some reduction of bandwidth is
mandatory, and in the circuit of Figure 25 it is due mostly to
a single-pole low-pass filter R5/C3, which provides a –3 dB
frequency of 458 kHz, which reduces the worst-case output
noise (at V

) to about 100 mV rms at a gain of 100 dB. Of

AGC

course, the bandwidth (and hence output noise) could be easily
reduced further, for example, in audio applications, merely by
increasing C3. The value chosen for this application is optimal
in minimizing the error in the V

output for small input signals.

LOG

The AD600 is dc-coupled, but even miniscule offset voltages at
the input would overload the output at high gains, so high-pass
filtering is also needed. To provide operation at low frequencies,
two simple zeros at about 12 Hz are provided by R1/C1 and
R4/C2; op amp sections U3A and U3B (AD713) are used to
provide impedance buffering, since the input resistance of the
AD600 is only 100 Ω. A further zero at 12 Hz is provided by C4
and the 6.7 kΩ input resistance of the AD636 rms converter.

The rms value of V

is generated at Pin 8 of the AD636; the

LOG

averaging time for this process is determined by C5, and the

value shown results in less than 1% rms error at 20 Hz. The
slowly varying V rms is compared with a fixed reference of
316 mV, derived from the positive supply by R10/R11. Any
difference between these two voltages is integrated in C6, in
conjunction with op amp U3C, the output of which is V

LOG

. A
fraction of this voltage, determined by R12 and R13, is returned
to the gain control inputs of all AD600 sections. An increase in

lowers the gain, because this voltage is connected to the

V

LOG

inverting polarity control inputs.

Now, in this case, the gains of all three VCA sections are being
varied simultaneously, so the scaling is not 32 dB/V but 96 dB/V,
or 10.42 mV/dB. The fraction of V

required to set its scaling

LOG

to 50 mV/dB is therefore 10.42/50, or 0.208. The resulting full­scale range of V

is nominally ±2.5 V. This scaling was chosen

LOG

to allow the circuit to operate from ±5 V supplies. Optionally,
the scaling could be altered to 100 mV/dB, which would be
more easily interpreted when V

is displayed on a DVM, by

LOG

increasing R12 to 25.5 kΩ. The full-scale output of ±5 V then
requires the use of supply voltages of at least ±7.5 V.

A simple attenuator of 16.6 ± 1.25 dB is formed by R2/R3 and
the 100 Ω input resistance of the AD600. This allows the refer­ence level of the decibel output to be precisely set to zero for an
input of 3.16 mV rms, and thus center the 100 dB range between
10 µV and 1 V. In many applications R2/R3 may be replaced
by a fixed resistor of 590 Ω. For example, in AGC applica­tions, neither the slope nor the intercept of the logarithmic output
is important.

A few additional components (R14–R16 and Q1) improve the
accuracy of V

at the top end of the signal range (that is, for

LOG

small gains). The gain starts rolling off when the input to the
first amplifier, U1A, reaches 0 dB. To compensate for this non­linearity, Q1 turns on at V

~ +1.5 V and increases the feedback

LOG

to the control inputs of the AD600s, thereby needing a smaller
voltage at V

to maintain the input to the AD636 to the set-

LOG

point of 316 mV rms.

A 120 dB RMS/AGC System with Optimal S/N Ratio
(Sequential Gain)

In the last case, all gains were adjusted simultaneously, resulting
in an output signal-to-noise ratio (S/N ratio) which is always less
than optimal. The use of sequential gain control results in a
major improvement in S/N ratio, with only a slight penalty in
the accuracy of V
of V

. The idea is simply to increase the gain of the earlier

AGC

, and no penalty in the stabilization accuracy

LOG

stages first (as the signal level decreases) and thus maintain the
highest S/N ratio throughout the amplifier chain. This can be
easily achieved with the AD600 because its gain is accurate even
when the control input is overdriven; that is, each gain con­trol “window” of 1.25 V is used fully before moving to the
next amplifier to the right.

Figure 29 shows the circuit for the sequential control scheme.
R6 to R9 with R16 provide offsets of 42.14 dB between the
individual amplifiers to ensure smooth transitions between the
gain of each successive X-AMP, with the sequence of gain
increase being U1A first, then U1B, and lastly U2A. The adjust­able attenuator provided by R3 + R17 and the 100 Ω input
resistance of U1A as well as the fixed 6 dB attenuation provided
by R2 and the input resistance of U1B are included both to set

to read 0 dB when VIN is 3.16 mV rms and to center the

V

LOG

100 dB range between 10 µV rms and 1 V rms input. R5 and
C3 provide a 3 dB noise bandwidth of 30 kHz. R12 to R15

REV. B

–17–

AD600/AD602

0dB

ADJUST

R3

R17

115

200

INPUT

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

+

7

8

A1

REF

A2

U1 AD600

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

+6V
DEC

–6V
DEC

C1

0.1F

133k

C2

0.1F

133k

R1

R4

R5

5.36k

0.001F

U3A

1/4

AD713

C3

R2

100

U3B

1/4

AD713

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

+

7

8

A1

REF

A2

U2 AD600

C1HI

16

A1CM

15

14

VPOS

13

12

A2OP

11

A2CM

10

C2HI

9

A1OP

VNEG

C4

2F

+5V
DEC

–5V
DEC

V

OUT

R6

R7

1k

R8

294

+6V

FB

+6V

DEC

–6V

DEC

FB

–6V

POWER SUPPLY

DECOUPLING

NETWORK

0.1F

0.1F

+6V DEC

R15

5.11k

R14

7.32k

+5V

R13

866

3.4k

R12

1k

Figure 29. 120 dB Dynamic Range RMS Responding Circuit Optimized for S/N Ratio

change the scaling from 625 mV/decade at the control inputs to
1 V/decade at the output and at the same time center the dynamic
range at 60 dB, which occurs if the V
These arrangements ensure that the V

of U1B is equal to zero.

G

will still fit within the

LOG

±6 V supplies.

5

4

3

2

1

0

–1

–2

–3

LOGARITHMIC OUTPUT – Volts

–4

–5

1V10V 10V1V100mV10mV1mV100V

Figure 30. V

Is Essentially Linear Over the Full 120 dB

LOG

INPUT SIGNAL – V RMS

Range

R9

R16

1k

287

C5

22F

–6V

DEC

NC

NC

NC

1

2

3

4

5

6

7

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

U4

AD636

COMM

VPOS

LDLO

V

Figure 30 shows V

14

NC

13

12

NC

11

NC

R10

3.16k

10

9

8

RMS

+316.2mV

NC = NO CONNECT

to be linear over a full 120 dB range.

LOG

R11

56.2k

C6

4.7F

U3C

AD713

+6V DEC

1/4

V

LOG

Figure 31 shows the error ripple due to the individual gain func­tions which is bounded by ±0.2 dB (dotted lines) from 6 µV to 2 V.
The small perturbations at about 200 µV and 20 mV, caused
by the impracticality of matching the gain functions perfectly,
are the only sign that the gains are now sequential. Figure 32 is
a plot of V

which remains very close to its set value of 316 mV

AGC

rms over the full 120 dB range.

To more directly compare the signal-to-noise ratios in the
“simultaneous” and “sequential” modes of operation, all inter­stage attenuation was eliminated (R2 and R3 in Figure 25, R2
in Figure 29), the input of U1A was shorted, R5 was selected to
provide a 20 kHz bandwidth (R5 = 7.87 kΩ), and only the gain
control was varied, using an external source. The rms value of
the noise was then measured at V

and expressed as an S/N

OUT

ratio relative to 0 dBV, this being almost the maximum output
capability of the AD600. Results for the simultaneous mode can
be seen in Figure 33. The S/ N ratio degrades uniformly as the
gain is increased. Note that since the inverting gain control was
used, the gain in this curve and in Figure 34 decreases for more
positive values of the gain-control voltage.

–18–

REV. B

AD600/AD602

2.0

1.5

1.0

0.5

0.2
0

–0.2

–0.5

GAIN ERROR – dB

–1.0

–1.5

–2.0

1V10V 10V1V100mV10mV1mV100V

INPUT SIGNAL – V RMS

Figure 31. The Error Ripple Due to the Individual Gain
Functions

400

350

300

GAIN ERROR – mV

250

200

1V10V 10V1V100mV10mV1mV100V

Figure 32. V

INPUT SIGNAL – V RMS

Remains Nose to Its Setpoint of

AGC

316 mV RMS Over the Full 120 dB Range

90

80

70

60

50

40

S/N RATIO – dB

30

20

10

0

–625.0–833.2

CONTROL VOLTAGE, V

(10.417mV/dB) – mV

C

833.2

625.0416.6208.30–208.3–416.6

Figure 33. S/N Ratio vs. Control Voltage for Parallel Gain
Control (Figure 25)

In contrast, the S/N ratio for the sequential mode is shown in
Figure 34. U1A always acts as a fixed noise source; varying its
gain has no influence on the output noise. (This is a feature of
the X-AMP technique.) Thus, for the first 40 dB of control
range (actually slightly more, as explained below), when only
this VCA section has its gain varied, the S/N ratio remains con­stant. During this time, the gains of U1B and U2A are at their
minimum value of –1.07 dB.

90

80

70

60

50

40

S/N RATIO – dB

30

20

10

0

–0.558–1.183

CONTROL VOLTAGE, VC (31.25mV/dB) – Volts

3.817

3.1922.5671.9421.3170.6920.067

Figure 34. S/N Ratio vs. Control Voltage for Sequential
Gain Control (Figure 29)

For the next 40 dB of control range, the gain of U1A remains
fixed at its maximum value of 41.07 dB and only the gain of
U1B is varied, while that of U2A remains at its minimum value
of –1.07 dB. In this interval, the fixed output noise of U1A is
amplified by the increasing gain of U1B and the S/N ratio pro­gressively decreases.

Once U1B reaches its maximum gain of 41.07 dB, its output
also becomes a gain independent noise source; this noise is
presented to U2A. As the control voltage is further increased,
the gains of both U1A and U1B remain fixed at their maximum
value of 41.07 dB, and the S/N ratio continues to decrease.
Figure 34 clearly shows this, because the maximum S/N ratio
of 90 dB is extended for the first 40 dB of input signal before it
starts to roll off.

This arrangement of staggered gains can be easily implemented
because, when the control inputs of the AD600 are overdriven,
the gain limits to its maximum or minimum values without side
effects. This eliminates the need for awkward nonlinear shaping
circuits that have previously been used to break up the gain
range of multistage AGC amplifiers. It is the precise values of the
AD600’s maximum and minimum gain (not 0 dB and 40 dB
but –1.07 dB and +41.07 dB) that explain the rather odd values
of the offset values that are used.

The optimization of the output S/N ratio is of obvious value in
AGC systems. However, in applications where these circuit are
considered for their wide range logarithmic measurements capa­bilities, the inevitable degradation of the S/N ratio at high gains
need not seriously impair their utility. In fact, the bandwidth of
the circuit shown in Figure 25 was specifically chosen so as to
improve measurement accuracy by altering the shape of the log
error curve (Figure 31) at low signal levels.

REV. B

–19–

AD600/AD602

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

0.012

(0.30)

0.31

(7.87)

0.18

(4.57)

16-Lead Plastic DIP Package

(N-16)

0.87 (22.1) MAX

16

18

0.100

0.018
(2.54)

(0.46)

9

0.033

(0.84)

0.25
(6.35)

0.035
(0.89)

0.125
(3.18)

MIN

16-Lead SOIC Package

(R-16)

0.413 (10.50)

0.019
(0.49)

9

0.299

0.419

(7.60)

(10.65)

8

0.104
(2.65)

16

1

0.05 (1.27)
REF

0.030
(0.75)

0.013
(0.32)

0.3 (7.62)

0.18 (4.57)
MAX

0.011
(0.28)

0.042

(1.07)

0.005 (0.13)
MIN

PIN 1

0.200 (5.08)
MAX

0.200 (5.08)

0.125 (3.18)

16-Lead Cerdip Package

0.840 (21.34) MAX

16

18

0.023 (0.58)

0.014 (0.36)

0.100

(2.54)

BSC

0.070 (1.78)

0.030 (0.76)

(Q-16)

0.080 (2.03) MAX

9

0.310 (7.87)

0.220 (5.59)

0.060 (1.52)

0.015 (0.38)

SEATING
PLANE

0.150
(3.81) MIN

0.320 (8.13)

0.290 (7.37)

0.015 (0.38)

15°

0.008 (0.20)

0°

C00538–0–3/01 (rev. B)

AD600/AD602–Revision History

Location Page

Data sheet changed from REV. A to REV. B.

Changes to Accuracy Section of AD600A/AD602A column . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 2

Changes to Ordering Guide . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3

Change to Figure 3 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8

–20–

REV. B

PRINTED IN U.S.A.

PRINTED IN U.S.A.