Datasheet AD600, AD602 Datasheet (ANALOG DEVICES)

A

查询AD602 供应商查询AD602 供应商

Dual, Low Noise, Wideband

FEATURES

2 channels with independent gain control

Linear in dB gain response

2 gain ranges

AD600: 0 dB to 40 dB

AD602: –10 dB to +30 dB
Accurate absolute gain: ±0.3 dB
Low input noise: 1.4 nV/√Hz
Low distortion: −60 dBc THD at ±1 V output
High bandwidth: dc to 35 MHz (−3 dB)
Stable group delay: ±2 ns
Low power: 125 mW (maximum) per amplifier
Signal gating function for each amplifier
Drives high speed ADCs
MIL-STD-883-compliant and DESC versions available

APPLICATIONS

Ultrasound and sonar time-gain controls
High performance audio and RF AGC systems
Signal measurement

GENERAL DESCRIPTION

The AD600/AD6021 dual channel, low noise, variable gain
amplifiers are optimized for use in ultrasound imaging systems
but are applicable to any application requiring precise gain, low
noise and distortion, and wide bandwidth. Each independent
channel provides a gain of 0 dB to +40 dB in the AD600 and

−10 dB to +30 dB in the AD602. The lower gain of the AD602
results in an improved signal-to-noise ratio (SNR) at the output.
However, both products have the same 1.4 nV/√Hz input noise
spectral density. The decibel gain is directly proportional to the
control voltage, accurately calibrated, and supply and
temperature stable.

To achieve the difficult performance objectives, a proprietary
circuit form, the X-AMP®, was developed. Each channel of the
X-AMP comprises a variable attenuator of 0 dB to −42.14 dB
followed by a high speed fixed gain amplifier. In this way, the
amplifier never has to cope with large inputs and can benefit
from the use of negative feedback to precisely define the gain
and dynamics. The attenuator is realized as a 7-stage R-2R
ladder network having an input resistance of 100 , laser
trimmed to ±2%. The attenuation between tap points is 6.02 dB;
the gain-control circuit provides continuous interpolation between
these taps. The resulting control function is linear in dB.

Rev. E

Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.

Variable Gain Amplifiers

AD600/AD602

FUNCTIONAL BLOCK DIAGRAM

G

T1

–18.06dB

PRECISION PASSIVE
INPUT ATTENUATOR

–22.08dB

–36.12dB

–30.1dB

–42.14dB

Figure 1.

62.5Ω

GATING

INTERFACE

RF2

2.24kΩ (AD600)
694Ω (AD602)

RF1
20Ω

FIXED-GAIN

AMPLIFIER

41.07dB (AD600)

31.07dB (AD602)

SCALING

REFERENCE

C1HI

V

G

C1LO

GAIN CONT ROL

INTERFACE

0dB

–12.04dB

–6.02dB

A1HI

A1LO

500Ω

R-2R LADDER NET WORK

The gain-control interfaces are fully differential, providing an
input resistance of ~15 M and a scale factor of 32 dB/V (that
is, 31.25 mV/dB) defined by an internal voltage reference. The
response time of this interface is less than 1 µs. Each channel
also has an independent gating facility that optionally blocks
signal transmission and sets the dc output level to within a few
millivolts of the output ground. The gating control input is
TTL- and CMOS-compatible.

The maximum gain of the AD600 is 41.07 dB, and the
maximum gain of the AD602 is 31.07 dB; the −3 dB bandwidth
of both models is nominally 35 MHz, essentially independent of
the gain. The SNR for a 1 V rms output and a 1 MHz noise
bandwidth is typically 76 dB for the AD600 and 86 dB for the
AD602. The amplitude response is flat within ±0.5 dB from
100 kHz to 10 MHz; over this frequency range, the group delay
varies by less than ±2 ns at all gain settings.

Each amplifier channel can drive 100  load impedances with
low distortion. For example, the peak specified output is ±2.5 V
minimum into a 500  load or ±1 V into a 100  load. For a
200  load in shunt with 5 pF, the total harmonic distortion for
a ±1 V sinusoidal output at 10 MHz is typically −60 dBc.

The AD600J/AD602J are specified for operation from 0°C to 70°C
and are available in 16-lead PDIP (N) and 16-lead SOIC_W
packages. The AD600A/AD602A are specified for operation from

−40°C to +85°C and are available in 16-lead CERDIP (Q) and
16-lead SOIC_W packages. The AD600S/ AD602S are specified
for operation from −55°C to +125°C, are available in a 16-lead
CERDIP (Q) package, and are MIL-STD-883-compliant. The
AD600S/AD602S are also available under DESC SMD 5962-94572.

1

Patented.

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: [phone] www.analog.com
Fax: [phone] ©2006 Analog Devices, Inc. All rights reserved.

A1OP

A1CM

00538-001

AD600/AD602

TABLE OF CONTENTS

Features .............................................................................................. 1

Applications..................................................................................... 15

Applications....................................................................................... 1

General Description......................................................................... 1

Functional Block Diagram .............................................................. 1

Revision History ............................................................................... 2

Specifications..................................................................................... 3

Absolute Maximum Ratings............................................................ 5

ESD Caution.................................................................................. 5

Pin Configuration and Function Descriptions............................. 6

Typical Performance Characteristics............................................. 7

Theory of Operation ...................................................................... 10

Noise Performance .....................................................................10

The Gain-Control Interface ......................................................11

Signal-Gating Inputs.................................................................. 11

Common-Mode Rejection........................................................ 11

Achieving 80 dB Gain Range.................................................... 12

Time-Gain Control (TGC) and Time-Variable

Gain (TVG)................................................................................. 15

Increasing Output Drive............................................................ 15

Driving Capacitive Loads.......................................................... 15

Realizing Other Gain Ranges ................................................... 16

An Ultralow Noise VCA............................................................ 16

A Low Noise, 6 dB Preamplifier............................................... 16

A Low Noise AGC Amplifier with 80 dB Gain Range.......... 17

A Wide Range, RMS-Linear dB Measurement System (2 MHz

AGC Amplifier with RMS Detector)....................................... 19

100 dB to 120 dB RMS Responding Constant Bandwidth

AGC Systems with High Accuracy dB Outputs..................... 21

A 100 dB RMS/AGC System with Minimal Gain Error

(Parallel Gain with Offset)........................................................ 22

A 120 dB RMS/AGC System with Optimal SNR

(Sequential Gain) ....................................................................... 23

Outline Dimensions .......................................................................27

Sequential Mode (Maximum SNR) .........................................12

Parallel Mode (Simplest Gain-Control Interface).................. 13

Low Ripple Mode (Minimum Gain Error)............................. 13

REVISION HISTORY

1/06—Rev. D to Rev. E

Updated Format..................................................................Universal

Changes to Table 2............................................................................ 5

Changes to The Gain-Control Interface Section........................ 11

Updated Outline Dimensions....................................................... 27

Changes to Ordering Guide.......................................................... 28

3/04—Rev. C to Rev. D

Changes to Specifications................................................................ 2

Changes to Ordering Guide............................................................ 3

Changes to Figure 3.......................................................................... 8

Changes to Figure 29...................................................................... 18

Updated Outline Dimensions....................................................... 20

Ordering Guide .......................................................................... 28

5/02—Rev. B to Rev. C

Changes to Specifications.................................................................2

Renumber Tables and TPCs...................................................Global

8/01—Rev. A to Rev. B

Changes to Accuracy Section of AD600A/AD602A column......2

Rev. E | Page 2 of 28

AD600/AD602

SPECIFICATIONS

Each amplifier section at TA = 25°C, VS = ±5 V, −625 mV ≤ VG ≤ +625 mV, RL = 500 Ω, and CL = 5 pF, unless otherwise noted.
Specifications for the AD600/AD602 are identical, unless otherwise noted.

Table 1.

AD600J/AD602J
Parameter Conditions Min Typ Max Min Typ Max Unit

INPUT CHARACTERISTICS

Input Resistance Pin 2 to Pin 3; Pin 6 to Pin 7

98

100
Input Capacitance 2 2 pF
Input Noise Spectral Density

2

1.4 1.4 nV/√Hz
Noise Figure RS = 50 Ω, maximum gain 5.3 5.3 dB
R

= 200 Ω, maximum gain 2 2 dB

S

Common-Mode Rejection Ratio f = 100 kHz 30 30 dB

OUTPUT CHARACTERISTICS

−3 dB Bandwidth V

= 100 mV rms 35 35 MHz

OUT

Slew Rate 275 275 V/µs
Peak Output

3

RL ≥ 500 Ω ±2.5 ±3 ±2.5 ±3 V
Output Impedance f ≤ 10 MHz 2 2 Ω
Output Short-Circuit Current 50 50 mA
Group Delay Change vs. Gain f = 3 MHz; full gain range ±2 ±2 ns
Group Delay Change vs. Frequency VG = 0 V, f = 1 MHz to 10 MHz ±2 ±2 ns
Total Harmonic Distortion RL= 200 Ω, V

= ±1 V peak, RPD = 1 kΩ −60 −60 dBc

OUT

ACCURACY

AD600

Gain Error 0 dB to 3 dB gain
3 dB to 37 dB gain
37 dB to 40 dB gain

0

−0.5

−1

+0.5
±0.2

−0.5
Maximum Output Offset Voltage4VG = –625 mV to +625 mV 10
Output Offset Variation VG = –625 mV to +625 mV 10

AD602

Gain Error –10 dB to –7 dB gain
–7 dB to +27 dB gain
27 dB to 30 dB gain

0

−0.5

−1

+0.5
±0.2

−0.5
Maximum Output Offset Voltage4VG = −625 mV to +625 mV 5
Output Offset Variation VG = −625 mV to +625 mV 5

GAIN CONTROL INTERFACE

Gain Scaling Factor

+3 dB to +37 dB (AD600);

31.7

32

−7 dB to +27 dB (AD602)
Common-Mode Range −0.75 +2.5 −0.75 +2.5 V
Input Bias Current 0.35 1 0.35 1 A
Input Offset Current 10 50 10 50 nA
Differential Input Resistance Pin 1 to Pin 16; Pin 8 to Pin 9 15 15 MΩ
Response Rate Full 40 dB gain change 40 40 dB/s

Rev. E | Page 3 of 28

1

AD600A/AD602A1

102 95

+1 −0.5
+0.5 −1.0
0 −1.5
50
50

10
10

+1 –0.5
+0.5 −1.0
0 −1.5
30
30

10
10

32.3 30.5

100

+0.5
±0.2

−0.5

+0.5
±0.2

−0.5

32

105

+1.5
+1.0
+0.5
65
65

+1.5
+1.0
+0.5
45
45

33.5

Ω

dB
dB
dB
mV
mV

dB
dB
dB
mV
mV

dB/V

AD600/AD602

AD600J/AD602J

1

AD600A/AD602A1

Parameter Conditions Min Typ Max Min Typ Max Unit

SIGNAL GATING INTERFACE

Logic Input LO (Output On) 0.8 0.8 V
Logic Input HI (Output Off) 2.4 2.4 V
Response Time On to off, off to on 0.3 0.3 µs
Input Resistance Pin 4 to Pin 3; Pin 5 to Pin 6 30 30 kΩ
Output Gated Off

Output Offset Voltage ±10

±100

±10

±400

mV
Output Noise Spectral Density 65 65 nV/√Hz
Signal Feedthrough @ 1 MHz

AD600 −80 −80 dB
AD602 −70 −70 dB

POWER SUPPLY

Specified Operating Range ±4.75 ±5.25 ±4.75 ±5.25 V
Quiescent Current 11

1

Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to calculate outgoing quality levels. All

minimum and maximum specifications guaranteed, although only those shown in boldface are tested on all production units.

2

Typical open- or short-circuited input; noise is lower when the system is set to maximum gain and the input is short-circuited. This figure includes the effects of both

voltage and current noise sources.

3

With an additional 1 kΩ pull-down resistor, if RL < 500 Ω.

4

The dc gain of the main amplifier in the AD600 is × 113; therefore, an input offset of only 100 V becomes an 11.3 mV output offset. In the AD602, the amplifier’s gain

is × 35.7; therefore, an input offset of 100 V becomes a 3.57 mV output offset.

12.5

11

14

mA

Rev. E | Page 4 of 28

AD600/AD602

ABSOLUTE MAXIMUM RATINGS

Table 2.

Parameter Rating

Supply Voltage ±V
Input Voltages

Pin 1, Pin 8, Pin 9, Pin 16 ±V
Pin 2, Pin 3, Pin 6, Pin 7 ±2 V continuous
±VS for 10 ms

Pin 4, Pin 5 ±V
Internal Power Dissipation 600 mW
Operating Temperature Range

J Grade 0°C to 70°C

A Grade −40°C to +85°C

S Grade −55°C to +125°C
Storage Temperature Range −65°C to +150°C
Lead Temperature (Soldering 60 sec) 300°C
θ

JA

16-Lead PDIP 85°C/W

16-Lead SOIC_W 100°C/W

16-Lead CERDIP 120°C/W

S

±7.5 V

S

S

Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.

Rev. E | Page 5 of 28

AD600/AD602

A

A

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

C1LO

A1HI

1LO

GAT1

GAT2

2LO

A2HI

C2LO

1

2

3

4

5

6

7

8

+

–

–

+

AD600 /
AD602

A1

REF

A2

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

11

A2OP

10

A2CM

9

C2HI

00538-002

Figure 2. Pin Configuration

Table 3. Pin Function Descriptions

Pin No. Mnemonic Description

1 C1LO CH1 Gain-Control Input LO (Positive Voltage Reduces CH1 Gain)
2 A1HI CH1 Signal Input HI (Positive Voltage Increases CH1 Output)
3 A1LO CH1 Signal Input LO (Usually Connected to CH1 Input Ground)
4 GAT1 CH1 Gating Input (A Logic HI Shuts Off CH1 Signal Path)
5 GAT2 CH2 Gating Input (A Logic HI Shuts Off CH2 Signal Path)
6 A2LO CH2 Signal Input LO (Usually Connected to CH2 Input Ground)
7 A2HI CH2 Signal Input HI (Positive Voltage Increases CH2 Output)
8 C2LO CH2 Gain-Control Input LO (Positive Voltage Reduces CH2 Gain)
9 C2HI CH2 Gain-Control Input HI (Positive Voltage Increases CH2 Gain)
10 A2CM CH2 Common (Usually Connected to CH2 Output Ground)
11 A2OP CH2 Output
12 VNEG Negative Supply for Both Amplifiers
13 VPOS Positive Supply for Both Amplifiers
14 A1OP CH1 Output
15 A1CM CH1 Common (Usually Connected to CH1 Output Ground)
16 C1HI CH1 Gain-Control Input HI (Positive Voltage Increases CH1 Gain)

Rev. E | Page 6 of 28

AD600/AD602

–

TYPICAL PERFORMANCE CHARACTERISTICS

10.0

9.8

9.6

9.4

9.2

9.0

8.8

GROUP DELAY (ns)

8.6

8.4

8.2

8.0
–0.7 0.5

GAIN CONTROL VOLTAGE (V)

Figure 6. AD600 and AD602 Typical Group Delay vs. V

VG=0V
10dB/DIV
CENTER
FREQ 1MHz
10kHz/DIV

00538-006

0.7–0.5 0.30.1–0.1–0.3

C

–0.05

–0.15

GAIN ERROR (dB)

–0.25

–0.35

–0.45

20dB

17dB

0.45

0.35

0.25

0.15

0.05

–0.5–0.7

GAIN CONTROL VOLTAGE (V)

Figure 3. Gain Error vs. Gain Control Voltage

00538-003

0.7

0.50.30.1–0.1–0.3

0°

–45°

–90°

FREQUENCY (Hz)

Figure 4. AD600 Frequency and Phase Response vs. Gain

10dB

7dB

0°

–45°

–90°

100k 1M 10M 100M

FREQUENCY (Hz)

Figure 5. AD602 Frequency and Phase Response vs. Gain

0538-007

00538-004

100M1M100k 10M

Figure 7. Third-Order Intermodulation Distortion, V

= 2 V p-p, RL = 500 Ω

OUT

1.0

–1.2

–1.4

–1.6

–1.8

–2.0

–2.2

–2.4

–2.6

–2.8

–3.0

NEGATIVE OUTPUT VOLTAGE LIMIT (V)

–3.2

00538-005

–3.4

50

0

LOAD RESISTANCE (Ω)

Figure 8. Typical Output Voltage vs. Load Resistance

20001000500200100

00538-008

(Negative Output Swing Limits First)

Rev. E | Page 7 of 28

AD600/AD602

102

101

100

99

98

97

96

95

INPUT IMPEDANCE (Ω)

94

93

92

100k 1M 10M 100M

GAIN = 40dB

GAIN = 20dB

GAIN = 0dB

FREQUENCY (Hz)

Figure 9. Input Impedance vs. Frequency

6

5

4

3

2

1

0

–1

–2

OUTPUT OFFSETVOLTAGE (mV)

–3

–4

–0.7

–0.3 –0.1 0.1 0.3

–0.5

GAIN CONTROL VO LTAGE (V)

AD600

AD602

0.5

Figure 10. Output Offset Voltage vs. Gain Control Voltage

(Control Channel Feedthrough)

0.7

100

90

OUTPUTINPUT

10

0%

00538-009

Figure 12. Gating Feedthrough to Output, Gating Off to On

100

90

OUTPUTINPUT

10

0%

00538-010

Figure 13. Gating Feedthrough to Output, Gating On to Off

50mV

5V 100ns

50mV

5V 100ns

00538-012

00538-013

1V VOUT

100

90

OUTPUT

10

INPUT

0%

1V VC

1µs

0538-011

Figure 11. Gain Control Channel Response Time. Top: Output Voltage, 2 V

max, Bottom: Gain Control Voltage V

= ±625 mV

C

Rev. E | Page 8 of 28

1V

100

90

OUTPUTINPUT

10

0%

100mV

500ns

Figure 14. Transient Response, Medium and High Gain

00538-014

AD600/AD602

T

R

T

10

500mV

100

90

OUTPUTINPU

10

0%

1V 200ns

Figure 15. Input Stage Overload Recovery Time

100

90

OUTPUTINPUT

10

0%

200mV 500ns

Figure 16. Output Stage Overload Recovery Time

100

90

OUTPUTINPUT

10

0%

1V 500n s

Figure 17. Transient Response Minimum Gain

1V

500mV

00538-015

00538-016

00538-017

AD600: G = 20dB

5

AD602: G = 10dB
BOTH: V

0

–5

–10

–15

CMRR (dB)

–20

–25

–30

–35

–40

1k 10k 100k 1M 10M 100M

= 100mV rms

CM

V

=±5V

S

R

= 500Ω

L

T

= 25°C

A

AD600

AD602

FREQUENCY (Hz)

Figure 18. CMRR vs. Frequency

20

10

0

–10

–20

–30

PSRR (dB)

–40

–50

–60

–70

–80

100k 1M 10M 100M

AD600

AD602

FREQUENCY (Hz)

AD600: G = 40dB
AD602: G = 30dB
BOTH: R

= 500Ω

L

VIN=0V
R

=50Ω

S

Figure 19. PSRR vs. Frequency

10

AD600: CH1 G = 40dB, VIN = 0

0

–10

–20

–30

ALK (dB)

–40

OSS

–50

C

–60

–70

–80

–90

100k 1M 10M 100M

CH2 G = 20dB, V

AD602: CH1 G = 30dB, V

CH2 G = 0dB, V

BOTH: V

CROSSTALK = 20log

= 1V rms1, RS = 50Ω

OUT

= 500Ω

R

L

= 100mV

IN

= 0

IN

= 316mV

IN

CH1 V

OUT

CH2 V

IN

AD600

FREQUENCY (Hz)

AD602

Figure 20. Crosstalk Between A1 and A2 vs. Frequency

00538-018

00538-019

00538-020

Rev. E | Page 9 of 28

AD600/AD602

THEORY OF OPERATION

The AD600/AD602 have the same general design and features.
They comprise two fixed gain amplifiers, each preceded by a
voltage-controlled attenuator of 0 dB to 42.14 dB with independent
control interfaces, each having a scaling factor of 32 dB per volt.
The AD600 amplifiers are laser trimmed to a gain of 41.07 dB
(×113), providing a control range of −1.07 dB to +41.07 dB
(0 dB to +40 dB with overlap). The AD602 amplifiers have a
gain of 31.07 dB (×35.8) and provide an overall gain of

−11.07 dB to +31.07 dB (−10 dB to +30 dB with overlap).

The advantage of this topology is that the amplifier can use
negative feedback to increase the accuracy of its gain. In
addition, because the amplifier does not have to handle large
signals at its input, the distortion can be very low. Another
feature of this approach is that the small-signal gain and phase
response, and thus the pulse response, are essentially
independent of gain.

Figure 21 is a simplified schematic of one channel. The input
attenuator is a 7-stage R-2R ladder network, using untrimmed
resistors of nominally R = 62.5 , which results in a characteristic
resistance of 125  ± 20%. A shunt resistor is included at the
input and laser trimmed to establish a more exact input
resistance of 100  ± 2%, which ensures accurate operation
(gain and HP corner frequency) when used in conjunction with
external resistors or capacitors.

GAT 1

SCALING

REFERENCE

C1HI

V

G

C1LO

GAIN CO NTROL

INTERFACE

0dB

–12.04dB

A1HI

A1LO

Figure 21. Simplified Block Diagram of Single Channel of the AD600/AD602

–6.02dB

500Ω

R-2R LADDER NETWORK

PRECISION PASSIVE
INPUT ATTENUATOR

–18.06dB

–22.08dB

–30.1dB

–36.12dB

–42.14dB

62.5Ω

GAT ING

INTERFACE

RF2

2.24kΩ (AD600)
694Ω (AD602)

RF1
20Ω

FIXED-GAIN

AMPLIFIER

41.07dB (AD600)

31.07dB (AD602)

A1OP

A1CM

The nominal maximum signal at input A1HI is 1 V rms (±1.4 V
peak) when using the recommended ±5 V supplies; although,
operation to ±2 V peak is permissible with some increase in HF
distortion and feedthrough. Each attenuator is provided with a
separate signal LO connection for use in rejecting common
mode, the voltage between input and output grounds. Circuitry
is included to provide rejection of up to ±100 mV.

00538-021

The signal applied at the input of the ladder network is
attenuated by 6.02 dB by each section; thus, the attenuation to
each of the taps is progressively 0 dB, 6.02 dB, 12.04 dB, 18.06 dB,

24.08 dB, 30.1 dB, 36.12 dB, and 42.14 dB. A unique circuit
technique is employed to interpolate between these tap points,
indicated by the slider in

Figure 21, providing continuous

attenuation from 0 dB to 42.14 dB.

To understand the AD600, it helps to think in terms of a
mechanical means for moving this slider from left to right; in
fact, it is voltage controlled. The details of the control interface
are discussed later. Note that the gain is exactly determined at
all times, and a linear decibel relationship is guaranteed
automatically between the gain and the control parameter that
determines the position of the slider. In practice, the gain
deviates from the ideal law by about ±0.2 dB peak (see

Figure 28).

Note that the signal inputs are not fully differential. A1LO,
A1CM (for CH1), A2LO, and A2CM (for CH2) provide
separate access to the input and output grounds. This recognizes
that even when using a ground plane, small differences arise in the
voltages at these nodes. It is important that A1LO and A2LO be
connected directly to the input ground(s). Significant impedance in
these connections reduces the gain accuracy. A1CM and A2CM
should be connected to the load ground(s).

NOISE PERFORMANCE

An important reason for using this approach is the superior
noise performance that can be achieved. The nominal resistance
seen at the inner tap points of the attenuator is 41.7  (one third of
125 ), which, at 27°C, exhibits a Johnson noise spectral density
(NSD) of 0.84 nV/√Hz (that is, √4kTR), a large fraction of the
total input noise. The first stage of the amplifier contributes
another 1.12 nV/√Hz, for a total input noise of 1.4 nV/√Hz.

The noise at the 0 dB tap depends on whether the input is
short-circuited or open-circuited. When shorted, the minimum
NSD of 1.12 nV/√Hz is achieved. When open, the resistance of
100  at the first tap generates 1.29 nV/√Hz, so the noise
increases to 1.71 nV/√Hz. This last calculation would be important
if the AD600 were preceded, for example, by a 900  resistor to
allow operation from inputs up to ±10 V rms. However, in most
cases, the low impedance of the source limits the maximum
noise resistance.

Rev. E | Page 10 of 28

AD600/AD602

It is apparent from the foregoing that it is essential to use a low
resistance in the design of the ladder network to achieve low
noise. In some applications, this can be inconvenient, requiring
the use of an external buffer or preamplifier. However, very few
amplifiers combine the needed low noise with low distortion at
maximum input levels, and the power consumption required to
achieve this performance is quite high (due to the need to
maintain very low resistance values while also coping with large
inputs). On the other hand, there is little value in providing a
buffer with high input impedance, because the usual reason for
this—the minimization of loading of a high resistance source—
is not compatible with low noise.

Apart from the small variations just mentioned, the SNR at the
output is essentially independent of the attenuator setting,
because the maximum undistorted output is 1 V rms and the
NSD at the output of the AD600 is fixed at 113 × 114 nV/√Hz,
or 158 nV/√Hz. Therefore, in a 1 MHz bandwidth, the output
SNR is 76 dB. The input NSD of the AD600/AD602 are the
same, but because of the 10 dB lower gain in the AD602’s fixed
amplifier, its output SNR is 10 dB better, or 86 dB in a 1 MHz
bandwidth.

THE GAIN-CONTROL INTERFACE

The attenuation is controlled through a differential, high
impedance (15 MΩ) input, with a scaling factor that is laser
trimmed to 32 dB per volt, that is, 31.25 mV/dB. Each of the
two amplifiers has its own control interface. An internal band
gap reference ensures stability of the scaling with respect to
supply and temperature variations and is the only circuitry
common to both channels.

When the differential input voltage V
slider is centered, providing an attenuation of +21.07 dB,
resulting in an overall gain of +20 dB (= –21.07 dB + +41.07 dB).
When the control input is −625 mV, the gain is lowered by
+20 dB (= +0.625 × +32) to 0 dB; when set to +625 mV, the
gain is increased by +20 dB to +40 dB. When this interface is
overdriven in either direction, the gain approaches either

−1.07 dB (= −42.14 dB + +41.07 dB) or +41.07 dB (= 0 +
+41.07 dB), respectively.

The gain of the AD600 can be calculated by

Gain (dB) = 32 V

where V

is in volts.

G

+ 20 (1)

G

For the AD602, the expression is

Gain (dB) = 32 VG + 10 (2)

Operation is specified for V

in the range from −625 mV dc to

G

+625 mV dc. The high impedance gain-control input ensures
minimal loading when driving many amplifiers in multiple­channel applications. The differential input configuration
provides flexibility in choosing the appropriate signal levels
and polarities for various control schemes.

= 0 V, the attenuator

G

Rev. E | Page 11 of 28

For example, the gain-control input can be fed differentially to
the inputs or single-ended by simply grounding the unused
input. In another example, if the gain is controlled by a DAC
providing a positive-only, ground-referenced output, the gain
control LO pin (either C1LO or C2LO) should be biased to a
fixed offset of 625 mV to set the gain to 0 dB when gain control
HI (C1HI or C2HI) is at zero and to set the gain to 40 dB when
at 1.25 V.

It is a simple matter to include a voltage divider to achieve other
scaling factors. When using an 8-bit DAC with an FS output of

2.55 V (10 mV/bit), a 1.6 divider ratio (generating 6.25 mV/bit)
results in a gain setting resolution of 0.2 dB/bit. Later in this
data sheet, cascading the two sections of an AD600 or AD602
when various options exist for gain control is explained (see the
Achieving 80 DB Gain Range section.)

SIGNAL-GATING INPUTS

Each amplifier section of the AD600/AD602 is equipped with a
signal gating function, controlled by a TTL or CMOS logic
input (GAT1 or GAT2). The ground references for these inputs
are the signal input grounds A1LO and A2LO, respectively.
Operation of the channel is unaffected when this input is LO or
left open-circuited. Signal transmission is blocked when this
input is HI. The dc output level of the channel is set to within a
few millivolts of the output ground (A1CM or A2CM) and
simultaneously the noise level drops significantly. The reduction
in noise and spurious signal feedthrough is useful in ultrasound
beam-forming applications, where many amplifier outputs are
summed.

COMMON-MODE REJECTION

A special circuit technique provides rejection of voltages
appearing between input grounds (A1LO and A2LO) and
output grounds (A1CM and A2CM). This is necessary because
of the op amp form of the amplifier, as shown in
The feedback voltage is developed across the RF1 resistor
(which, to achieve low noise, has a value of only 20 ). The
voltage developed across this resistor is referenced to the input
common, so the output voltage is also referred to that node.

For zero differential signal input between A1HI and A1LO, the
output A1OP simply follows the voltage at A1CM. Note that the
range of voltage differences that can exist between A1LO and
A1CM (or A2LO and A2CM) is limited to about ±100 mV.
Figure 18 shows the typical common-mode rejection ratio vs.
frequency.

Figure 21.

AD600/AD602

ACHIEVING 80 dB GAIN RANGE

The two amplifier sections of the X-AMP can be connected in
series to achieve higher gain. In this mode, the output of A1
(A1OP and A1CM) drives the input of A2 via a high-pass
network (usually just a capacitor) that rejects the dc offset.
The nominal gain range is now –2 dB to +82 dB for the AD600
or −22 dB to +62 dB for the AD602.

There are several options in connecting the gain-control inputs.
The choice depends on the desired SNR and gain error (output
ripple). The following examples feature the AD600; the
arguments generally apply to the AD602, with appropriate
changes to the gain values.

SEQUENTIAL MODE (MAXIMUM SNR)

In the sequential mode of operation, the SNR is maintained at
its highest level for as much of the gain control range as
possible, as shown in
0 dB to 80 dB.
general connections to accomplish this. Both gain-control
inputs, C1HI and C2HI, are driven in parallel by a positive-only,
ground-referenced source with a range of 0 V to 2.5 V.

Figure 22. Note here that the gain range is

Figure 23, Figure 24, and Figure 25 show the

85

80

75

70

65

60

55

SNR (dB)

50

45

40

35

30

–0.5

0

V

G

2.52.01.51.00.5

00538-022

3.0

Figure 22. SNR vs. Control Voltage Sequential Control (1 MHz Bandwidth)

An auxiliary amplifier that senses the voltage difference
between input and output commons is provided to reject this
common voltage.

V

G2

VO2 = 1.908V

A2

–41.07dB

41.07dB

OUTPUT
0dB

00538-023

INPUT

0dB

VC = 0V

A1

–40.00dB

–40.00dB

C1HI C1LO

V

G1

V

O1

= 0.592V

41.07dB

1.07dB

–42.14dB

C2HI C2LO

Figure 23. AD600 Gain Control Input Calculations for Sequential Control Operation (A)

A2

–1.07dB

41.07dB

V

G2

= 1.908V

V

O2

OUTPUT
40dB

00538-055

INPUT

0dB

VC = 1.25V

A1

–0.51dB

–0.51dB

C1HI C1LO

V

G1

VO1 = 0.592V

41.07dB

40.56dB

–41.63dB

C2HI C2LO

Figure 24. AD600 Gain Control Input Calculations for Sequential Control Operation (B)

V

G2

VO2 = 1.908V

A2

38.93dB

41.07dB

OUTPUT
80dB

00538-056

INPUT

0dB

VC = 2.5V

A1

0dB

0dB

C1HI C1LO

V

G1

VO1 = 0.592V

41.07dB

41.07dB

–2.14dB

C2HI C2LO

Figure 25. AD600 Gain Control Input Calculations for Sequential Control Operation (C)

Rev. E | Page 12 of 28

AD600/AD602

The gains are offset such that A2’s gain is increased only after
A1’s gain has reached its maximum value (see

Figure 26). Note
that for a differential input of −700 mV or less, the gain of a
single amplifier (A1 or A2) is at its minimum value of −1.07 dB;
for a differential input of +700 mV or more, the gain is at its
maximum value of +41.07 dB. Control inputs beyond these
limits do not affect the gain and can be tolerated without damage or
foldover in the response. See the

Specifications section for more

details on the allowable voltage range. The gain is now

Gain (dB) = 32 V

(3)

C

PARALLEL MODE (SIMPLEST GAIN-CONTROL
INTERFACE)

In this mode, the gain-control voltage is applied to both inputs
in parallel—C1HI and C2HI are connected to the control
voltage, and C1LO and C2LO are optionally connected to an
offset voltage of 0.625 V. The gain scaling is then doubled to
64dB/V, requiring only 1.25 V for an 80 dB change of gain. In
this case, the amplitude of the gain ripple is also doubled, as is
shown in
A2 decreases linearly as the gain is increased (see

Figure 29, and the instantaneous SNR at the output of

Figure 30).

where V

GAIN

(dB)

is the applied control voltage.

C

+41.07dB

A1 A2

+20dB

+1.07dB

0

–2.14

Figure 26. Explanation of Offset Calibration for Sequential Control

0

–0.56dB

0.625
20

*

GAIN OFFSET O F 1.07dB, OR 33.44mV

+40.56dB

*

*

809.1295.0

1.25

1.875

40

60

2.5
80

+38.93dB

–1.07dB

V

(V)

C

82.14

When VC is set to zero, VG1 = −0.592 V and the gain of A1 is

1.07 dB (recall that the gain of each amplifier section is 0 dB for
V

= 625 mV); meanwhile, VG2 = −1.908 V, so the gain of A2 is

G

−1.07 dB. The overall gain is thus 0 dB (see
= 1.25 V, VG1 = 1.25 V – 0.592 V = 0.658 V, which sets the

V

C

gain of A1 to 40.56 dB, while V

= 1.25 V – 1.908 V = −0.658 V,

G2

Figure 23). When

which sets A2’s gain at −0.56 dB. The overall gain is now 40 dB
(see

Figure 24). When VC = 2.5 V, the gain of A1 is 41.07 dB and
the gain of A2 is 38.93 dB, resulting in an overall gain of 80 dB
(see

Figure 25). This mode of operation is further clarified by
Figure 27, which is a plot of the separate gains of A1 and A2 and
the overall gain vs. the control voltage.

Figure 28 is a plot of the

gain error of the cascaded amplifiers vs. the control voltage.

LOW RIPPLE MODE (MINIMUM GAIN ERROR)

As can be seen in Figure 28 and Figure 29, the output ripple is
periodic. By offsetting the gains of A1 and A2 by half the
period of the ripple, or 3 dB, the residual gain errors of the two
amplifiers can be made to cancel.
lower gain ripple when configured in this manner.
plots the SNR as a function of gain; it is very similar to that in
the parallel mode.

00538-024

Figure 31 shows the much

Figure 32

Rev. E | Page 13 of 28

AD600/AD602

R

R

90

80

70

60

50

OVERALL G AIN (dB)

–10

40

30

20

10

0

–0.5

A1

COMBINED

0

V

C

A2

3.0

2.52.01.51.00.5

Figure 27. Plot of Separate and Overall Gains in Sequential Control

5

4

3

2

1

0

–1

OR (dB)

–2

–3

GAIN ER

–4

–5

–6

–7

–8

–0.5

0

1.51.00.5

V

C

2.0 2.5

3.0

Figure 28. Gain Error for Cascaded Stages—Sequential Control

5

4

3

2

1

0

–1

–2

GAIN ERROR (dB)

–3

–4

–5

–6

–0.1

0

0.2 0.4 0.6

V

C

1.21.00.8

Figure 29. Gain Error for Cascaded Stages—Parallel Control

00538-025

00538-026

00538-027

75

70

65

60

55

50

SNR (dB)

45

40

35

30

0.20
V

C

1.21.00.80. 60.4

Figure 30. SNR for Cascaded Stages—Parallel Control

1.2

1.0

0.8

0.6

0.4

0.2

OR (dB)

0.0

–0.2

GAIN ER

–0.4

–0.6

–0.8

–1.0

–1.2

0.1

0

0.2 0.8

V

C

1.21.11.00. 90.70.60.50.40.3

Figure 31. Gain Error for Cascaded Stages—Low Ripple Mode

80

75

70

65

60

55

SNR (dB)

50

45

40

35

0.20
V

C

1.21.00.80.60.4

Figure 32. SNR vs. Control Voltage—Low Ripple Mode

1.4

1.4

00538-028

00538-029

1.3

00538-030

Rev. E | Page 14 of 28

AD600/AD602

T

A

APPLICATIONS

The full potential of any high performance amplifier can only
be realized by careful attention to details in its applications. The
following pages describe fully tested circuits in which many
such details have already been considered. However, as is always
true of high accuracy, high speed analog circuits, the schematic
is only part of the story; this is no less true for the AD600/
AD602. Appropriate choices in the overall board layout and the
type and placement of power supply decoupling components
are very important. As explained previously, the input grounds
A1LO and A2LO must use the shortest possible connections.

The following circuits show examples of time-gain control for
ultrasound and sonar, methods for increasing the output drive,
and AGC amplifiers for audio and RF/IF signal processing
using both peak and rms detectors. These circuits also illustrate
methods of cascading X-AMPs for either maintaining the
optimal SNR or maximizing the accuracy of the gain-control
voltage for use in signal measurement. These AGC circuits can
be modified for use as voltage-controlled amplifiers in sonar
and ultrasound applications by removing the detector and
substituting a DAC or other voltage source for supplying the
control voltage.

TIME-GAIN CONTROL (TGC) AND TIME-VARIABLE
GAIN (TVG)

Ultrasound and sonar systems share a similar requirement: both
need to provide an exponential increase in gain in response to a
linear control voltage, that is, a gain control that is linear in dB.
Figure 33 shows the AD600/AD602 configured for a control
voltage ramp starting at −625 mV and ending at +625 mV for a
gain-control range of 40 dB. The polarity of the gain-control
voltage can be reversed, and the control voltage inputs C1HI
and C1LO can be reversed to achieve the same effect. The gain­control voltage can be supplied by a voltage-output DAC, such
as the

AD7244, which contains two complete DACs, operates

from ±5 V supplies, has an internal reference of +3 V, and
provides ±3 V of output swing. As such, it is well suited for use
with the AD600/AD602, needing only a few resistors to scale
the output voltage of the DACs to the levels needed by the
AD600/AD602.

CONTROLVOL

+625mV

0dB 40dB

Figure 33. The Simplest Application of the X-AMP Is as a TGC or TVG Amplifier

in Ultrasound or Sonar. Only the A1 connections are shown for simplicity.

INCREASING OUTPUT DRIVE

The AD600/AD602’s output stage has limited capability for
negative-load driving capability. For driving loads less than
500 , the load drive can be increased by approximately 5 mA
by connecting a 1 k pull-down resistor from the output to the
negative supply (see

DRIVING CAPACITIVE LOADS

For driving capacitive loads of greater than 5 pF, insert a 10 
resistor between the output and the load. This lowers the
possibility of oscillation.

V

IN

Figure 34. Adding a 1 kΩ Pull-Down Resistor Increases the X-AMP’s Output

Drive by About 5 mA. Only the A1 connections are shown for simplicity.

V

G

–625mV

C1LO

A1LO

GAT1

GAT2

A2LO

C2LO

A1HI

A2HI

GE,

A1
GAIN

1

2

3

4

5

6

7

8

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

Figure 34).

GAIN-CONTROL

VOLTAG E

+

A1

–

REF

–

A2

+

AD600/

AD602

VOLTAGE-OUTPUT

DAC

1

2

+

3

–

4

5

–

6

7

+

8

AD600 OR

AD602

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

9

C2HI

A1

A2

REF

+5V

V

G

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

1kΩ

–5V

+5V

–5V

ADDED

PULL-DOWN

RESISTOR

00538-031

00538-032

Rev. E | Page 15 of 28

AD600/AD602

V

V

REALIZING OTHER GAIN RANGES

Larger gain ranges can be accommodated by cascading
amplifiers. Combinations built by cascading two amplifiers
include −20 dB to +60 dB (using one AD602), −10 dB to +70 dB
(using ½ of an AD602 followed by ½ of an AD600), and 0 dB to
80 dB (using one AD600). In multiple-channel applications,
extra protection against oscillation can be provided by using
amplifier sections from different packages.

AN ULTRALOW NOISE VCA

The two channels of the AD600 or AD602 can operate in
parallel to achieve a 3 dB improvement in noise level, providing
1 nV/√Hz without any loss of gain accuracy or bandwidth.

In the simplest case, as shown in
A1HI and A2HI are tied directly together. The outputs A1OP
and A2OP are summed via R1 and R2 (100  each), and the
control inputs C1HI/C2HI and C1LO/C2LO operate in parallel.
Using these connections, both the input and output resistances
are 50 . Thus, when driven from a 50  source and terminated
in a 50  load, the gain is reduced by 12 dB, so the gain range
becomes –12 dB to +28 dB for the AD600 and −22 dB to
+18 dB for the AD602. The peak input capability remains
unaffected (1 V rms at the IC pins, or 2 V rms from an
unloaded 50  source). The loading on each output, with a
50  load, is effectively 200 , because the load current is
shared between the two channels, so the overall amplifier still
meets its specified maximum output and distortion levels for a
200  load. This amplifier can deliver a maximum sine wave
power of 10 dBm to the load.

GAIN-CONTROL

VOLTAG E

C1LO

1

A1HI

2

A1LO

GAT1

IN

GAT2

A2LO

A2HI

C2LO

Figure 35. An Ultralow Noise VCA Using the AD600 or AD602

+

3

–

4

5

–

6

+

7

8

AD600 OR

AD602

Figure 35, the signal inputs

V

G

–+

C1HI

16

A1CM

15

A1

REF

A2

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

+5V

–5V

100Ω

100Ω

V

OUT

50Ω

00538-033

A LOW NOISE, 6 dB PREAMPLIFIER

In some ultrasound applications, a high input impedance
preamplifier is needed to avoid the signal attenuation that
results from loading the transducer by the 100  input resistance
of the X-AMP. High gain cannot be tolerated because the
peak transducer signal is typically ±0.5 V, while the peak input
capability of the AD600 or AD602 is only slightly more than
±1 V. A gain of 2 is a suitable choice. It can be shown that if the
preamplifier’s overall referred-to-input (RTI) noise is the same
as that due to the X-AMP alone (1.4 nV/√Hz), the input noise
of nX2 preamplifier must be √(3/4) times as large, that is,

1.2 nV/√Hz.

+5

R1

49.9Ω

R2

174Ω

1µF

R4

42.2Ω

V

IN

R5

42.2Ω

1µF

R7

174Ω

R8

49.9Ω

Figure 36. A Low Noise Preamplifier for the AD600/AD602

An inexpensive circuit using complementary transistor types
chosen for their low r
determined by the ratio of the net collector load resistance to
the net emitter resistance. It is an open-loop amplifier. The gain
is ×2 (6 dB) only into a 100  load, assumed to be provided by
the input resistance of the X-AMP; R2 and R7 are in shunt
with this load, and their value is important in defining the gain.
For small-signal inputs, both transistors contribute an equal
transconductance that is rendered less sensitive to signal level
by the emitter resistors, R4 and R5. They also play a dominant
role in setting the gain.

1µF

Q1
MRF904

R3
562Ω

–5V

INPUT
GROUND

+5V

R6
562Ω

Q2
MM4049

1µF

–5V

is shown in Figure 36. The gain is

bb

0.1µF

0.1µF

OUTPUT

GROUND

100Ω
R

OF X-AMP

IN

00538-034

Rev. E | Page 16 of 28

AD600/AD602

V

This is a Class AB amplifier. As VIN increases in a positive
direction, Q1 conducts more heavily and its r

becomes lower

e

while Q2 increases. Conversely, increasingly negative values of
V

result in the re of Q2 decreasing, while the re of Q1 increases.

IN

The design is chosen such that the net emitter resistance is
essentially independent of the instantaneous value of V

,

IN

resulting in moderately low distortion. Low values of resistance
and moderately high bias currents are important in achieving
the low noise, wide bandwidth, and low distortion of this
preamplifier. Heavy decoupling prevents noise on the power
supply lines from being conveyed to the input of the X-AMP.

Table 4. Measured Preamplifier Performance

Measurement

Value Unit

Gain (f = 30 MHz) 6 dB
Bandwidth (−3 dB) 250 MHz
Input Signal for 1 dB Compression 1 V p-p
Distortion

VIN = 200 mV p-p HD2 0.27 %

HD3 0.14 %

VIN = 500 mV p-p HD2 0.44 %
HD3 0.58 %
System Input Noise 1.03 nV/√Hz

Spectral Density (NSD)

(Preamp plus X-AMP)
Input Resistance 1.4 kΩ
Input Capacitance 15 pF
Input Bias Current ±150 µA
Power Supply Voltage ±5 V
Quiescent Current 15 mA

+5

R3

46.4kΩ

R4

3.74kΩ

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

C4

0.1µF

+5V DEC

–5V DEC

100Ω

Figure 37. This Accurate HF AGC Amplifier Uses Three Active Components

RF

INPUT

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

–

–

+

A1

REF

A2

AD600

3

4

5

6

7

8

R1

100pF

A LOW NOISE AGC AMPLIFIER WITH 80 dB GAIN
RANGE

Figure 37 provides an example of the ease with which the
AD600 can be connected as an AGC amplifier. A1 and A2 are
cascaded, with 6 dB of attenuation introduced by the 100 
resistor R1, while a time constant of 5 ns is formed by C1 and
the 50  of net resistance at the input of A2. This has the dual
effect of lowering the overall gain range from 0 dB to 80 dB to

−6 dB to 74 dB and introducing a single-pole, low-pass filter
with a −3 dB frequency of about 32 MHz. This ensures stability
at the maximum gain for a slight reduction in the overall
bandwidth. The Capacitor C4 blocks the small dc offset voltage
at the output of A1 (which may otherwise saturate A2 at its
maximum gain) and introduces a high-pass corner at about
8 kHz, useful in eliminating low frequency noise and spurious
signals that can be present at the input.

+5V

VG´

C1

1µF

C3

15pF

AD590

C2

806Ω

1%

300µA
(AT 300K)

RF
OUTPUT

+5V DEC

–5V DEC

DECOUPLING NETWORK

Q1
2N3904

+

R2

V

PTAT

–

+5V

FB

FB

–5V

POWER SUPPLY

0.1µF

0.1µF

00538-035

Rev. E | Page 17 of 28

AD600/AD602

A simple half-wave detector is used based on Q1 and R2. The
average current into Capacitor C2 is the difference between the
current provided by the
collector current of Q1. In turn, the control voltage V
time integral of this error current. When V

AD590 (300 µA at 300 K, 27°C) and the

is the

G

(thus the gain) is

G

stable, the rectified current in Q1 must, on average, balance
exactly the current in the
small to do this, V

AD590. If the output of A2 is too

ramps up, causing the gain to increase until

G

Q1 conducts sufficiently. The operation of this control system
follows.

First, consider the particular case where R2 is zero and the
output voltage V

is a square wave at, for example, 100 kHz,

OUT

well above the corner frequency of the control loop. During the
time V

is negative, Q1 conducts. When V

OUT

is positive, it is

OUT

cut off. Since the average collector current is forced to be
300 A and the square wave has a 50% duty-cycle, the current
when conducting must be 600 A. With R2 omitted, the peak
value of V
about 700 mV) or 2 V

would be just the VBE of Q1 at 600 A (typically

OUT

p-p. This voltage, thus the amplitude at

BE

which the output stabilizes, has a strong negative temperature
coefficient (TC), typically –1.7 mV/°C. While this may not be
troublesome in some applications, the correct value of R2
renders the output stable with temperature.

To understand this, first note that the current in the

AD590 is

closely proportional to absolute temperature (PTAT). In fact,
this IC is intended for use as a thermometer. For the moment,
assume that the signal is a square wave. When Q1 is conducting,
V

is the sum of VBE. V

OUT

is also a voltage that is PTAT and

OUT

that can be chosen to have a TC equal but opposite to the TC of
the base-to-emitter voltage. This is actually nothing more than
the band gap voltage reference principle in thinly veiled
disguise. When R2 is chosen so that the sum of the voltage
across it and the V
about 1.2 V, V
provided Q1 and the

of Q1 is close to the band gap voltage of

BE

is stable over a wide range of temperatures,

OUT

AD590 share the same thermal environment.

An offset of 375 mV is applied to the inverting gain-control
inputs C1LO and C2LO. Therefore, the nominal –625 mV to
+625 mV range for V

is translated upwards (at VG´) to –0.25 V

G

for minimum gain to +1 V for maximum gain. This prevents
Q1 from going into heavy saturation at low gains and leaves
sufficient headroom of 4 V for the

AD590 to operate correctly

at high gains when using a 5 V supply.

In fact, the 6 dB interstage attenuator means that the overall
gain of this AGC system actually runs from –6 dB to +74 dB.
Thus, an input of 2 V rms would be required to produce a
1 V rms output at the minimum gain, which exceeds the 1 V rms
maximum input specification of the AD600. The available gain
range is therefore 0 dB to 74 dB (or X1 to X5000). Since the gain
scaling is 15.625 mV/dB (because of the cascaded stages), the
minimum value of V

´ is actually increased by 6 × +15.625 mV,

G

or about 94 mV, to −156 mV, so the risk of saturation in Q1 is
reduced.

The emitter circuit of Q1 is somewhat inductive (due its finite f
and base resistance). Consequently, the effective value of R2
increases with frequency. This results in an increase in the
stabilized output amplitude at high frequencies, but for the
addition of C3, determined experimentally to be 15 pF for the
2N3904 for maximum response flatness. Alternatively, a faster
transistor can be used here to reduce HF peaking.

Figure 38

shows the ac response at the stabilized output level of about

1.3 rms.

Figure 39 demonstrates the output stabilization for the
sine wave inputs of 1 mV to 1 V rms at frequencies of 100 kHz,
1 MHz, and 10 MHz.

t

Since the average emitter current is 600 A during each half­cycle of the square wave, a resistor of 833  would add a PTAT
voltage of 500 mV at 300 K, increasing by 1.66 mV/°C. In
practice, the optimum value of R2 depends on the transistor
used and, to a lesser extent, on the waveform for which the
temperature stability is to be optimized; for the devices shown
and sine wave signals, the recommended value is 806 . This
resistor also serves to lower the peak current in Q1 and the
200 Hz LP filter it forms with C2 helps to minimize distortion
due to ripple in V

. Note that the output amplitude under sine

G

wave conditions is higher than for a square wave because the
average value of the current for an ideal rectifier would be

0.637 times as large, causing the output amplitude to be 1.88 V
(= 1.2/0.637), or 1.33 V rms. In practice, the somewhat nonideal
rectifier results in the sine wave output being regulated to about

1.275 V rms.

Rev. E | Page 18 of 28

3dB

AGC OUTPUT CHANG E (dB)

0.1

Figure 38. AC Response at the Stabilized Output Level of 1.3 V rms

1

FREQUENCY (MHz)

10

100

00538-036

AD600/AD602

A

V

F

V

+0.2

0

–0.2

TIVE OUTPUT (dB)

–0.4

REL

0.001 0.01 0.1 1

Figure 39. Output Stabilization vs. rms Input for

Sine Wave Inputs at 100 kHz, 1 MHz, and 10 MHz

INPUT AMPLITUDE (V rms)

100kHz

1MHz

10MHz

00538-037

While the band gap principle used here sets the output
amplitude to 1.2 V (for the square wave case), the stabilization
point can be set to any higher amplitude, up to the maximum
output of ±(V

− 2) V that the AD600 can support. It is only

S

necessary to split R2 into two components of appropriate ratio
whose parallel sum remains close to the zero-TC value of
806 .

Figure 40 shows this and how the output can be raised

without altering the temperature stability.

5

R2A

300µA
(AT 300K)

Q1
2N3904

R2B

+

R2 = R2A || R2B ≈ 806Ω

V

PTAT

–

RF
OUTPUT

00538-038

AD590

TO A D600 PI N 16

TO AD600 PIN 11

Figure 40. Modification in Detector to Raise Output to 2 V rms

1µF

15pF

C2

C3

A WIDE RANGE, RMS-LINEAR dB MEASUREMENT
SYSTEM (2 MHz AGC AMPLIFIER WITH RMS
DETECTOR)

Monolithic rms-dc converters provide an inexpensive means to
measure the rms value of a signal of arbitrary waveform; they
can also provide a low accuracy logarithmic (decibel-scaled)
output. However, they have certain shortcomings. The first of
these is their restricted dynamic range, typically only 50 dB.
More troublesome is that the bandwidth is roughly proportional
to the signal level; for example, the
bandwidth of 900 kHz for an input of 100 mV rms but has a
bandwidth of only 100 kHz for a 10 mV rms input. Its
logarithmic output is unbuffered, uncalibrated, and not stable
over temperature. Considerable support circuitry, including at
least two adjustments and a special high TC resistor, is required
to provide a useful output.

AD636 provides a 3 dB

Rev. E | Page 19 of 28

These problems can be eliminated using an
detector element in an AGC loop, in which the difference
between the rms output of the amplifier and a fixed dc reference
are nulled in a loop integrator. The dynamic range and the
accuracy with which the signal can be determined are now
entirely dependent on the amplifier used in the AGC system.
Since the input to the rms-dc converter is forced to a constant
amplitude, close to its maximum input capability, the bandwidth is
no longer signal dependent. If the amplifier has an exactly
exponential (linear-dB) gain-control law, its control voltage V
is forced by the AGC loop to have the general form

()

RMSIN

OUT

VV 10log=

SCALE

(4)

V

RE

Figure 41 shows a practical wide dynamic range rms­responding measurement system using the AD600. Note that
the signal output of this system is available at A2OP, and the
circuit can be used as a wideband AGC amplifier with an rms­responding detector. This circuit can handle inputs from
100 V to 1 V rms with a constant measurement bandwidth of
20 Hz to 2 MHz, limited primarily by the AD636 rms converter.
Its logarithmic output is a loadable voltage accurately calibrated
to 100 mV/dB or 2 V per decade, which simplifies the
interpretation of the reading when using a DVM and is
arranged to be −4 V for an input of 100 V rms input, zero for
10 mV, and +4 V for a 1 V rms input. In terms of Equation 4,
V

is 10 mV and V

REF

SCALE

is 2 V.

Note that the peak log output of ±4 V requires the use of ±6 V
supplies for the dual op amp U3 (

AD712) although lower

supplies would suffice for the AD600 and
supplies are available, it is necessary to either use a reduced
value for V

(say 1 V, in which case the peak output would

SCALE

be only ±2 V) or restrict the dynamic range of the signal to
about 60 dB.

As in the previous case, the two amplifiers of the AD600 are
used in cascade. However, the 6 dB attenuator and low-pass
filter found in

Figure 21 are replaced by a unity gain buffer
amplifier U3A, whose 4 MHz bandwidth eliminates the risk of
instability at the highest gains. The buffer also allows the use of
a high impedance coupling network (C1/R3) that introduces a
high-pass corner at about 12 Hz. An input attenuator of 10 dB
(X0.316) is now provided by R1 + R2 operating in conjunction
with the AD600’s input resistance of 100 . The adjustment
provides exact calibration of the logarithmic intercept V
critical applications, but R1 and R2 can be replaced by a fixed
resistor of 215  if very close calibration is not needed, because
the input resistance of the AD600 (and all other key parameters
of it and the

AD636) is already laser trimmed for accurate

operation. This attenuator allows inputs as large as ±4 V to be
accepted, that is, signals with an rms value of 1 V combined
with a crest factor of up to 4.

AD636 as the

AD636. If only ±5 V

in

REF

G

AD600/AD602

V

C1

0.1µF

CAL

INPUT

1V rms

MAX

(SINE WAVE)

0dB

R1

115Ω

R2 200Ω

133kΩ

U3A

1/2

AD712

V

G

15.625mV/dB

C1LO

1

A1HI

2

+

A1LO

3

GAT1

4

R3

GAT2

A2LO

A2HI

C2LO

R4

3.01kΩ

5

6

7

8

–

–

+

R5

16.2kΩ

A1

A2

U1

AD600

REF

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

Figure 41. The Output of This Three-IC Circuit Is Proportional to the Decibel Value of the rms Input

The output of A2 is ac-coupled via another 12 Hz high-pass
filter formed by C2 and the 6.7 k input resistance of the
AD636. The averaging time constant for the rms-dc converter
is determined by C4. The unbuffered output of the AD636 (at
Pin 8) is compared with a fixed voltage of 316 mV set by the
positive supply voltage of 6 V and Resistors R6 and R7. V

REF

is
proportional to this voltage, and systems requiring greater
calibration accuracy should replace the supply dependent
reference with a more stable source.

Any difference in these voltages is integrated by the op amp
U3B, with a time constant of 3 ms formed by the parallel sum
of R6/R7 and C3. Now, if the output of the AD600 is too high,
V rms is greater than the setpoint of 316 mV, causing the output
of U3B—that is, V
noninverting). A fraction of V

—to ramp up (note that the integrator is

OUT

is connected to the inverting

OUT

gain-control inputs of the AD600, so causing the gain to be
reduced, as required, until V rms is exactly equal to 316 mV, at
which time the ac voltage at the output of A2 is forced to be
exactly 316 mV rms. This fraction is set by R4 and R5 such that
a 15.625 mV change in the control voltages of A1 and A2—
which would change the gain of the cascaded amplifiers by
1 dB—requires a change of 100 mV at V

. Notice here that

OUT

since A2 is forced to operate at an output level well below its
capacity, waveforms of high crest factor can be tolerated
throughout the amplifier.

+6V
DEC

–6V
DEC

rms

C2
2µF

–6V

DEC

AF/RF
OUTPUT

NC

NC

NC

C4

4.7µF

1

2

3

4

5

6

7

NC = NO CONNECT

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

U2

AD636

VPOS

COMM

LDLO

V

RMS

14

13

12

11

10

9

8

1/2

AD712

NC

NC

NC

R7

56.2kΩ

R6

3.16kΩ

+316.2mV

U3B

C3
1µF

+100mV/dB

0V = 0d B (AT 10mV rms)

+6V DEC

+6V

DEC

–6V

DEC

V

+6V

FB

0.1µF

0.1µF

FB

–6V

POWER SUPPLY

DECOUPLING

NETWORK

OUT

00538-039

To check the operation, assume an input of 10 mV rms is
applied to the input, which results in a voltage of 3.16 mV rms
at the input to A1, due to the 10 dB loss in the attenuator. If the
system operates as claimed, V

(and hence VG) should be 0.

OUT

This being the case, the gain of both A1 and A2 is 20 dB and the
output of the AD600 is therefore 100 times (40 dB) greater than
its input, which evaluates to 316 mV rms, the input required at
the

AD636 to balance the loop. Finally, note that unlike most

AGC circuits that need strong temperature compensation for
the internal kT/q scaling, these voltages, and thus the output of
this measurement system, are temperature stable, arising
directly from the fundamental and exact exponential
attenuation of the ladder networks in the AD600.

Typical results are presented for a sine wave input at 100 kHz.
Figure 42 shows that the output is held close to the setpoint of
316 mV rms over an input range in excess of 80 dB.

450

425

400

375

350

325

(mV)

300

OUT

275

V

250

225

200

175

150

10µ 100µ 101100m10m1m

INPUT SIGNAL (V rms)

Figure 42. RMS Output of A2 Held Close to the Setpoint 316 mV

for an Input Range of over 80 dB

00538-040

Rev. E | Page 20 of 28

AD600/AD602

R

This system can, of course, be used as an AGC amplifier in
which the rms value of the input is leveled.
decibel output voltage. More revealing is

Figure 43 shows the

Figure 44, which
shows that the deviation from the ideal output predicted by
Equation 1 over the input range 80 V to 500 mV rms is within
±0.5 dB, and within ±1 dB for the 80 dB range from 80 V to
800 mV. By suitable choice of the input attenuator R1 + R2, this
can be centered to cover any range from a low of 25 mV to
250 mV to a high of 1 mV to 10 V, with appropriate correction
to the value of V

. Note that V

REF

is not affected by the

SCALE

changes in the range. The gain ripple of ±0.2 dB seen in this
curve is the result of the finite interpolation error of the
X-AMP. Note that it occurs with a periodicity of 12 dB, twice
the separation between the tap points (because of the two
cascaded stages).

5

4

3

2

1

(V)

0

OUT

V

–1

–2

–3

–4

–5

10µ 100µ 101100m10m1m

Figure 43. The dB Output of

2.5

2.0

1.5

1.0

0.5

OR (dB)

0

–0.5

OUTPUT ER

–1.0

–1.5

–2.0

–2.5

10µ 100µ 101100m10m1m

Figure 44. Data from

from the Ideal Output Given in Equation 4

INPUT SIGNAL (V rms)

Figure 41’s Circuit is Linear over an 80 dB Range

INPUT SIGNAL (V rms)

Figure 42 Presented as the Deviation

00538-041

00538-042

Rev. E | Page 21 of 28

This ripple can be canceled whenever the X-AMP stages are
cascaded by introducing a 3 dB offset between the two pairs of
control voltages. A simple means to achieve this is shown in
Figure 45: the voltages at C1HI and C2HI are split by
±46.875 mV, or ±1.5 dB. Alternatively, either one of these pins
can be individually offset by 3 dB and a 1.5 dB gain adjustment
made at the input attenuator (R1 + R2).

C1HI

U1

AD600

NC = NO CONNECT

16

15

14

13

12

11

10

9

–6V

DEC

A1CM

A1OP
VPOS

VNEG

A2OP

A2CM

C2HI

10kΩ

+6V DEC

–6V DEC

–46.875mV

78.7Ω 78.7Ω

–6V DEC

C2
2µF

+46.875mV

+6V
DEC

10kΩ

3dB OFFSET

MODIFICATION

NC

NC

NC

1

2

3

4

5

6

7

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

U2

AD636

Figure 45. Reducing the Gain Error Ripple

The error curve shown in Figure 46 demonstrates that over the
central portion of the range the output voltage can be maintained
close to the ideal value. The penalty for this modification is the
higher errors at the extremities of the range. The next two
applications show how three amplifier sections can be cascaded
to extend the nominal conversion range to 120 dB, with the
inclusion of simple LP filters of the type shown in

Figure 37.

Very low errors can then be maintained over a 100 dB range.

2.5

2.0

1.5

1.0

0.5

0

–0.5

OUTPUT ERROR (dB)

–1.0

–1.5

–2.0

–2.5

10µ 100µ 101100m10m1m

INPUT SIGNAL (V rms)

00538-044

Figure 46. Using a 3 dB Offset Network Reduces Ripple

100 dB TO 120 dB RMS RESPONDING CONSTANT
BANDWIDTH AGC SYSTEMS WITH HIGH
ACCURACY dB OUTPUTS

The next two applications double as both AGC amplifiers and
measurement systems. In both, precise gain offsets are used to
achieve either a high gain linearity of ±0.1 dB over the full
100 dB range or the optimal SNR at any gain.

00538-043

AD600/AD602

INPUT

1V rms

MAX

(SINE WAVE)

R13

3.01kΩ

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

7

+

8

U1 AD600

+5V

FB

+5V

DEC

–5V

DEC

FB

–5V

POWER SUPPLY

DECOUPLING

NETWORK

Q1

2N3906

A1

A2

0.1µF

0.1µF

R14

301kΩ

R12

11.3kΩ

REF

16

15

14

13

12

11

10

9

+5V DEC

C1HI

A1CM

A1OP

VPOS

VNEG

A2OP

A2CM

C2HI

R15

19.6kΩ

R16

6.65kΩ

+5V
DEC

–5V
DEC

R6

10kΩ

C1

0.1µF

R1

133kΩ

C2

0.1µF

R4

133kΩ

–2dB
–62.5mV

R7

127Ω

Figure 47. RMS Responding AGC Circuit with 100 dB Dynamic Range

A 100 dB RMS/AGC SYSTEM WITH MINIMAL GAIN
ERROR (PARALLEL GAIN WITH OFFSET)

Figure 47 shows an rms-responding AGC circuit that can be
used equally well as an accurate measurement system. It accepts
inputs of 10 V to 1 V rms (−100 dBV to 0 dBV) with generous
overrange.
is accurately scaled 1 V per decade, that is, 50 mV/dB, with an
intercept (V
±2 dB were introduced between the amplifiers, provided by the
±62.5 mV introduced by R6 to R9. These offsets cancel a small
gain ripple that arises in the X-AMP from its finite interpolation
error, which has a period of 18 dB in the individual VCA
sections. The gain ripple of all three amplifier sections without
this offset (in which case, the gain errors simply add) is shown
in

Figure 49; it is still a remarkably low ±0.25 dB over the
108 dB range from 6 V to 1.5 V rms. However, with the gain
offsets connected, the gain linearity remains under ±0.1 dB over
the specified 100 dB range (see

Figure 48 shows the logarithmic output, V

= 0) at 3.16 mV rms (−50 dBV). Gain offsets of

LOG

LOG

Figure 50).

, which

U3A

1/4

AD713

R5

1.58kΩ

C3

220pF

0dB

R8

127ΩR910kΩ

R2

487Ω

U3B

+2dB
+62.5mV

NC

–5V

DEC

NC

NC

Figure 48. V

C1LO

1

A1HI

2

+

A1LO

R3

200Ω

1/4

AD713

+5V–5V

C5

22µF

1

VINP

U4

2

AD636

3

VNEG

4

CAVG

VLOG

5

BFOP

6

BFIN

7

5

4

3

2

1

0

–1

–2

LOGARIT HMIC OUTPUT (V)

–3

–4

–5

1µ 10µ 101100m10m1m100µ

Plotted vs. VIN for Figure 47’s Circuit Showing 120 dB AGC Range

LOG

GAT1

GAT2

A2LO

A2HI

C2LO

VPOS

COMM

LDLO

V

RMS

3

4

5

6

7

8

14

13

12

11

10

9

8

A1

–

REF

–

A2

+

U2 AD600

NC

NC

NC

R10

3.16kΩ

+316.2mV

NC = NO CO NNECT

INPUT SIGNAL (V rms)

16

15

14

13

12

11

10

R11

46.4kΩ

4.7µF

U3C

C1HI

A1CM

A1OP

VPOS

VNEG

A2OP

A2CM

C2HI

9

+5V DEC

C6

1/4

AD713

C4

2µF

+5V
DEC

–5V
DEC

V

OUT

V

LOG

00538-045

00538-046

Rev. E | Page 22 of 28

AD600/AD602

R

R

2.0

1.5

1.0

0.5

0.1

OR (dB)

0

–0.1

–0.5

GAIN ER

–1.0

–1.5

–2.0

1µ 10µ 101100m10m1m100µ

Figure 49. Gain Error for

2.0

1.5

1.0

0.5

0.1

OR (dB)

0

–0.1

–0.5

GAIN ER

–1.0

–1.5

–2.0

1µ 10µ 101100m10m1m100µ

Figure 50. Adding the 2 dB Offsets Improves the Linearization

INPUT SIGNAL (V rms)

Figure 41 Without the 2 dB Offset Modification

INPUT SIGNAL (V rms)

00538-047

00538-048

The maximum gain of this circuit is 120 dB. If no filtering was
used, the noise spectral density of the AD600 (1.4 nV/√Hz)
would amount to an input noise of 8.28 V rms in the full
bandwidth (35 MHz). At a gain of one million, the output noise
would dominate. Consequently, some reduction of bandwidth is
mandatory, and in the circuit of

Figure 47, it is due mostly to a
single-pole, low-pass filter R5/C3, which provides a −3 dB
frequency of 458 kHz, which reduces the worst-case output
noise (at V

) to about 100 mV rms at a gain of 100 dB. Of

AGC

course, the bandwidth (and therefore the output noise) could be
further reduced, for example, in audio applications, merely by
increasing C3. The value chosen for this application is optimal
in minimizing the error in the V

output for small input signals.

LOG

The AD600 is dc-coupled, but even miniscule offset voltages at
the input would overload the output at high gains; thus, high­pass filtering is also needed. To provide operation at low
frequencies, two simple 0s at about 12 Hz are provided by
R1/C1 and R4/C2; op amp sections U3A and U3B (

AD713)

are used to provide impedance buffering, because the input
resistance of the AD600 is only 100 . A further 0 at 12 Hz is
provided by C4 and the 6.7 k input resistance of the AD636
rms converter.

The rms value of V
averaging time for this process is determined by C5, and the
value shown results in less than 1% rms error at 20 Hz. The
slowly varying V rms is compared with a fixed reference of
316 mV, derived from the positive supply by R10/R11. Any
difference between these two voltages is integrated in C6, in
conjunction with Op Amp U3C, the output of which is V
fraction of this voltage, determined by R12 and R13, is returned
to the gain control inputs of all AD600 sections. An increase in
V

lowers the gain because this voltage is connected to the

LOG

inverting polarity control inputs.

In this case, the gains of all three VCA sections are being varied
simultaneously, so the scaling is not 32 dB/V but 96 dB/V or

10.42 mV/dB. The fraction of V
50 mV/dB is therefore 10.42/50 or 0.208. The resulting full­scale range of V
circuit to operate from ±5 V supplies.

Optionally, the scaling can be altered to 100 mV/dB, which
would be more easily interpreted when V
DVM by increasing R12 to 25.5 k. The full-scale output of
±5 V then requires the use of supply voltages of at least ±7.5 V.

A simple attenuator of 16.6 ± 1.25 dB is formed by R2/R3
and the 100  input resistance of the AD600. This allows the
reference level of the decibel output to be precisely set to 0 for
an input of 3.16 mV rms, and thus center the 100 dB range
between 10 V and 1 V. In many applications, R2/R3 can be
replaced by a fixed resistor of 590 . For example, in AGC
applications, neither the slope nor the intercept of the
logarithmic output is important.

A few additional components (R14 to R16 and Q1) improve the
accuracy of V

LOG

small gains). The gain starts rolling off when the input to the
first amplifier, U1A, reaches 0 dB. To compensate for this
nonlinearity, Q1 turns on at V
feedback to the control inputs of the AD600s, thereby needing a
smaller voltage at V
the setpoint of 316 mV rms.

A 120 dB RMS/AGC SYSTEM WITH OPTIMAL SNR
(SEQUENTIAL GAIN)

In the last case, all gains were adjusted simultaneously, resulting
in an output SNR that is always less than optimal. The use of
sequential gain control results in a major improvement in SNR,
with only a slight penalty in the accuracy of V
penalty in the stabilization accuracy of V
increase the gain of the earlier stages first (as the signal level
decreases) and maintain the highest SNR throughout the
amplifier chain. This can be easily achieved with the AD600
because its gain is accurate even when the control input is
overdriven. That is, each gain control window of 1.25 V is
used fully before moving to the next amplifier to the right.

is generated at Pin 8 of the AD636; the

LOG

LOG

required to set its scaling to

LOG

is nominally ±2.5 V. This scaling allows the

LOG

is displayed on a

LOG

at the top end of the signal range (that is, for

~ 1.5 V and increases the

LOG

to maintain the input to the AD636 to

LOG

, and no

LOG

. The idea is to

AGC

. A

Rev. E | Page 23 of 28

AD600/AD602

Figure 51 shows the circuit for the sequential control scheme.
R6 to R9 with R16 provide offsets of 42.14 dB between the
individual amplifiers to ensure smooth transitions between the
gain of each successive X-AMP, with the sequence of gain
increase being U1A, then U1B, and then U2A. The adjustable
attenuator provided by R3 + R17 and the 100  input resistance
of U1A, as well as the fixed 6 dB attenuation provided by R2
and the input resistance of U1B, are included both to set V

LOG

to

0dB

ADJUST

INPUT

R17

115Ω

R3

200Ω

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

7

+

8

A1

REF

A2

U1 AD600

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

+6V
DEC

–6V
DEC

C1

0.1µF

133kΩ

C2

0.1µF

133kΩ

R1

R4

read 0 dB when V

is 3.16 mV rms and to center the 100 dB

IN

range between 10 µV rms and 1 V rms input. R5 and C3
provide a 3 dB noise bandwidth of 30 kHz. R12 to R15 change
the scaling from 625 mV/decade at the control inputs to
1 V/decade at the output. At the same time, R12 to R15 center
the dynamic range at 60 dB, which occurs if the V
equal to 0. These arrangements ensure that the V
within the ±6 V supplies.

R5

5.36kΩ

0.001µF

U3A

1/4

AD713

C3

R2

100Ω

U3B

1/4

AD713

C1LO

A1HI

A1LO

GAT1

GAT2

A2LO

A2HI

C2LO

1

2

+

3

–

4

5

–

6

7

+

8

A1

REF

A2

U2 AD600

C1HI

16

A1CM

15

A1OP

14

VPOS

13

VNEG

12

A2OP

11

A2CM

10

C2HI

9

of U1B is

G

still fits

LOG

C4

2µF

+5V
DEC

–5V
DEC

V

OUT

+6V

FB

+6V

DEC

–6V

DEC

0.1µF

0.1µF

FB

–6V

POWER SUPPLY

DECOUPLING

NETWORK

+6V DEC

R15

5.11kΩ

R14

7.32kΩ

+6V

R13

866Ω

R6

3.4kΩ

R12
1kΩ

R7

1kΩ

R8

294Ω

R9

1kΩ

–6V

DEC

R16

287Ω

NC

NC

NC

1

2

3

4

5

6

7

VINP

VNEG

CAVG

VLOG

BFOP

BFIN

C5

22µF

U4

AD636

COMM

VPOS

LDLO

V

RMS

14

NC

13

12

NC

NC

11

10

9

8

Figure 51. 120 dB Dynamic Range RMS Responding Circuit Optimized for SNR

R10

3.16kΩ

+316.2mV

NC = NO CONNECT

R11

56.2kΩ

4.7µF

U3C

AD713

+6V DEC

C6

1/4

V

LOG

00538-049

Rev. E | Page 24 of 28

AD600/AD602

5

4

3

2

1

0

–1

–2

LOGARITHMIC OUTPUT (V)

–3

–4

–5

1µ 10µ 101100m10m1m100µ

Figure 52. V

Figure 52 shows V

INPUT SIGNAL (V rms)

Is Linear over the Full 120 dB Range

LOG

to be linear over a full 120 dB range.

LOG

00538-050

Figure 53 shows the error ripple due to the individual gain
functions bounded by ±0.2 dB (dotted lines) from 6 V to 2 V.
The small perturbations at about 200 V and 20 mV, caused by
the impracticality of matching the gain functions perfectly, are
the only sign that the gains are now sequential.
plot of V

that remains very close to its set value of 316 mV

AGC

Figure 54 is a

rms over the full 120 dB range.

To compare the SNRs in the simultaneous and sequential
modes of operation more directly, all interstage attenuation was
eliminated (R2 and R3 in

Figure 47 and R2 in Figure 51), the
input of U1A was shorted, R5 was selected to provide a 20 kHz
bandwidth (R5 = 7.87 k), and only the gain control was
varied, using an external source. The rms value of the noise was
then measured at V

and expressed as an SNR relative to

OUT

0 dBV, which is almost the maximum output capability of the
AD600. Results for the simultaneous mode can be seen in
Figure 55. The SNR degrades uniformly as the gain is increased.
Note that since the inverting gain control was used, the gain in
this curve and in

Figure 56 decreases for more positive values of

the gain-control voltage.

2.0

1.5

1.0

0.5

0.2
0

–0.2

–0.5

GAIN ERRO R (dB)

–1.0

–1.5

–2.0

1µ 10µ 101100m10m1m100µ

INPUT SIGNAL (V rms)

Figure 53. Error Ripple due to the Individual Gain Functions

00538-051

400

350

300

GAIN ERROR (mV)

250

200

1µ 10µ 101100m10m1m100µ

Figure 54. V

INPUT SIGNAL (V rms)

Remains Close to Its Setpoint of

AGC

00538-052

316 mV rms over the Full 120 dB Range

90

VC SCALE = 10.417mV/dB

80

70

60

50

40

SNR (dB)

30

20

10

0

–625.0–833.2

V

(mV)

C

625.0416.6208.30–208.3–416.6

Figure 55. SNR vs. Control Voltage for Parallel Gain Control (See

00538-053

833.2

Figure 47)

In contrast, the SNR for the sequential mode is shown in Figure 56.
U1A always acts as a fixed noise source; varying its gain has no
influence on the output noise. This is a feature of the X-AMP
technique. Therefore, for the first 40 dB of control range
(actually slightly more, as is explained later), when only this
VCA section has its gain varied, the SNR remains constant.
During this time, the gains of U1B and U2A are at their
minimum value of −1.07 dB.

90

VC SCALE = 31. 25mV/dB

80

70

60

50

40

SNR (dB)

30

20

10

0

–0.558–1.183

VC (V)

3.1922.5671.9421.3170.6920. 067

Figure 56. SNR vs. Control Voltage for Sequential Gain Control (See

00538-054

3.817

Figure 51)

Rev. E | Page 25 of 28

AD600/AD602

For the next 40 dB of control range, the gain of U1A remains
fixed at its maximum value of 41.07 dB and only the gain of
U1B is varied, while that of U2A remains at its minimum value
of −1.07 dB. In this interval, the fixed output noise of U1A is
amplified by the increasing gain of U1B and the SNR
progressively decreases.

Once U1B reaches its maximum gain of 41.07 dB, its output
also becomes a gain independent noise source; this noise is
presented to U2A. As the control voltage is further increased,
the gains of both U1A and U1B remain fixed at their maximum
value of 41.07 dB, and the SNR continues to decrease.
clearly shows this, because the maximum SNR of 90 dB is
extended for the first 40 dB of input signal before it starts to roll off.

Figure 56

This arrangement of staggered gains can be easily implemented
because when the control inputs of the AD600 are overdriven,
the gain limits to its maximum or minimum values without side
effects. This eliminates the need for awkward nonlinear shaping
circuits that have previously been used to break up the gain
range of multistage AGC amplifiers. The precise values of the
AD600’s maximum and minimum gain (not 0 dB and +40 dB
but −1.07 dB and +41.07 dB) explain the rather odd values of
the offset values that are used.

The optimization of the output SNR is of obvious value in AGC
systems. However, in applications where these circuits are
considered for their wide range logarithmic measurement
capabilities, the inevitable degradation of the SNR at high gains
need not seriously impair their utility. In fact, the bandwidth of
the circuit shown in
measurement accuracy by altering the shape of the log error
curve at low signal levels (see

Figure 47 was specifically chosen to improve

Figure 53).

Rev. E | Page 26 of 28

AD600/AD602

0

OUTLINE DIMENSIONS

0.800 (20.32)

0.790 (20.07)

0.780 (19.81)

0.210
(5.33)

0.150 (3.81)

0.130 (3.30)

0.115 (2.92)

0.022 (0.56)

0.018 (0.46)

0.014 (0.36)

16

1

PIN 1

0.100 (2.54)
BSC

MAX

0.070 (1.78)

0.060 (1.52)

0.045 (1.14)

CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.

COMPLIANT TO JEDEC STANDARDS MS-001-AB

Figure 57. 16-Lead Plastic Dual In-Line Package [PDIP]

Dimensions shown in inches and (millimeters)

9

0.280 (7.11)

0.250 (6.35)

0.240 (6.10)

8

0.015
(0.38)
MIN

SEATING
PLANE

0.005 (0.13)
MIN

Narrow Body (N-16)

0.060 (1.52)
MAX

0.015 (0.38)
GAUGE

PLANE

0.325 (8.26)

0.310 (7.87)

0.300 (7.62)

0.430 (10.92)
MAX

0.195 (4.95)

0.130 (3.30)

0.115 (2.92)

0.014 (0.36)

0.010 (0.25)

0.008 (0.20)

0.005 (0.13) MIN

16

1

PIN 1

0.840 (21.34) MAX

.200 (5.08)

MAX

0.200 (5.08)

0.125 (3.18)

0.023 (0.58)

0.014 (0.36)

CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.

Figure 58. 16-Lead Ceramic Dual In-Line Package [CERDIP]

Dimensions shown in inches and (millimeters)

0.098 (2.49) MAX

0.100 (2.54) BSC

0.070 (1.78)

0.030 (0.76)

9

8

(Q-16)

0.310 (7.87)

0.220 (5.59)

0.060 (1.52)

0.015 (0.38)

0.150
(3.81)
MIN

SEATING
PLANE

15°

0.320 (8.13)

0.290 (7.37)

0°

0.015 (0.38)

0.008 (0.20)

10.50 (0.4134)

10.10 (0.3976)

16

1

1.27 (0.0500)
BSC

0.30 (0.0118)

0.10 (0.0039)

COPLANARITY

0.10

CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS
(IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR
REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.

0.51 (0.0201)

0.31 (0.0122)

COMPLIANT TO JEDEC STANDARDS MS-013-AA

9

7.60 (0.2992)

7.40 (0.2913)

8

2.65 (0.1043)

2.35 (0.0925)

SEATING
PLANE

10.65 (0.4193)

10.00 (0.3937)

0.33 (0.0130)

0.20 (0.0079)

0.75 (0.0295)

0.25 (0.0098)

8°
0°

× 45°

1.27 (0.0500)

0.40 (0.0157)

Figure 59. 16-Lead Standard Small Outline Package [SOIC_W]

Wide Body (RW-16)

Dimensions shown in millimeters and (inches)

Rev. E | Page 27 of 28

AD600/AD602

ORDERING GUIDE

Model Gain Range Temperature Range Package Description Package Option

AD600AQ 0 dB to 40 dB −40°C to +85°C
AD600AR 0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD600AR-REEL 0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD600AR-REEL7 0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD600ARZ1 0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD600ARZ-R7
AD600ARZ-RL

1

1

0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16

0 dB to 40 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD600JN 0 dB to 40 dB 0°C to 70°C 16-Lead PDIP N-16
AD600JNZ

1

0 dB to 40 dB 0°C to 70°C 16-Lead PDIP N-16
AD600JR 0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD600JR-REEL 0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD600JR-REEL7 0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD600JRZ
AD600JRZ-R7
AD600JRZ-RL
AD600SQ/883B

1

1

1

2

0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16

0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16

0 dB to 40 dB 0°C to 70°C 16-Lead SOIC_W RW-16

0 dB to 40 dB −55°C to +125°C 16-Lead CERDIP Q-16
AD602AQ −10 dB to +30 dB −40°C to +85°C 16-Lead CERDIP Q-16
AD602AR −10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD602AR-REEL −10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD602AR-REEL7 −10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD602ARZ
AD602ARZ-R7
AD602ARZ-RL

1

1

1

−10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16

−10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16

−10 dB to +30 dB −40°C to +85°C 16-Lead SOIC_W RW-16
AD602JCHIPS DIE
AD602JN −10 dB to +30 dB 0°C to 70°C 16-Lead PDIP N-16
AD602JNZ

1

−10 dB to +30 dB 0°C to 70°C 16-Lead PDIP N-16
AD602JR −10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD602JR-REEL –10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD602JR-REEL7 −10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16
AD602JRZ
AD602JRZ-R7
AD602JRZ-RL
AD602SQ/883B3

1

Z = Pb-free part.

2

Refer to AD600/AD602 military data sheet. Also available as 5962-9457201MEA.

3

Refer to AD600/AD602 military data sheet. Also available as 5962-9457202MEA.

1

1

1

−10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16

–10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16

−10 dB to +30 dB 0°C to 70°C 16-Lead SOIC_W RW-16

−10 dB to +30 dB −55°C to +150°C

16-Lead CERDIP

16-Lead CERDIP

Q-16

Q-16

©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
C00538-0-1/06(E)

Rev. E | Page 28 of 28