67 4
3
2
1
138
9 10 11
125
14
15
16
A3
A4
A1
A2
R3
R
B
R1
R2
R4
R5
R6
AD588
GAIN
ADJ
GND
SENSE
+IN
GND
SENSE
–IN
V
LOW
BAL
ADJ
VCTA4 IN
–V
S
+V
S
A4 OUT
FORCE
A4 OUT
SENSE
A3 OUT
FORCE
A3 OUT
SENSE
A3 IN
V
HIGH
NOISE
REDUCTION
a
High Precision Voltage Reference
AD588*
FEATURES
Low Drift: 1.5 ppm/8C
Low Initial Error: 1 mV
Pin-Programmable Output
+10 V, +5 V, 65 V Tracking, –5 V, –10 V
Flexible Output Force and Sense Terminals
High Impedance Ground Sense
Machine-lnsertable DIP Packaging
MIL-STD-883 Compliant Versions Available
PRODUCT DESCRIPTION
The AD588 represents a major advance in the state-of-the-art in
monolithic voltage references. Low initial error and low temperature drift give the AD588 absolute accuracy performance
previously not available in monolithic form. The AD588 uses a
proprietary ion-implanted buried Zener diode, and laser-waferdrift trimming of high stability thin-film resistors to provide outstanding performance at low cost.
The AD588 includes the basic reference cell and three additional amplifiers which provide pin-programmable output
ranges. The amplifiers are laser-trimmed for low offset and low
drift to maintain the accuracy of the reference. The amplifiers
are configured to allow Kelvin connections to the load and/or
boosters for driving long lines or high-current loads, delivering
the full accuracy of the AD588 where it is required in the application circuit.
The low initial error allows the AD588 to be used as a system
reference in precision measurement applications requiring 12-bit
absolute accuracy. In such systems, the AD588 can provide a
known voltage for system calibration in software and the low
drift allows compensation for the drift of other components in
a system. Manual system calibration and the cost of periodic
recalibration can therefore be eliminated. Furthermore, the
mechanical instability of a trimming potentiometer and the
potential for improper calibration can be eliminated by using the
AD588 in conjunction autocalibration software.
The AD588 is available in seven versions. The AD588 JQ and
KQ grades are packaged in a 16-pin cerdip and are specified for
0°C to +70°C operation. AD588AQ and BQ grades are packaged
in a 16-pin cerdip and are specified for the –25°C to +85°C industrial temperature range. The ceramic AD588SQ and TQ
grades are specified for the full military/aerospace temperature
range. For military surface mount applications, the AD588SE
and TE grades are also available in 20-pin LCC packages.
*
Covered by Patent Number 4,644,253.
REV. B
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
FUNCTIONAL BLOCK DIAGRAM
PRODUCT HIGHLIGHTS
1. The AD588 offers 12-bit absolute accuracy without any user
adjustments. Optional fine-trim connections are provided for
applications requiring higher precision. The fine-trimming
does not alter the operating conditions of the Zener or the
buffer amplifiers and thus does not increase the temperature
drift.
2. Output noise of the AD588 is very low—typically 6 µV p-p.
A pin is provided for additional noise filtering using an external capacitor.
3. A precision ±5 V tracking mode with Kelvin output connections is available with no external components. Tracking
error is less than one millivolt and a fine-trim is available for
applications requiring exact symmetry between the +5 V and
–5 V outputs.
4. Pin strapping capability allows configuration of a wide variety
of outputs: ±5 V, +5 V and +10 V, –5 V & –10 V dual outputs or +5 V, –5 V, +10 V, –10 V single outputs.
5. Extensive temperature testing at –55°C, –25°C, 0°C, +25°C,
+50°C, +70°C, +85°C and +125°C ensures that the specified temperature coefficient is truly representative of device
performance.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700 Fax: 617/326-8703
AD588–SPECIFICA TIONS
(typical @ + 258C, +10 V output, VS = 615 V unless otherwise noted1)
AD588SQ AD588JQ/AQ/TQ AD588KQ/BQ
Min Typ Max Min Typ Max Min Typ Max Units
OUTPUT VOLTAGE ERROR
+10 V, –10 V Outputs –5 +5 –3 +3 –1 +1 mV
+5 V, –5 V Outputs –5 +5 –3 +3 –1 +1 mV
±5 V TRACKING MODE
Symmetry Error –1.5 +1.5 –1.5 +1.5 –0.75 +0.75 mV
OUTPUT VOLTAGE DRIFT
0°C to +70°C (J, K, B) –3 ±2 +3 –1.5 +1.5 ppm/°C
–25°C to +85°C (A, B) –3 +3 –3 +3 ppm/°C
–55°C to +125°C (S, T) –6 +6 –4 +4 ppm/°C
GAIN ADJ AND BAL ADJ
2
Trim Range ±4 ±4 ±4mV
Input Resistance 150 150 150 kΩ
LINE REGULATION
T
to T
MIN
MAX
3
6200 6200 6200 µV/V
LOAD REGULATION
to T
T
MIN
+10 V Output, 0 < I
–10 V Output, –10 < I
MAX
< 10 mA 650 650 650 µV/mA
OUT
< 0 mA 650 650 650 µV/mA
OUT
SUPPLY CURRENT
to T
T
MIN
MAX
6 10 6 10 6 10 mA
Power Dissipation 180 300 180 300 180 300 mW
OUTPUT NOISE (Any Output)
0.1 Hz to 10 Hz 6 6 6 µV p-p
Spectral Density, 100 Hz 100 100 100 nV/√Hz
LONG-TERM STABILITY (@ +25°C) 15 15 15 ppm/1000 hr
BUFFER AMPLIFIERS
Offset Voltage 100 100 10 µV
Offset Voltage Drift 1 1 1 µV/°C
Bias Current 20 20 20 nA
Open Loop Gain 110 110 110 dB
Output Current A3, A4 –10 +10 –10 +10 –10 +10 mA
Common-Mode Rejection (A3, A4)
= 1 V p-p 100 100 100 dB
V
CM
Short-Circuit Current 50 50 50 mA
TEMPERATURE RANGE
Specified Performance
J, K Grades 0 +70 0 +70 °C
A, B Grades –25 +85 –25 +85 °C
S, T Grades –55 +125 –55 +125 °C
NOTES
1
Output Configuration
+10 V Figure 2a
–10 V Figure 2c
+5 V, –5 V, ±5 V Figure 2b
Specifications tested using +10 V configuration unless otherwise indicated.
2
Gain and balance adjustments guaranteed capable of trimming output voltage
error and symmetry error to zero.
3
Test Conditions:
+10 V Output –VS = –15 V, 13.5 V ≤ +VS ≤ 18 V
–10 V Output –18 V ≤ –VS ≤ –13.5 V, +VS = 15 V
±5 V Output +VS = +18 V, –VS = –18 V
Specifications subject to change without notice
Specifications shown in boldface are tested on all production units at final
electrical test. Results from those tests are used to calculate outgoing quality
levels. All min and max specifications are guaranteed, although only those
shown in boldface are tested on all production units.
+VS = +10.8 V, –VS = –10.8 V
Part Initial Temperature Temperature Package
Number1Error Coefficient Range °C Option
AD588AQ 3 mV 3 ppm/°C –25 to +85 Cerdip (Q-16)
AD588BQ 1 mV 1.5 ppm/°C –25 to +85
AD588SQ 5 mV 6 ppm/°C –55 to +125 Cerdip (Q-16)
AD588TQ 3 mV 4 ppm/°C –55 to +125 Cerdip (Q-16)
AD588JQ 3 mV 3 ppm/°C 0 to +70 Cerdip (Q-16)
AD588KQ 1 mV 1.5 ppm/°C 0 to +70 Cerdip (Q-16)
N
OTES
1
For details on grade and package offerings screened in accordance with MIL-STD-883,
refer to the Analog Devices Military Products Databook or current AD588/883B.
2
Temperature coefficient specified from 0°C to +70°C.
ORDERING GUIDE
2
Cerdip (Q-16)
–2–
REV. B
AD588
ABSOLUTE MAXIMUM RATINGS*
+VS to –VS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .36 V
Power Dissipation
(+25°C)
Q Package . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .600 mW
Storage Temperature Range . . . . . . . . . . . . .–65°C to +150°C
Lead Temperature Range (Soldering 10 sec) . . . . . . . . +300°C
Package Thermal Resistance
Q (θ
) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .90/25°C/W
JA/θJC
Output Protection: All Outputs Safe If Shorted to Ground
*Stresses above those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. This is a stress rating only and functional
operation of the device at these or any other conditions above those indicated in the
operational sections of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
PIN CONFIGURATIONS
THEORY OF OPERATION
The AD588 consists of a buried Zener diode reference, amplifiers used to provide pin programmable output ranges, and associated thin-film resistors as shown in the block diagram of
Figure 1. The temperature compensation circuitry provides the
device with a temperature coefficient of 1.5 ppm/°C or less.
Figure 1. AD588 Functional Block Diagram
Amplifier A1 performs several functions. A1 primarily acts to
amplify the Zener voltage from 6.5 V to the required 10 V output. In addition, A1 also provides for external adjustment of the
10 V output through Pin 5, the GAIN ADJUST. Using the bias
compensation resistor between the Zener output and the noninverting input to A1, a capacitor can be added at the NOISE
REV. B
–3–
REDUCTION pin (Pin 7) to form a low-pass filter and reduce
the noise contribution of the Zener to the circuit. Two matched
10 kΩ nominal thin-film resistors (R4 and R5) divide the 10 V
output in half. Pin V
(Pin 11) provides access to the center of
CT
the voltage span and Pin 12 (BALANCE ADJUST) can be used
for fine adjustment of this division.
Ground sensing for the circuit is provided by amplifier A2. The
noninverting input (Pin 9) senses the system ground which will
be transferred to the point on the circuit where the inverting
input (Pin 10) is connected. This may be Pin 6, 8 or 11. The
output of A2 drives Pin 8 to the appropriate voltage. Thus, if
Pin 10 is connected to Pin 8, the V
pin will be the same
LOW
voltage as the system ground. Alternatively, if Pin 10 is connected to the V
pin, it will be ground and Pin 6 and Pin 8 will
CT
be +5 V and –5 V respectively.
Amplifiers A3 and A4 are internally compensated and are used
to buffer the voltages at Pins 6, 8, and 11 as well as to provide a
full Kelvin output. Thus, the AD588 has a full Kelvin capability
by providing the means to sense a system ground and provide
forced and sensed outputs referenced to that ground.
APPLYING THE AD588
The AD588 can be configured to provide +10 V and –10 V reference outputs as shown in Figures 2a and 2c respectively. It
can also be used to provide +5 V, –5 V or a ±5 V tracking reference as shown in Figure 2b. Table I details the appropriate pin
connections for each output range. In each case, Pin 9 is connected to system ground and power is applied to Pins 2 and 16.
The architecture of the AD588 provides ground sense and
uncommitted output buffer amplifiers which offer the user a
great deal of functional flexibility. The AD588 is specified and
tested in the configurations shown in Figure 2. The user may
choose to take advantage of the many other configuration options available with the AD588. However, performance in these
configurations is not guaranteed to meet the extremely stringent
data sheet specifications.
As indicated in Table I, a +5 V buffered output can be provided
using amplifier A4 in the +10 V configuration (Figure 2a). A
–5 V buffered output can be provided using amplifier A3 in the
–10 V configuration (Figure 2c). Specifications are not guaranteed for the +5 V or –5 V outputs in these configurations. Performance will be similar to that specified for the +10 V or –10 V
outputs.
As indicated in Table I, unbuffered outputs are available at Pins
6, 8 and 11. Loading of these unbuffered outputs will impair
circuit performance.
Amplifiers A3 and A4 can be used interchangeably. However,
the AD588 is tested (and the specifications are guaranteed) with
the amplifiers connected as indicated in Figure 2 and Table I.
When either A3 or A4 is unused, its output force and sense pins
should be connected and the input tied to ground.
Two outputs of the same voltage may be obtained by connecting
both A3 and A4 to the appropriate unbuffered output on Pins 6,
8 or 11. Performance in these dual output configurations will
typically meet data sheet specifications.
CALIBRATION
Generally, the AD588 will meet the requirements of a precision
system without additional adjustment. Initial output voltage
error of 1 mV and output noise specs of 10 µV p-p allow for
Table I. AD588 Connections
AD588
Connect Buffered
Pin 10 Unbuffered
1
Output on Pins Output Buffered Output on Pins
Range To Pin: –10 V –5 V 0 V +5 V +10 V Connections –10 V –5 V 0 V +5 V +10 V
+10 V 8 – – 8 11 6 11–13 & 14–15 – – – 15 –
6–4 & 3–1 – – – – 1
–5 V or +5 V 11 – 18 11 6 – 8–13 & 14–15 – 15 – – –
6–4 & 3–1 – – – 1
–10 V 6 8 11 6 – – 8–13 & 14–15 – 15 – – –
11–4 & 3–1 1 – – – –
+5 V –––6–6–4 & 3–1 – – – 1 –
11
–5 V – 8 – – – 8–13 & 14–15 – 15 – – –
1
“Unbuffered” outputs should not be loaded.
accuracies of 12–16 bits. However, in applications where an
even greater level of accuracy is required, additional calibration
may be called for. Provision for trimming has been made
through the use of the GAIN ADJUST and BALANCE ADJUST pins (Pins 5 and 12 respectively).
The AD588 provides a precision 10 V span with a center tap
(V
) which is used with the buffer and ground sense amplifiers
CT
to achieve the voltage output configurations in Table I. GAIN
ADJUST and BALANCE ADJUST can be used in any of these
configurations to trim the magnitude of the span voltage and the
position of the center tap within the span. The GAIN ADJUST
should be performed first. Although the trims are not interactive
within the device, the GAIN trim will move the BALANCE trim
point as it changes the magnitude of the span.
Figure 2b shows GAIN and BALANCE trims in a +5 V and
–5 V tracking configuration. A 100 kΩ 20-turn potentiometer is
Figure 2a. +10 V Output
used for each trim. The potentiometer for GAIN trim is connected between Pins 6 (V
) and 8 (V
HIGH
) with the wiper
LOW
connected to Pin 5 (GAIN ADJ). The potentiometer is adjusted
to produce exactly 10 V between Pins 1 and 15, the amplifier
outputs. The BALANCE potentiometer, also connected between Pins 6 and 8 with the wiper to Pin 12 (BAL ADJ), is then
adjusted to center the span from +5 V to –5 V.
Trimming in other configurations works in exactly the same
manner. When producing +10 V and +5 V, GAIN ADJ is used
to trim +10 V and BAL ADJ is used to trim +5 V. In the –10 V
and –5 V configuration, GAIN ADJ is again used to trim the
magnitude of the span, –10 V, while BAL ADJ is used to trim
the center tap, –5 V.
In single output configurations, GAIN ADJ is used to trim outputs utilizing the full span (+10 V or –10 V) while BAL ADJ is
used to trim outputs using half the span (+5 V or –5 V).
Input impedance on both the GAIN ADJUST and BALANCE
ADJUST pins is approximately 150 kΩ. The GAIN ADJUST
trim network effectively attenuates the 10 V across the trim
potentiometer by a factor of about 1500 to provide a trim range
of –3.5 mV to +7.5 mV with a resolution of approximately
550 µV/turn (20 turn potentiometer). The BALANCE ADJUST
trim network attenuates the trim voltage by a factor of about
Figure 2b. +5 V and –5 V Outputs
1400, providing a trim range of ±4.5 mV with resolution of
450 µV/turn.
REV. B
–4–
Figure 2c. –10 V Output
Trimming the AD588 introduces no additional errors over temperature so precision potentiometers are not required.
For single output voltage ranges, or in cases when BALANCE
ADJUST is not required, Pin 12 should be connected to Pin 11.
If GAIN ADJUST is not required, Pin 5 should be left floating.
AD588
Figure 4. Effect of 1 µF Noise Reduction Capacitor on
Broadband Noise
TURN-ON TIME
Upon application of power (cold start), the time required for the
output voltage to reach its final value within a specified error
band is the turn-on settling time. Two components normally associated with this are: time for active circuits to settle and time
for thermal gradients on the chip to stabilize. Figure 5 shows the
turn-on characteristics of the AD588. It shows the settling to be
about 600 µs. Note the absence of any thermal tails when the
horizontal scale is expanded to 2 ms/cm in Figure 5b.
NOISE PERFORMANCE AND REDUCTION
The noise generated by the AD588 is typically less than 6 µV p-p
over the 0.1 Hz to 10 Hz band. Noise in a 1 MHz bandwidth is
approximately 600 µV p-p. The dominant source of this noise is
the buried Zener which contributes approximately 100 nV/√
In comparison, the op amp’s contribution is negligible. Figure 3
shows the 0. 1 Hz to 10 Hz noise of a typical AD588.
Figure 3. 0.1 Hz to 10 Hz Noise
If further noise reduction is desired, an optional capacitor may
be added between the NOISE REDUCTION pin and ground
as shown in Figure 2b. This will form a low-pass filter with the
4 kΩ R
have a 3 dB point at 40 Hz and will reduce the high frequency
(to 1 MHz) noise to about 200 µV p-p. Figure 4 shows the
1 MHz noise of a typical AD588 both with and without a
1 µF capacitor.
Note that a second capacitor is needed in order to implement
the NOISE REDUCTION feature when using the AD588 in
the –10 V mode (Figure 2c.). The NOISE REDUCTION
capacitor is limited to 0.1 µF maximum in this mode.
on the output of the Zener cell. A 1 µF capacitor will
B
Hz.
a. Electrical Turn-On
b. Extended Time Scale
Figure 5. Turn-On Characteristics
Output turn-on time is modified when an external noise reduction capacitor is used. When present, this capacitor presents an
additional load to the internal Zener diode’s current source, resulting in a somewhat longer turn-on time. In the case of a 1 µF
capacitor, the initial turn-on time is approximately 60 ms (see
Figure 6).
Note: If the NOISE REDUCTION feature is used in the ±5 V
configuration, a 39 kΩ resistor between Pins 6 and 2 is required
for proper start up.
REV. B
–5–
Figure 6. Turn-On with 1 µF C
TEMPERATURE PERFORMANCE
N
The AD588 is designed for precision reference applications
where temperature performance is critical. Extensive temperature testing ensures that the device’s high level of performance is
maintained over the operating temperature range.
Figure 7 shows typical output voltage drift for the AD588BD
and illustrates the test methodology. The box in Figure 7 is
bounded on the sides by the operating temperature extremes
and on top and bottom by the maximum and minimum output
voltages measured over the operating temperature range. The
slope of the diagonal drawn from the lower left corner of the box
determines the performance grade of the device.
AD588
Figure 8. Maximum Output Change—mV
KELVIN CONNECTIONS
Force and sense connections, also referred to as Kelvin connections, offer a convenient method of eliminating the effects of
voltage drops in circuit wires. As seen in Figure 9a, the load current and wire resistance produce an error (V
the load. The Kelvin connection of Figure 9b overcomes the
problem by including the wire resistance within the forcing loop
of the amplifier and sensing the load voltage. The amplifier corrects for any errors in the load voltage. In the circuit shown, the
output of the amplifier would actually be at 10 volts + V
and the voltage at the load would be the desired 10 volts.
The AD588 has three amplifiers which can be used to implement Kelvin connections. Amplifier A2 is dedicated to the
ground force-sense function while uncommitted amplifiers A3
and A4 are free for other force-sense chores.
In some single-output applications, one amplifier may be unused.
ERROR
= R × IL) at
ERROR
Figure 7. Typical AD588BD Temperature Drift
Each AD588A and B grade unit is tested at –25°C, 0°C, +25°C,
+50°C, +70°C and +85°C. Each AD588S and T grade unit is
tested at –55°C, –25°C, 0°C, +25°C, +50°C, +70°C and
+125°C. This approach ensures that the variations of output
voltage that occur as the temperature changes within the specified range will be contained within a box whose diagonal has a
slope equal to the maximum specified drift. The position of the
box on the vertical scale will change from device to device as
initial error and the shape of the curve vary. Maximum height of
the box for the appropriate temperature range is shown in Figure 8. Duplication of these results requires a combination of
high accuracy and stable temperature control in a test system.
Evaluation of the AD588 will produce a curve similar to that in
Figure 7, but output readings may vary depending on the test
methods and equipment utilized.
REV. B
–6–
Open Loop Frequency Response (A3, A4)
Power Supply Rejection vs. Frequency (A3, A4)
AD588
67 4
3
2
1
138
9 10 11
125
14
15
16
A3
A4
A1
A2
R3
R
B
R1
R2
R4
R5
R6
AD588
GAIN
ADJ
GND
SENSE
+IN
GND
SENSE
–IN
V
LOW
BAL
ADJ
VCTA4 IN
–V
S
+V
S
A4 OUT
FORCE
A4 OUT
SENSE
A3 OUT
FORCE
A3 OUT
SENSE
A3 IN
V
HIGH
NOISE
REDUCTION
Unity Gain Follower Pulse Response (Large Signal)
Figure 9. Advantage of Kelvin Connection
In such cases, the unused amplifier should be connected as a
unity-gain follower (force + sense pin tied together) and the
input should be connected to ground.
An unused amplifier section may be used for other circuit functions as well. The curves on this page show the typical performance of A3 and A4.
Unity Gain Follower Pulse Response (Small Signal)
DYNAMIC PERFORMANCE
The output buffer amplifiers (A3 and A4) are designed to
provide the AD588 with static and dynamic load regulation
superior to less complete references.
Many A/D and D/A converters present transient current loads
to the reference, and poor reference response can degrade the
converter’s performance.
Figure 10 displays the characteristics of the AD588 output
amplifier driving a 0 mA to 10 mA load.
Figure 10a. Transient Load Test Circuit
REV. B
Common-Mode Rejection vs. Frequency (A3, A4)
Input Noise Voltage Spectral Density
Figure 10b. Large-Scale Transient Response
Figure 10c. Fine Scale Settling for Transient Load
–7–
AD588
Figure 11 displays the output amplifier characteristics driving a
5 mA to 10 mA load, a common situation found when the reference is shared among multiple converters or is used to provide a
bipolar offset current.
Figure 11a. Transient and Constant Load Test Circuit
Figure 13 displays the crosstalk between output amplifiers. The
top trace shows the output of A4, dc-coupled and offset by 10
volts, while the output of A3 is subjected to a 0 mA-to-10 mA
load current step. The transient at A4 settles in about 1 µs, and
the load-induced offset is about 100 µV.
Figure 13a. Load Crosstalk Test Circuit
Figure 11b. Transient Response 5 mA–10 mA Load
In some applications, a varying load may be both resistive and
capacitive in nature, or be connected to the AD588 by a long
capacitive cable.
Figure 12 displays the output amplifier characteristics driving a
1,000 pF, 0 mA-to-10 mA load.
Figure 12a. Capacitive Load Transient Response Test
Circuit
Figure 13b. Load Crosstalk
Attempts to drive a large capacitive load (in excess of 1,000 pF)
may result in ringing or oscillation, as shown in the step response
photo (Figure 14a). This is due to the additional pole formed by
the load capacitance and the output impedance of the amplifier,
which consumes phase margin. The recommended method of
driving capacitive loads of this magnitude is shown in Figure
14b. The 150 Ω resistor isolates the capacitive load from the
output stage, while the 10 kΩ resistor provides a dc feedback
path and preserves the output accuracy The 1 µF capacitor pro-
vides a high frequency feedback loop. The performance of this
circuit is shown in Figure 14c.
Figure 12b. Output Response with Capacitive Load
REV. B
Figure 14a. Output Amplifier Step Response, CL = 1 µF
–8–
Figure 14b. Compensation for Capacitive Loads
Figure 14c. Output Amplifier Step Response Using Figure
14b Compensation
AD588
USING THE AD588 WITH CONVERTERS
The AD588 is an ideal reference for a wide variety of A/D and
D/A converters. Several representative examples follow.
14-Bit Digital-to-Analog Converter—AD7535
High resolution CMOS D/A converters require a reference voltage of high precision to maintain rated accuracy. The combination of the AD588 and AD7535 takes advantage of the initial
accuracy, drift and full Kelvin output capability of the AD588 as
well as the resolution, monotonicity and accuracy of the AD7535
to produce a subsystem with outstanding characteristics.
16-Bit Digital-to-Analog Converter—AD569
Another application which fully utilizes the capabilities of the
AD588 is supplying a reference for the AD569, as shown in Figure 16. Amplifier A2 senses system common and forces V
assume this value, producing +5 V and –5 V at Pins 6 and 8
respectively. Amplifiers A3 and A4 buffer these voltages out to
the appropriate reference force-sense pins of the AD569. The
full Kelvin scheme eliminates the effect of the circuit traces or
wires and the wire bonds of the AD588 and AD569 themselves,
which would otherwise degrade system performance.
SUBSTITUTING FOR INTERNAL REFERENCES
Many converters include built-in references. Unfortunately,
such references are the major source of drift in these converters.
By using a more stable external reference like the AD588, drift
performance can be improved dramatically.
CT
to
REV. B
Figure 15. AD588/AD7535 Connections
–9–
AD588
Figure 16. High Accuracy ±5 V Tracking Reference for AD569
12-Bit Analog-to-Digital Converter—AD574A
The AD574A i s specified for gain drift from 10 ppm/°C to
50 ppm/°C, (depending on grade) using the on-chip reference.
The reference contributes typically 75% of this drift. Therefore,
the total drift using an AD588 to supply the reference can be
improved by a factor of 3 to 4.
Using this combination may result in apparent increases in fullscale error due to the difference between the on-board reference
by which the device is laser trimmed and the external reference
with which the device is actually applied. The on-board reference is specified to be 10 V ± 100 mV while the external reference is specified to be 10 V ± 1 mV. This may result in up to
101 mV of apparent full-scale error beyond the ± 25 mV specified AD574 gain error. Resistors R2 and R3 allow this error to
be nulled. Their contribution to full-scale drift is negligible.
The high output drive capability allows the AD588 to drive up
to 6 converters in a multi-converter system. All converters will
have gain errors that track to better than ± 5 ppm/°C.
REV. B
Figure 17. AD588/AD574A Connections
–10–
RTD EXCITATION
The Resistance Temperature Detector (RTD) is a circuit element whose resistance is characterized by a positive temperature
coefficient. A measurement of resistance indicates the measured
temperature. Unfortunately, the resistance of the wires leading
to the RTD often adds error to this measurement. The 4-wire
ohms measurement overcomes this problem. This method uses
two wires to bring an excitation current to the RTD and two
additional wires to tap off the resulting RTD voltage. If these
additional two wires go to a high input impedance measurement
circuit, the effect of their resistance is negligible. Therefore, they
transmit the true RTD voltage.
Figure 18. 4-Wire Ohms Measurement
A practical consideration when using the 4-wire ohms technique
with an RTD is the self-heating effect that the excitation current
has on the temperature of the RTD. The designer must choose
the smallest practical excitation current that still gives the desired resolution. RTD manufactures usually specify the
self-heating, effect of each of their models or types of RTDs.
Figure 19 shows an AD588 providing the precision excitation
current for a 100 Ω RTD. The small excitation current of 1 mA
dissipates a mere 0.1 mW of power in the RTD.
AD588
Figure 20. Boosted Precision Current Source
BRIDGE DRIVER CIRCUITS
The Wheatstone bridge is a common transducer. In its simplest
form, a bridge consists of 4 two terminal elements connected to
form a quadrilateral, a source of excitation connected along one
of the diagonals and a detector comprising the other diagonal.
Figure 21a shows a simple bridge driven from a unipolar excitation supply. E
viation of the element from the initial bridge values. Unfortunately,
this bridge output voltage is riding on a common-mode voltage
equal to approximately V
may necessarily be limited to high common-mode rejection
techniques such as instrumentation or isolation amplifiers.
Figure 21b shows the same bridge transducer, but this time it is
driven from pair of bipolar supplies. This configuration ideally
eliminates the common-mode voltage and relaxes the restrictions on any processing elements that follow.
, a differential voltage, is proportional to the de-
O
/2. Further processing of this signal
IN
Figure 19. Precision Current Source for RTD
BOOSTED PRECISION CURRENT SOURCE
In the RTD current-source application the load current is limited to ±10 mA by the output drive capability of amplifier A3.
In the event that more drive current is needed, a series pass
transistor can be inserted inside the feedback loop to provide
higher current. Accuracy and drift performance are unaffected
by the pass transistor.
REV. B
–11–
a. Unipolar Drive
b. Bipolar Drive
Figure 21. Bridge Transducer Excitation
Figure 22. Bipolar Bridge Drive
AD588
As shown in Figure 22, the AD588 is an excellent choice for the
control element in a bipolar bridge driver scheme. Transistors
Q1 and Q2 serve as series pass elements to boost the current
drive capability to the 28 mA required by a typical 350 Ω
bridge. A differential gain stage may still be required if the
bridge balance is not perfect. Such gain stages can be expensive.
Additional common-mode voltage reduction is realized by using
the circuit illustrated in Figure 23. A1, the ground sense amplifier, servo’s the supplies on the bridge to maintain a virtual
ground at one center tap. The voltage which appears on the opposite center tap is now single-ended (referred to ground) and
can be amplified by a less expensive circuit.
C1016b–10–10/86
REV. B
Figure 23. Floating Bipolar Bridge Drive with Minimum CMV
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
Cerdip (Q) Package
PRINTED IN U.S.A.
–12–
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