Datasheet AD1990 Datasheet (ANALOG DEVICES)

Page 1

Audio Switching Amplifier

FEATURES

Integrated stereo modulator and power stage <0.002% THD + N 101 dB dynamic range (A-weighted) 2 × 5 W output power (4 Ω, <0.01% THD + N)

< 0.3 Ω (per transistor)

DS-ON

PSRR > 65 dB On-off-mute pop noise suppression EMI optimized modulator Short-circuit protection Overtemperature protection Low cost DMOS process

APPLICATIONS

Advanced televisions Compact multimedia systems Minicomponents

PGA1 PGA0

GENERAL DESCRIPTION

The AD1990 is a 2-channel, bridge tied load (BTL), switching audio power amplifier with integrated Σ-Δ modulator. The modulator accepts a single-ended, analog input signal and converts it to a switching waveform to drive speakers directly. A digital, microprocessor-compatible interface provides control of reset, mute, and PGA gain, as well as feedback signals for thermal and overcurrent error conditions. The output stage can operate over a power supply voltages range of 8 V to 12 V. The analog modulator and digital logic operate from a 5 V supply.

FUNCTIONAL BLOCK DIAGRAM

NFL–

NFL+

FEEDBACK

NETWO RK

AD1990

AINL

AINR

CLKI

CLKO

REF_FILT

AD1990

OSCILLATOR

REFERENCE

VOLTAGE

AGND

PGA

MODE CONT ROL

LOGI C AND POP/CLICK

SUPPRESSION

ERR2

MUTE

ERR1

RESET

MODULAT OR

ERR0

Σ-Δ

LEVEL

SHIFTER

AND

DEAD TIME

CONTROL

Σ-Δ

NFR–

NFR+

DCTRL2

DCTRL1

DCTRL0

H-BRIDGE

PGND

Figure 1.

FEEDBACK

NETWO RK

OUTL+

OUTL–

OUTR+

OUTR–

05380-001

Rev. 0

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

Page 2

AD1990

Features .............................................................................................. 1

Overview ..................................................................................... 11

Applications....................................................................................... 1

General Description ......................................................................... 1

Functional Block Diagram .............................................................. 1

Revision History ............................................................................... 2

Specifications..................................................................................... 3

Absolute Maximum Ratings............................................................ 5

ESD Caution.................................................................................. 5

Pin Configuration and Function Descriptions............................. 6

Typical Performance Characteristics ............................................. 8

Theory of Operation ...................................................................... 11

REVISION HISTORY

4/06—Revision 0: Initial Version

Σ-Δ Modulator............................................................................ 11

and

RESET

..................................................................... 11

MUTE

Gain Structure............................................................................. 11

Power Stage ................................................................................. 13

Clocking....................................................................................... 13

Protection Circuits and Error Reporting ................................ 14

Application Circuits ....................................................................... 15

Outline Dimensions ....................................................................... 16

Ordering Guide .......................................................................... 16

Rev. 0 | Page 2 of 16

Page 3

AD1990

SPECIFICATIONS

Test conditions, unless otherwise specified.

Table 1.

Parameter Ratings

SUPPLY VOLTAGES

AMBIENT TEMPERATURE 25°C LOAD IMPEDANCE 6 Ω CLOCK FREQUENCY 12.288 MHz PGA GAIN 0 dB MEASUREMENT BANDWIDTH 20 Hz to 20 kHz

Table 2.

Parameter Min Typ Max Unit Test Conditions/Comments

DS-ON

Per High-Side Transistor 260 355 mΩ T = 25°C

Per Low-Side Transistor 210 265 mΩ T = 25°C MAXIMUM CURRENT THROUGH OUTx 5 A Peak THERMAL WARNING ACTIVE 135 °C Die temperature THERMAL SHUTDOWN ACTIVE 150 °C Die temperature RESTORE TEMPERATURE AFTER THERMAL SHUTDOWN 120 °C Die temperature

Table 3. Performance Specifications

Parameter Typ Unit Test Conditions/Comments

TOTAL HARMONIC DISTORTION AND NOISE (THD + N) 0.003 % PGA = 0 dB, PO = 1 W, 1 kHz

0.006 % PGA = 6 dB, PO = 1 W, 1 kHz

0.01 % PGA = 12 dB, PO = 1 W, 1 kHz

0.02 % PGA = 18 dB, PO = 1 W, 1 kHz SIGNAL-TO-NOISE RATIO (SNR) 102 dB 1 kHz, A-weighted, 0 dB referred to 1% THD + N output DYNAMIC RANGE (DNR) 102 dB 1 kHz, A-weighted, −60 dB referred to 1% THD + N output CROSSTALK (LEFT-TO-RIGHT OR RIGHT-TO-LEFT) −100 dB PGA = 0 dB, PO = 5 W, 1 kHz

Table 4. DC Specifications

Parameter Typ Unit Test Conditions/Comments

INPUT IMPEDANCE 20 kΩ AINL, AINR input pins OUTPUT DC OFFSET ±4 mV Independent of PGA setting

5 V 5 V 12 V

Rev. 0 | Page 3 of 16

Page 4

AD1990

Table 5. Power Supplies

Parameter Min Typ Max Unit Test Conditions/Comments

ANALOG SUPPLY, AV DIGITAL SUPPLY, DV POWER TRANSISTOR SUPPLY, PV

RESET/POWER-DOWN CURRENT

QUIESCENT CURRENT Inputs grounded, nonoverlap = minimum

OPERATING CURRENT VIN = 1 V rms, RL = 6 Ω, PO = 1 W

Table 6. Digital I/O

Parameter Min Typ Max Unit Test Conditions/Comments

INPUT LOGIC HIGH 2.0 V INPUT LOGIC LOW 0.8 V OUTPUT LOGIC HIGH 2.4 V @ 4 mA OUTPUT LOGIC LOW 0.4 V @ 4 mA LEAKAGE CURRENT ON DIGITAL OUTPUTS 10 μA

Table 7. Digital Timing

Parameter Typ Unit Test Conditions/Comments

10 μs 34 μs

4.5 5.0 5.5 V

6.5 8 to 12 15 V RESET held low

0.6 1 μA 5 V

7.5 11 μA 5 V

19 40 μA 12 V

20 mA 5 V

5.5 mA 5 V

30 mA 12 V

20 27 mA 5 V

5.5 7 mA 5 V

218 260 mA 12 V

Delay after Delay after

MUTE is asserted until output stops switching MUTE is deasserted until output starts switching

05380-002

MUTE

OUTx

Figure 2. Mute and Unmute Delay Timing

Rev. 0 | Page 4 of 16

Page 5

AD1990

ABSOLUTE MAXIMUM RATINGS

Table 8.

Parameter Rating

AVDD, DVDD to AGND, DGND −0.3 V to +6.5 V PVDDx to PGNDx AGND to DGND to PGNDx −0.3 V to +0.3 V AVDD, to DVDD −0.5 V to +0.5 V Operating Temperature Range –40°C to +85°C Storage Temperature Range –65°C to +150°C Maximum Junction Temperature 150°C Thermal Resistance

θJC (at the Exposed Pad Surface) 0.9°C/W θJB (on JEDEC Standard PCB) 9.7°C/W

Including any induced voltage due to inductive load.

−0.3 V to +22.5 V

19.2°C/W

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.

Rev. 0 | Page 5 of 16

Page 6

AD1990

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

AVDD

AGND

PGND1

NFL+

NFL–NCAINLNCMOD_FIL

646362616059585756555453525150

REF_FILTNCAINRNCNFR–

NFR+

PGND2

PGND1

OUTL+

PVDD1

OUTL–

PGND1

NC = NO CONNECT

PIN 1 INDICATO R

AD1990

TOP VIEW

(Not to Scale)

171819202122232425262728293031

ERR2

ERR1

ERR0

DCTRL2

DCTRL1

DCTRL0

CLKI

DVDD

CLKO

DGND

MUTE

DGND

RESET

PGND2

OUTR+

PVDD2

OUTR–

PGND2

PGA1

PGA0

05380-003

Figure 3. Pin Configuration

Table 9. Pin Function Descriptions

Pin No. Mnemonic In/Out Description

1, 2, 3, 64 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors. 4, 5, 6 OUTL+ O Output of Transistor Pair A1 and A2. 7, 8, 9, 10 PVDD1 Positive Power Supply. Used for the A1 and B1 high power transistors. 11, 12, 13 OUTL− O Output of Transistor Pair B1 and B2. 14, 15, 16 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors. 17 18 19

ERR2 ERR1 ERR0

O Active Low Thermal Shutdown Error Output. O Active Low Thermal Warning Error Output. O Active Low Overcurrent Error Output.

20 DCTRL2 I/O Nonoverlap Time Setting MSB. 21 DCTRL1 I Nonoverlap Time Setting. 22 DCTRL0 I Nonoverlap Time Setting LSB. 23, 26 DGND Negative Power Supply for Low Power Digital Circuitry. 24, 25 DVDD Positive Power Supply for Low Power Digital Circuitry. 27 CLKI I Clock Input for 256 × fS Audio Modulator Clock. 28 CLKO O Inverted Version of CLKI for Use with an External XTAL Oscillator. 29 30

MUTE RESET

I Active Low Mute Input. I Active Low Reset Input.

31 PGA1 I PGA Gain Control MSB. 32 PGA0 I PGA Gain Control LSB. 33, 34, 35 PGND2 Negative Power Supply for High Power Transistors C2 and D2. 36, 37, 38 OUTR− O Output of Transistor Pair D1 and D2. 39, 40, 41, 42 PVDD2 Positive Power Supply for High Power Transistors C1 and D1. 43, 44, 45 OUTR+ O Output of Transistor Pair C1 and C2. 46, 47, 48, 49 PGND2 Negative Power Supply for High Power Transistors C2 and D2. 50 NFR+ I Right Channel Negative Feedback—Noninverting Input. 51 NFR− I Right Channel Negative Feedback—Inverting Input.

Rev. 0 | Page 6 of 16

Page 7

AD1990

Pin No. Mnemonic In/Out Description

52, 54, 59, 61 NC No Connection—Should Be Left Floating. 53 AINR I Analog Input for Right Channel. 55 REF_FILT O Filter Pin for Band Gap Reference—Should Be Bypassed to AGND. 56 AGND Negative Power Supply for Low Power Analog Circuitry. 57 AVDD Positive Power Supply for Low Power Analog Circuitry. 58 MOD_FILT O Modulator Filter Pin—Used to Set Time Constant of Modulator Order Reduction Circuit. 60 AINL O Analog Input for Left Channel. 62 NFL− I Left Channel Negative Feedback—Inverting Input. 63 NFL+ I Left Channel Negative Feedback—Noninverting Input. 64 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors.

Rev. 0 | Page 7 of 16

Page 8

AD1990

TYPICAL PERFORMANCE CHARACTERISTICS

–20

–40

–60

–80

–100

THD = 1% (5.9W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

–160

2 4 6 8 1012141618

FREQUENCY (kHz)

05380-004

Figure 4. 1 W Output Power into 4 Ω Load

–20

–40

–60

–80

–100

THD = 1% (5.9W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

–160

020

2 4 6 8 10 12 14 16 18

Figure 7. −60 dBFS Output Power into 4 Ω Load

–20

–40

–60

–80

–100

THD = 1% (4.0W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

–20

–40

–60

–80

–100

THD = 1% (4.0W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

FREQUENCY (kHz)

05380-007

–160

2 4 6 8 10 12 14 16 18

FREQUENCY (kHz)

05380-005

Figure 5. 1 W Output Power into 6 Ω Load

–160

020

2 4 6 8 10 12 14 16 18

Figure 8. −60 dBFS Output Power into 6 Ω Load

–20

–40

–60

–80

–100

THD = 1% (3.0W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

–160

2 4 6 8 10 12 14 16 18

FREQUENCY (kHz)

05380-006

Figure 6. 1 W Output Power into 8 Ω Load

–20

–40

–60

–80

–100

THD = 1% (3.0W))

–120

POWER (d BFS: 0dB = Power at Wh ich

–140

–160

020

2 4 6 8 10 12 14 16 18

Figure 9. −60 dBFS Output Power into 8 Ω Load

Rev. 0 | Page 8 of 16

FREQUENCY (kHz)

05380-008

05380-009

Page 9

AD1990

–

–20

–40

–60

–80

–100

–120

POWER (d B, Relati ve to 500mW O utpu t Power)

–140

100 1k 10k

FREQUENCY (Hz)

Figure 10. IMD for 19 kHz/20 kHz Twin-Tone Stimulus with

500 mW Power in Each Tone

AMPLIFIER GAIN (dB)

PGA GAIN = 18dB

PGA GAIN = 12dB

PGA GAIN = 6dB

PGA GAIN = 0dB

100 1k 10k

FREQUENCY (Hz)

Figure 11. Amplifier Gain vs. Frequency, 6 Ω Load, PVDD = 12 V

0.1

0.01

THD (%)

0.001

0.0001

05380-010

100 1k 10k

FREQUENCY (Hz)

–50

–60

–70

–80

–90

–100

–110

–120

THD (dB, Rel ative to F undamen tal)

05380-013

Figure 13. THD vs. Frequency, 1 W Output Power into 4 Ω Load, PVDD = 12 V

0.1

0.01

THD (%)

0.001

0.0001

05380-011

100 1k 10k

FREQUENCY (Hz)

–50

–60

–70

–80

–90

–100

–110

–120

THD (dB, Rel ative to F undamen tal)

05380-014

Figure 14. THD vs. Frequency, 1 W Output Power into 6 Ω Load, PVDD = 12 V

–20

–40

–60

–80

Driven Chan nel Sig nal)

–100

SIGNAL IN I DLE CHANNEL (d B, Relative to

–120

100 1k 10k

L CHANNEL IDLE, R CHANNEL DRIVEN

FREQUENCY (Hz)

L CHANNEL DRIVEN, R CHANNEL IDLE

Figure 12. Channel Separation vs. Frequency, Driven Channel Has

1 W Output Power into 6 Ω Load

0.1

0.01

THD (%)

0.001

0.0001

05380-012

100 1k 10k

FREQUENCY (Hz)

Figure 15. THD vs. Frequency, 1 W Output Power into 8 Ω Load, PVDD = 12 V

–50

–60

–70

–80

–90

–100

–110

–120

THD (dB, Rel ative to F undamen tal)

05380-015

Rev. 0 | Page 9 of 16

Page 10

AD1990

100

0.1

THD OR THD + N (%)

0.01

0.001

THD + N

THD

0.1 1 10

OUTPUT PO WER (W)

Figure 16. THD and THD + N vs. Output Power, 1 kHz Sine, 4 Ω Load, PVDD = 12 V

100

0.1

THD OR THD + N (%)

0.01

0.001

THD + N

THD

0.1 1 10

OUTPUT PO WER (W)

Figu re 17. THD and THD + N vs. Output Power, 1 kHz Sine, 6 Ω Load, PVDD = 12 V

100

0.1

THD OR THD + N (%)

0.01

0.001

THD + N

THD

0.1 1 10

OUTPUT PO WER (W)

Figu re 18. THD and THD + N vs. Output Power, 1 kHz Sine, 8 Ω Load, PVDD = 12 V

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

–10

–20

–30

–40

–50

–60

–70

–80

–90

–100

OUTPUT PO WER PER CHANNEL (W)

THD OR THD + N (dB, Relati ve to Fun damental)

05380-016

8.0 12.0

8.5 9.0 9.5 10.0 10.5 11. 0 11.5

THD = 10%

THD = 1%

PVDD VOLTAGE (V)

05380-019

Figure 19. Maximum Output Power vs. PVDD, 4 Ω Load

OUTPUT PO WER PER CHANNEL (W)

THD OR THD + N (dB, Relati ve to Fun damental)

05380-017

8.0 12.0

8.5 9.0 9.5 10.0 10.5 11. 0 11.5

THD = 10%

THD = 1%

PVDD VOLTAGE (V)

05380-020

Figure 20. Maximum Output Power vs. PVDD, 6 Ω Load

OUTPUT PO WER PER CHANNEL (W)

THD OR THD + N (dB, Relati ve to Fun damental)

05380-018

8.0 12.0

8.5 9.0 9.5 10.0 10.5 11. 0 11.5

THD = 10%

PVDD VOLTAGE (V)

THD = 1%

05380-021

Figure 21. Maximum Output Power vs. PVDD, 8 Ω Load

Rev. 0 | Page 10 of 16

Page 11

AD1990

THEORY OF OPERATION

OVERVIEW

The AD1990 is a 2-channel, high performance, switching, audio power amplifier. Each of the two Σ-Δ modulators converts a single-ended analog input into a 2-level pulse stream that controls the differential, full H-bridge, power output stage. The combination of an Σ-Δ modulator and a switching power stage provides an inherently linear and efficient means of amplifying the entire range of audio frequencies. The AD1990 also offers warning and protection circuits for overcurrent and overtemperature conditions, as well as silent turn-on and turn-off transitions.

Σ-Δ MODULATOR

The AD1990 is a switching type, also known as a Class-D, audio power amplifier. This class of amplifiers maximizes efficiency by only using its power output devices in full-on or full-off states. While most Class-D amplifiers use some variation of pulse-width modulation (PWM), the AD1990 uses Σ-Δ modulation to determine the switching pattern of the output devices. This provides a number of important benefits. Σ-Δ modulators do not produce a sharp peak with many harmonics in the AM frequency band as pulse-width modulators (PWM) often do. In addition, the 1-bit quantizer produces excellent linearity across the full amplitude range.

Σ-Δ modulators require feedback to generate an error signal with respect to the input. The feedback voltages for the AD1990 modulators come from the outputs of the power devices and before the passive low-pass filters (see for nonlinear behavior in the power stage, such as nonoverlap time, mismatched rise and fall times, and propagation delays. It also reduces sensitivity to both dc and transient changes of the power supply voltage.

Σ-Δ modulators operate in discrete time. As with all timequantized systems, the Nyquist frequency is equal to half of the sampling frequency and input signals above that point aliases back into the base band. The AD1990 sampling frequency (master clock) is equal to half the frequency of the input clock, approximately 6 MHz, so images only alias for input frequencies above approximately 3 MHz. This is far enough above the audio band that bandwidth and aliasing are not a problem in real applications.

The modulator has a noise shaping effect, and SNR is increased in the audio band by shifting the quantization noise upward in frequency. For a nominal input clock frequency of 12.288 MHz, the noise floor rises sharply above 20 kHz. The actual clock frequency used in an application circuit can deviate from this rate by as much as ±10%, and the corner frequency of the noise scales proportionately. The frequency at which the quantization noise dominates the output determines the amplifier’s practical bandwidth.

Figure 23). This compensates

MUTE AND RESET

When power is applied and the the AD1990 is in its lowest power consumption mode. The analog modulator is not running, and the power stage is tristated. On deasserting the start-up sequence that includes initialization of the modulator, the protection circuits, and other functions.

Once the start-up sequence is complete, the amplifier is in a state in which the modulator is running, but the output stage is not driven. When using a soft-start sequence that avoids any audible pop or click noise in the output signal.

The output power transistors do not switch while remains asserted. Unlike the analog mute circuits found on some amplifiers that can be limited in their attenuation by the control logic or crosstalk, the mute attenuation on the AD1990 is greater than its dynamic range. The noise floor of the output signal also drops while in are not switching.

MUTE

Power-Up Sequencing

Careful power-up is necessary when using the AD1990 to ensure correct operation and to avoid possible latch-up issues. The AD1990 should be powered up with held low until all the power supplies have stabilized. Once the supplies have stabilized, bring the AD1990 out of bringing

Begin the soft unmute sequence by bringing least 1 sec after the audio using a shorter start-up sequence (as shown in but the amplifier can produce an audible pop or click noise as the output starts switching. This is because the ac coupling capacitors at the analog input have a long time constant. If MUTE RESET state. They need ample time to settle at a bias voltage of V the reference voltage for the single-ended inputs, or the

amplifier starts with a slight dc offset.

RESET

high.

RESET

is deasserted substantially less than 1 sec after deasserting

, then these capacitors may not have charged to a steady

RESET

pin remains asserted,

RESET

pin, the modulator begins a

is deasserted, the output is started

MUTE

because the output transistors

RESET

MUTE

rising edge. The amplifier produces

and

RESET

high at

MUTE

Tabl e 7),

REF

GAIN STRUCTURE

Analog Input Levels

The AD1990 has single-ended inputs for the left and right channels. The analog input section uses an internal amplifier to bias the input signal to the reference level, V equal to AV prevents this bias voltage from affecting the signal source. In combination with the nominal 20 kΩ input impedance, the value of this capacitor should be large enough to produce a flat frequency response at the lowest input frequency of interest.

/2. A dc-blocking capacitor, as shown in Figure 22,

, which is nominally

REF

Rev. 0 | Page 11 of 16

Page 12

AD1990

Note that the amplifier is capable of dc-coupled operation if the circuit includes some means to account for this bias voltage.

AINL/ AINR

This fixed total resistance to ground eliminates the last free variable and gives the following equations for the resistors:

21810

R4R2

R1 = R3 = 6000 − R2

05380-028

Figure 22. AC-Coupled Input Signal

Setting the Modulator Gain

The AD1990 modulator uses a combination of the input signal and feedback from the power output stage to calculate its twostate output pattern. The feedback input nodes are part of the internal analog circuit that operates from the AV

(nominal

5 V) power supply. Because the voltage measured at the power outputs is nominally between 0 V and PV the 0 V to AV

range, a voltage divider is required to scale the

, and thus beyond

feedback to an appropriate level.

Resistor voltage dividers should sense the voltage on each side of the differential output and provide these feedback signals to the modulator, as shown in

OUTx+ OUTx–

PGND PGND

NFx+ NFx–

D1 D3

D2 D4

R1 R3

R2 R4

Figure 23. H-Bridge Configuration

Figure 23.

EXTERNAL COMPONENTS

LLR

The resistor values should satisfy the following equation to maintain modulator stability.

Gain =

635.34

Selecting a gain that meets this criterion ensures that the modulator remains in a stable operating condition.

The ratio of the resistances sets the gain rather than the absolute values. However, the dividers provide a path from the high voltage supply to ground; therefore, the values should be large enough to produce negligible loss due to quiescent current.

Note that the gain previously mentioned applies to each side of the differential output pair. Therefore, the total forward gain for the modulator and output stage is twice that value. Recommended resistor values for some common supply voltages are shown in

Tabl e 1 0 .

Table 10. Recommended Feedback Resistor Values

PVDD (V) R1 (kΩ) R2 (kΩ)

Voltage Divider Gain

Differential System Gain

8 3.27 2.73 2.2 4.4 (13.8 dB) 10 3.82 2.18 2.8 5.6 (17.6 dB) 12 4.18 1.82 3.3 6.6 (20.8 dB)

Programmable Gain Amplifier (PGA)

The Σ-Δ modulator itself requires a fixed gain for a given value of PV

to maintain optimal stability. This gain can be appropriate,

but many applications require more gain to account for low source signal levels. The AD1990 includes a programmable gain amplifier (PGA) to boost the overall amplifier gain. The total gain for the amplifier is the product of the modulator gain and the PGA gain. PGA1 (Pin 31) and PGA0 (Pin 32) select one of four PGA gain values, as shown in

Tabl e 11 .

Table 11. PGA Gain Settings

05380-029

PGA1 PGA0 PGA Gain

0 0 1 (0 dB) 0 1 2 (6 dB) 1 0 4 (12 dB) 1 1 8 (18 dB)

The AD1990 incorporates a single-ended-to-differential converter for each channel in the analog front-end section. The PGA is also part of this analog front-end, and it affects the analog input signal before it enters the Σ-Δ modulator. The PGA1 and PGA0 pins are continuously monitored and allow the gain to be changed at any time.

The chip contains a calibration circuit to minimize voltage offsets at the speaker, which helps to minimize clicks and pops when muting or unmuting. Optimal performance is achieved for the offset calibration circuit when the feedback divider resistors sum to 6 kΩ, that is, (R1 + R2) = 6 kΩ, and (R3 + R4) = 6 kΩ.

Rev. 0 | Page 12 of 16

Page 13

AD1990

POWER STAGE

The H-Bridge

The output stage of the AD1990 includes four integrated MOSFET devices arranged in a full H-bridge, as shown in Figure 23. The P-Type, high-side transistor of one leg and the N-Type, low-side transistor of the opposite leg switch on and off as a pair producing a total voltage swing across the load of

to +PVDD. The drive is floating and differential, and it is

−PV

important that neither output terminal be shorted to ground.

The power supply for the output stage of the AD1990, PV should be in the 8 V to 20 V range and should be capable of supplying enough current to drive the load. Connect the power supply across the PVDD and PGND pins. The feedback pins, NFR+, NFR−, NFL+, and NFL−, supply negative feedback to the modulator as described in the

Setting the Modulator Gain section.

Table 12. Nonoverlap Time Settings

DCTRL2 DCTRL1 DCTRL0 Nonoverlap Time (ns)

0 0 0 62 0 0 1 49 0 1 0 37 0 1 1 24 1 0 0 15 1 0 1 13.5 1 1 0 12 1 1 1 9

Values are typical and are not production tested.

HIGH-SIDE

GATE DRIVE

LOW-SIDE

GATE DRIVE

For reactive loads, the impedance can only be below the recommended threshold over a small portion of the amplifier’s bandwidth. In these cases, the amplifier can enter overcurrent shutdown in response to even small input signals in those frequency bands. When designing a system, use the minimum load impedance over the entire range of amplified frequencies when calculating current output rather than the average or nominal load impedance ratings often cited by loudspeaker driver manufacturers.

Output Transistor Nonoverlap Time

The AD1990 allows the user to select from one of eight different nonoverlap times, as shown in

Figure 24. Nonoverlap time prevents or minimizes the period during which both the highside and low-side devices are on simultaneously due to propagation delays and nonzero rise and fall times. If both the upper and lower portions of a half-bridge conduct simultaneously, there is a path directly from the power supply to ground and an induced current flow known as shoot-through. However, introducing this delay increases distortion by pushing the switching pattern further from an ideal two-state waveform. Selecting the nonoverlap delay requires a compromise between distortion and efficiency. The logic levels on the three delay control pins, DCTRL2, DCTRL1, and DCTRL0, set the nonoverlap time according to rising edge of

Tabl e 1 2 . The state of DCTRL[2:0] is read on the

RESET

and should not be changed while

RESET

is logic high.

NOL

Figure 24. Half-Bridge Nonoverlap Delay Timing

NOL

05380-030

The shortest setting (DCTRL[2:0] = 111) or the second shortest setting (DCTRL[2:0] = 111) is recommended for most applications. These two settings allow a small trade-off between efficiency and distortion. Longer nonoverlap times generally increase distortion while providing little or no decrease in shootthrough current.

CLOCKING

The AD1990 Σ-Δ modulator requires an external clock source with a nominal frequency of 12.288 MHz. This clock can come from a crystal or from an existing clock signal in the application circuit. The discrete time portions of the modulator run internally at 6.144 MHz, corresponding to 128 × f

As mentioned in the

Σ-Δ Modulator section, the modulator has a noise-shaping effect such that SNR is increased within the audio band by shifting modulator quantization noise upward in frequency. For an external clock frequency of 12.288 MHz, the modulator’s noise-shaping works in a manner that results in a flat noise floor at the amplifier output for frequencies 20 kHz and below. Above 20 kHz, the amplifier noise rises due to the spectral shaping of the modulator quantization noise. At very high frequencies, the noise floor levels off and decreases due to poles in the modulator noise-transfer function and in the external LC filter.

The clock frequency does not have to be exactly equal to

12.288 kHz and can vary by up to ±10%. For other rates, the noise corner scales linearly with frequency. When the modulator runs at a rate lower than nominal, the average power stage switching frequency decreases, the efficiency increases slightly, and the noise floor begins to rise at a slightly lower frequency. Likewise, a faster clock gives slightly increased bandwidth and slightly lower efficiency.

, where fS = 48 kHz.

Rev. 0 | Page 13 of 16

Page 14

AD1990

Using a Crystal Oscillator Clocking Multiple Amplifiers in Parallel

The AD1990 can use a crystal connected to the CLKI and CLKO pins as a master clock source, as shown in

Figure 25. The CLKI and CLKO pins connect to an internal inverter to create a full resonator. The typical values shown work in many applications, but the crystal manufacturer should provide the exact type and value of the capacitors and the resistor.

22pF 22pFXTAL

47Ω

CLKI

Figure 25. Crystal Connection

CLKO

05380-031

Using an External Clock Source

If a clock signal of the appropriate frequency already exists in the application circuit, connect it directly to CLKI and leave CLKO floating. The logic levels of the square wave should be compatible with those defined in

Specifications section.

Large amounts of jitter on the clock input degrade performance. Whenever possible, avoid passing the clock signal through programmable logic and other circuits with unknown or variable propagation delay. In general, clock signals suitable for audio ADCs or DACs are also appropriate for use with the AD1990.

If there are multiple AD199x family amplifiers connected to the same PV

supply, use the same clock source (or synchronous

derivatives) for each amplifier as previously described. Avoid clocking amplifiers from similar but asynchronous clocks if they use the same power supply because this can result in beat frequencies.

PROTECTION CIRCUITS AND ERROR REPORTING

Thermal Protection

The AD1990 features thermal protection. When the die temperature exceeds approximately 135°C, the thermal warning

ERR1

error output ( approximately 150°C, the thermal shutdown error output

ERR2

) is asserted. If this occurs, the part shuts down to

( prevent damage to the part. When the die temperature drops below approximately 120°C, the part returns to normal operation automatically and negates both error outputs.

Overcurrent Protection

The AD1990 features over current or short-circuit protection. If the current through any power transistors exceeds approximately 4 A, the part enters a mute state and the overcurrent error output (

ERR0 clear automatically. Restore normal operation and clear the error condition by either asserting and then negating by asserting and then negating

) is asserted. If the die temperature exceeds

) is asserted. This is a latched error and does not

RESET

MUTE

Rev. 0 | Page 14 of 16

Page 15

AD1990

APPLICATION CIRCUITS

0.1µF

10µF

4.7µF 0.1µF

47µF

AINL

AINR

REF_FILT

47µF

AVDD

DVDD

AD1990

1000µF

PVDD1

OUTL+

NFL+

NFL–

OUTL–

PVDD2

1000µF

0.1µF

0.1µF0.1µF

R1 = 4.2kΩ R2 = 1.8kΩ L = 18µH C = 1µF LOAD = 6Ω

05380-032

DIGITAL

INPUTS

HERMAL SHUTDOW N

THERMAL WARNI NG

OVERCURRENT

PGA0

PGA1

DCTRL2

DCTRL1

DCTRL0

MUTE

RESET

ERR2

ERR1

ERR0

CLKI

CLKO

AGND

DGND

OUTR+

NFR+

NFR–

OUTR–

PGND1

PGND2

Figure 26. Typical Application Circuit

Rev. 0 | Page 15 of 16

Page 16

AD1990

OUTLINE DIMENSIONS

7.50 REF

0.30

0.25

0.18 PIN 1

INDICATOR

7.25

7.10 SQ

6.95

0.25 MIN

1.00

0.85

0.80

SEATING

PLANE

12° MAX

9.00

BSC SQ

PIN 1 INDICATOR

VIEW

TOP

0.80 MAX

0.65 TYP

0.50 BSC

8.75

BSC SQ

0.20 REF

0.60 MAX

0.45

0.40

0.35

0.05 MAX

0.02 NOM

0.60 MAX

EXPOSED PAD

(BOTTOM VIEW)

COMPLIANT TO JEDEC S TANDARDS MO-220- VMMD-4

122105-0

Figure 27. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]

9 mm × 9 mm Body, Very Thin Quad

(CP-64-3)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD1990ACPZ AD1990ACPZRL AD1990ACPZRL71−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ), 7” Tape and Reel CP-64-3 EVAL-AD1990EB Evaluation Board

Z = Pb-free part.

−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) CP-64-3

−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ), 13” Tape and Reel CP-64-3

Rev. 0 | Page 16 of 16

Datasheet AD1990 Datasheet (ANALOG DEVICES)

FEATURES

APPLICATIONS

GENERAL DESCRIPTION

FUNCTIONAL BLOCK DIAGRAM

TABLE OF CONTENTS

REVISION HISTORY

SPECIFICATIONS

ABSOLUTE MAXIMUM RATINGS

ESD CAUTION

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

TYPICAL PERFORMANCE CHARACTERISTICS

THEORY OF OPERATION

OVERVIEW

Σ-Δ MODULATOR

MUTE AND RESET

Power-Up Sequencing

GAIN STRUCTURE

Analog Input Levels

Setting the Modulator Gain

Programmable Gain Amplifier (PGA)

POWER STAGE

The H-Bridge

Output Transistor Nonoverlap Time

CLOCKING

Using a Crystal Oscillator Clocking Multiple Amplifiers in Parallel

Using an External Clock Source

PROTECTION CIRCUITS AND ERROR REPORTING

Thermal Protection

Overcurrent Protection

APPLICATION CIRCUITS

OUTLINE DIMENSIONS

ORDERING GUIDE