Datasheet AD1990 Datasheet (ANALOG DEVICES)

Page 1
Audio Switching Amplifier

FEATURES

Integrated stereo modulator and power stage <0.002% THD + N 101 dB dynamic range (A-weighted) 2 × 5 W output power (4 Ω, <0.01% THD + N)
< 0.3 Ω (per transistor)
R
DS-ON
PSRR > 65 dB On-off-mute pop noise suppression EMI optimized modulator Short-circuit protection Overtemperature protection Low cost DMOS process

APPLICATIONS

Advanced televisions Compact multimedia systems Minicomponents
PGA1 PGA0
AV

GENERAL DESCRIPTION

The AD1990 is a 2-channel, bridge tied load (BTL), switching audio power amplifier with integrated Σ-Δ modulator. The modulator accepts a single-ended, analog input signal and converts it to a switching waveform to drive speakers directly. A digital, microprocessor-compatible interface provides control of reset, mute, and PGA gain, as well as feedback signals for thermal and overcurrent error conditions. The output stage can operate over a power supply voltages range of 8 V to 12 V. The analog modulator and digital logic operate from a 5 V supply.

FUNCTIONAL BLOCK DIAGRAM

NFL–
DD
NFL+
DV
DD
FEEDBACK
NETWO RK
PV
DD
AD1990
AINL
AINR
CLKI
CLKO
REF_FILT
AD1990
OSCILLATOR
REFERENCE
VOLTAGE
AGND
PGA
PGA
MODE CONT ROL
LOGI C AND POP/CLICK
SUPPRESSION
ERR2
MUTE
ERR1
RESET
MODULAT OR
MODULAT OR
ERR0
Σ-Δ
LEVEL
SHIFTER
AND
DEAD TIME
CONTROL
Σ-Δ
NFR–
NFR+
DCTRL2
DCTRL1
DCTRL0
A1
A2
B1
B2
H-BRIDGE
C1
C2
D1
D2
PGND
Figure 1.
FEEDBACK
NETWO RK
OUTL+
OUTL–
OUTR+
OUTR–
05380-001
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2006 Analog Devices, Inc. All rights reserved.
Page 2
AD1990

TABLE OF CONTENTS

Features .............................................................................................. 1
Overview ..................................................................................... 11
Applications....................................................................................... 1
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Revision History ............................................................................... 2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Pin Configuration and Function Descriptions............................. 6
Typical Performance Characteristics ............................................. 8
Theory of Operation ...................................................................... 11

REVISION HISTORY

4/06—Revision 0: Initial Version
Σ-Δ Modulator............................................................................ 11
and
RESET
..................................................................... 11
MUTE
Gain Structure............................................................................. 11
Power Stage ................................................................................. 13
Clocking....................................................................................... 13
Protection Circuits and Error Reporting ................................ 14
Application Circuits ....................................................................... 15
Outline Dimensions ....................................................................... 16
Ordering Guide .......................................................................... 16
Rev. 0 | Page 2 of 16
Page 3
AD1990

SPECIFICATIONS

Test conditions, unless otherwise specified.
Table 1.
Parameter Ratings
SUPPLY VOLTAGES
AV
DD
DV
DD
PV
DD
AMBIENT TEMPERATURE 25°C LOAD IMPEDANCE 6 Ω CLOCK FREQUENCY 12.288 MHz PGA GAIN 0 dB MEASUREMENT BANDWIDTH 20 Hz to 20 kHz
Table 2.
Parameter Min Typ Max Unit Test Conditions/Comments
R
DS-ON
Per High-Side Transistor 260 355 T = 25°C
Per Low-Side Transistor 210 265 T = 25°C MAXIMUM CURRENT THROUGH OUTx 5 A Peak THERMAL WARNING ACTIVE 135 °C Die temperature THERMAL SHUTDOWN ACTIVE 150 °C Die temperature RESTORE TEMPERATURE AFTER THERMAL SHUTDOWN 120 °C Die temperature
Table 3. Performance Specifications
Parameter Typ Unit Test Conditions/Comments
TOTAL HARMONIC DISTORTION AND NOISE (THD + N) 0.003 % PGA = 0 dB, PO = 1 W, 1 kHz
0.006 % PGA = 6 dB, PO = 1 W, 1 kHz
0.01 % PGA = 12 dB, PO = 1 W, 1 kHz
0.02 % PGA = 18 dB, PO = 1 W, 1 kHz SIGNAL-TO-NOISE RATIO (SNR) 102 dB 1 kHz, A-weighted, 0 dB referred to 1% THD + N output DYNAMIC RANGE (DNR) 102 dB 1 kHz, A-weighted, −60 dB referred to 1% THD + N output CROSSTALK (LEFT-TO-RIGHT OR RIGHT-TO-LEFT) −100 dB PGA = 0 dB, PO = 5 W, 1 kHz
Table 4. DC Specifications
Parameter Typ Unit Test Conditions/Comments
INPUT IMPEDANCE 20 AINL, AINR input pins OUTPUT DC OFFSET ±4 mV Independent of PGA setting
5 V 5 V 12 V
Rev. 0 | Page 3 of 16
Page 4
AD1990
Table 5. Power Supplies
Parameter Min Typ Max Unit Test Conditions/Comments
ANALOG SUPPLY, AV DIGITAL SUPPLY, DV POWER TRANSISTOR SUPPLY, PV
DD
DD
DD
RESET/POWER-DOWN CURRENT
AV
DD
DV
DD
PV
DD
QUIESCENT CURRENT Inputs grounded, nonoverlap = minimum
AV
DD
DV
DD
PV
DD
OPERATING CURRENT VIN = 1 V rms, RL = 6 Ω, PO = 1 W
AV
DD
DV
DD
PV
DD
Table 6. Digital I/O
Parameter Min Typ Max Unit Test Conditions/Comments
INPUT LOGIC HIGH 2.0 V INPUT LOGIC LOW 0.8 V OUTPUT LOGIC HIGH 2.4 V @ 4 mA OUTPUT LOGIC LOW 0.4 V @ 4 mA LEAKAGE CURRENT ON DIGITAL OUTPUTS 10 μA
Table 7. Digital Timing
Parameter Typ Unit Test Conditions/Comments
t
MD
t
UD
10 μs 34 μs
4.5 5.0 5.5 V
4.5 5.0 5.5 V
6.5 8 to 12 15 V RESET held low
0.6 1 μA 5 V
7.5 11 μA 5 V
19 40 μA 12 V
20 mA 5 V
5.5 mA 5 V
30 mA 12 V
20 27 mA 5 V
5.5 7 mA 5 V
218 260 mA 12 V
Delay after Delay after
MUTE is asserted until output stops switching MUTE is deasserted until output starts switching
t
UD
05380-002
MUTE
OUTx
t
MD
Figure 2. Mute and Unmute Delay Timing
Rev. 0 | Page 4 of 16
Page 5
AD1990

ABSOLUTE MAXIMUM RATINGS

Table 8.
Parameter Rating
AVDD, DVDD to AGND, DGND −0.3 V to +6.5 V PVDDx to PGNDx AGND to DGND to PGNDx −0.3 V to +0.3 V AVDD, to DVDD −0.5 V to +0.5 V Operating Temperature Range –40°C to +85°C Storage Temperature Range –65°C to +150°C Maximum Junction Temperature 150°C Thermal Resistance
θ
JA
θJC (at the Exposed Pad Surface) 0.9°C/W θJB (on JEDEC Standard PCB) 9.7°C/W
1
Including any induced voltage due to inductive load.
1
−0.3 V to +22.5 V
19.2°C/W
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

ESD CAUTION

ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. 0 | Page 5 of 16
Page 6
AD1990
T

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

AVDD
AGND
PGND1
NFL+
NFL–NCAINLNCMOD_FIL
646362616059585756555453525150
REF_FILTNCAINRNCNFR–
NFR+
PGND2
49
PGND1
1
PGND1
2
PGND1
3
OUTL+
4
OUTL+
5
OUTL+
6
PVDD1
7
PVDD1
8
PVDD1
9
PVDD1
10
OUTL–
11
OUTL–
12
OUTL–
13
PGND1
14
PGND1
15
PGND1
16
NC = NO CONNECT
PIN 1 INDICATO R
AD1990
TOP VIEW
(Not to Scale)
171819202122232425262728293031
ERR2
ERR1
ERR0
DCTRL2
DCTRL1
DCTRL0
CLKI
DVDD
DVDD
CLKO
DGND
MUTE
DGND
RESET
48
PGND2
47
PGND2
46
PGND2
45
OUTR+
44
OUTR+
43
OUTR+
42
PVDD2
41
PVDD2
40
PVDD2
39
PVDD2
38
OUTR–
37
OUTR–
36
OUTR–
35
PGND2
34
PGND2
33
PGND2
32
PGA1
PGA0
05380-003
Figure 3. Pin Configuration
Table 9. Pin Function Descriptions
Pin No. Mnemonic In/Out Description
1, 2, 3, 64 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors. 4, 5, 6 OUTL+ O Output of Transistor Pair A1 and A2. 7, 8, 9, 10 PVDD1 Positive Power Supply. Used for the A1 and B1 high power transistors. 11, 12, 13 OUTL− O Output of Transistor Pair B1 and B2. 14, 15, 16 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors. 17 18 19
ERR2 ERR1 ERR0
O Active Low Thermal Shutdown Error Output. O Active Low Thermal Warning Error Output. O Active Low Overcurrent Error Output.
20 DCTRL2 I/O Nonoverlap Time Setting MSB. 21 DCTRL1 I Nonoverlap Time Setting. 22 DCTRL0 I Nonoverlap Time Setting LSB. 23, 26 DGND Negative Power Supply for Low Power Digital Circuitry. 24, 25 DVDD Positive Power Supply for Low Power Digital Circuitry. 27 CLKI I Clock Input for 256 × fS Audio Modulator Clock. 28 CLKO O Inverted Version of CLKI for Use with an External XTAL Oscillator. 29 30
MUTE RESET
I Active Low Mute Input. I Active Low Reset Input.
31 PGA1 I PGA Gain Control MSB. 32 PGA0 I PGA Gain Control LSB. 33, 34, 35 PGND2 Negative Power Supply for High Power Transistors C2 and D2. 36, 37, 38 OUTR− O Output of Transistor Pair D1 and D2. 39, 40, 41, 42 PVDD2 Positive Power Supply for High Power Transistors C1 and D1. 43, 44, 45 OUTR+ O Output of Transistor Pair C1 and C2. 46, 47, 48, 49 PGND2 Negative Power Supply for High Power Transistors C2 and D2. 50 NFR+ I Right Channel Negative Feedback—Noninverting Input. 51 NFR− I Right Channel Negative Feedback—Inverting Input.
Rev. 0 | Page 6 of 16
Page 7
AD1990
Pin No. Mnemonic In/Out Description
52, 54, 59, 61 NC No Connection—Should Be Left Floating. 53 AINR I Analog Input for Right Channel. 55 REF_FILT O Filter Pin for Band Gap Reference—Should Be Bypassed to AGND. 56 AGND Negative Power Supply for Low Power Analog Circuitry. 57 AVDD Positive Power Supply for Low Power Analog Circuitry. 58 MOD_FILT O Modulator Filter Pin—Used to Set Time Constant of Modulator Order Reduction Circuit. 60 AINL O Analog Input for Left Channel. 62 NFL− I Left Channel Negative Feedback—Inverting Input. 63 NFL+ I Left Channel Negative Feedback—Noninverting Input. 64 PGND1 Negative Power Supply. Used for the A2 and B2 high power transistors.
Rev. 0 | Page 7 of 16
Page 8
AD1990

TYPICAL PERFORMANCE CHARACTERISTICS

0
0
–20
–40
–60
–80
–100
THD = 1% (5.9W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
–160
02
2 4 6 8 1012141618
FREQUENCY (kHz)
0
05380-004
Figure 4. 1 W Output Power into 4 Ω Load
–20
–40
–60
–80
–100
THD = 1% (5.9W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
–160
020
2 4 6 8 10 12 14 16 18
Figure 7. −60 dBFS Output Power into 4 Ω Load
0
–20
–40
–60
–80
–100
THD = 1% (4.0W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
0
–20
–40
–60
–80
–100
THD = 1% (4.0W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
FREQUENCY (kHz)
05380-007
–160
02
2 4 6 8 10 12 14 16 18
FREQUENCY (kHz)
0
05380-005
Figure 5. 1 W Output Power into 6 Ω Load
–160
020
2 4 6 8 10 12 14 16 18
Figure 8. −60 dBFS Output Power into 6 Ω Load
0
–20
–40
–60
–80
–100
THD = 1% (3.0W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
–160
02
2 4 6 8 10 12 14 16 18
FREQUENCY (kHz)
0
05380-006
Figure 6. 1 W Output Power into 8 Ω Load
0
–20
–40
–60
–80
–100
THD = 1% (3.0W))
–120
POWER (d BFS: 0dB = Power at Wh ich
–140
–160
020
2 4 6 8 10 12 14 16 18
Figure 9. −60 dBFS Output Power into 8 Ω Load
Rev. 0 | Page 8 of 16
FREQUENCY (kHz)
FREQUENCY (kHz)
05380-008
05380-009
Page 9
AD1990
20
0
–20
–40
–60
–80
–100
–120
POWER (d B, Relati ve to 500mW O utpu t Power)
–140
100 1k 10k
FREQUENCY (Hz)
Figure 10. IMD for 19 kHz/20 kHz Twin-Tone Stimulus with
500 mW Power in Each Tone
40
35
30
25
20
15
AMPLIFIER GAIN (dB)
10
5
0
PGA GAIN = 18dB
PGA GAIN = 12dB
PGA GAIN = 6dB
PGA GAIN = 0dB
100 1k 10k
FREQUENCY (Hz)
Figure 11. Amplifier Gain vs. Frequency, 6 Ω Load, PVDD = 12 V
0
1
0.1
0.01
THD (%)
0.001
0.0001
05380-010
100 1k 10k
FREQUENCY (Hz)
40
–50
–60
–70
–80
–90
–100
–110
–120
THD (dB, Rel ative to F undamen tal)
05380-013
Figure 13. THD vs. Frequency, 1 W Output Power into 4 Ω Load, PVDD = 12 V
1
0.1
0.01
THD (%)
0.001
0.0001
05380-011
100 1k 10k
FREQUENCY (Hz)
40
–50
–60
–70
–80
–90
–100
–110
–120
THD (dB, Rel ative to F undamen tal)
05380-014
Figure 14. THD vs. Frequency, 1 W Output Power into 6 Ω Load, PVDD = 12 V
1
40
–20
–40
–60
–80
Driven Chan nel Sig nal)
–100
SIGNAL IN I DLE CHANNEL (d B, Relative to
–120
100 1k 10k
L CHANNEL IDLE, R CHANNEL DRIVEN
FREQUENCY (Hz)
L CHANNEL DRIVEN, R CHANNEL IDLE
Figure 12. Channel Separation vs. Frequency, Driven Channel Has
1 W Output Power into 6 Ω Load
0.1
0.01
THD (%)
0.001
0.0001
05380-012
100 1k 10k
FREQUENCY (Hz)
Figure 15. THD vs. Frequency, 1 W Output Power into 8 Ω Load, PVDD = 12 V
–50
–60
–70
–80
–90
–100
–110
–120
THD (dB, Rel ative to F undamen tal)
05380-015
Rev. 0 | Page 9 of 16
Page 10
AD1990
100
10
1
0.1
THD OR THD + N (%)
0.01
0.001
THD + N
THD
0.1 1 10
OUTPUT PO WER (W)
Figure 16. THD and THD + N vs. Output Power, 1 kHz Sine, 4 Ω Load, PVDD = 12 V
100
10
1
0.1
THD OR THD + N (%)
0.01
0.001
THD + N
THD
0.1 1 10
OUTPUT PO WER (W)
Figu re 17. THD and THD + N vs. Output Power, 1 kHz Sine, 6 Ω Load, PVDD = 12 V
100
10
1
0.1
THD OR THD + N (%)
0.01
0.001
THD + N
THD
0.1 1 10
OUTPUT PO WER (W)
Figu re 18. THD and THD + N vs. Output Power, 1 kHz Sine, 8 Ω Load, PVDD = 12 V
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
0
–10
–20
–30
–40
–50
–60
–70
–80
–90
–100
10
9
8
7
6
5
4
3
2
OUTPUT PO WER PER CHANNEL (W)
THD OR THD + N (dB, Relati ve to Fun damental)
05380-016
1
0
8.0 12.0
8.5 9.0 9.5 10.0 10.5 11. 0 11.5
THD = 10%
THD = 1%
PVDD VOLTAGE (V)
05380-019
Figure 19. Maximum Output Power vs. PVDD, 4 Ω Load
10
9
8
7
6
5
4
3
2
OUTPUT PO WER PER CHANNEL (W)
THD OR THD + N (dB, Relati ve to Fun damental)
05380-017
1
0
8.0 12.0
8.5 9.0 9.5 10.0 10.5 11. 0 11.5
THD = 10%
THD = 1%
PVDD VOLTAGE (V)
05380-020
Figure 20. Maximum Output Power vs. PVDD, 6 Ω Load
10
9
8
7
6
5
4
3
2
OUTPUT PO WER PER CHANNEL (W)
THD OR THD + N (dB, Relati ve to Fun damental)
05380-018
1
0
8.0 12.0
8.5 9.0 9.5 10.0 10.5 11. 0 11.5
THD = 10%
PVDD VOLTAGE (V)
THD = 1%
05380-021
Figure 21. Maximum Output Power vs. PVDD, 8 Ω Load
Rev. 0 | Page 10 of 16
Page 11
AD1990

THEORY OF OPERATION

OVERVIEW

The AD1990 is a 2-channel, high performance, switching, audio power amplifier. Each of the two Σ-Δ modulators converts a single-ended analog input into a 2-level pulse stream that controls the differential, full H-bridge, power output stage. The combination of an Σ-Δ modulator and a switching power stage provides an inherently linear and efficient means of amplifying the entire range of audio frequencies. The AD1990 also offers warning and protection circuits for overcurrent and over­temperature conditions, as well as silent turn-on and turn-off transitions.

Σ-Δ MODULATOR

The AD1990 is a switching type, also known as a Class-D, audio power amplifier. This class of amplifiers maximizes efficiency by only using its power output devices in full-on or full-off states. While most Class-D amplifiers use some variation of pulse-width modulation (PWM), the AD1990 uses Σ-Δ modulation to determine the switching pattern of the output devices. This provides a number of important benefits. Σ-Δ modulators do not produce a sharp peak with many harmonics in the AM frequency band as pulse-width modulators (PWM) often do. In addition, the 1-bit quantizer produces excellent linearity across the full amplitude range.
Σ-Δ modulators require feedback to generate an error signal with respect to the input. The feedback voltages for the AD1990 modulators come from the outputs of the power devices and before the passive low-pass filters (see for nonlinear behavior in the power stage, such as nonoverlap time, mismatched rise and fall times, and propagation delays. It also reduces sensitivity to both dc and transient changes of the power supply voltage.
Σ-Δ modulators operate in discrete time. As with all time­quantized systems, the Nyquist frequency is equal to half of the sampling frequency and input signals above that point aliases back into the base band. The AD1990 sampling frequency (master clock) is equal to half the frequency of the input clock, approximately 6 MHz, so images only alias for input frequencies above approximately 3 MHz. This is far enough above the audio band that bandwidth and aliasing are not a problem in real applications.
The modulator has a noise shaping effect, and SNR is increased in the audio band by shifting the quantization noise upward in frequency. For a nominal input clock frequency of 12.288 MHz, the noise floor rises sharply above 20 kHz. The actual clock frequency used in an application circuit can deviate from this rate by as much as ±10%, and the corner frequency of the noise scales proportionately. The frequency at which the quantization noise dominates the output determines the amplifier’s practical bandwidth.
Figure 23). This compensates

MUTE AND RESET

When power is applied and the the AD1990 is in its lowest power consumption mode. The analog modulator is not running, and the power stage is tri­stated. On deasserting the start-up sequence that includes initialization of the modulator, the protection circuits, and other functions.
Once the start-up sequence is complete, the amplifier is in a state in which the modulator is running, but the output stage is not driven. When using a soft-start sequence that avoids any audible pop or click noise in the output signal.
The output power transistors do not switch while remains asserted. Unlike the analog mute circuits found on some amplifiers that can be limited in their attenuation by the control logic or crosstalk, the mute attenuation on the AD1990 is greater than its dynamic range. The noise floor of the output signal also drops while in are not switching.
MUTE

Power-Up Sequencing

Careful power-up is necessary when using the AD1990 to ensure correct operation and to avoid possible latch-up issues. The AD1990 should be powered up with held low until all the power supplies have stabilized. Once the supplies have stabilized, bring the AD1990 out of bringing
Begin the soft unmute sequence by bringing least 1 sec after the audio using a shorter start-up sequence (as shown in but the amplifier can produce an audible pop or click noise as the output starts switching. This is because the ac coupling capacitors at the analog input have a long time constant. If MUTE RESET state. They need ample time to settle at a bias voltage of V the reference voltage for the single-ended inputs, or the
amplifier starts with a slight dc offset.
RESET
high.
RESET
is deasserted substantially less than 1 sec after deasserting
, then these capacitors may not have charged to a steady
RESET
pin remains asserted,
RESET
pin, the modulator begins a
is deasserted, the output is started
MUTE
MUTE
because the output transistors
RESET
MUTE
rising edge. The amplifier produces
and
RESET
high at
MUTE
by
Tabl e 7),
REF
,

GAIN STRUCTURE

Analog Input Levels

The AD1990 has single-ended inputs for the left and right channels. The analog input section uses an internal amplifier to bias the input signal to the reference level, V equal to AV prevents this bias voltage from affecting the signal source. In combination with the nominal 20 kΩ input impedance, the value of this capacitor should be large enough to produce a flat frequency response at the lowest input frequency of interest.
/2. A dc-blocking capacitor, as shown in Figure 22,
DD
, which is nominally
REF
Rev. 0 | Page 11 of 16
Page 12
AD1990
0V
Note that the amplifier is capable of dc-coupled operation if the circuit includes some means to account for this bias voltage.
+
AINL/ AINR
This fixed total resistance to ground eliminates the last free variable and gives the following equations for the resistors:
21810
R4R2
==
PV
DD
R1 = R3 = 6000 − R2
05380-028
Figure 22. AC-Coupled Input Signal

Setting the Modulator Gain

The AD1990 modulator uses a combination of the input signal and feedback from the power output stage to calculate its two­state output pattern. The feedback input nodes are part of the internal analog circuit that operates from the AV
(nominal
DD
5 V) power supply. Because the voltage measured at the power outputs is nominally between 0 V and PV the 0 V to AV
range, a voltage divider is required to scale the
DD
, and thus beyond
DD
feedback to an appropriate level.
Resistor voltage dividers should sense the voltage on each side of the differential output and provide these feedback signals to the modulator, as shown in
PV
DD
OUTx+ OUTx–
PGND PGND
NFx+ NFx–
D1 D3
D2 D4
R1 R3
R2 R4
Figure 23. H-Bridge Configuration
Figure 23.
EXTERNAL COMPONENTS
LLR
L
CC
PV
DD
The resistor values should satisfy the following equation to maintain modulator stability.
PV
21
RR
Gain =
+
=
=
2
R
43
RR
+
R
DD
635.34
Selecting a gain that meets this criterion ensures that the modulator remains in a stable operating condition.
The ratio of the resistances sets the gain rather than the absolute values. However, the dividers provide a path from the high voltage supply to ground; therefore, the values should be large enough to produce negligible loss due to quiescent current.
Note that the gain previously mentioned applies to each side of the differential output pair. Therefore, the total forward gain for the modulator and output stage is twice that value. Recommended resistor values for some common supply voltages are shown in
Tabl e 1 0 .
Table 10. Recommended Feedback Resistor Values
PVDD (V) R1 (kΩ) R2 (kΩ)
Voltage Divider Gain
Differential System Gain
8 3.27 2.73 2.2 4.4 (13.8 dB) 10 3.82 2.18 2.8 5.6 (17.6 dB) 12 4.18 1.82 3.3 6.6 (20.8 dB)

Programmable Gain Amplifier (PGA)

The Σ-Δ modulator itself requires a fixed gain for a given value of PV
to maintain optimal stability. This gain can be appropriate,
DD
but many applications require more gain to account for low source signal levels. The AD1990 includes a programmable gain amplifier (PGA) to boost the overall amplifier gain. The total gain for the amplifier is the product of the modulator gain and the PGA gain. PGA1 (Pin 31) and PGA0 (Pin 32) select one of four PGA gain values, as shown in
Tabl e 11 .
Table 11. PGA Gain Settings
05380-029
PGA1 PGA0 PGA Gain
0 0 1 (0 dB) 0 1 2 (6 dB) 1 0 4 (12 dB) 1 1 8 (18 dB)
The AD1990 incorporates a single-ended-to-differential converter for each channel in the analog front-end section. The PGA is also part of this analog front-end, and it affects the analog input signal before it enters the Σ-Δ modulator. The PGA1 and PGA0 pins are continuously monitored and allow the gain to be changed at any time.
The chip contains a calibration circuit to minimize voltage offsets at the speaker, which helps to minimize clicks and pops when muting or unmuting. Optimal performance is achieved for the offset calibration circuit when the feedback divider resistors sum to 6 kΩ, that is, (R1 + R2) = 6 kΩ, and (R3 + R4) = 6 kΩ.
Rev. 0 | Page 12 of 16
Page 13
AD1990

POWER STAGE

The H-Bridge

The output stage of the AD1990 includes four integrated MOSFET devices arranged in a full H-bridge, as shown in Figure 23. The P-Type, high-side transistor of one leg and the N-Type, low-side transistor of the opposite leg switch on and off as a pair producing a total voltage swing across the load of
to +PVDD. The drive is floating and differential, and it is
−PV
DD
important that neither output terminal be shorted to ground.
The power supply for the output stage of the AD1990, PV should be in the 8 V to 20 V range and should be capable of supplying enough current to drive the load. Connect the power supply across the PVDD and PGND pins. The feedback pins, NFR+, NFR−, NFL+, and NFL−, supply negative feedback to the modulator as described in the
Setting the Modulator Gain section.
DD
,
Table 12. Nonoverlap Time Settings
DCTRL2 DCTRL1 DCTRL0 Nonoverlap Time (ns)
0 0 0 62 0 0 1 49 0 1 0 37 0 1 1 24 1 0 0 15 1 0 1 13.5 1 1 0 12 1 1 1 9
1
Values are typical and are not production tested.
HIGH-SIDE
GATE DRIVE
LOW-SIDE
GATE DRIVE
1
For reactive loads, the impedance can only be below the recommended threshold over a small portion of the amplifier’s bandwidth. In these cases, the amplifier can enter overcurrent shutdown in response to even small input signals in those frequency bands. When designing a system, use the minimum load impedance over the entire range of amplified frequencies when calculating current output rather than the average or nominal load impedance ratings often cited by loudspeaker driver manufacturers.

Output Transistor Nonoverlap Time

The AD1990 allows the user to select from one of eight different nonoverlap times, as shown in
Figure 24. Nonoverlap time prevents or minimizes the period during which both the high­side and low-side devices are on simultaneously due to propagation delays and nonzero rise and fall times. If both the upper and lower portions of a half-bridge conduct simultaneously, there is a path directly from the power supply to ground and an induced current flow known as shoot-through. However, introducing this delay increases distortion by pushing the switching pattern further from an ideal two-state waveform. Selecting the nonoverlap delay requires a compromise between distortion and efficiency. The logic levels on the three delay control pins, DCTRL2, DCTRL1, and DCTRL0, set the nonoverlap time according to rising edge of
Tabl e 1 2 . The state of DCTRL[2:0] is read on the
RESET
and should not be changed while
RESET
is logic high.
t
NOL
Figure 24. Half-Bridge Nonoverlap Delay Timing
t
NOL
05380-030
The shortest setting (DCTRL[2:0] = 111) or the second shortest setting (DCTRL[2:0] = 111) is recommended for most applications. These two settings allow a small trade-off between efficiency and distortion. Longer nonoverlap times generally increase distortion while providing little or no decrease in shoot­through current.

CLOCKING

The AD1990 Σ-Δ modulator requires an external clock source with a nominal frequency of 12.288 MHz. This clock can come from a crystal or from an existing clock signal in the application circuit. The discrete time portions of the modulator run internally at 6.144 MHz, corresponding to 128 × f
As mentioned in the
Σ-Δ Modulator section, the modulator has a noise-shaping effect such that SNR is increased within the audio band by shifting modulator quantization noise upward in frequency. For an external clock frequency of 12.288 MHz, the modulator’s noise-shaping works in a manner that results in a flat noise floor at the amplifier output for frequencies 20 kHz and below. Above 20 kHz, the amplifier noise rises due to the spectral shaping of the modulator quantization noise. At very high frequencies, the noise floor levels off and decreases due to poles in the modulator noise-transfer function and in the external LC filter.
The clock frequency does not have to be exactly equal to
12.288 kHz and can vary by up to ±10%. For other rates, the noise corner scales linearly with frequency. When the modulator runs at a rate lower than nominal, the average power stage switching frequency decreases, the efficiency increases slightly, and the noise floor begins to rise at a slightly lower frequency. Likewise, a faster clock gives slightly increased bandwidth and slightly lower efficiency.
, where fS = 48 kHz.
S
Rev. 0 | Page 13 of 16
Page 14
AD1990

Using a Crystal Oscillator Clocking Multiple Amplifiers in Parallel

The AD1990 can use a crystal connected to the CLKI and CLKO pins as a master clock source, as shown in
Figure 25. The CLKI and CLKO pins connect to an internal inverter to create a full resonator. The typical values shown work in many applications, but the crystal manufacturer should provide the exact type and value of the capacitors and the resistor.
22pF 22pFXTAL
47
CLKI
Figure 25. Crystal Connection
CLKO
05380-031

Using an External Clock Source

If a clock signal of the appropriate frequency already exists in the application circuit, connect it directly to CLKI and leave CLKO floating. The logic levels of the square wave should be compatible with those defined in
Specifications section.
Large amounts of jitter on the clock input degrade performance. Whenever possible, avoid passing the clock signal through programmable logic and other circuits with unknown or variable propagation delay. In general, clock signals suitable for audio ADCs or DACs are also appropriate for use with the AD1990.
If there are multiple AD199x family amplifiers connected to the same PV
supply, use the same clock source (or synchronous
DD
derivatives) for each amplifier as previously described. Avoid clocking amplifiers from similar but asynchronous clocks if they use the same power supply because this can result in beat frequencies.

PROTECTION CIRCUITS AND ERROR REPORTING

Thermal Protection

The AD1990 features thermal protection. When the die temperature exceeds approximately 135°C, the thermal warning
ERR1
error output ( approximately 150°C, the thermal shutdown error output
ERR2
) is asserted. If this occurs, the part shuts down to
( prevent damage to the part. When the die temperature drops below approximately 120°C, the part returns to normal operation automatically and negates both error outputs.

Overcurrent Protection

The AD1990 features over current or short-circuit protection. If the current through any power transistors exceeds approximately 4 A, the part enters a mute state and the overcurrent error output (
ERR0 clear automatically. Restore normal operation and clear the error condition by either asserting and then negating by asserting and then negating
) is asserted. If the die temperature exceeds
) is asserted. This is a latched error and does not
RESET
MUTE
.
or
Rev. 0 | Page 14 of 16
Page 15
AD1990
T

APPLICATION CIRCUITS

0.1µF
10µF
10µF
+
4.7µF 0.1µF
+
+
AV
DD
+
47µF
AINL
AINR
REF_FILT
+
47µF
AVDD
DV
DD
DVDD
AD1990
PV
DD
1000µF
PVDD1
PV
DD
OUTL+
NFL+
NFL–
OUTL–
+
PVDD2
+
1000µF
0.1µF
0.1µF0.1µF
L
C
R1
R2
R2
R1
L
C
R1
R2
R2
R1
R1 = 4.2k R2 = 1.8k L = 18µH C = 1µF LOAD = 6
L
C
L
C
05380-032
DIGITAL
INPUTS
HERMAL SHUTDOW N
THERMAL WARNI NG
OVERCURRENT
PGA0
PGA1
DCTRL2
DCTRL1
DCTRL0
MUTE
RESET
ERR2
ERR1
ERR0
CLKI
CLKO
AGND
DGND
OUTR+
NFR+
NFR–
OUTR–
PGND1
PGND2
Figure 26. Typical Application Circuit
Rev. 0 | Page 15 of 16
Page 16
AD1990

OUTLINE DIMENSIONS

7.50 REF
0.30
0.25
0.18 PIN 1
1
16
INDICATOR
7.25
7.10 SQ
6.95
0.25 MIN
64
17
1.00
0.85
0.80
SEATING
PLANE
12° MAX
9.00
BSC SQ
PIN 1 INDICATOR
VIEW
TOP
0.80 MAX
0.65 TYP
0.50 BSC
8.75
BSC SQ
0.20 REF
0.60 MAX
0.45
0.40
0.35
0.05 MAX
0.02 NOM
49
48
33
32
0.60 MAX
EXPOSED PAD
(BOTTOM VIEW)
COMPLIANT TO JEDEC S TANDARDS MO-220- VMMD-4
122105-0
Figure 27. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
9 mm × 9 mm Body, Very Thin Quad
(CP-64-3)
Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option
AD1990ACPZ AD1990ACPZRL AD1990ACPZRL71−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ), 7” Tape and Reel CP-64-3 EVAL-AD1990EB Evaluation Board
1
Z = Pb-free part.
1
−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ) CP-64-3
1
−40°C to +85°C 64-Lead Lead Frame Chip Scale Package (LFCSP_VQ), 13” Tape and Reel CP-64-3
©2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D05380-0-4/06(0)
Rev. 0 | Page 16 of 16
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