Analog Devices TMP01 Datasheet

Low Power, Programmable

VPTAT

V+

TEMPERATURE

SENSOR &

VOLTAGE

REFERENCE

2.5V

SENSOR

1

2

3

4

8

7

6

5

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

R1

TMP01

VREF

SET

HIGH

SET

LOW

GND

R2

R3

UNDER

OVER

a

FEATURES
–558C to +1258C (–678F to +2578F) Operation

61.08C Accuracy Over Temperature (typ)
Temperature-Proportional Voltage Output
User Programmable Temperature Trip Points
User Programmable Hysteresis
20 mA Open Collector Trip Point Outputs
TTL/CMOS Compatible
Single-Supply Operation (4.5 V to 13.2 V)
Low Cost 8-Pin DIP and SO Packages

APPLICATIONS
Over/Under Temperature Sensor and Alarm
Board Level Temperature Sensing
Temperature Controllers
Electronic Thermostats
Thermal Protection
HVAC Systems
Industrial Process Control
Remote Sensors

GENERAL DESCRIPTION

The TMP01 is a temperature sensor which generates a voltage
output proportional to absolute temperature and a control signal
from one of two outputs when the device is either above or
below a specific temperature range. Both the high/low tempera­ture trip points and hysteresis (overshoot) band are determined
by user-selected external resistors. For high volume production,
these resistors are available on-board.

The TMP01 consists of a bandgap voltage reference combined
with a pair of matched comparators. The reference provides
both a constant 2.5 V output and a voltage proportional to abso­lute temperature (VPTAT) which has a precise temperature co­efficient of 5 mV/K and is 1.49 V (nominal) at +25°C. The
comparators compare VPTAT with the externally set tempera­ture trip points and generate an open-collector output signal
when one of their respective thresholds has been exceeded.

Temperature Controller

TMP01*

FUNCTIONAL BLOCK DIAGRAM

Hysteresis is also programmed by the external resistor chain and
is determined by the total current drawn out of the 2.5 V refer­ence. This current is mirrored and used to generate a hysteresis
offset voltage of the appropriate polarity after a comparator has
been tripped. The comparators are connected in parallel, which
guarantees that there is no hysteresis overlap and eliminates
erratic transitions between adjacent trip zones.

The TMP01 utilizes proprietary thin-film resistors in conjunc­tion with production laser trimming to maintain a temperature
accuracy of ±1°C (typ) over the rated temperature range, with
excellent linearity. The open-collector outputs are capable of
sinking 20 mA, enabling the TMP01 to drive control relays di­rectly. Operating from a +5 V supply, quiescent current is only
500 µA (max).

The TMP01 is available in the low cost 8-pin epoxy mini-DIP
and SO (small outline) packages, and in die form.

*Protected by U.S. Patent No. 5,195,827.

REV. C

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

© Analog Devices, Inc., 1995

One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: [phone] Fax: [phone]

TMP01EP/FP, TMP01ES/FS–SPECIFICATIONS

Plastic DIP and Surface Mount Packages

(V+ = +5 V, GND = O V, –408C ≤ TA ≤ +858C unless otherwise noted)

Parameter Symbol Conditions Min Typ Max Units

INPUTS SET HIGH, SET LOW

Offset Voltage V
Offset Voltage Drift TCV
Input Bias Current, “E” I
Input Bias Current, “F” I

OUTPUT VPTAT

1

OS

OS
B
B

0.25 mV
3 µV/°C
25 50 nA
25 100 nA

Output Voltage VPTAT TA = +25°C, No Load 1.49 V
Scale Factor TC
Temperature Accuracy, “E” T
Temperature Accuracy, “F” T

VPTAT

= +25°C, No Load –1.5 ±0.5 1.5 °C

A

= +25°C, No Load –3 ±1.0 3 °C

A

Temperature Accuracy, “E” 10°C < T
Temperature Accuracy, “F” 10°C < T
Temperature Accuracy, “E” –40°C < T
Temperature Accuracy, “F” –40°C < T
Temperature Accuracy, “E” –55°C < T
Temperature Accuracy, “F” –55°C < T

< 40°C, No Load ±0.75 °C

A

< 40°C, No Load ±1.5 °C

A

< 85°C, No Load –3.0 ±1 3.0 °C

A

< 85°C, No Load –5.0 ±2 5.0 °C

A

< 125°C, No Load ±1.5 °C

A

< 125°C, No Load ±2.5 °C

A

5 mV/K

Repeatability Error ∆VPTAT Note 4 0.25 Degree
Long Term Drift Error Notes 2 and 6 0.25 0.5 Degree
Power Supply Rejection Ratio PSRR TA = +25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ±0.1 %/V

OUTPUT VREF

Output Voltage, “E” VREF T
Output Voltage, “F” VREF T
Output Voltage, “E” VREF –40°C < T
Output Voltage, “F” VREF –40°C < T
Output Voltage, “E” VREF –55°C < T
Output Voltage, “F” VREF –55°C < T
Drift TC

VREF

= +25°C, No Load 2.495 2.500 2.505 V

A

= +25°C, No Load 2.490 2.500 2.510 V

A

< 85°C, No Load 2.490 2.500 2.510 V

A

< 85°C, No Load 2.485 2.500 2.515 V

A

< 125°C, No Load 2.5 ± 0.01 V

A

< 125°C, No Load 2.5 ± 0.015 V

A

–10 ppm/°C
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ±0.05 %/V
Load Regulation 10 µA ≤ I
Output Current, Zero Hysteresis I
Hysteresis Current Scale Factor SF

VREF

HYS

(Note 1) 5.0 µA/°C

≤ 500 µA ±0.1 ±0.25 %/mA

VREF

7 µA
Turn-On Settling Time To Rated Accuracy 25 µs

OPEN-COLLECTOR OUTPUTS OVER, UNDER

I

Output Low Voltage V
Output Low Voltage V
Output Leakage Current I
Fall Time t

OL

OL
OH
HL

= 1.6 mA 0.25 0.4 V

SINK

I

= 20 mA 0.6 V

SINK

V+ = 12 V 1 100 µA
See Test Load 40 ns

POWER SUPPLY

Supply Range V+ 4.5 13.2 V
Supply Current I
Supply Current I
Power Dissipation P

NOTES

1

K = °C + 273.15.

2

Guaranteed but not tested.

3

Does not consider errors caused by heating due to dissipation of output load currents.

4

Maximum deviation between +25°C readings after temperature cycling between –55°C and +125°C.

5

Typical values indicate performance measured at TA = +25°C.

6

Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

SY
SY

DISS

Unloaded, +V = 5 V 400 500 µA
Unloaded, +V = 13.2 V 450 800 µA
+V = 5 V 2.0 2.5 mW

Test Load

V+

1kΩ

20pF

–2–

REV. C

TMP01

TO-99 Metal Can Package (V+ = +5 V, GND = O V, –408C ≤ TA ≤ +858C

TMP01FJ–SPECIFICA TIONS

Parameter Symbol Conditions Min Typ Max Units

INPUTS SET HIGH, SET LOW

Offset Voltage V
Offset Voltage Drift TCV
Input Bias Current, “F” I

OUTPUT VPTAT

1

Output Voltage VPTAT TA = +25°C, No Load 1.49 V
Scale Factor TC
Temperature Accuracy, “F” T
Temperature Accuracy, “F” 10°C < T
Temperature Accuracy, “F” –40°C < T
Temperature Accuracy, “F” –55°C < T
Repeatability Error ∆VPTAT Note 4 0.25 Degree
Long Term Drift Error Notes 2 and 6 0.25 0.5 Degree
Power Supply Rejection Ratio PSRR TA = +25°C, 4.5 V ≤ V+ ≤ 13.2 V ±0.02 ±0.1 %/V

OUTPUT VREF

Output Voltage, “F” VREF T
Output Voltage, “F” VREF –40°C < T
Output Voltage, “F” VREF –55°C < T
Drift TC
Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.01 ±0.05 %/V
Load Regulation 10 µA ≤ I
Output Current, Zero Hysteresis I
Hysteresis Current Scale Factor SF
Turn-On Settling Time To Rated Accuracy 25 µs

OPEN-COLLECTOR OUTPUTS OVER, UNDER

Output Low Voltage V
Output Low Voltage V
Output Leakage Current I
Fall Time t

POWER SUPPLY

Supply Range V+ 4.5 13.2 V
Supply Current I
Supply Current I
Power Dissipation P

NOTES

1

K = °C + 273.15.

2

Guaranteed but not tested.

3

Does not consider errors caused by heating due to dissipation of output load currents.

4

Maximum deviation between +25°C readings after temperature cycling between –55°C and +125°C.

5

Typical values indicate performance measured at TA = +25°C.

6

Observed in a group sample over an accelerated life test of 500 hours at 150°C.

Specifications subject to change without notice.

OS

OS

B

VPTAT

VREF

VREF

HYS

OL

OL
OH
HL

SY
SY

DISS

unless otherwise noted)

0.25 mV
3 µV/°C
25 100 nA

5 mV/K

= +25°C, No Load –3 ±1.0 3 °C

A

= +25°C, No Load 2.490 2.500 2.510 V

A

< 40°C, No Load ±1.5 °C

A

< 85°C, No Load –5.0 ±2 5.0 °C

A

< 125°C, No Load ±2.5 °C

A

< 85°C, No Load 2.480 2.500 2.520 V

A

< 125°C, No Load 2.5 ± 0.015 V

A

–10 ppm/°C

≤ 500 µA ±0.1 ±0.25 %/mA

VREF

7 µA

(Note 1) 5.0 µA/°C

I

= 1.6 mA 0.25 0.4 V

SINK

I

= 20 mA 0.6 V

SINK

V+ = 12 V 1 100 µA
See Test Load, Note 2 40 ns

Unloaded, +V = 5 V 400 500 µA
Unloaded, +V = 13.2 V 450 800 µA
+V = 5 V 2.0 2.5 mW

REV. C

–3–

TMP01
W AFER TEST LIMITS

(VDD = +5.0 V, GND = 0 V, TA = +258C, unless otherwise noted)

Parameter Symbol Conditions Min Typ Max Units

INPUTS SET HIGH, SET LOW

Input Bias Current I

B

100 nA

OUTPUT VPTAT

Temperature Accuracy TA = +25°C, No Load 1.5 °C

OUTPUT VREF

Nominal Value VREF T

= +25°C, No Load 2.490 2.510 V

A

Line Regulation 4.5 V ≤ V+ ≤ 13.2 V ±0.05 %/V
Load Regulation 10 µA ≤ I

≤ 500 µA ±0.25 %/mA

VREF

OPEN-COLLECTOR OUTPUTS OVER, UNDER

Output Low Voltage V
Output Low Voltage V
Output Leakage Current I

OL
OL

OH

I

= 1.6 mA 0.4 mV

SINK

I

= 20 mA 1.0 V

SINK

100 µA

POWER SUPPLY

Supply Range V+ 4.5 13.2 V
Supply Current I

NOTES
Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed
for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing.

SY

Unloaded 600 µA

DICE CHARACTERISTICS

Die Size 0.078 × 0.071 inch, 5,538 sq. mils

(1.98 × 1.80 mm, 3.57 sq. mm)

Transistor Count: 105

8

1

7

2 3

6

5

1. VREF

2. SETHIGH

3. SETLOW

4. GND (TWO PLACES)
(CONNECTED TO SUBSTRATE)

5. VPTAT

4

4

6. UNDER

7. OVER

8. V+

For additional DICE ordering information, refer to databook.

–4–

REV. C

TMP01

ABSOLUTE MAXIMUM RATINGS

Maximum Supply Voltage . . . . . . . . . . . . . . . . –0.3 V to +15 V

Maximum Input Voltage

(SETHIGH, SETLOW) . . . . . . . . .–0.3 V to [(V+) +0.3 V]

Maximum Output Current (VREF, VPTAT) . . . . . . . . . 2 mA

Maximum Output Current (Open Collector Outputs) . . 50 mA

Maximum Output Voltage (Open Collector Outputs) . . . .15 V

Operating Temperature Range . . . . . . . . . . . .–55°C to +150°C

Dice Junction Temperature . . . . . . . . . . . . . . . . . . . . . +150°C

Storage Temperature Range . . . . . . . . . . . . – 65°C to +150°C

Lead Temperature (Soldering, 60 sec) . . . . . . . . . . . . . +300°C

Package Type θ

8-Pin Plastic DIP (P) 103
8-Lead SOIC (S) 158
8-Lead TO-99 Can (J) 150

NOTES

1

θJA is specified for device in socket (worst case conditions).

2

θJA is specified for device mounted on PCB.

JA

1
2
1

θ

JC

Units

43 °C/W
43 °C/W
18 °C/W

CAUTION

1. Stresses above those listed under “Absolute Maximum Rat-

ings” may cause permanent damage to the device. This is a
stress rating only and functional operation at or above this
specification is not implied. Exposure to the above maximum
rating conditions for extended periods may affect device
reliability.

2. Digital inputs and outputs are protected, however, permanent

damage may occur on unprotected units from high energy
electrostatic fields. Keep units in conductive foam or packag­ing at all times until ready to use. Use proper antistatic han­dling procedures.

3. Remove power before inserting or removing units from their

sockets.

ORDERING GUIDE

Temperature Package Package

Model/Grade Range

l

Description Option

GENERAL DESCRIPTION

The TMP01 is a very linear voltage-output temperature sensor,
with a window comparator that can be programmed by the user
to activate one of two open-collector outputs when a predeter­mined temperature setpoint voltage has been exceeded. A low
drift voltage reference is available for setpoint programming.

The temperature sensor is basically a very accurately tempera­ture compensated, bandgap-type voltage reference with a buff­ered output voltage proportional to absolute temperature
(VPTAT), accurately trimmed to a scale factor of 5 mV/K. See
the Applications Information following.

The low drift 2.5 V reference output VREF is easily divided ex­ternally with fixed resistors or potentiometers to accurately es­tablish the programmed heat/cool setpoints, independent of
temperature. Alternatively, the setpoint voltages can be supplied
by other ground referenced voltage sources such as user­programmed DACs or controllers. The high and low setpoint
voltages are compared to the temperature sensor voltage, thus
creating a two-temperature thermostat function. In addition,
the total output current of the reference (I

) determines the

VREF

magnitude of the temperature hysteresis band. The open collec­tor outputs of the comparators can be used to control a wide va­riety of devices.

HYSTERESIS

VREF

SET

HIGH

SET

LOW

GND

2

3

4

CURRENT

VOLTAGE

REFERENCE

SENSOR

CURRENT

MIRROR

AND

ENABLE

WINDOW

COMPARATOR

1kΩ

I

HYS

HYSTERESIS
VOLTAGE

TEMPERATURE

OUTPUT

TMP01

81

7

6

5

V+

OVER

UNDER

VPTAT

TMP01EP XIND Plastic DIP N-8
TMP01FP XIND Plastic DIP N-8
TMP01ES XIND SOIC SO-8
TMP01FS XIND SOIC SO-8
TMP01FJ

2

XIND TO-99 Can H-08A

TMP01GBC +25°C Die

NOTES

1

XIND = –40°C to +85°C.

2

Consult factory for availability of MIL/883 version in TO-99 can.

REV. C

Figure 1. Detailed Block Diagram

–5–

TMP01

Temperature Hysteresis

The temperature hysteresis is the number of degrees beyond the
original setpoint temperature that must be sensed by the TMP01
before the setpoint comparator will be reset and the output dis­abled. Figure 2 shows the hysteresis profile. The hysteresis is
programmed by the user by setting a specific load on the refer­ence voltage output VREF. This output current I

VREF

is also
called the hysteresis current, which is mirrored internally and
fed to a buffer with an analog switch.

HYSTERESIS

HIGH

T

SETHIGH

OUTPUT

VOLTAGE

OVER, UNDER

LO

HI

HYSTERESIS

LOW

T

SETLOW

HYSTERESIS HIGH =
HYSTERESIS LOW

TEMPERATURE

Figure 2. TMP01 Hysteresis Profile

After a temperature setpoint has been exceeded and a compara­tor tripped, the buffer output is enabled. The output is a cur­rent of the appropriate polarity which generates a hysteresis
offset voltage across an internal 1000 Ω resistor at the compara­tor input. The comparator output remains “on” until the volt­age at the comparator input, now equal to the temperature
sensor voltage VPTAT summed with the hysteresis offset, has
returned to the programmed setpoint voltage. The comparator
then returns LOW, deactivating the open-collector output and
disabling the hysteresis current buffer output. The scale factor
for the programmed hysteresis current is:

I

HYS

= I

= 5 µA/°C + 7 µA

VREF

Thus since VREF = 2.5 V, with a reference load resistance of
357 kΩ or greater (output current 7 µA or less), the temperature
setpoint hysteresis will be zero degrees. See the temperature
programming discussion below. Larger values of load resistance
will only decrease the output current below 7 µA and will have
no effect on the operation of the device. The amount of hyster­esis is determined by selecting a value of load resistance for
VREF, as shown below.

Programming the TMP01

In the basic fixed-setpoint application utilizing a simple resistor
ladder voltage divider, the desired temperature setpoints are
programmed in the following sequence:

1. Select the desired hysteresis temperature.

2. Calculate the hysteresis current I

VREF

.

3. Select the desired setpoint temperatures.

4. Calculate the individual resistor divider ladder values needed

to develop the desired comparator setpoint voltages at
SETHIGH and SETLOW.

The hysteresis current is readily calculated, as shown. For
example, for 2 degrees of hysteresis, I
setpoint voltages V

SETHIGH

and V

SETLOW

= 17 µA. Next, the

VREF

are determined using
the VPTAT scale factor of 5 mV/K = 5 mV/(°C + 273.15),
which is 1.49 V for +25°C. We then calculate the divider resis­tors, based on those setpoints. The equations used to calculate
the resistors are:

V
V
R1 (kΩ) = (V

R2 (kΩ) = (V
R3 (kΩ) = V

SETHIGH
SETLOW

(V

(V

SETHIGH

= (T

SETHIGH

= (T

SETLOW

– V

VREF

= (2.5 V – V

SETHIGH

SETLOW/IVREF

– V

VREF

SETHIGH

– V

SETLOW

V

SETLOW

+ 273.15)(5 mV/°C)

+ 273.15) (5 mV/°C)

)/I

)/I

VREF

VREF

)/I

I

VREF

=

VREF

1

2

3

4

V

VREF

)/I

)/I

/I

– V

VREF

V

VREF

VREF

SETHIGH

SETHIGH

SETLOW

= 2.5V

= R1

SETHIGH

= R2

V

SETLOW

= R3

GND

TMP01

8

7

6

5

V+

OVER

UNDER

VPTAT

Figure 3. TMP01 Setpoint Programming

The total R1 + R2 + R3 is equal to the load resistance needed
to draw the desired hysteresis current from the reference, or
I

.

VREF

The formulas shown above are also helpful in understanding the
calculation of temperature setpoint voltages in circuits other
than the standard two-temperature thermostat. If a setpoint
function is not needed, the appropriate comparator should be
disabled. SETHIGH can be disabled by tying it to V+, SET­LOW by tying it to GND. Either output can be left unconnected.

218 248 273 298 323 348 373 398

K

–55 –25

–18

0 25 50 75 100 125

°C

–67 –25 0 32 50 77 100 150 200 212 257

°F

1.09 1.24 1.365 1.49 1.615 1.74 1.865 1.99

VPTAT

Figure 4. Temperature—VPTAT Scale

–6–

REV. C

TMP01

Understanding Error Sources

The accuracy of the VPTAT sensor output is well characterized
and specified, however preserving this accuracy in a heating or
cooling control system requires some attention to minimizing
the various potential error sources. The internal sources of
setpoint programming error include the initial tolerances and
temperature drifts of the reference voltage VREF, the setpoint
comparator input offset voltage and bias current, and the hys­teresis current scale factor. When evaluating setpoint program­ming errors, remember that any VREF error contribution at the
comparator inputs is reduced by the resistor divider ratios. The
comparator input bias current (inputs SETHIGH, SETLOW)
drops to less than 1 nA (typ) when the comparator is tripped.
This can account for some setpoint voltage error, equal to the
change in bias current times the effective setpoint divider ladder
resistance to ground.

The thermal mass of the TMP01 package and the degree of
thermal coupling to the surrounding circuitry are the largest
factors in determining the rate of thermal settling, which ulti­mately determines the rate at which the desired temperature
measurement accuracy may be reached. Thus, one must allow
sufficient time for the device to reach the final temperature.
The typical thermal time constant for the plastic package is
approximately 140 seconds in still air! Therefore, to reach the
final temperature accuracy within 1%, for a temperature change
of 60 degrees, a settling time of 5 time constants, or 12 min­utes, is necessary.

The setpoint comparator input offset voltage and zero hyster­esis current affect setpoint error. While the 7 µA zero hysteresis
current allows the user to program the TMP01 with moderate
resistor divider values, it does vary somewhat from device to de­vice, causing slight variations in the actual hysteresis obtained

in practice. Comparator input offset directly impacts the pro­grammed setpoint voltage and thus the resulting hysteresis
band, and must be included in error calculations.

External error sources to consider are the accuracy of the pro­gramming resistors, grounding error voltages, and the overall
problem of thermal gradients. The accuracy of the external
programming resistors directly impacts the resulting setpoint
accuracy. Thus in fixed-temperature applications the user
should select resistor tolerances appropriate to the desired
programming accuracy. Resistor temperature drift must be
taken into account also. This effect can be minimized by select­ing good quality components, and by keeping all components in
close thermal proximity. Applications requiring high measure­ment accuracy require great attention to detail regarding
thermal gradients. Careful circuit board layout, component
placement, and protection from stray air currents are necessary
to minimize common thermal error sources.

Also, the user should take care to keep the bottom of the
setpoint programming divider ladder as close to GND (Pin 4)
as possible to minimize errors due to IR voltage drops and cou­pling of external noise sources. In any case, a 0.1 µF capacitor
for power supply bypassing is always recommended at the chip.

Safety Considerations In Heating And Cooling System Design

Designers should anticipate potential system fault conditions
which may result in significant safety hazards which are outside
the control of and cannot be corrected by the TMP01-based
circuit. Governmental and industrial regulations regarding
safety requirements and standards for such designs should be
observed where applicable.

REV. C

550

525

500

475

+125°C

450

425

SUPPLY CURRENT – µA

400

375

350

+85°C

–55°C

+25°C

–40°C

SUPPLY VOLTAGE – Volts

Figure 5. Supply Current vs. Supply Voltage

5.0

4.5

4.0

3.5

MINIMUM SUPPLY VOLTAGE – Volts

3.0

20501510

–75 125–50 1007550250–25

TEMPERATURE – °C

Figure 6. Minimum Supply Voltage vs. Temperature

–7–

TMP01

+2.0

+1.5

+1.0

C

°

+0.5

0

–0.5

VPTAT ERROR –

–1.0

–1.5

–3.0

–75 125–50 1007550250–25

TEMPERATURE –

V+ = +5V

°

C

Figure 7. VPTAT Accuracy vs. Temperature

2.508

2.506

2.504

2.502

VREF – Volts

2.500

V+ = +5V

2.510

2.508
CURVES NOT NORMALIZED

2.506
EXTRAPOLATED FROM OPERATING LIFE DATA

2.504

2.502

2.500

2.498

VREF – Volts

2.496

2.494

2.492

2.490

T = HOURS OF OPERATION AT 125°C; V+ = +5V

X + 3σ

X

X – 3σ

10002000 800400 600

Figure 10. VREF Long Term Drift Accelerated by Burn-In

100

80

60

40

PSRR – dB

20

V+ = +5V
I

= 10µA

VREF

2.498

2.496
–75 125–50 1007550250–25

TEMPERATURE – °C

Figure 8. VREF Accuracy vs. Temperature

6.0
VC = +15V

V+ = +5V

5.0
TA = +25°C

4.0

3.0

– Volts

CE

V

2.0

1.0

0

IC – mA

Figure 9. Open-Collector Output (
tion Voltage vs. Output Current

OVER, UNDER

50100 403020

) Satura-

0

–20

100

1k 1M100k10k

FREQUENCY – Hz

Figure 11. VREF Power Supply Rejection vs. Frequency

1.0

0.1

OFFSET VOLTAGE – mV

V+ = +5V
I

= 7.5µA

VREF

0.01

–75 –50 1251007550250–25

TEMPERATURE – °C

Figure 12. Set High, Set Low Input Offset Voltage vs.
Temperature

–8–

REV. C

8

7.26.2 7

6.8

6.66.4 87.87.67.4

REFERENCE CURRENT – µA

NUMBER OF DEVICES

10

0

2

1

4

3

5

6

7

8

9

V+ = +5V
T

A

= +25°C

7

6

5

4

3

NUMBER OF DEVICES

2

1

0

–0.32

–0.4 –0.24

OFFSET – mV

V+ = +5V

= +25°C

T

A

I

VREF

0–0.08–0.16 0.160.08

TMP01

= 5µA

APPLICATIONS INFORMATION
Self-Heating Effects

In some applications the user should consider the effects of self­heating due to the power dissipated by the open-collector out­puts, which are capable of sinking 20 mA continuously. Under full
load, the TMP01 open-collector output device is dissipating

which in a surface-mount SO package accounts for a tempera­ture increase due to self-heating of

This will of course directly affect the accuracy of the TMP01
and will for example cause the device to switch the heating out­put “OFF” 2 degrees early. Alternatively, bonding the same
package to a moderate heatsink limits the self-heating effect to
approximately

which is a much more tolerable error in most systems. The
VREF and VPTAT outputs are also capable of delivering suffi­cient current to contribute heating effects and should not be
ignored.

Buffering the Voltage Reference

As mentioned before, the reference output VREF is used to gen­erate the temperature setpoint programming voltages for the
TMP01 and also is used to determine the hysteresis temperature
band by the reference load current I
buffer amplifier is typically capable of 500 µA output drive into
as much as 50 pF load (max). Exceeding this load will affect the
accuracy of the reference voltage, could cause thermal sensing
errors due to dissipation, and may induce oscillations. Selection
of a low drift buffer functioning as a voltage follower with high
input impedance will ensure optimal reference accuracy, and
will not affect the programmed hysteresis current. Amplifiers
which offer the low drift, low power consumption, and low cost
appropriate to this application include the OP295, and members
of the OP90, OP97, OP177 families, and others as shown in the
following applications circuits.

REV. C

Figure 13. Comparator Input Offset Distribution

P

= 0.6 V × .020A = 12 mW

DISS

∆T = P

∆T = P

× θJA = .012 W × 158°C/W = 1.9°C.

DISS

× θJC = .012 W × 43°C/W = 0.52°C.

DISS

. The on-board output

VREF

Figure 14. Zero Hysteresis Current Distribution

With excellent drift and noise characteristics, VREF offers a
good voltage reference for data acquisition and transducer exci­tation applications as well. Output drift is typically better than
–10 ppm/°C, with 315 nV/√

Hz (typ) noise spectral density at

1 kHz.

Preserving Accuracy Over Wide Temperature Range
Operation

The TMP01 is unique in offering both a wide-range temperature
sensor and the associated detection circuitry needed to imple­ment a complete thermostatic control function in one mono­lithic device. While the voltage reference, setpoint comparators,
and output buffer amplifiers have been carefully compensated to
maintain accuracy over the specified temperature range, the user
has an additional task in maintaining the accuracy over wide op­erating temperature ranges in this application. Since the TMP01
is both sensor and control circuit, in many applications it is pos­sible that the external components used to program and inter­face the device may be subjected to the same temperature
extremes. Thus it may be necessary to locate components in
close thermal proximity to minimize large temperature differen­tials, and to account for thermal drift errors where appropriate,
such as resistor matching tempcos, amplifier error drift, and
the like. Circuit design with the TMP01 requires a slightly dif­ferent perspective regarding the thermal behavior of electronic
components.

Thermal Response Time

The time required for a temperature sensor to settle to a speci­fied accuracy is a function of the thermal mass of the sensor,
and the thermal conductivity between the sensor and the object
being sensed. Thermal mass is often considered equivalent to
capacitance. Thermal conductivity is commonly specified using
the symbol Q, and can be thought of as the reciprocal of thermal
resistance. It is commonly specified in units of degrees per watt
of power transferred across the thermal joint. Thus, the time re­quired for the TMP01 to settle to the desired accuracy is depen­dent on the package selected, the thermal contact established in
that particular application, and the equivalent power of the heat
source. In most applications, the settling time is probably best
determined empirically.

–9–

TMP01

Switching Loads With The Open-Collector Outputs

In many temperature sensing and control applications some type
of switching is required. Whether it be to turn on a heater when
the temperature goes below a minimum value or to turn off a
motor that is overheating, the open-collector outputs Over and
Under can be used. For the majority of applications, the switches
used need to handle large currents on the order of 1 amp and
above. Because the TMP01 is accurately measuring tempera­ture, the open-collector outputs should handle less than 20 mA
of current to minimize self-heating. Clearly, the Over-temp and
Under-temp outputs should not drive the equipment directly.
Instead, an external switching device is required to handle the
large currents. Some examples of these are relays, power
MOSFETs, thyristors, IGBTs, and Darlingtons.

Figure 15 shows a variety of circuits where the TMP01 controls
a switch. The main consideration in these circuits, such as the
relay in Figure 15a, is the current required to activate the
switch.

+12V

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

8

IN4001

OR EQUIV.

7

6

5

2604-12-311

COTO

MOTOR
SHUTDOWN

Figure 15a. Reed Relay Drive

It is important to check the particular relay you choose to ensure
that the current needed to activate the coil does not exceed the
TMP01’s recommended output current of 20 mA. This is easily
determined by dividing the relay coil voltage by the specified
coil resistance. Keep in mind that the inductance of the relay
will create large voltage spikes that can damage the TMP01 out­put unless protected by a commutation diode across the coil, as
shown. The relay shown has a contact rating of 10 watts maxi­mum. If a relay capable of handling more power is desired, the
larger contacts will probably require a commensurately larger
coil, with lower coil resistance and thus higher trigger current.
As the contact power handling capability increases, so does the
current needed for the coil. In some cases an external driving
transistor should be used to remove the current load on the
TMP01 as explained in the next section.

Power FETs are popular for handling a variety of high current
DC loads. Figure 15b shows the TMP01 driving a p-channel
MOSFET transistor for a simple heater circuit. When the out­put transistor turns on, the gate of the MOSFET is pulled down
to approximately 0.6 V, turning it on. For most MOSFETs a
gate-to-source voltage or Vgs on the order of –2 V to –5 V is suf­ficient to turn the device on. Figure 15c shows a similar circuit
for turning on an n-channel MOSFET, except that now the gate
to source voltage is positive. Because of this reason an external
transistor must be used as an inverter so that the MOSFET will
turn on when the “Under Temp” output pulls down.

TEMPERATURE

1

R1

2

R2

3

R3

4

VREF

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

NC = NO CONNECT

VPTAT

TMP01

V+

8

2.4kΩ (12V)

1.2kΩ (6V)
5%

7

NC

6

NC

5

IRFR9024
OR EQUIV.

HEATING
ELEMENT

Figure 15b. Driving a P-Channel MOSFET

TEMPERATURE

R1

R2

R3

VREF

1

2

3

4

NC = NO CONNECT

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

V+

8

7

NC

6

NC

5

4.7kΩ 4.7kΩ

2N1711

HEATING
ELEMENT

IRF130

Figure 15c. Driving a N-Channel MOSFET

Isolated Gate Bipolar Transistors (IGBT) combine many of the
benefits of power MOSFETs with bipolar transistors, and are
used for a variety of high power applications. Because IGBTs
have a gate similar to MOSFETs, turning on and off the devices
is relatively simple as shown in Figure 15d. The turn on voltage
for the IGBT shown (IRGBC40S) is between 3.0 and 5.5 volts.
This part has a continuous collector current rating of 50 A and a
maximum collector to emitter voltage of 600 V, enabling it to
work in very demanding applications.

TEMPERATURE

R1

R2

R3

VREF

1

2

3

4

NC = NO CONNECT

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

V+

8

7

NC

6

NC

5

4.7kΩ 4.7kΩ

2N1711

MOTOR
CONTROL

IRGBC40S

Figure 15d. Driving an IGBT

–10–

REV. C

TMP01

Q1

Q2

2N1711

V+

R1

R2

R3

4.7kΩ

TEMPERATURE

SENSOR &

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS
GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

I

C

4.7kΩ

2N1711

The last class of high power devices discussed here are Thyris­tors, which includes SCRs and Triacs. Triacs are a useful alter­native to relays for switching ac line voltages. The 2N6073A
shown in Figure 15e is rated to handle 4A (rms). The
optoisolated MOC3011. Triac shown features excellent electri­cal isolation from the noisy ac line and complete control over
the high power Triac with only a few additional components.

TEMPERATURE

1

R1

2

R2

3

R3

4

VREF

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

NC = NO CONNECT

VPTAT

TMP01

V+ = 5V

8

300Ω

7

NC

1
2

6

5

MOC3011

34

NC

LOAD

150Ω

6

5

2N6073A

AC

Figure 15e. Controlling the 2N6073A Triac

High Current Switching

As mentioned above, internal dissipation due to large loads on
the TMP01 outputs will cause some temperature error due to
self-heating. External transistors remove the load from the
TMP01, so that virtually no power is dissipated in the internal
transistors and no self-heating occurs. Figure 16 shows a few ex­amples using external transistors. The simplest case, using a
single transistor on the output to invert the output signal is
shown in Figure 16a. When the open-collector of the TMP01
turns “ON” and pulls the output down, the external transistor
Q1’s base will be pulled low, turning off the transistor. Another
transistor can be added to reinvert the signal as shown in Figure
16b. Now, when the output of the TMP01 is pulled down, the
first transistor, Q1, turns off and its collector goes high, which
turns Q2 on, pulling its collector low. Thus, the output taken
from the collector of Q2 is identical to the output of the
TMP01. By picking a transistor that can accommodate large
amounts of current, many high power devices can be switched.

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

V+

8

4.7kΩ

7

6

5

2N1711

Q1

I

C

Figure 16a. An External Resistor Minimizes Self-Heating

Figure 16b. Second Transistor Maintains Polarity of
TMP01 Output

An example of a higher power transistor is a standard Darling­ton configuration as shown in Figure 16c. The part chosen,
TIP-110, can handle 2A continuous which is more than enough
to control many high power relays. In fact the Darlington itself
can be used as the switch, similar to MOSFETs and IGBTs.

+12V

RELAY

MOTOR
SWITCH

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

REV. C

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS

GENERATOR

Figure 16c. Darlington Transistor Can Handle Large Currents

VPTAT

V+

8

4.7kΩ

7

6

5

4.7kΩ

2N1711

TIP-110

TMP01

–11–

I

C

TMP01

Buffering the Temperature Output Pin

The VPTAT sensor output is a low impedance dc output volt­age with a 5 mV/K temperature coefficient, and is useful in a
number of measurement and control applications. In many ap­plications, this voltage needs to be transmitted to a central loca­tion for processing. The buffered VPTAT voltage output is
capable of 500 µA drive into 50 pF (max). As mentioned in the
discussion above regarding buffering circuits for the VREF out­put, it is useful to consider external amplifiers for interfacing
VPTAT to external circuitry to ensure accuracy, and to mini­mize loading which could create dissipation-induced tempera­ture sensing errors. An excellent general-purpose buffer circuit
using the OP177 is shown in Figure 17 which is capable of driv­ing over 10 mA, and will remain stable under capacitive loads of
up to 0.1 µF. Other interfacing ideas are shown below.

Differential Transmitter

In noisy industrial environments, it is difficult to send an accu­rate analog signal over a significant distance. However, by send­ing the signal differentially on a wire pair, these errors can be
significantly reduced. Since the noise will be picked up equally
on both wires, a receiver with high common-mode input rejec­tion can be used to cancel out the noise very effectively at the

8

7

6

5

V+

VPTAT

10kΩ

10kΩ

10kΩ

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS

GENERATOR

VPTAT

TMP01

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

V+

8

7

6

VPTAT

5

0.1µF

V+

V–

10kΩ

OP177

100Ω

V

OUT

C

L

Figure 17. Buffer VPTAT to Handle Difficult Loads

receiving end. Figure 18 shows two amplifiers being used to
send the signal differentially, and an excellent differential
receiver, the AMP03, which features a common-mode rejection
ratio of 95 dB at dc and very low input and drift errors.

50Ω

1/2

OP297

1/2

OP297

50Ω

V+

V–

AMP03

V

OUT

Figure 18. Send the Signal Differentially for Noise Immunity

–12–

REV. C

TMP01

4

3

7

6

8

1

2

5

AD654

VPTAT

V+

R1

R2

R3

TEMPERATURE

SENSOR &

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS

GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

R1

1.8kΩ

OSC

V+

F

OUT

C

T

0.1µF
5kΩ

V+

R2

500Ω

4 mA-20 mA Current Loop

Another, very common method of transmitting a signal over
long distances is to use a 4 mA-20 mA Loop, as shown in Fig­ure 19. An advantage of using a 4 mA-20 mA loop is that the
accuracy of a current loop is not compromised by voltage drops
across the line. One requirement of 4 mA-20 mA circuits is that
the remote end must receive all of its power from the loop,
meaning that the circuit must consume less than 4 mA. Operat­ing from +5 V, the quiescent current of the TMP01 is 500 µA
max, and the OP90s is 20 µA max, totaling less than 4 mA.
Although not shown, the open collector outputs and tempera­ture setting pins can be connected to do any local control of
switching.

The current is proportional to the voltage on the VPTAT out­put, and is calibrated to 4 mA at a temperature of –40°C, to
20 mA for +85°C. The main equation governing the operation
of this circuit gives the current as a function of VPTAT:

I

OUT

1

=

R6



VPTAT × R5



R2



VREF × R3

–

R3 + R1



1+




R5
R2












The resulting temperature coefficient of the output current is
128 µA/°C.

243kΩ

100kΩ

VREF

TMP01

4

R2

GNDV+VPTAT

2

OP90

3

R1

R3

39.2kΩ

81

5

7

6

4

+5V TO +13.2V

2N1711

high accuracy. For initial accuracy, a 10 kΩ trim potentiometer
can be included in series with R3, and the value of R3 lowered
to 95 kΩ. The potentiometer should be adjusted to produce an
output current of 12.3 mA at 25°C.

Temperature-to-Frequency Converter

Another common method of transmitting analog information is
to convert a voltage to the frequency domain. This is easily
done with any of the low cost monolithic Voltage-to-Frequency
Converters (VFCs) available, which feature a robust, open-col­lector digital output. A digital signal is very immune to noise
and voltage drops because the only important information is the
frequency. As long as the conversions between temperature and
frequency are done accurately, the temperature data can be suc­cessfully transmitted.

A simple circuit to do this combines the TMP01 with an
AD654 VFC, as shown in Figure 20. The AD654 outputs a
square wave that is proportional to the dc input voltage accord­ing to the following equation:

V

F

=

OUT

10(R1+ R2)C

IN

T

By simply connecting the VPTAT output to the input of the
AD654, the 5 mV/°C temperature coefficient gives a sensitivity
of 25 Hz/°C, centered around 7.5 kHz at 25°C. The trimming
resistor R2 is needed to calibrate the absolute accuracy of the
AD654. For more information on that part, please consult the
AD654 data sheet. Finally, the AD650 can be used to accu­rately convert the frequency back to a dc voltage on the receiv­ing end.

R6

R5

100kΩ

100Ω

4–20mA

R

L

Figure 19. 4-20 mA Current Loop

To determine the resistor values in this circuit, first note that
VREF remains constant over temperature. Thus the ratio of R5
over R2 must give a variation of I

from 4 mA to 20 mA as

OUT

VPTAT varies from 1.165 V at –40°C to 1.79 V at +85°C. The
absolute value of the resistors is not important, only the ratio.
For convenience, 100 kΩ is chosen for R5. Once R2 is calcu­lated, the value of R3 and R1 is determined by substituting
4 mA for I
values are shown in the circuit. The OP90 is chosen for this cir­cuit because of its ability to operate on a single supply and its

REV. C

and 1.165 V for VPTAT and solving. The final

OUT

Figure 20. Temperature-to-Frequency Converter

–13–

TMP01

LED

VPTAT

V+

R1

47.5kΩ

R2

4.99kΩ

R3

71.5kΩ

200Ω

TEMPERATURE

SENSOR &

VOLTAGE

REFERENCE

1

2

3

4

HYSTERESIS

GENERATOR

WINDOW

COMPARATOR

TMP01

VPTAT

VREF

7

8

5

6

OP290

8

7

6

5

V+

R1

470kΩ

3

OP290

2

V+

7

4

100kΩ

V+
2
4

1.16V TO 1.7V

6

IL300XC

1

V+

7

4

6

680pF

100Ω

IN4148

2
3

4

I

I

1

2

ISOLATION

BARRIER

6

REF43

2.5V

6

5

3

OP90

2

604kΩ

680pF

TEMPERATURE

VREF

1

R1

2

R2

3

R3

4

SENSOR &

VOLTAGE

REFERENCE

WINDOW

COMPARATOR

HYSTERESIS
GENERATOR

VPTAT

TMP01

Figure 21. Isolation Amplifier

Isolation Amplifier

In many industrial applications the sensor is located in an envi­ronment that needs to be electrically isolated from the central
processing area. Figure 21 shows a simple circuit that uses an
8-pin optoisolator (IL300XC) that can operate across a 5,000 V
barrier. IC1 (an OP290 single-supply amplifier) is used to drive
the LED connected between Pins 1 to 2. The feedback actually
comes from the photodiode connected from Pins 3 to 4. The
OP290 drives the LED such that there is enough current gener­ated in the photodiode to exactly equal the current derived from
the VPTAT voltage across the 470 kΩ resistor. On the receiving
end, an OP90 converts the current from the second photodiode
to a voltage through its feedback resistor R2. Note that the other
amplifier in the dual OP290 is used to buffer the 2.5 V reference
voltage of the TMP01 for an accurate, low drift LED bias level
without affecting the programmed hysteresis current. A REF43
(a precision 2.5 V reference) provides an accurate bias level at
the receiving end.

To understand this circuit, it helps to examine the overall equa­tion for the output voltage. First, the current (I1) in the photo­diode is set by:

2.5V – VPTAT

I1=

470 kΩ

Note that the IL300XC has a gain of 0.73 (typical) with a min
and max of 0.693 and 0.769 respectively. Since this is less than

1.0, R2 must be larger than R1 to achieve overall unity gain. To
show this the full equation is:

example, at room temperature, VPTAT = 1.49 V, so adjust R2
until V

= 1.49 V as well. Both the REF43 and the OP90

OUT

operate from a single supply, and contribute no significant error
due to drift.

In order to avoid the accuracy trim, and to reduce board space,
complete isolation amplifiers are available, such as the high
accuracy AD202.

Out-of-Range Warning

By connecting the two open collector outputs of the TMP01
together into a “wired-OR” configuration, a temperature “out­of-range” warning signal is generated. This can be useful in sen­sitive equipment calibrated to work over a limited temperature
range. R1, R2, and R3 in Figure 22 are chosen to give a tem­perature range of 10°C around room temperature (25°C). Thus,
if the temperature in the equipment falls below +15°C or rises
above +35°C, the Undertemp Output or Overtemp Output re­spectively will go low and turn the LED on. The LED may be
replaced with a simple pull-up resistor to give a logic output for
controlling the instrument, or any of the switching devices dis­cussed above can be used.



2.5 V – VPTAT



470 kΩ



V

= 2.5 V – I2R2= 2.5 V –0.7

OUT

A trim is included for R2 to correct for the initial gain accuracy
of the IL300XC. To perform this trim, simply adjust for an out­put voltage equal to VPTAT at any particular temperature. For



644 kΩ=VPTAT




–14–

Figure 22. Out-of-Range Warning

REV. C

TMP01

Translating 5 mV/K to 10 mV/°C

A useful circuit is shown in Figure 23 that translates the VPTAT
output voltage, which is calibrated in Kelvins, into an output
that can be read directly in degrees Celsius on a voltmeter
display. To accomplish this, an external amplifier is configured
as a differential amplifier. The resistors are scaled so the VREF
voltage will exactly cancel the VPTAT voltage at 0.0°C.

10pF

105kΩ

4.22kΩ

VREF

TMP01

VPTAT

1

5

4.12kΩ

100kΩ

487Ω

100kΩ

2

OP177

+15V

7

–15V

V

(10mV/°C)

6

43

OUT

= 0.0V @ T = 0.0°C)

(V

OUT

Figure 23. Translating 5 mV/K to 10 mV/°C

10pF

90.9kΩ

1.0kΩ

+15V

100kΩ

VREF

TMP01

VPTAT

1

5

6.49kΩ

121Ω

100kΩ

7

2

6

4

3

1/2

OP297

–15V

However, the gain from VPTAT to the output is two, so that
5 mV/K becomes 10 mV/°C. Thus, for a temperature of +80°C,
the output voltage is 800 mV. Circuit errors will be due prima­rily to the inaccuracies of the resistor values. Using 1% resistors
the observed error was less than 10 mV, or 1°C. The 10 pF
feedback capacitor helps to ensure against oscillations. For bet­ter accuracy, a adjustment potentiometer can be added in series
with either 100 kΩ resistor.

Translating VPTAT to the Fahrenheit Scale

A very similar circuit to the one shown in Figure 23 can be used
to translate VPTAT into an output that can be read directly in
degrees Fahrenheit, with a scaling of 10 mV/°F. Only unity gain
or less is available from the first stage differentiating circuit, so
the second amplifier provides a gain of two to complete the con­version to the Fahrenheit scale. Using the circuit in Figure 24, a
temperature of 0.0°F gives an output of 0.00 V. At room tem­perature (70°F) the output voltage is 700 mV. A –40°C to
+85°C operating range translates into –40°F to +185°F. The
errors are essentially the same as for the circuit in Figure 23.

100kΩ

100kΩ

6

5

7

1/2

V

= 0.0V @ T = 0.0°F

OUT

°

F)

(10mV/

OP297

Figure 24. Translating 5 mV/K to 10 mV/°F

REV. C

–15–

TMP01

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm).

8-Pin Epoxy DIP

0.160 (4.06)

0.115 (2.93)

0.2440 (6.20)

0.2284 (5.80)

0.0098 (0.25)

0.0040 (0.10)

0.210
(5.33)

MAX

0.022 (0.558)

0.014 (0.356)

1

0.1968 (5.00)

0.1890 (4.80)

0.0500 (1.27) BSC

8

1

0.430 (10.92)

0.348 (8.84)

0.100
(2.54)

BSC

58

4

0.0192 (0.49)

0.0138 (0.35)

5

0.280 (7.11)

0.240 (6.10)

4

0.070 (1.77)

0.045 (1.15)

0.015
(0.381) TYP

0.130
(3.30)
MIN

SEATING
PLANE

8-Pin SOIC

0.1574 (4.00)

0.1497 (3.80)

0.102 (2.59)

0.094 (2.39)

SEATING

PLANE

0°- 15°

0.0196 (0.50)

0.0099 (0.25)

0.0098 (0.25)

0.0075 (0.19)

0.325 (8.25)

0.300 (7.62)

0.015 (0.381)

0.008 (0.204)

× 45°

0.195 (4.95)

0.115 (2.93)

0.0500 (1.27)

0.0160 (0.41)

C1802b–5–7/95

0°-8°

0.335 (8.51)

0.305 (7.75)

0.370 (9.40)

0.335 (8.51)

0.185 (4.70)

0.165 (4.19)

0.040 (1.02) MAX

0.045 (1.14)

0.010 (0.25)

0.050

(1.27)

MAX

8-Pin TO-99

REFERENCE PLANE

0.750 (19.05)

0.500 (12.70)

0.250 (6.35)
MIN

0.019 (0.48)

0.016 (0.41)

0.021 (0.53)

0.016 (0.41)

BASE & SEATING PLANE

0.230
(5.84)

BSC

–16–

0.115

(2.92)

BSC

0.115
(2.92)

BSC

4

3

2

5

1

0.034 (0.86)

0.027 (0.69)

6

8

BSC

45

0.160 (4.06)

0.110 (2.79)

7

°

0.045 (1.14)

0.027 (0.69)

PRINTED IN U.S.A.

REV. C