ANALOG DEVICES LT 1074 CT Datasheet

LT1074/LT1076

Step-Down Switching

Regulator

EATU

F

■

5A On-Board Switch (LT1074)

■

100kHz Switching Frequency

■

Greatly Improved Dynamic Behavior

■

Available in Low Cost 5 and 7-Lead Packages

■

Only 8.5mA Quiescent Current

■

Programmable Current Limit

■

Operates Up to 60V Input

■

Micropower Shutdown Mode

PPLICATI

A

■

Buck Converter with Output Voltage Range of 2.5V

RE

S

O

U
S

to 50V

■

Tapped-Inductor Buck Converter with 10A Output
at 5V

■

Positive-to-Negative Converter

■

Negative Boost Converter

■

Multiple Output Buck Converter

DUESCRIPTIO

The LT1074 is a 5A (LT1076 is rated at 2A) monolithic
bipolar switching regulator which requires only a few
external parts for normal operation. The power switch, all
oscillator and control circuitry, and all current limit com­ponents, are included on the chip. The topology is a classic
positive “buck” configuration but several design innova-

tions allow this device to be used as a positive-to-negative
converter, a negative boost converter, and as a flyback
converter. The switch output is specified to swing 40V
below ground, allowing the LT1074 to drive a tapped­inductor in the buck mode with output currents up to 10A.

The LT1074 uses a true analog multiplier in the feedback
loop. This makes the device respond nearly instanta­neously to input voltage fluctuations and makes loop gain
independent of input voltage. As a result, dynamic behav­ior of the regulator is significantly improved over previous
designs.

On-chip pulse by pulse current limiting makes the LT1074
nearly bust-proof for output overloads or shorts. The input
voltage range as a buck converter is 8V to 60V, but a self­boot feature allows input voltages as low as 5V in the
inverting and boost configurations.

The LT1074 is available in low cost TO-220 or TO-3
packages with frequency pre-set at 100kHz and current
limit at 6.5A (LT1076 = 2.6A). A 7-pin TO-220 package is
also available which allows current limit to be adjusted
down to zero. In addition, full micropower shutdown can
be programmed. See Application Note 44 for design
details.

A fixed 5V output, 2A version is also available. See LT1076-5.

10V TO 40V

U

O

A

PPLICATITYPICAL

Basic Positive Buck Converter

L1**

50 H (LT1074)

µ

100 H (LT1076)

µ

MBR745*

R1

2.8k
1%

R2

2.21k
1%

LT1074

V

SW

FB

V

C

R3

2.7k

C2

0.01 F

V

IN

GND

+

†

C3
200 F

µµ µ

5V
5A

+

C1
500 F

25V

*USE MBR340 FOR LT1076
**COILTRONICS #50-2-52 (LT1074)
#100-1-52 (LT1076)

PULSE ENGINEERING, INC.
#PE-92114 (LT1074)
#PE-92102 (LT1076)

HURRICANE #HL-AK147QQ (LT1074)
#HL-AG210LL (LT1076)

†

RIPPLE CURRENT RATING I / 2

≥

OUT

LT1074 • TA01

Buck Converter Efficiency

100

90

80

EFFICIENCY (%)

70

60

50

0

LT1074

V = 12V, V = 20V

OUT IN

V = 5V, V = 15V

OUT IN

L = 50 H TYPE 52 CORE

µ

DIODE = MBR735

1234

OUTPUT LOAD CURRENT (A)

5

LT1074 • TPC27

1

6

LT1074/LT1076

5
4
3
2
1

V

GND
FB

V
V

C

FRONT VIEW

T PACKAGE, 5-LEAD T0-220

LEADS ARE FORMED STANDARD FOR
STRAIGHT LEADS, ORDER FLOW 06

IN

SW

LT1074: θJC = 2.5°C/W, θJA = 50°C/W
LT1076: θ

JC

= 4°C/W, θJA = 50°C/W

A

W

O

LUTEXI T

S

A

WUW

ARB

U
G

I

S

Input Voltage

LT1074/ LT1076 .................................................. 45V

LT1074HV/76HV.................................................. 64V

Switch Voltage with Respect to Input Voltage

LT1074/ 76 .......................................................... 64V

LT1074HV/76HV.................................................. 75V

Switch Voltage with Respect to Ground Pin (V

Negative)

SW

LT1074/76 (Note 6) ............................................. 35V

LT1074HV/76HV (Note 6).................................... 45V

Feedback Pin Voltage..................................... –2V, +10V

Shutdown Pin Voltage (Not to Exceed VIN).............. 40V

WU

/

PACKAGE

LT1076: θ

LT1076: θ

Y PACKAGE, 7-LEAD TO-220

LT1074: θ
LT1076: θ

* Assumes package is soldered to 0.5 IN2 of 1 oz. copper over internal ground

plane or over back side plane.

O

RDER I FOR ATIO

FRONT VIEW

5
4
3
2
1

Q PACKAGE

5-LEAD PLASTIC DD

= 4°C/W, θJA = 30°C/W*

JC

FRONT VIEW

7
6
5
4
3
2
1

R PACKAGE

7-LEAD PLASTIC DD

= 4°C/W, θJA = 30°C/W*

JC

FRONT VIEW

7

6

5

4

3

2

1

= 2.5°C/W, θ

JC

= 4°C/W, θ

JC

JA

= 50°C/W

JA

VIN
V

SW

GND
V

C

FB/SENSE

SHDN



V

C

FB/SENSE
GND
I

LIM

V

SW

V

IN

V

C

GND
V

SW

= 50°C/W









SHUTDOWN
FB
I

LIM

V

IN

ORDER PART

NUMBER

LT1076CQ

LT1076CR
LT1076HVCR

LT1074CY
LT1074HVCY
LT1074IY
LT1074HVIY
LT1076CY
LT1076HVCY

U

I

Pin Voltage (Forced) ............................................ 5.5V

LIM

Maximum Operating Ambient Temperature Range

LT1074C/76C, LT1074HVC/76HVC ............ 0°C to 70°C

LT1074I/76I, LT1074HVI/76HVI ............. –40°C to 85°C

LT1074M/76M, LT1074HVM/76HVM ... –55°C to 125°C

Maximum Operating Junction Temperature Range

LT1074C/76C, LT1074HVC/76HVC .......... 0°C to 125°C

LT1074I/76I, LT1074HVI/76HVI ........... –40°C to 125°C

LT1074M/76M, LT1074HVM/76HVM ... –55°C to 150°C

Maximum Storage Temperature................ –65°C to 150°C

Lead Temperature (Soldering, 10 sec) ..................... 300°C

ORDER PART

NUMBER

BOTTOM VIEW

V

C

1

2
3

4

FB

K PACKAGE, 4-LEAD TO-3 METAL CAN
LT1074: θJC = 2.5°C/W, θ

LT1076: θ

= 4°C/W, θJA = 35°C/W

JC

V

V

= 35°C/W

JA

IN

SW

CASE IS GND

LT1074CK
LT1074HVCK
LT1074MK
LT1074HVMK
LT1076CK
LT1076HVCK
LT1076MK
LT1076HVMK

LT1074CT
LT1074HVCT
LT1074IT
LT1074HVIT
LT1076CT
LT1076HVCT
LT1076IT

LECTRICAL C CHARA TERIST

E

PARAMETER CONDITIONS MIN TYP MAX UNITS

Switch “On” Voltage (Note 1) LT1074 I

2

ICS

SW

I

SW

I

SW

I

SW

LT1076 I

I

SW
SW

Tj = 25°C, VIN = 25V, unless otherwise noted.

= 1A, Tj ≥ 0°C 1.85 V
= 1A, Tj < 0°C 2.1 V
= 5A, Tj ≥ 0°C 2.3 V
= 5A, Tj < 0°C 2.5 V

= 0.5A ● 1.2 V
= 2A ● 1.7 V

LT1074/LT1076

LECTRICAL C CHARA TERIST

E

PARAMETER CONDITIONS MIN TYP MAX UNITS

Switch “Off” Leakage LT1074 V

LT1076 V

Supply Current (Note 2) V

Minimum Supply Voltage Normal Mode ● 7.3 8 V

Switch Current Limit (Note 4) LT1074 I

Maximum Duty Cycle ● 85 90 %
Switching Frequency 90 100 110 kHz

Switching Frequency Line Regulation 8V ≤ V
Error Amplifier Voltage Gain (Note 6) 1V ≤ VC ≤ 4V 2000 V/V
Error Amplifier Transconductance 3700 5000 8000 µmho
Error Amplifier Source and Sink Current Source (V

Feedback Pin Bias Current V
Reference Voltage VC = 2V ● 2.155 2.21 2.265 V
Reference Voltage Tolerance V

Reference Voltage Line Regulation 8V ≤ VIN ≤ V
VC Voltage at 0% Duty Cycle 1.5 V

Multiplier Reference Voltage 24 V
Shutdown Pin Current V

Shutdown Thresholds Switch Duty Cycle = 0 ● 2.2 2.45 2.7 V

Thermal Resistance Junction to Case LT1074 2.5 °C/W

FB

40V < VIN < 60V ● 912 mA
V

SHUT

Startup Mode (Note 3) ● 3.5 4.8 V

LT1076 I

Tj ≤ 125°C ● 85 120 kHz
Tj > 125°C ● 85 125 kHz
V

FB

Sink (V

FB

REF

All Conditions of Input Voltage, Output ● ±1 ± 2.5 %
Voltage, Temperature and Load Current

Over Temperature ● – 4 mV/°C

SH

V

SH

Fully Shut Down ● 0.1 0.3 0.5 V

LT1076 4.0 °C/W

ICS

V

V

= 2.5V, V

= 0.1V (Device Shutdown) (Note 8) ● 140 300 µA

LIM

R
R

LIM

R
R

= 0V through 2kΩ (Note 4) 20 kHz

≤ V

IN

FB

= 2.5V) 0.7 1 1.6 mA

FB

= V

REF

(Nominal) = 2.21V ±0.5 ± 1.5 %

= 5V ● 51020 µA

≤ V

THRESHOLD

Tj = 25°C, VIN = 25V, unless otherwise noted.

≤ 25V, V

IN

= V

IN

= 25V, V

IN

= V

IN

≤ 40V ● 8.5 11 mA

IN

Open ● 5.5 6.5 8.5 A

= 10k (Note 5) 4.5 A

LIM

= 7k (Note 5) 3 A

LIM

Open ● 2 2.6 3.2 A

= 10k (Note 5) 1.8 A

LIM

= 7k (Note 5) 1.2 A

LIM

(Note 7) ● 0.03 0.1 %/V

MAX

= 2V) 100 140 225 µA

(Note 7) ● 0.005 0.02 %/V

MAX

= 0 5 300 µA

SW

= 0 (Note 7) 10 500 µA

MAX, VSW

= 0 150 µA

SW

= 0 (Note 7) 250 µA

MAX, VSW

● 0.5 2 µA

(≅2.5V) ● 50 µA

The ● denotes the specifications which apply over the full operating temperature
range.

Note 1: To calculate maximum switch “on” voltage at currents between low and
high conditions, a linear interpolation may be used.

Note 2: A feedback pin voltage (VFB) of 2.5V forces the VC pin to its low clamp
level and the switch duty cycle to zero. This approximates the zero load condition
where duty cycle approaches zero.

Note 3: Total voltage from VIN pin to ground pin must be ≥ 8V after startup for
proper regulation.

Note 4: Switch frequency is internally scaled down when the feedback pin voltage
is less than 1.3V to avoid extremely short switch on times. During testing, VFB is
adjusted to give a minimum switch on time of 1µs.

R

– 1k

Note 5: I

Note 6: Switch to input voltage limitation must also be observed.
Note 7: V
Note 8: Does not include switch leakage.

LIM

≈ (LT1074), I

LIM

2k

= 40V for the LT1074/76 and 60V for the LT1074HV/76HV.

MAX

R

– 1k

LIM

≈ (LT1076).

LIM

5.5k

3

LT1074/LT1076

BLOCK

IDAGRA

W

INPUT SUPPLY

10 Aµ

SHUTDOWN*

2.21V

0.3V
+

-POWER

µ

SHUTDOWN
–

2.35V
+

–

+

ERROR

–

FB V

CURRENT

LIMIT

SHUTDOWN

A1

AMP

6V

REGULATOR

AND BIAS

I *

LIM

MULTIPLIER

X

24V (EQUIVALENT)

C

FREQ SHIFT

SYNC

V

IN

Z

ANALOG

XY

Z
Y

320 Aµ

6V TO ALL
CIRCUITRY

100kHz

OSCILLATOR

3V(p-p)

+

C1

–

4.5V

10k

PULSE WIDTH

COMPARATOR

S

R

R/S

LATCH

R

CURRENT

LIMIT
COMP

Q

LT1076

LT1074

500

Ω

15

Ω

SWITCH 

OUTPUT

(V )

SW

0.04

+

C2

–

G1

250

Ω

400

Ω

4

*AVAILABLE ON PACKAGES WITH PIN
COUNTS GREATER THAN 5.

0.1

100

Ω

SWITCH

OUTPUT (V )

SW

Ω

LT1074 • BD01

LT1074/LT1076

W

BLOCK

A switch cycle in the LT1074 is initiated by the oscillator
setting the R/S latch. The pulse that sets the latch also
locks out the switch via gate G1. The effective width of this
pulse is approximately 700ns, which sets the maximum
switch duty cycle to approximately 93% at 100kHz switch­ing frequency. The switch is turned off by comparator C1,
which resets the latch. C1 has a sawtooth waveform as one
input and the output of an analog multiplier as the other
input. The multiplier output is the product of an internal
reference voltage, and the output of the error amplifier, A1,
divided by the regulator input voltage. In standard buck
regulators, this means that the output voltage of A1
required to keep a constant regulated output is indepen­dent of regulator input voltage. This greatly improves line
transient response, and makes loop gain independent of
input voltage. The error amplifier is a transconductance
type with a GM at null of approximately 5000µmho. Slew
current going positive is 140µA, while negative slew
current is about 1.1mA. This asymmetry helps prevent
overshoot on start-up. Overall loop frequency compensa­tion is accomplished with a series RC network from VC to
ground.

Switch current is continuously monitored by C2, which
resets the R/S latch to turn the switch off if an overcurrent
condition occurs. The time required for detection and
switch turn off is approximately 600ns. So minimum
switch “on” time in current limit is 600ns. Under dead
shorted output conditions, switch duty cycle may have to
be as low as 2% to maintain control of output current. This
would require switch on time of 200ns at 100kHz switch­ing frequency, so frequency is reduced at very low output

IDAGRA

ESCRIPTIO

D

U

voltages by feeding the FB signal into the oscillator and
creating a linear frequency downshift when the FB signal
drops below 1.3V. Current trip level is set by the voltage on
the I
source. When this pin is left open, it self-clamps at about

4.5V and sets current limit at 6.5A for the LT1074 and 2.6A
for the LT1076. In the 7-pin package an external resistor
can be connected from the I
current limit. A capacitor in parallel with this resistor will
soft start the current limit. A slight offset in C2 guarantees
that when the I
C2 output will stay high and force switch duty cycle to zero.

The “Shutdown” pin is used to force switch duty cycle to
zero by pulling the I
the regulator. Threshold for the former is approximately

2.35V, and for complete shutdown, approximately 0.3V.
Total supply current in shutdown is about 150µA. A 10µA
pull-up current forces the shutdown pin high when left
open. A capacitor can be used to generate delayed start­up. A resistor divider will program “undervoltage lockout”
if the divider voltage is set at 2.35V when the input is at the
desired trip point.

The switch used in the LT1074 is a Darlington NPN (single
NPN for LT1076) driven by a saturated PNP. Special
patented circuitry is used to drive the PNP on and off very
quickly even from the saturation state. This particular
switch arrangement has no “isolation tubs” connected to
the switch output, which can therefore swing to 40V below
ground.

pin which is driven by an internal 320µA current

LIM

pin to ground to set a lower

LIM

pin is pulled to within 200mV of ground,

LIM

pin low, or to completely shut down

LIM

5

LT1074/LT1076

0

–400

CURRENT ( A)

–200

0

1

234

LT1074 • TPC03

–500

–300

5

6

78 109

–100

100

200

300

400

500

VOLTAGE (V)

µ

START OF
FREQUENCY SHIFTING

Y

PICA

LPER

F

O

R

AT

UW

CCHARA TERIST

E

C

ICS

VC Pin Characteristics VC Pin Characteristics Feedback Pin Characteristics

200

150

100

50

–50

CURRENT (mA)

–100

–150

–200

V ADJUSTED FOR

FB

I = 0 AT V = 2V

0

1234

0

CC

≈

SLOPE 400k

V 2V

≤

FB

VOLTAGE (V)

Ω

5

789

6

LT1074 • TPC01

Shutdown Pin Characteristics Shutdown Pin Characteristics I

40

30

20

V

= 50V

WITH V

DETAILS OF THIS
AREA SHOWN IN
OTHER GRAPH

20 40

10

IN

IN

30

VOLTAGE (V)

50 60 70 80

LT1074 • TPC04

10

µ

THIS POINT MOVES

0

–10

CURRENT ( A)

–20

–30

–40

0

2.0

1.5

1.0

0.5

0

–0.5

CURRENT (mA)

–1.0

–1.5

–2.0

1234

0

0

–5

–10

–15

µ

–20

–25

CURRENT ( A)

–30

–35

–40

0.5

0

V 2.5V

≥

FB

5

VOLTAGE (V)

°

= 25 CT

j

CURRENT FLOWS OUT

OF SHUTDOWN PIN

SHUTDOWN
THRESHOLD

1.0 2.0

1.5

VOLTAGE (V)

789

6

LT1074 • TPC02

2.5 3.0 3.5 4.0

LT1074 • TPC05

Pin Characteristics

LIM

100

50

0

–50

–100

µ

–150
–200

CURRENT ( A)

–250
–300
–350
–400

–1

–2

T = 25 C

j

0

°

12

VOLTAGE (V)

3

4

56

5

LT1074 • TPC06

7

8

6

Supply Current

20
18
16
14
12
10

8
6

INPUT CURRENT (mA)

4
2
0

DEVICE NOT SWITCHING

10 40 50

0

V

= 1V

C

20 30 60

INPUT VOLTAGE (V)

LT1074 • TPC11

LT1074/LT1076

JUNCTION TEMPERATURE ( C)

–50

80

FREQUENCY (kHz)

95

100

105

110

115

120

–25 0 25 50

LT1074 • TPC18

°

90

85

75

100

125

150

0

"ON" VOLTAGE (V)

3.0

2.5

2.0

1.5

1.0

0.5
123

LT1074 • TPC28

4 56

SWITCH CURRENT (A)

T = 25 C

j

°

LT1074

LT1076

UW

Y

PICA

300

250

µ

200

150

100

INPUT CURRENT ( A)

50

0

20
10

0
–10
–20
–30
–40

–50
–60

–70

CHANGE IN REFERENCE VOLTAGE (mV)

–80

LPER

F

O

R

AT

CCHARA TERIST

E

C

ICS

Reference Voltage vs

Supply Current (Shutdown) Temperature Switch “On” Voltage

2.25

2.24

2.23

2.22

2.21

2.20

VOLTAGE (V)

2.19

2.18

10 20 40

0

INPUT VOLTAGE (V)

30 50 60

LT1074 • TPC13

2.17
–25 0 25 50

–50

75 100 125 150

JUNCTION TEMPERATURE (°C)

LT1074 • TPC14

Reference Shift with Ripple Switching Frequency vs
Voltage Error Amplifier Phase and G

8k

7k

6k

µ

TRI WAVE

SQUARE

WAVE

100 120

0

20 40 60 80

PEAK-TO-PEAK RIPPLE AT FB PIN (mV)

140

160 180 200

LT1074 • TPC16

( mho)

5k

4k

3k

2k

TRANSCONDUCTANCE

1k

0

1k 100k 1M 10M

10k

FREQUENCY (Hz)

G

M

LT1074 • TPC17

M

200

150

θ

100

PHASE ( )

50

0

°

–50

–100

–150

–200

Temperature

SWITCHING FREQUENCY (kHz)

Feedback Pin Frequency Shift Current Limit vs Temperature*

160

140

120

100

80

60

–55°C

40

20

0

0.5 2.0 2.5

0

FEEDBACK PIN VOLTAGE (V)

150°C

25°C

1.0 1.5 3.0

LT1074 • TPC19

8

7

I PIN OPEN

LIM

6

5

4

3

Ω

R = 5k

2

OUTPUT CURRENT LIMIT (A)

1

*MULTIPLY CURRENTS BY 0.4 FOR LT1076

0

–50 –25 0 25 50

LIM

JUNCTION TEMPERATURE ( C)

R = 10k

LIM

75

100

°

LT1074 • TPC22

Ω

125

150

7

LT1074/LT1076

U

PI

DUESCRIPTIO

VIN PIN

S

The VIN pin is both the supply voltage for internal control
circuitry and one end of the high current switch. It is
important,

especially at low input voltages

, that this pin be
bypassed with a low ESR, and low inductance capacitor to
prevent transient steps or spikes from causing erratic
operation. At full switch current of 5A, the switching
transients at the regulator input can get very large as
shown in Figure 1. Place the input capacitor very close to
the regulator and connect it with wide traces to avoid extra
inductance. Use radial lead capacitors.

dI

L

( )( )

P

dt

STEP =

I

( )( )

ESR

SW

Figure 1. Input Capacitor Ripple

RAMP =

TI

( )( )

ON

SW

C

LT1074 • PD01

VV

()()

∆∆V

OUT

=

GND OUT

221.

To ensure good load regulation, the ground pin must be
connected directly to the proper output node, so that no
high currents flow in this path. The output divider resistor
should also be connected to this low current connection
line as shown in Figure 2.

LT1074

FB

GND

R2

HIGH CURRENT
RETURN PATH

Figure 2. Proper Ground Pin Connection

NEGATIVE OUTPUT NODE
WHERE LOAD REGULATION
WILL BE MEASURED

LT1074 • PD02

LP = Total inductance in input bypass connections
and capacitor.

“Spike” height (dI/dt • LP) is approximately 2V

inch

of lead length for LT1074 and 0.8V

per

per inch

for

LT1076.
“Step” for ESR = 0.05Ω and I
“Ramp” for C = 200µF, T

ON

= 5A is 0.25V.

SW

= 5µs, and I

SW

= 5A,

is 0.12V.

Input current on the VIN Pin in shutdown mode is the sum
of actual supply current (≈140µA, with a maximum of
300µA), and switch leakage current. Consult factory for
special testing if shutdown mode input current is critical.

GROUND PIN

It might seem unusual to describe a ground pin, but in the
case of regulators, the ground pin must be connected
properly to ensure good load regulation. The internal
reference voltage is referenced to the ground pin; so any
error in ground pin voltage will be multiplied at the output;

FEEDBACK PIN

The feedback pin is the inverting input of an error amplifier
which controls the regulator output by adjusting duty
cycle. The non-inverting input is internally connected to a
trimmed 2.21V reference. Input bias current is typically

0.5µA when the error amplifier is balanced (I

OUT

= 0). The
error amplifier has asymmetrical GM for large input sig­nals to reduce startup overshoot. This makes the amplifier
more sensitive to large ripple voltages at the feedback pin.
100mVp-p ripple at the feedback pin will create a 14mV
offset in the amplifier, equivalent to a 0.7% output voltage
shift. To avoid output errors, output ripple (P-P) should be
less than 4% of DC output voltage at the point where the
output divider is connected.

See the “Error Amplifier” section for more details.

Frequency Shifting at the Feedback Pin

The error amplifier feedback pin (FB) is used to downshift
the oscillator frequency when the regulator output voltage

8

U

VOLTAGE (V)

0

–40

CURRENT ( A)

1.0 2.0

LT1074 • TPC05

1.5

2.5 3.0 3.5 4.0

–35

–30

–25

–20

–15

–10

–5

0

SHUTDOWN
THRESHOLD

CURRENT FLOWS OUT

OF SHUTDOWN PIN

µ

= 25 CT

°

j

0.5

PI

DUESCRIPTIO

LT1074/LT1076

S

is low. This is done to guarantee that output short circuit
current is well controlled even when switch duty cycle
must be extremely low. Theoretical switch “on” time for a
buck converter in continuous mode is;

VV

+

t

ON

OUT D

=

Vf

•

IN

VD = Catch diode forward voltage ( ≈ 0.5V)
f = Switching frequency

At f = 100kHz, tON must drop to 0.2µs when V
and the output is shorted (V

= 0V). In current limit,

OUT

IN

= 25V

the LT1074 can reduce tON to a minimum value of ≈

0.6µs, much too long to control current correctly for
V

= 0. To correct this problem, switching frequency

OUT

is lowered from 100kHz to 20kHz as the FB pin drops
from 1.3V to 0.5V. This is accomplished by the circuitry
shown in Figure 3.

+

ERROR

V

AMPLIFIER

C

–

+2V

2.21V

TO
OSCILLATOR

Q1

R3
3k

FB

R1

R2

2.21k

V

OUT

EXTERNAL
DIVIDER

SHUTDOWN PIN

The shutdown pin is used for undervoltage lockout, mi­cropower shutdown, soft start, delayed start, or as a
general purpose on/off control of the regulator output. It
controls switching action by pulling the I

pin low, which

LIM

forces the switch to a continuous “off” state. Full
micropower shutdown is initiated when the shutdown pin
drops below 0.3V.

The V/I characteristics of the shutdown pin are shown in
Figure 4. For voltages between 2.5V and ≈VIN, a current
of 10µA flows

out

of the shutdown pin. This current

increases to ≈25µA as the shutdown pin moves through
the 2.35V threshold. The current increases further to ≈
30µA at the 0.3V threshold, then drops to ≈15µA as the
shutdown voltage falls below 0.3V. The 10µA current
source is included to pull the shutdown pin to its high or
default state when left open. It also provides a convenient
pullup for delayed start applications with a capacitor on
the shutdown pin.

When activated, the typical collector current of Q1 in
Figure 5, is ≈ 2mA. A soft start capacitor on the I

LIM

pin

will delay regulator shutdown in response to C1, by
≈(5V)(C

)/2mA. Soft start after full micropower shut-

LIM

down is ensured by coupling C2 to Q1.

LT1074 • PD03

Figure 3. Frequency Shifting

Q1 is off when the output is regulating (VFB = 2.21V). As
the output is pulled down by an overload, VFB will eventu­ally reach 1.3V, turning on Q1. As the output continues to
drop, Q1 current increases proportionately and lowers the
frequency of the oscillator. Frequency shifting starts when
the output is ≈ 60% of normal value, and is down to its
minimum value of ≅ 20kHz when the output is ≅ 20% of
normal value. The rate at which frequency is shifted is
determined by both the internal 3k resistor R3 and the
external divider resistors. For this reason, R2 should not
be increased to more than 4kΩ, if the LT1074 will be
subjected to the simultaneous conditions of high input
voltage and output short circuit.

Figure 4. Shutdown Pin Characteristics

9

LT1074/LT1076

Trip Po V V

R

R

TP

int .== +











235 1

1
2

R

VVR

VV

R
R

SH

UTP SH

3

08 1

1

1

2

=

()()

+











–.

–

Rk

VV

V

k

Rk

1232

15 2 35

235

12 5

3

235 08 125

20 2 35 1

12 5
232

39

=

()

()

=

=

()()

+











=

.

–.
.

.

.–. .

–.

.

.

.

U

PI

DUESCRIPTIO

10 Aµ

SHUTDOWN

PIN

–

2.3V

+

–

0.3V

+

Figure 5. Shutdown Circuitry

S

V

IN

300 Aµ

I

LIM

Q1

TO TOTAL
REGULATOR
SHUTDOWN

PIN

6V

LT1074 • PD07

EXTERNAL
C

LIM

C1

C2

Undervoltage Lockout

Undervoltage lockout point is set by R1 and R2 in Figure

6. To avoid errors due to the 10µA shutdown pin current,
R2 is usually set at 5k, and R1 is found from:

–

VV

()

RR

12=

TP SH

V

SH

Hysteresis in undervoltage lockout may be accomplished
by connecting a resistor (R3) from the I

pin to the

LIM

shutdown pin as shown in Figure 7. D1 prevents the
shutdown divider from altering current limit.

V

R1

D1*

R3

R2

*1N4148

Figure 7. Adding Hysteresis

IN

SHUT

LT1074

I

LIM

OPTIONAL CURRENT
LIMIT RESISTOR

LT1074 • PD09

If R3 is added, the lower trip point (VIN descending) will be
the same. The upper trip point (V



R

R

1

VV

=++

UTP SH

1





R

1

R

2

3

) will be;

UTP



08

–.







R

1

V



R

3








If R1 and R2 are chosen, R3 is given by

V

= Desired undervoltage lockout voltage.

TP

V

= Threshold for lockout on the shutdown

SH

pin = 2.45V.

If quiescent supply current is critical, R2 may be increased
up to 15kΩ, but the denominator in the formula for R2
should replace VSH with VSH – (10µA)(R2).

R1

R2

5k

Figure 6. Undervoltage Lockout

10

V

SHUT

LT1074

GND

IN

LT1074 • PD08

Example: An undervoltage lockout is required such that
the output will not start until VIN = 20V, but will continue
to operate until VIN drops to 15V. Let R2 = 2.32k.

U

R

15 0 44 9k

05 9k 1k 15

FB

=

−

()

()

−

()

−

..

..

Ω

PI

DUESCRIPTIO

LT1074/LT1076

S

I

PIN

LIM

The I

pin is used to reduce current limit below the

LIM

preset value of 6.5A. The equivalent circuit for this pin is
shown in Figure 8.

When I

TO LIMIT
CIRCUIT

R1
8K

is left open, the voltage at Q1 base clamps at 5V

LIM

V

Q1

I

LIM

Figure 8. I

IN

320 Aµ

D1

LIM

D2

4.3V

D3
6V

LT1047 • PD12

Pin Circuit

through D2. Internal current limit is determined by the
current through Q1. If an external resistor is connected
between I

and ground, the voltage at Q1 base can be

LIM

reduced for lower current limit. The resistor will have a
voltage across it equal to (320µA) (R), limited to ≈ 5V
when clamped by D2. Resistance required for a given
current limit is

R

= I

(2kΩ) + 1kΩ (LT1074)

LIM

= I

(5.5kΩ) + 1kΩ (LT1076)

LIM

R

LIM

LIM

As an example, a 3A current limit would require 3A (2k) +
1k = 7kΩ for the LT1074. The accuracy of these formulas
is ±25% for 2A ≤ I
(LT1076), so I

peak

switch current required.

≤ 5A (LT1074) and 0.7A ≤ I

LIM

should be set at least 25% above the

LIM

LIM

≤ 1.8A

Foldback current limiting can be easily implemented by
adding a resistor from the output to the I

pin as shown

LIM

in Figure 9. This allows full desired current limit (with or
without R

) when the output is regulating, but reduces

LIM

current limit under short circuit conditions. A typical value
for RFB is 5kΩ, but this may be adjusted up or down to set
the amount of foldback. D2 prevents the output voltage

from forcing current back into the I
value for RFB, first calculate R

I 0.44* R

−

()()

R

FB

SC L

=

05.* – –

R1

LSC

()

LIM

Ω

kI

pin. To calculate a

LIM

, then R

FB;

Rink

()

Ω

L

*Change 0.44 to 0.16, and 0.5 to 0.18 for LT1076.

Example: I

= 4A, ISC = 1.5A, R

LIM

= (4)(2k) + 1k = 9k

LIM

V

OUT

LT1074

I

LIM

R

LIM

Figure 9. Foldback Current Limit

FB

R

D2

FB

1N4148

LT1074 • PD13

ERROR AMPLIFIER

The error amplifier in Figure 10 is a single stage design
with added inverters to allow the output to swing above
and below the common mode input voltage. One side of
the amplifier is tied to a trimmed internal reference voltage
of 2.21V. The other input is brought out as the FB (feed­back) pin. This amplifier has a GM (voltage “in” to current
“out”) transfer function of ≈5000µmho. Voltage gain is
determined by multiplying GM times the total equivalent
output loading, consisting of the output resistance of Q4
and Q6 in parallel with the series RC external frequency
compensation network. At DC, the external RC is ignored,
and with a parallel output impedance for Q4 and Q6 of
400kΩ, voltage gain is ≈ 2000. At frequencies above a few
hertz, voltage gain is determined by the external compen­sation, RC and CC.

11

LT1074/LT1076

U

PI

DUESCRIPTIO

S

5.8V

Q4

90 Aµ

Q3

50 Aµ

FB

50 Aµ

D2

Ω

300

Figure 10. Error Amplifier

A

=

V

AG

=•

Vm

G

2fC

••

π

R at high frequencies

m

C

C

Q2

Q1

X1.8

2.21V

140 Aµ

ALL CURRENTS SHOWN ARE AT NULL CONDITION

at midfrequencies

Phase shift from the FB pin to the VC pin is 90° at mid­frequencies where the external CC is controlling gain, then
drops back to 0° (actually 180° since FB is an inverting
input) when the reactance of CC is small compared to RC.
The low frequency “pole” where the reactance of CC is
equal to the output impedance of Q4 and Q6 (rO), is

f

POLE

Although f

=≈

POLE

1
rC

2

π

••

0

rk

400

0

Ω

varies as much as 3:1 due to rO variations,
mid-frequency gain is dependent only on GM, which is
specified much tighter on the data sheet. The higher
frequency “zero” is determined solely by RC and CC.

µ

90 A

D1

The error amplifier has

V

C

90 Aµ

Q6

EXTERNAL
FREQUENCY
COMPENSATION

R

C

C

C

LT1074 • PD11

asymmetrical

peak output current.
Q3 and Q4 current mirrors are unity gain, but the Q6 mirror
has a gain of 1.8 at output null and a gain of 8 when the FB
pin is high (Q1 current = 0). This results in a maximum
positive output current of 140µA and a maximum negative
(sink) output current of ≅ 1.1mA. The asymmetry is
deliberate —it results in much less regulator output over­shoot during rapid start-up or following the release of an
output overload. Amplifier offset is kept low by area scaling
Q1 and Q2 at 1.8:1.

Amplifier swing is limited by the internal 5.8V supply for
positive outputs and by D1 and D2 when the output goes
low. Low clamp voltage is approximately one diode drop
(≈ 0.7V – 2mV/°C).

Note that both the FB pin and the VC pin have other internal
connections. Refer to the frequency shifting and
sychronizing discussions.

f

ZERO

12

=

••

RC

2π

1

CC

LT1074/LT1076

U

O

PPLICATITYPICAL

SA

Tapped-Inductor Buck Converter

V

IN

†

20V - 35V

+

* = 1% FILM RESISTORS
D1 = MOTOROLA-MBR745
C1 = NICHICON-UPL1C221MRH6
C2 = NICHICON-UPL1A102MRH6
L1 = COILTRONICS-CTX25-5-52

L2

µ

5 H

+

C1
4400 F

µ

(2 EA
2200 F,

µ

16V)

+

C4
390 F
16V

LT1074 • TA02

V

IN

LT1074HV

GND

C3
200 F

µ

50V

*PULSE ENGINEERING #PE–65282

**MOTOROLA MBR2030CTL

†

IF INPUT VOLTAGE IS BELOW 20V,
MAXIMUM OUTPUT CURRENT WILL BE REDUCED. SEE AN44

V

SW

FB

V

C

R3
1k

C2

0.2 F

µ

31

D2
35V
5W

D3
1N5819

L1*

D1**

0.01 F

R1

2.8k

R2

2.21k

µ

Positive-to-Negative Converter with 5V Output

V

IN

4.5V to
40V

+

LT1074

GND

✝

LOWER REVERSE VOLTAGE RATING MAY BE USED FOR LOWER INPUT VOLTAGES.
LOWER CURRENT RATING IS ALLOWED FOR LOWER OUTPUT CURRENT. SEE AN44.

✝✝

LOWER CURRENT RATING MAY BE USED FOR LOWER OUTPUT CURRENT. SEE AN44.
R1, R2, AND C4 ARE USED FOR LOOP FREQUENCY COMPENSATION WITH LOW INPUT VOLTAGE,

BUT R1 AND R2 MUST BE INCLUDED IN THE CALCULATION FOR OUTPUT VOLTAGE DIVIDER VALUES.
FOR HIGHER OUTPUT VOLTAGES, INCREASE R1, R2, AND R3 PROPORTIONATELY.
FOR INPUT VOLTAGE > 10V, R1, R2, AND C4 CAN BE ELIMINATED, AND COMPENSATION IS
DONE TOTALLY ON THE V PIN.
R3 = –2.37 (K )

V
R1 = (R3) (1.86)
R2 = (R3) (3.65)

*****

MAXIMUM OUTPUT CURRENT OF 1A IS DETERMINED BY MINIMUM INPUT
VOLTAGE OF 4.5V. HIGHER MINIMUM INPUT VOLTAGE WILL ALLOW MUCH HIGHER
OUTPUT CURRENTS. SEE AN44.

V

OUT

+

C1

µ

220 F
50V

L1
25 H

µ

✝✝

5A

IN

V

SW

V

FB

V

C

C3

0.1 Fµ

C

Ω

R1**

5.1k

R2**
10k

✝

D1
MBR745

C4**

0.01 Fµ

R3*

2.74k

R4

1.82k*

+

C2

µ

1000 F
10V

OPTIONAL FILTER

5 H

–5V,1A***

LT1074 • TA03

V

OUT

5V, 10A

µ

µ

†

–

200 F

µ

10V

+

13

LT1074/LT1076

DD7 0693

0.022 ± 0.005

(0.559 ± 0.127)

0.105 ± 0.008

(2.667 ± 0.203)

0.004

+0.008
–0.004

()

0.102

+0.203
–0.102

0.050 ± 0.012

(1.270 ± 0.305)

0.059

(1.499)

TYP

0.050 ± 0.008

(1.270 ± 0.203)

0.175 ± 0.008

(4.445 ± 0.203)

0.060

(1.524)



0.401 ± 0.015

(10.185 ± 0.381)

15° TYP

0.030 ± 0.008

(0.762 ± 0.203)

0.331

+0.012
–0.020

()

8.407

+0.305
–0.508

0.143

+0.012
–0.020

()

3.632

+0.305
–0.508

0.050 ± 0.010

(1.270 ± 0.254)

PPLICATITYPICAL

O

U
SA

Negative Boost Converter

200 F

µ

15V

–V

IN

–5 TO –15V

*MBR735

** I (MAX) = 1A-3A DEPENDING ON INPUT VOLTAGE.

OUT

SEE AN44

PACKAGEDESCRIPTI

+

R3
750

C2
1nF

100pF

L1

µ

25 H

Ω

V

IN

FB

LT1074

V

SW

GND

+

C3

V

C

0.01 F

µ

D1*

R1

12.7k

R2

2.21k

+

5 Hµ

OPTIONAL OUTPUT FILTER

U

O

Dimensions in inches (milimeters) unless otherwise noted.

+

100 F

µ

C1
1000 F
25V

LT1074 • TA04

µ

V **

OUT

–15V

Q Package, 5-Lead PLASTIC DD

(1.702 ± 0.254)

0.032 ± 0.008

0.401 ± 0.015

(10.185 ± 0.381)

15° TYP

0.067 ± 0.010

0.060

(1.524)

+0.012

0.331
–0.020

+0.305

8.407

()

–0.508

+0.012

0.143
–0.020

+0.305

3.632

()

–0.508

(0.813 ± 0.203)

14

(4.445 ± 0.203)

0.175 ± 0.008

0.059

(1.499)

TYP

0.022 ± 0.005

(0.559 ± 0.127)

R Package, 7-Lead PLASTIC DD

0.050 ± 0.008

(1.270 ± 0.203)

+0.008

0.004
–0.004

+0.203

0.102

()

–0.102

0.105 ± 0.008

(2.667 ± 0.203)

0.050 ± 0.012

(1.270 ± 0.305)

DD5 0693

PACKAGEDESCRIPTI

0.320 – 0.350
(8.13 – 8.89)

0.420 – 0.480

(10.67 – 12.19)

0.380 – 0.420

(9.652 – 10.668)

0.139 – 0.153

(3.531 – 3.886)

U

O

Dimensions in inches (milimeters) unless otherwise noted.

K Package, 4-Lead TO-3 Metal Can

0.169 – 0.185

(4.293 – 4.699)

1.177 – 1.197

(29.90 – 30.40)

72°

18°

0.470 TP
P.C.D.

0.760 – 0.775

(19.30 – 19.69)

DIA

0.060 – 0.135

(1.524 – 3.429)

0.038 – 0.043

(0.965 – 1.09)

T Package, 5-Lead TO-220

0.079 – 0.135

(2.007 – 3.429)

0.655 – 0.675

(16.64 – 19.05)

0.151 – 0.161

0.167 – 0.177
(4.24 – 4.49)

R TYP

0.495 – 0.525

(12.57 – 13.34)

R

0.035 – 0.055

(0.889 – 1.397)

LT1074/LT1076

(3.84 – 4.09)

DIA 2 PLC

K4 0592

0.560 – 0.650

(14.224 – 16.510)

0.057 – 0.077

(1.448 – 1.956)

0.390 – 0.410
(9.91 – 10.41)

0.103 – 0.113
(2.62 – 2.87)

0.460 – 0.500

0.866 – 0.913

(21.996 – 23.190)

0.028 – 0.035

(0.711 – 0.889)

0.970 – 1.050

(24.64 – 26.67)

(11.68 – 12.70)

0.015 – 0.025

(0.381 – 0.635)

0.210 – 0.240

(5.334 – 6.096)

0.304 – 0.380

(7.722 – 9.652)

Y Package, 7-Lead Molded TO-220

0.147 – 0.155
(3.73 – 3.94)

0.235 – 0.258
(5.97 – 6.55)

DIA

0.169 – 0.185
(4.29 – 4.70)

0.560 – 0.590

(14.22 – 14.99)

0.152 – 0.202
(3.86 – 5.13)

0.620 ± 0.020

(15.75 ± 0.508)



0.700 – 0.728

(17.780 – 18.491)

0.055 – 0.090

(1.397 – 2.286)

0.079 – 0.115 

(2.007 – 2.921)

0.149 – 0.230

(3.785 – 5.842)

0.045 – 0.055
(1.14 – 1.40)

0.620

(15.75)

TYP



0.700 – 0.728

(17.78 – 18.49)



T5 (FORMED) 0993



0.260 – 0.320
(6.60 – 8.13)

0.026 – 0.036
(0.66 – 0.91)

0.045 – 0.055
(1.14 – 1.40)

Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-

0.016 – 0.022
(0.41 – 0.56)

0.135 – 0.165
(3.43 – 4.19)

0.095 – 0.115 
(2.41 – 2.92)

0.155 – 0.195
(3.94 – 4.95)

tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.

Y7 0893

15

LT1074/LT1076

U.S. Area Sales Offices

NORTHEAST REGION
Linear Technology Corporation

One Oxford Valley
2300 E. Lincoln Hwy.,Suite 306
Langhorne, PA 19047
Phone: [phone]
FAX: [phone]

Linear Technology Corporation

266 Lowell St., Suite B-8
Wilmington, MA 01887
Phone: [phone]
FAX: [phone]

FRANCE
Linear Technology S.A.R.L.

Immeuble "Le Quartz"
58 Chemin de la Justice
92290 Chatenay Malabry
France
Phone: [phone]
FAX: [phone]

SOUTHEAST REGION
Linear Technology Corporation

17060 Dallas Parkway
Suite 208
Dallas, TX 75248
Phone: [phone]
FAX: [phone]

CENTRAL REGION
Linear Technology Corporation

Chesapeake Square
229 Mitchell Court, Suite A-25
Addison, IL 60101
Phone: [phone]
FAX: [phone]

International Sales Offices

KOREA
Linear Technology Korea Branch

Namsong Building, #505
Itaewon-Dong 260-199
Yongsan-Ku, Seoul
Korea
Phone: [phone]
FAX: [phone]

SOUTHWEST REGION
Linear Technology Corporation

22141 Ventura Blvd.
Suite 206
Woodland Hills, CA 91364
Phone: [phone]
FAX: [phone]

NORTHWEST REGION
Linear Technology Corporation

782 Sycamore Dr.
Milpitas, CA 95035
Phone: [phone]
FAX: [phone]

TAIWAN
Linear Technology Corporation

Rm. 801, No. 46, Sec. 2
Chung Shan N. Rd.
Taipei, Taiwan, R.O.C.
Phone: [phone]
FAX: [phone]

GERMANY
Linear Techonolgy GmbH

Untere Hauptstr. 9
D-85386 Eching
Germany
Phone: [phone]
FAX: [phone]

JAPAN
Linear Technology KK

5F YZ Bldg.
4-4-12 Iidabashi, Chiyoda-Ku
Tokyo, 102 Japan
Phone: [phone]
FAX: [phone]

SINGAPORE
Linear Technology Pte. Ltd.

101 Boon Keng Road
#02-15 Kallang Ind. Estates
Singapore 1233
Phone: [phone]
FAX: [phone]

World Headquarters

Linear Technology Corporation

1630 McCarthy Blvd.
Milpitas, CA 95035-7487
Phone: [phone]
FAX: [phone]

UNITED KINGDOM
Linear Technology (UK) Ltd.

The Coliseum, Riverside Way
Camberley, Surrey GU15 3YL
United Kingdom
Phone: [phone]
FAX: [phone]

0294

16

Linear Technology Corporation

1630 McCarthy Blvd., Milpitas, CA 95035-7487

[phone]

●

FAX

: [phone]

●

TELEX

: 499-3977

BA/GP 0494 2K REV B • PRINTED IN USA

 LINEAR TECHNOLOGY CORPORATION 1994