14-Bit, 40 MSPS
FEATURES
Integrated dual 14-bit ADC
Single 3 V supply operation: 2.7 V to 3.6 V
Differential input with 500 MHz, 3 dB bandwidth
Flexible analog input: 1 V p-p to 2 V p-p range
Offset binary or twos complement data format
Clock duty cycle stabilizer
Output datamux option
APPLICATIONS
Ultrasound equipment
Direct conversion or IF sampling receivers
WB-CDMA, CDMA2000, WiMAX
Battery-powered instruments
Hand-held scopemeters
Low cost digital oscilloscopes
GENERAL DESCRIPTION
The ADW12001 is a dual, 3 V, 14-bit, 40 MSPS analog-todigital converter (ADC). It features dual high performance
sample-and-hold amplifiers (SHAs) and an integrated voltage
reference. The ADW12001 uses a multistage differential
pipelined architecture with output error correction logic to
provide 14-bit accuracy and to guarantee no missing codes
over the full operating temperature. The wide bandwidth
differential SHA allows for a variety of user-selectable input
ranges and offsets, including single-ended applications. It is
suitable for various applications, including multiplexed systems
that switch full-scale voltage levels in successive channels and
for sampling inputs at frequencies well beyond the Nyquist rate.
Dual single-ended clock inputs are used to control all internal
conversion cycles. A duty cycle stabilizer is available and can
compensate for wide variations in the clock duty cycle, allowing
the converter to maintain excellent performance. The digital
output data is presented in either straight binary or twos complement format. Out-of-range signals indicate an overflow
condition, which can be used with the most significant bit to
determine low or high overflow.
Dual Analog-to-Digital Converter
ADW12001
FUNCTIONAL BLOCK DIAGRAM
AVDD AGND
VIN+_A
VIN–_A
REFT_A
REFB_A
VREF
SENSE
AGND
REFT_B
REFB_B
VIN+_B
VIN–_B
SHA
0.5V
SHA
ADW12001
14
ADC
BUFFERS
DUTY CYCLE
STABILIZER
CONTROL
ADC
BUFFERS
DRVDD DRGND
Figure 1.
OUTPUT
MUX/
CLOCK
MODE
OUTPUT
MUX/
Fabricated on an advanced CMOS process, the ADW12001
is available in a Pb-free, space saving, 64-lead LFCSP and
is specified over the industrial temperature range (−40°C
to +115°C).
PRODUCT HIGHLIGHTS
1. Pin compatible with the AD9238, 12-bit 40 MSPS ADC.
2. Low power consumption: 40 MSPS = 330 mW.
3. Typical channel isolation of 85 dB @ f
4. The clock duty cycle stabilizer maintains performance over
a wide range of clock duty cycles.
5. Multiplexed data output option enables single port
operation from either Data Port A or Data Port B.
IN
OTR_A
14
D13_A TO D0_A
OEB_A
MUX_SELECT
CLK_A
CLK_B
DCS
SHARED_REF
PWDN_A
PWDN_B
DFS
OTR_B
1414
D13_B TO D0_B
OEB_B
= 10 MHz.
07737-001
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700 www.analog.com
Fax: 781.461.3113 ©2009 Analog Devices, Inc. All rights reserved.
ADW12001
TABLE OF CONTENTS
Features .............................................................................................. 1
Applications ....................................................................................... 1
General Description ......................................................................... 1
Functional Block Diagram .............................................................. 1
Product Highlights ........................................................................... 1
Table of Contents .............................................................................. 2
Revision History ........................................................................... 2
Specifications ..................................................................................... 3
DC Specifications ......................................................................... 3
AC Specifications .......................................................................... 5
Digital Specifications ................................................................... 6
Switching Specifications .............................................................. 7
Absolute Maximum Ratings ............................................................ 8
Thermal Resistance ...................................................................... 8
ESD Caution .................................................................................. 8
Pin Configuration and Function Descriptions ..............................9
Ter minology .................................................................................... 11
Typical Performance Characteristics ........................................... 12
Equivalent Circuits ......................................................................... 15
Theory of Operation ...................................................................... 16
Analog Input ............................................................................... 16
Clock Input and Considerations .............................................. 17
Power Dissipation and Standby Mode .................................... 18
Digital Outputs ........................................................................... 18
Timing ......................................................................................... 18
Data Format ................................................................................ 19
Voltage Reference ....................................................................... 19
Thermal Considerations ............................................................ 20
Outline Dimensions ....................................................................... 21
Ordering Guide .......................................................................... 21
REVISION HISTORY
1/09—Revision 0: Initial Version
Rev. 0 | Page 2 of 24
ADW12001
SPECIFICATIONS
DC SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, maximum sample rate, CLK_A = CLK_B; AIN = −0.5 dBFS differential input, 1.0 V internal reference,
T
to T
MIN
Table 1.
Parameter Te mp Min Typ Max Min Typ Max Unit
RESOLUTION Full 14 14 Bits
ACCURACY
No Missing Codes Guaranteed Full 14 14 Bits
Offset Error 25°C ±0.2 ±1.3 ±0.2 ±1.3 % FSR
Gain Error
Differential Nonlinearity (DNL)
25°C ±0.6 ±1.0 ±0.65 ±1.0 LSB
Integral Nonlinearity (INL)
25°C ±2.3 ±4.5 ±2.4 ±4.5 LSB
TEMPERATURE DRIFT
Offset Error Full ±2 ±2 ppm/°C
Gain Error
INTERNAL VOLTAGE REFERENCE
Output Voltage Error (1 V Mode) Full ±5 ±35 ±5 ±35 mV
Load Regulation @ 1.0 mA Full 0.8 0.8 mV
Output Voltage Error (0.5 V Mode) Full ±2.5 ±2.5 mV
Load Regulation @ 0.5 mA Full 0.1 0.1 mV
INPUT REFERRED NOISE
Input Span = 1 V 25°C 2.1 2.1 LSB rms
Input Span = 2.0 V 25°C 1.05 1.05 LSB rms
ANALOG INPUT
Input Span = 1.0 V Full 1 1 V p-p
Input Span = 2.0 V Full 2 2 V p-p
Input Capacitance
REFERENCE INPUT RESISTANCE Full 7 7 kΩ
POWER SUPPLIES
Supply Voltages
Supply Current
PSRR Full ±0.01 ±0.01 % FSR
POWER CONSUMPTION
DC Input
Sine Wave Input
Standby Power
, DCS enabled, unless otherwise noted.
MAX
25°C 115°C
1
Full ±0.3 ±2.4 ±0.5 ±2.5 % FSR
1
Full ±12 ±12 ppm/°C
3
Full 7 7 pF
2
Full ±0.65 ±0.7 LSB
2
Full ±2.7 ±2.8 LSB
AVDD Full 2.7 3.0 3.6 2.7 3.0 3.6 V
DRVDD Full 2.25 3.0 3.6 2.25 3.0 3.6 V
2
IAVDD
Full 110 115 mA
IDRVDD
2
Full 10 11 mA
4
Full 330 350 mW
2
Full 360 400 315 412 mW
5
Full 7.0 7.0 mW
Rev. 0 | Page 3 of 24
ADW12001
25°C 115°C
Parameter Te mp Min Typ Max Min Typ Max Unit
MATCHING CHARACTERISTICS
Offset Error (Nonshared Reference Mode) 25°C ±0.19 ±1.56 ±0.25 ±1.74 % FSR
Offset Error (Shared Reference Mode) 25°C ±0.19 ±1.56 ±0.25 ±1.74 % FSR
Gain Error (Nonshared Reference Mode) 25°C ±0.07 ±1.43 ±0.07 ±1.47 % FSR
Gain Error (Shared Reference Mode) 25°C ±0.01 ±0.06 ±0.01 ±0.10 % FSR
1
Gain error and gain temperature coefficients are based on the ADC only (with a fixed 1.0 V external reference).
2
Measured at the maximum clock rate with a low frequency sine wave input and approximately 5 pF loading on each output bit.
3
Input capacitance refers to the effective capacitance between one differential input pin and AVSS. Refer to Figure for the equivalent analog input structure. 18
4
Measured with dc input at the maximum clock rate.
5
Standby power is measured with the CLK_A and CLK_B pins inactive (that is, set to AVDD or AGND).
Rev. 0 | Page 4 of 24
ADW12001
AC SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, maximum sample rate, CLK_A = CLK_B; AIN = −0.5 dBFS differential input, 1.0 V external reference,
T
to T
MIN
Table 2.
Parameter Te mp Min Ty p Max Min Ty p Max Unit
SIGNAL-TO-NOISE RATIO (SNR)
fIN = 2.4 MHz Full
25°C 72.8 73.4 70.0 72.5 dB
fIN = 19.6 MHz Full
25°C 72.3 72.9 70.5 71.8 dB
SIGNAL-TO-NOISE AND DISTORTION RATIO (SINAD)
fIN = 2.4 MHz Full dB
25°C 72.0 73.0 69.5 72.0 dB
fIN = 19.6 MHz Full dB
25°C 71.0 72.3 69.5 71.5 dB
EFFECTIVE NUMBER OF BITS (ENOB)
fIN = 2.4 MHz Full 11.6 Bits
25°C 11.7 11.8 Bits
WORST HARMONIC (SECOND or THIRD)
fIN = 2.4 MHz Full 86.0 dBc
25°C 77.5 86.0 77 dBc
fIN = 19.6 MHz Full dBc
25°C 76.0 84.0 dBc
85 dBc
75 dBc
WORST OTHER SPUR (NONSECOND or THIRD)
fIN = 2.4 MHz Full 88.0 84 dBc
25°C 83.5 89.0 dBc
fIN = 19.6 MHz Full 88.0 85 dBc
25°C 82.6 88.5 dBc
SPURIOUS-FREE DYNAMIC RANGE (SFDR)
fIN = 2.4 MHz Full 85.0 84 92 dBc
25°C 77.5 86.0 77.5 86.0 dBc
fIN = 19.6 MHz Full 83.0 85 90 dBc
25°C 76.0 84.0 dBc
CROSSTALK Full −85.0 −85.0 dB
, DCS enabled, unless otherwise noted.
MAX
25°C 115°C
Rev. 0 | Page 5 of 24
ADW12001
DIGITAL SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, maximum sample rate, CLK_A = CLK_B; AIN = −0.5 dBFS differential input, 1.0 V internal reference,
T
to T
MIN
Table 3.
Parameter Tem p Min Typ Max Unit
LOGIC INPUTS
High Level Input Voltage Full 2.0 V
Low Level Input Voltage Full 0.8 V
High Level Input Current Full −10 +10 μA
Low Level Input Current Full −10 +10 μA
Input Capacitance Full 2 pF
LOGIC OUTPUTS
High Level Output Voltage Full DRVDD − 0.05 V
Low Level Output Voltage Full 0.05 V
1
Output voltage levels measured with capacitive load only on each output.
, DCS enabled, unless otherwise noted.
MAX
1
25°C/115°C
Rev. 0 | Page 6 of 24
ADW12001
A
SWITCHING SPECIFICATIONS
AVDD = 3 V, DRVDD = 2.5 V, maximum sample rate, CLK_A = CLK_B; AIN = −0.5 dBFS differential input, 1.0 V internal reference,
T
to T
MIN
Table 4.
Parameter Te mp Min Typ Max Unit
SWITCHING PERFORMANCE
Maximum Conversion Rate Full 40 MSPS
Minimum Conversion Rate Full 1 MSPS
CLK Period Full 25.0 ns
CLK Pulse Width High
CLK Pulse Width Low
DATA OUTPUT PARAMETER
Output Delay2 (tPD) Full 2 3.5 6 ns
Pipeline Delay (Latency) Full 7 Cycles
Aperture Delay (tA) Full 1.0 ns
Aperture Uncertainty (tJ) Full 0.5 ps rms
Wake-Up Time
OUT-OF-RANGE RECOVERY TIME Full 2 Cycles
1
This model has a duty cycle stabilizer circuit that, when enabled, corrects for a wide range of duty cycles (see Figure 16).
2
Output delay is measured from clock 50% transition to data 50% transition with a 5 pF load on each output.
3
Wake-up time is dependent on the value of the decoupling capacitors; typical values shown with 0.1 μF and 10 μF capacitors on REFT and REFB.
Timing Diagram
, DCS enabled, unless otherwise noted.
MAX
1
1
Full 8.8 ns
3
Full 2.5 ms
N
NALOG
INPUT
N – 1
N + 1
N + 2
Full 8.8 ns
N + 8
N + 3
N + 4
N + 5
N + 6
N + 7
CLOCK
DATA
OUT
N – 9 N – 8
N – 7
N – 6
N – 5
Figure 2. Timing Diagram
N – 4
N – 3
N – 2
N – 1
N
t
PD
MIN 2.0ns,
=
MAX 6.0ns
07737-002
Rev. 0 | Page 7 of 24
ADW12001
ABSOLUTE MAXIMUM RATINGS
Table 5.
Parameter1 Rating
Electrical
AVDD to AGND −0.3 V to +3.9 V
DRVDD to DRGND −0.3 V to +3.9 V
AGND to DRGND −0.3 V to +0.3 V
AVDD to DRVDD −3.9 V to +3.9 V
Digital Outputs CLK_A, CLK_B, DCS,
MUX_SELECT, SHARED_REF to DRGND
OEB, DFS to AGND
VIN±_A, VIN±_B to AGND
VREF to AGND
SENSE to AGND
REFB_A, REFB_B, REFT_A, REFT_B to AGND
PDWN_A, PDWN_B to AGND
Environmental2
Operating Temperature Range −45°C to +115 °C
Junction Temperature 150°C
Lead Temperature (10 sec) 300°C
Storage Temperature Range −65°C to +150°C
1
Absolute maximum ratings are limiting values to be applied individually, and
beyond which the serviceability of the circuit may be impaired. Functional
operability is not necessarily implied.
2
Typical thermal impedances: 64-lead LFCSP, θJA = 26.4°C/W with heat slug
soldered to the ground plane. These measurements were taken on a 4-layer
board in still air, in accordance with EIA/JESD51-7.
−0.3 V to
DRVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
−0.3 V to
AVDD + 0.3 V
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
THERMAL RESISTANCE
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.
Table 6. Thermal Resistance
Package Type θJA Unit
64-Lead LFCSP 26.4 °C/W
ESD CAUTION
Rev. 0 | Page 8 of 24
ADW12001
A
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
AVDD63CLK_A62SHARED_REF61MUX_SELECT60PDWN_A59OEB_A58OTR_A57D13_A (MSB)56D12_A55D11_A54D10_A53DRGND52DRVDD51D9_A50D8_A49D7_A
64
AGND
VIN+_A
VIN–_A
AGND
AVDD
REFT_A
REFB_
VREF
SENSE
REFB_B
REFT_B
AVDD
AGND
VIN–_B
VIN+_B
AGND
1
PIN 1
2
3
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
19
20
ADW12001
(Not to Scale)
21
22
TOP VIEW
23
24
28
29
30
48
D6_A
47
D5_A
46
D4_A
45
D3_A
44
D2_A
43
D1_A
42
D0_A (LSB)
41
DRVDD
40
DRGND
39
OTR_B
38
D13_B (MSB)
37
D12_B
36
D11_B
35
D10_B
34
D9_B
33
D8_B
DFS
DCS
AVDD
CLK_B
NOTES
1. THE ADW12001 L FCSP HAS AN EXP OSED PAD ON T HE UNDERSIDE
OF THE PACKAG E THAT MUST BE CONNECTED TO PCB GND.
WN_BPD
OEB_B
LSB)D0_B (
D1_B25D2_B26D3_B27D4_B
D5_B31D6_B32D7_B
DRVDD
DRGND
07737-003
Figure 3. Pin Configuration
Table 7. Pin Function Descriptions
Pin No. Mnemonic Description
1, 4, 13, 16 AGND Analog Ground.
2 VIN+_A Analog Input Pin (+) for Channel A.
3 VIN−_A Analog Input Pin (−) for Channel A.
5, 12, 17, 64 AVDD Analog Power Supply.
6 REFT_A Differential Reference (+) for Channel A.
7 REFB_A Differential Reference (−) for Channel A.
8 VREF Voltage Reference Input/Output.
9 SENSE Reference Mode Selection.
10 REFB_B Differential Reference (−) for Channel B.
11 REFT_B Differential Reference (+) for Channel B.
14 VIN−_B Analog Input Pin (−) for Channel B.
15 VIN+_B Analog Input Pin (+) for Channel B.
18 CLK_B Clock Input Pin for Channel B.
19 DCS Enable Duty Cycle Stabilizer (DCS) Mode.
20 DFS Data Output Format Select Pin. Low for offset binary, high for twos complement.
21 PDWN_B
Power-Down Function Selection for Channel B. Logic 0 enables Channel B. Logic 1 powers down
Channel B (outputs static, not high-Z).
22 OEB_B Output Enable Pin for Channel B. Logic 0 enables Data Bus B. Logic 1 sets outputs to high-Z.
23 to 27 D0_B (LSB) to D4_B Channel B Data Output Bits.
30 to 37 D5_B to D12_B Channel B Data Output Bits.
38 D13_B (MSB) Channel B Data Output Bits.
28, 40, 53 DRGND Digital Output Ground.
Rev. 0 | Page 9 of 24
ADW12001
Pin No. Mnemonic Description
29, 41, 52 DRVDD
39 OTR_B Out-of-Range Indicator for Channel B.
42 to 51 D0_A (LSB) to D9_A Channel A Data Output Bits.
54 to 56 D10_A to D12_A Channel A Data Output Bits.
57 D13_A (MSB) Channel A Data Output Bits.
58 OTR_A Out-of-Range Indicator for Channel A.
59 OEB_A Output Enable Pin for Channel A. Logic 0 enables Data Bus A. Logic 1 sets outputs to high-Z.
60 PDWN_A
61 MUX_SELECT
62 SHARED_REF Shared Reference Control Pin. Low for independent reference mode, high for shared reference mode.
63 CLK_A Clock Input Pin for Channel A.
EP
Digital Output Driver Supply. Must be decoupled to DRGND with a minimum 0.1 μF capacitor.
Recommended decoupling is 0.1 μF capacitor in parallel with 10 μF capacitor.
Power-Down Function Selection for Channel A. Logic 0 enables Channel A. Logic 1 powers down
Channel A (outputs static, not high-Z).
Data Multiplexed Mode. See Data Format section for how to enable; high setting disables output
data multiplexed mode.
Exposed Pad. This part has an exposed pad on the underside of the package that must be connected
to PCB GND.
Rev. 0 | Page 10 of 24
ADW12001
TERMINOLOGY
Aperture Delay
SHA performance measured from the rising edge of the clock
input to when the input signal is held for conversion.
Aperture Jitter
The variation in aperture delay for successive samples, which is
manifested as noise on the input to the ADC.
Integral Nonlinearity (INL)
Deviation of each individual code from a line drawn from
negative full scale through positive full scale. The point used as
negative full scale occurs ½ LSB before the first code transition.
Positive full scale is defined as a level 1½ LSB beyond the last
code transition. The deviation is measured from the middle of
each particular code to the true straight line.
Differential Nonlinearity (DNL, No Missing Codes)
An ideal ADC exhibits code transitions that are exactly 1 LSB
apart. DNL is the deviation from this ideal value. Guaranteed
no missing codes to 14-bit resolution indicates that all 16,384
codes must be present over all operating ranges.
Offset Error
The major carry transition should occur for an analog value
½ LSB below VIN+ = VIN−. Offset error is defined as the
deviation of the actual transition from that point.
Gain Error
The first code transition should occur at an analog value ½ LSB
above negative full scale. The last transition should occur at an
analog value 1½ LSB below the nominal full scale. Gain error is
the deviation of the actual difference between first and last code
transitions and the ideal difference between first and last code
transitions.
Temp er at u re D ri ft
The temperature drift for zero error and gain error specifies the
maximum change from the initial (25°C) value to the value at
T
MIN
or T
MAX
.
Power Supply Rejection
The specification shows the maximum change in full scale from
the value with the supply at the minimum limit to the value
with the supply at its maximum limit.
Total Harmonic Distortion (THD)
The ratio of the rms sum of the first six harmonic components
to the rms value of the measured input signal, expressed as a
percentage or in decibels relative to the peak carrier signal (dBc).
Signal-to-Noise and Distortion (SINAD) Ratio
The ratio of the rms value of the measured input signal to the
rms sum of all other spectral components below the Nyquist
frequency, including harmonics but excluding dc. The value for
SINAD is expressed in dB.
Effective Number of Bits (ENOB)
Using the following formula:
ENOB = (SINAD − 1.76)/6.02
ENOB for a device for sine wave inputs at a given input
frequency can be calculated directly from its measured SINAD.
Signal-to-Noise Ratio (SNR)
The ratio of the rms value of the measured input signal to the
rms sum of all other spectral components below the Nyquist
frequency, excluding the first six harmonics and dc. The value
for SNR is expressed in dB.
Spurious-Free Dynamic Range (SFDR)
The difference in dB between the rms amplitude of the input
signal and the peak spurious signal.
Nyquist Sampling
When the frequency components of the analog input are below
the Nyquist frequency (f
/2), this is often referred to as
CLOCK
Nyquist sampling.
IF Sampling
Due to the effects of aliasing, an ADC is not limited to Nyquist
sampling. Higher sampled frequencies are aliased down into the
first Nyquist zone (dc − f
/2) on the output of the ADC. The
CLOCK
bandwidth of the sampled signal should not overlap Nyquist
zones and alias onto itself. Nyquist sampling performance is
limited by the bandwidth of the input SHA and clock jitter
(jitter adds more noise at higher input frequencies).
Two -Tone SFDR
The ratio of the rms value of either input tone to the rms value
of the peak spurious component. The peak spurious component
may or may not be an IMD product.
Out-of-Range Recovery Time
The time it takes for the ADC to reacquire the analog input
after a transient from 10% above positive full scale to 10% above
negative full scale or from 10% below negative full scale to 10%
below positive full scale.
Crosstalk
Coupling onto one channel being driven by a (−0.5 dBFS) signal
when the adjacent interfering channel is driven by a full-scale
signal. Measurement includes all spurs resulting from both
direct coupling and mixing components.
Rev. 0 | Page 11 of 24
ADW12001
TYPICAL PERFORMANCE CHARACTERISTICS
AVDD, DRVDD = 3.0 V, T = 25°C, AIN differential drive, full scale = 2 V, unless otherwise noted.
100
95
90
85
80
75
70
SFDR/S NR (dBc)
65
60
55
50
40
45 50 55 60 65
ADC SAMPLE RATE (MSPS)
SNR
SFDR
Figure 7. Single-Tone SFDR/SNR vs. FS with fIN = 20 MHz
100
90
80
SNR
SFDR
07737-007
MAGNITUDE (dBFS)
–20
–40
–60
–80
–100
–120
0
CROSSTALK
10 15 20 25 3050
FREQUENCY ( MHz)
SECOND
HARMONIC
SNR = 72.6dB
SINAD = 71.9dB
H2 = –81.5dBc
H3 = –86.8dBc
SFDR = 81.5d B
THIRD
HARMONIC
Figure 4. Single-Tone FFT of Channel A Digitizing fIN = 12.5 MHz
While Channel B Is Digitizing f
0
–20
–40
= 10 MHz
IN
SNR = 70.5dB
SINAD = 69.4dB
H2 = –92.3dBc
H3 = –80.1dBc
SFDR = 80.1dBc
07737-004
MAGNITUDE (d BFS)
–60
–80
–100
–120
SECOND
HARMONIC
10 15 20 25 3050
FREQUENCY ( MHz)
THIRD HARMONIC
CROSSTALK
Figure 5. Single-Tone FFT of Channel A Digitizing fIN = 70 MHz
= 76 MHz
IN
SNR = 68.1dB
SINAD = 68.0dB
H2 = –83.4dBc
H3 = –83.1dBc
SFDR = 75.1dBc
SECOND
HARMONIC
THIRD
HARMONIC
MAGNITUDE (d BFS)
–100
–120
While Channel B Is Digitizing f
0
–20
–40
–60
CROSSTALK
–80
10 15 20 25 3050
FREQUENCY (MHz )
Figure 6. Single-Tone FFT of Channel A Digitizing fIN = 120 MHz
While Channel B Is Digitizing f
= 126 MHz
IN
70
SFDR/SNR (dBc)
60
50
–3540–30 –25 –20 –15 –10 –5
07737-005
INPUT AMPLITUDE (dBFS)
SNR
0
7737-008
Figure 8. Single-Tone SFDR/SNR vs. AIN with fIN = 20 MHz
95
90
85
80
SFDR/SNR (dBc)
75
70
65
0
20 40 60 80 100 120
07737-006
Figure 9. Single-Tone SFDR/SNR vs. f
SFDR
SNR
INPUT FREQUENCY (MHz)
140
07737-009
IN
Rev. 0 | Page 12 of 24
ADW12001
0
–20
–40
100
SNR
95
90
85
SFDR
–60
IMD = –85dBc
–80
MAGNITUDE (dBFS)
–100
–120
10 15 20 25 3050
FREQUENCY ( MHz)
Figure 10. Dual-Tone FFT with f
0
–20
–40
IMD =
–83dBc
–60
–80
MAGNITUDE (dBFS)
–100
–120
10 15 20 25 3050
FREQUENCY ( MHz)
Figure 11. Dual-Tone FFT with f
0
= 39 MHz and f
IN1
= 70 MHz and f
IN1
= 40 MHz
IN2
= 71 MHz
IN2
80
75
SFDR/SNR (dBF S)
70
65
60
–24
07737-010
Figure 13. Dual-Tone SFDR/SNR vs. A
100
95
90
85
80
75
SFDR/SNR (dBF S)
65
60
–2470–21 –18 –15 –12 –9 –6
07737-011
Figure 14. Dual-Tone SFDR/SNR vs. AIN with f
100
SNR
–21 –18 –15 –12 –9 –6
INPUT AMPLITUDE (dBFS)
with f
IN
SFDR
SNR
INPUT AMPLITUDE (dBFS)
= 45 MHz and f
IN1
= 70 MHz and f
IN1
= 46 MHz
IN2
= 71 MHz
IN2
07737-013
07737-014
–20
–40
–60
–80
MAGNITUDE (dBFS)
–100
–120
10 15 20 25 3050
FREQUENCY (MHz)
Figure 12. Dual-Tone FFT with f
= 200 MHz and f
IN1
= 201 MHz
IN2
07737-012
Rev. 0 | Page 13 of 24
95
90
85
80
75
SFDR/SNR (dBF S)
70
65
60
–24 –21 –18 –15 –12 –9 –6
SFDR
SNR
INPUT AMPLITUDE (dBFS)
Figure 15. Dual-Tone SFDR/SNR vs.
A
with f
IN
= 200 MHz and f
IN1
= 201 MHz
IN2
07737-015
ADW12001
95
90
85
80
75
70
65
SINAD/SFDR (dBc)
60
55
50
30 40 45 50 55 60 65
DCS OFF (SFDR)
DCS ON (SINAD)
35
DCS ON (S FDR)
DCS OFF (SINAD)
DUTY CYCLE (%)
Figure 16. SINAD/SFDR vs. Clock Duty Cycle
7737-016
84
82
80
78
76
74
72
SINAD/SFDR (d B)
70
68
66
–50 0 50 100
SFDR
SINAD
TEMPERATURE (°C)
Figure 17. SINAD/SFDR vs. Temperature with fIN = 32.5 MHz
07737-017
Rev. 0 | Page 14 of 24
ADW12001
A
V
V
V
A
EQUIVALENT CIRCUITS
DD
VDD
IN+_A, VIN–_A,
IN+_B, VIN–_B
Figure 18. Equivalent Analog Input Circuit
DRVDD
07737-019
Figure 19. Equivalent Digital Output Circuit
CLK_A, CLK_B
DCS, DFS,
MUX_SELECT,
SHARED_REF
737-018
07
07737-020
Figure 20. Equivalent Digital Input Circuit
Rev. 0 | Page 15 of 24
ADW12001
V
V
THEORY OF OPERATION
The ADW12001 consists of two high performance ADCs that
are based on the AD9235 converter core. The dual ADC paths
are independent, except for a shared internal band gap reference
source, VREF. Each of the ADC paths consists of a proprietary
front-end SHA followed by a pipelined switched capacitor ADC.
The pipelined ADC is divided into three sections, consisting of
a 4-bit first stage, followed by eight 1.5-bit stages, and a final
3-bit flash. Each stage provides sufficient overlap to correct for
flash errors in the preceding stages. The quantized outputs from
each stage are combined through the digital correction logic
block into a final 12-bit result. The pipelined architecture permits
the first stage to operate on a new input sample, whereas the
remaining stages operate on preceding samples. Sampling
occurs on the rising edge of the respective clock.
Each stage of the pipeline, excluding the last, consists of a low
resolution flash ADC and a residual multiplier to drive the next
stage of the pipeline. The residual multiplier uses the flash ADC
output to control a switched capacitor digital-to-analog converter
(DAC) of the same resolution. The DAC output is subtracted
from the input signal of the stage, and the residual is amplified
(multiplied) to drive the next pipeline stage. The residual multiplier
stage is also called a multiplying DAC (MDAC). One bit of
redundancy is used in each one of the stages to facilitate digital
correction of flash errors. The last stage simply consists of a
flash ADC.
The input stage contains a differential SHA that can be configured as ac- or dc-coupled in differential or single-ended modes.
The output staging block aligns the data, carries out the error
correction, and passes the data to the output buffers. The output
buffers are powered from a separate supply, allowing adjustment
of the output voltage swing.
ANALOG INPUT
The analog input to the ADW12001 is a differential switched
capacitor SHA designed for optimum performance while
processing a differential input signal. The SHA input accepts
inputs over a wide common-mode range. An input commonmode voltage of midsupply is recommended to maintain
optimal performance.
The SHA input is a differential switched capacitor circuit. In
Figure 21, the clock signal alternatively switches the SHA between
sample mode and hold mode. When the SHA is switched into
sample mode, the signal source must be capable of charging the
sample capacitors and settling within one-half of a clock cycle.
A small resistor in series with each input can help reduce the
peak transient current required from the output stage of the
driving source. Also, a small shunt capacitor can be placed
across the inputs to provide dynamic charging currents. This
passive network creates a low-pass filter at the ADC input;
therefore, the precise values are dependent on the application.
In IF under sampling applications, any shunt capacitors should
be removed. In combination with the driving source impedance,
they limit the input bandwidth. For the best dynamic performance,
match the source impedances driving VIN+ and VIN− such that
common-mode settling errors are symmetrical. These errors are
reduced by the common-mode rejection of the ADC.
H
T
T
H
7737-021
IN+_A,
VIN+_B
IN–_A,
VIN–_B
T
C
PAR
T
C
PAR
Figure 21. Switched Capacitor Input
5pF
5pF
An internal differential reference buffer creates positive and
negative reference voltages, REFT and REFB, respectively, that
define the span of the ADC core. The output common mode of
the reference buffer is set to midsupply, and the REFT and
REFB voltages and span are defined as:
REFT = ½(AVDD + V
REFB = ½(AVDD + V
Span = 2 × (REFT − REFB) = 2 × V
REF
REF
)
)
REF
These equations show that the REFT and REFB voltages are
symmetrical around the midsupply voltage and, by definition,
the input span is twice the value of the V
voltage.
REF
The internal voltage reference can be pin-strapped to fixed
values of 0.5 V or 1.0 V or adjusted within the same range as
discussed in the Internal Reference Connection section. Maximum SNR performance is achieved with the ADW12001 set to
the largest input span of 2 V p-p. The relative SNR degradation
is 3 dB when changing from 2 V p-p mode to 1 V p-p mode.
The SHA may be driven from a source that keeps the signal
peaks within the allowable range for the selected reference
voltage. The minimum and maximum common-mode input
levels are defined as
VCM
VCM
= V
MIN
REF
= (AVDD + V
MAX
/2
REF
)/2
The minimum common-mode input level allows the ADW12001
to accommodate ground referenced inputs. Although optimum
performance is achieved with a differential input, a single-ended
source may be driven into VIN+ or VIN−. In this configuration,
one input accepts the signal, while the opposite input is set to
midscale by connecting it to an appropriate reference. For
Rev. 0 | Page 16 of 24
ADW12001
2
p
example, a 2 V p-p signal may be applied to VIN+, while a 1 V
reference is applied to VIN−. The ADW12001 then accepts an
input signal varying between 2 V and 0 V. In the single-ended
configuration, distortion performance may degrade significantly
as compared to the differential case. However, the effect is less
noticeable at lower input frequencies.
Differential Input Configurations
Optimum performance is achieved while driving the
ADW12001 in a differential input configuration. For baseband
applications, the AD8138 differential driver provides excellent
performance and a flexible interface to the ADC. The output
common-mode voltage of the AD8138 is easily set to AVDD/2,
and the driver can be configured in a Sallen-Key filter topology
to provide band limiting of the input signal.
At input frequencies in the second Nyquist zone and above, the
performance of most amplifiers is not adequate to achieve the
true performance of the ADW12001. This is especially true in
IF under sampling applications where frequencies in the 70 MHz
to 200 MHz range are being sampled. For these applications,
differential transformer coupling is the recommended input
configuration, as shown in Figure 22.
AVDD
VIN+_A,
VIN−_A
ADW12001
VIN+_B,
VIN−_B
AGND
07737-022
V p-
50Ω
49.9Ω
0.1µF
Figure 22. Differential Transformer Coupling
10pF
50Ω
10pF
1kΩ
1kΩ
The signal characteristics must be considered when selecting a
transformer. Most RF transformers saturate at frequencies
below a few megahertz, and excessive signal power can also
cause core saturation, which leads to distortion.
Single-Ended Input Configuration
A single-ended input may provide adequate performance in
cost-sensitive applications. In this configuration, there is a
degradation in SFDR and distortion performance due to the
large input common-mode swing. However, if the source
impedances on each input are matched, there should be little
effect on SNR performance.
CLOCK INPUT AND CONSIDERATIONS
Typical high speed ADCs use both clock edges to generate
a variety of internal timing signals and, as a result, may be
sensitive to the clock duty cycle. Commonly, a 5% tolerance
is required on the clock duty cycle to maintain dynamic
performance characteristics.
The ADW12001 provides separate clock inputs for each channel.
The optimum performance is achieved with the clocks operating
at the same frequency and phase. Clocking the channels asynchronously may degrade performance significantly. In some
applications, it is desirable to skew the clock timing of adjacent
channels. The separate clock inputs of the ADW12001 allow for
clock timing skew (typically ±1 ns) between the channels without
significant performance degradation.
The ADW12001 contains two clock duty cycle stabilizers, one
for each converter, that retime the nonsampling edge, providing
an internal clock with a nominal 50% duty cycle. When proper
track-and-hold times for the converter are required to maintain
high performance, maintaining a 50% duty cycle clock is particularly important in high speed applications. It may be difficult
to maintain a tightly controlled duty cycle on the input clock on
the PCB (see Figure 16). DCS can be enabled by tying the DCS
pin high.
The duty cycle stabilizer uses a delay-locked loop to create the
nonsampling edge. As a result, any changes to the sampling
frequency require approximately 2 μs to 3 μs to allow the DLL
to acquire and settle to the new rate.
High speed, high resolution ADCs are sensitive to the quality
of the clock input. The degradation in SNR at a given full-scale
input frequency (f
calculated as
SNR
In the equation, the rms aperture jitter, t
sum square of all jitter sources, which includes the clock input,
analog input signal, and ADC aperture jitter specification. Under
sampling applications are particularly sensitive to jitter.
For optimal performance, especially in cases where aperture
jitter may affect the dynamic range of the ADW12001, it is
important to minimize input clock jitter. The clock input
circuitry should use stable references; for example, use analog
power and ground planes to generate the valid high and low
digital levels for the ADW12001 clock input. Separate power
supplies for clock drivers from the ADC output driver supplies
to avoid modulating the clock signal with digital noise. Low
jitter, crystal-controlled oscillators make the best clock sources.
If the clock is generated from another type of source (by gating,
dividing, or other methods), it should be retimed by the
original clock at the last step.
) due only to aperture jitter (tJ) can be
IN
⎡
×=
log20
⎢
⎢
⎣
1
()
2
⎤
⎥
×××
tfπ
⎥
jIN
⎦
, represents the root-
J
Rev. 0 | Page 17 of 24
ADW12001
POWER DISSIPATION AND STANDBY MODE
The power dissipated by the ADW12001 is proportional to
its sampling rates. The digital (DRVDD) power dissipation is
determined primarily by the strength of the digital drivers and
the load on each output bit. The digital drive current can be
calculated by
= V
I
DRVDD
where N is the number of bits changing and C
load on the digital pins that changed.
The analog circuitry is optimally biased so that each speed
grade provides excellent performance while affording reduced
power consumption. Each speed grade dissipates a baseline
power at low sample rates that increases with clock frequency.
Either channel of the ADW12001 can be placed into standby mode
independently by asserting the PDWN_A or PDWN_B pins.
It is recommended that the input clock(s) and analog input(s)
remain static during either independent or total standby, which
results in a typical power consumption of 1 mW for the ADC.
Note that if DCS is enabled, it is mandatory to disable the clock
of an independently powered-down channel. Otherwise, significant distortion results on the active channel. If the clock inputs
remain active while in total standby mode, typical power dissipation of 12 mW results.
The minimum standby power is achieved when both channels
are placed into full power-down mode (PDWN_A = PDWN_B
= high). Under this condition, the internal references are powered
down. When either or both of the channel paths are enabled
after a power-down, the wake-up time is directly related to the
recharging of the REFT and REFB decoupling capacitors and
to the duration of the power-down. Typically, it takes approximately 5 ms to restore full operation with fully discharged
0.1 μF and 10 μF decoupling capacitors on REFT and REFB.
A
–1
DRVDD
× C
× f
CLOCK
A
1
× N
LOAD
A
2
is the average
A
3
LOAD
A
0
A
4
A single channel can be powered down for moderate power
savings. The powered-down channel shuts down internal
circuits, but both the reference buffers and shared reference
remain powered on. Because the buffer and voltage reference
remain powered on, the wake-up time is reduced to several
clock cycles.
DIGITAL OUTPUTS
The ADW12001 output drivers can be configured to interface
with 2.5 V or 3.3 V logic families by matching DRVDD to
the digital supply of the interfaced logic. The output drivers
are sized to provide sufficient output current to drive a wide
variety of logic families. However, large drive currents tend to
cause current glitches on the supplies that may affect converter
performance. Applications requiring the ADC to drive large
capacitive loads or large fanouts may require external buffers
or latches.
The data format can be selected for either offset binary or twos
complement. See the Data Format section for more information.
TIMING
The ADW12001 provides latched data outputs with a pipeline
delay of seven clock cycles. Data outputs are available one propagation delay (t
to Figure 2 for a detailed timing diagram.
The internal duty cycle stabilizer can be enabled on the ADW12001
using the DCS pin. This provides a stable 50% duty cycle to internal
circuits.
Minimize the length of the output data lines and the loads
placed on them to reduce transients within the ADW12001.
These transients can detract from the dynamic performance of
the converter. The lowest typical conversion rate of the ADW12001
is 1 MSPS. At clock rates below 1 MSPS, dynamic performance
may degrade.
A
5
) after the rising edge of the clock signal. Refer
PD
8
ANALOG INPUT
ADC A
A
A
6
7
A
B
B
–1
0
B
–8
t
PD
B
1
A
B
–7
–7
B
2
A–6B–6A
–5
t
PD
B
3
B–5A
B
B
4
B
–4
–4
B
5
A
B
–3
–3
B
6
A
–2
A
B
–2
7
B
–1
–1
B
8
A
B
0
A
0
Figure 23. Multiplexed Data Format Using the Channel A Output and the Same Clock Tied to CLK_A, CLK_B, and MUX_SELECT
Rev. 0 | Page 18 of 24
ANALOG INPUT
ADC B
CLK_A = CLK_B =
MUX_SELECT
1
D0_A TO
D11_A
07737-023
ADW12001
DATA FORMAT
The ADW12001 data output format can be configured for
either twos complement or offset binary. This is controlled by
the data format select pin (DFS). Connecting DFS to AGND
produces offset binary output data. Conversely, connecting DFS
to AVDD formats the output data as twos complement.
The output data from the dual ADCs can be multiplexed onto a
single 14-bit output bus. The multiplexing is accomplished by
toggling the MUX_SELECT bit, which directs channel data to
the same or opposite channel data port. When MUX_SELECT
is logic high, the Channel A data is directed to the Channel A
output bus, and the Channel B data is directed to the Channel B
output bus. When MUX_SELECT is logic low, the channel data
is reversed, that is, the Channel A data is directed to the Channel B
output bus, and the Channel B data is directed to the Channel A
output bus. By toggling the MUX_SELECT bit, multiplexed
data is available on either of the output data ports.
If the ADCs run with synchronized timing, this same clock
can be applied to the MUX_SELECT pin. Any skew between
CLK_A, CLK_B, and MUX_SELECT can degrade ac performance. It is recommended to keep the clock skew <100 ps.
After the MUX_SELECT rising edge, either data port has the
data for its respective channel; after the falling edge, the data of
the alternate channel is placed on the bus. Typically, the other
unused bus is disabled by setting the appropriate OEB higher
to reduce power consumption and noise. Figure 23 shows an
example of multiplex mode. When multiplexing data, the data
rate is two times the sample rate. Note that both channels must
remain active in this mode and that each power-down pin of
the channel must remain low.
VOLTAGE REFERENCE
A stable and accurate 0.5 V voltage reference is built into the
ADW12001. The input range can be adjusted by varying the
reference voltage applied to the ADW12001, using either the
internal reference with different external resistor configurations
or an externally applied reference voltage. The input span of the
ADC tracks reference voltage changes linearly. If the ADC is
being driven differentially through a transformer, the reference
voltage can be used to bias the center tap (common-mode voltage).
The shared reference mode allows the user to connect the references from the dual ADCs together externally for superior gain
and offset matching performance. If the ADCs are to function
independently, the reference decoupling can be treated independently and can provide superior isolation between the dual
channels. To enable shared reference mode, the SHARED_REF
pin must be tied high and the external differential references
must be externally shorted. (REFT_A must be externally shorted to
REFT_B, and REFB_A must be shorted to REFB_B.)
Internal Reference Connection
A comparator within the ADW12001 detects the potential at
the SENSE pin and configures the reference into four possible
states, which are summarized in Tab l e 8. If SENSE is grounded,
the reference amplifier switch is connected to the internal resistor
divider (see Figure 24), setting VREF to 1 V. Connecting the
SENSE pin to VREF switches the reference amplifier output
to the SENSE pin, completing the loop and providing a 0.5 V
reference output. If a resistor divider is connected, as shown in
Figure 25, the switch is again set to the SENSE pin. This puts
the reference amplifier in a noninverting mode with the VREF
output defined as
V
= 0.5 × (1 + R2/R1)
REF
In all reference configurations, REFT and REFB drive the ADC
core and establish its input span. The input range of the ADC
always equals twice the voltage at the reference pin for either an
internal or an external reference.
VIN+_A, VIN+ _B
VIN–_A, VIN–_B
10µF
CORE
VREF
0.1µF
SENSE
Figure 24. Internal Reference Configuration
SELECT
LOGIC
0.5V
ADW12001
ADC
REFT
REFB
0.1µF
0.1µF
0.1µF
10µF
07737-024
Table 8. Reference Configuration Summary
Selected Mode SENSE Voltage Resulting V
External Reference AVDD N/A
Internal Fixed Reference V
Programmable Reference 0.2 V to V
0.5 1.0
REF
0.5 × (1 + R2/R1) 2 × V
REF
1
2 × External Reference
(V) Resulting Differential Span (V p-p)
REF
Internal Fixed Reference AGND to 0.2 V 1.0 2.0
1
N/A means not applicable.
Rev. 0 | Page 19 of 24
(See Figure 25)
REF
ADW12001
External Reference Operation
The use of an external reference may be necessary to enhance
the gain accuracy of the ADC or to improve thermal drift
characteristics. When multiple ADCs track one another, a single
reference (internal or external) may be necessary to reduce gain
matching errors to an acceptable level. A high precision external
reference may also be selected to provide lower gain and offset
temperature drift. Figure 26 shows the typical drift characteristics of the internal reference in both 1 V and 0.5 V modes. When
the SENSE pin is tied to AVDD, the internal reference is disabled,
allowing the use of an external reference. An internal reference
buffer loads the external reference with an equivalent 7 kΩ load.
The internal buffer still generates the positive and negative fullscale references, REFT and REFB, for the ADC core. The input
span is always twice the value of the reference voltage; therefore,
the external reference must be limited to a maximum of 1 V. If
the internal reference of the ADW12001 is used to drive multiple
converters to improve gain matching, the loading of the reference
by the other converters must be considered. Figure 27 depicts
how the internal reference voltage is affected by loading.
VIN+_A, VI N+_B
VIN–_A, VI N–_B
VREF
ADC
CORE
REFT
REFB
0.1µF
0.1µF
0.1µF
10µF
0.05
0
–0.05
–0.10
ERROR (%)
–0.15
–0.20
–0.25
0 0.5 1.0 1.5 2.0 2.5 3.0
1V ERROR
LOAD (mA)
0.5V ERROR
07737-027
Figure 27. VREF Accuracy vs. Load
THERMAL CONSIDERATIONS
The ADW12001 LFCSP has an integrated heat slug that improves
the thermal and electrical properties of the package when locally
attached to a ground plane at the PCB. A thermal (filled) via
array to a ground plane beneath the part provides a path for heat
to escape the package, lowering junction temperature. Improved
electrical performance also results from the reduction in package
parasitics due to proximity of the ground plane. The recommended
array consists of 0.3 mm vias on a 1.2 mm pitch. θ
with this recommended configuration. Soldering the slug to the
PCB is a requirement for this package.
= 26.4°C/W
JA
10µF
10µF
SENSE
R2
R1
SELECT
LOGIC
ADW12001
Figure 25. Programmable Reference Configuration
1.2
1.0
0.8
0.6
VREF ERROR (%)
0.4
0.2
0
–40 –30 –20 –10 0 10 20 30 40 50 60
TEMPERATURE (°C)
Figure 26. Typical VREF Drift
0.5V
VREF = 1V
VREF = 0.5V
70 80
7737-025
07737-028
Figure 28. Thermal Via Array
07737-026
Rev. 0 | Page 20 of
24
ADW12001
OUTLINE DIMENSIONS
9.20
PIN 1
TOP VIEW
(PINS DOWN)
0.40
BSC
LEAD PITCH
9.00 SQ
8.80
0.23
0.18
0.13
4964
48
7.20
7.00 SQ
6.80
33
32
1.45
1.40
1.35
0.15
SEATING
0.05
PLANE
VIEW A
ROTATED 90° CCW
0.75
0.60
0.45
0.20
0.09
7°
3.5°
0°
0.08
COPLANARIT Y
1.60
MAX
1
16
17
VIEW A
COMPLIANT TO JEDEC STANDARDS MS-026-BBD
051706-A
Figure 29. 64-Lead Low Profile Quad Flat Package [LQFP]
(ST-64-1)
Dimensions shown in millimeters (mm)
0.30
0.25
0.18
64
17
FOR PROPER CONNECTION O F
THE EXPOSE D PAD, REFER T O
THE PIN CONF IGURATIO N AND
FUNCTION DESCRIPTIO NS
SECTION OF THIS DATA SHEET.
PIN 1
INDICATOR
1
*
4.85
4.70 SQ
4.55
16
082908-B
1.00
0.85
0.80
SEATING
PLANE
12° MAX
9.00
BSC SQ
PIN 1
INDICATOR
VIEW
0.60 MAX
TOP
0.80 MAX
0.65 TYP
0.50 BSC
*
COMPLIANT TO JEDEC STANDARDS MO-220-VMMD-4
EXCEPT FO R EXPOSED PAD DI MENSION
8.75
BSC SQ
0.20 REF
0.50
0.40
0.30
0.05 MAX
0.02 NOM
49
48
33
32
0.60 MAX
Figure 30. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
9 mm × 9 mm Body, Very Thin Quad
(CP-64-1)
Dimensions shown in millimeters (mm)
EXPOSED PAD
(BOTTOM VIEW)
7.50
REF
ORDERING GUIDE
Model Temperature Range Package Description Package Option
ADW12001BCPZ-40
ADW12001BCPZRL-40
ADW12001BSTZ-40
ADW12001BSTZRL-40
1
Z = RoHS Compliant Part.
1
−40°C to +115°C 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-64-1
1
−40°C to +115°C 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-64-1
1
−40°C to +115°C 64-Lead Low Profile Quad Flat Package [LQFP] ST-64-1
1
−40°C to +115°C 64-Lead Low Profile Quad Flat Package [LQFP] ST-64-1
Rev. 0 | Page 21 of 24
ADW12001
NOTES
Rev. 0 | Page 22 of 24
ADW12001
NOTES
Rev. 0 | Page 23 of 24
ADW12001
NOTES
©2009 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D07737-0-1/09(0)
Rev. 0 | Page 24 of 24
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