Datasheet ADP1610 Datasheet (ANALOG DEVICES)

1.2 MHz DC-to-DC Step-Up Switching

S

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FEATURES

Fully integrated 1.2 A, 0.2 Ω, power switch Pin-selectable 700 kHz or 1.2 MHz PWM frequency 92% efficiency Adjustable output voltage up to 12 V 3% output regulation accuracy Adjustable soft start Input undervoltage lockout MSOP 8-lead package

APPLICATIONS

TFT LC bias supplies Portable applications Industrial/instrumentation equipment

FUNCTIONAL BLOCK DIAGRAM

REF

FB

2

RAMP

GEN

7

RT

OSC

ERROR

g

m

COMP

1 6

AMP

COMPARATOR

Converter

ADP1610

GENERAL DESCRIPTION

The ADP1610 is a dc-to-dc step-up switching converter with an integrated 1.2 A, 0.2  power switch capable of providing an output voltage as high as 12 V. With a package height of less than 1.1 mm, the ADP1610 is optimal for space-constrained applications such as portable devices or thin film transistor (TFT) liquid crystal displays (LCDs).

The ADP1610 operates in pulse-width modulation (PWM) current mode with up to 92% efficiency. Adjustable soft start prevents inrush currents at startup. The pin-selectable switching frequency and PWM current mode architecture allow excellent transient response, easy noise filtering, and the use of small, cost-saving external inductors and capacitors.

The ADP1610 is offered in the RoHS compliant 8-lead MSOP and operates over the temperature range of −40°C to +85°C.

IN

ADP1610

BIAS

SW

F/F

QSR

DRIVER

5

8

SOFT START

SS

3

D

Rev. A

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

CURRENT

SENSE

AMPLIFIER

4

Figure 1.

GND

04472-001

ADP1610

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REVISION HISTORY

9/08—Rev. 0 to Rev. A

Changes to Table 4 .......................................................................... 11

Changes to Table 5 .......................................................................... 14

10/04—Revision 0: Initial Version

Rev. A | Page 2 of 20

ADP1610

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SPECIFICATIONS

VIN = 3.3 V, TA = −40°C to +85°C, unless otherwise noted. All limits at temperature extremes are guaranteed by correlation and characterization using standard statistical quality control (SQC), unless otherwise noted.

Table 1.

Parameter Symbol Conditions Min Typ Max Unit SUPPLY

Input Voltage VIN 2.5 5.5 V Quiescent Current

Nonswitching State IQ V Shutdown I

Switching State

1

V

SD

Q

I

F

QSW

OUTPUT

Output Voltage V

V

OUT

Load Regulation I Overall Regulation Line, load, temperature ±3 %

REFERENCE

Feedback Voltage VFB 1.212 1.230 1.248 V Line Regulation VIN = 2.5 V to 5.5 V −0.15 +0.15 %/V

ERROR AMPLIFIER

Transconductance gm ∆I = 1 A 100 µA/V Voltage Gain AV 60 dB FB Input Bias Current VFB= 1.23 V 10 nA

SWITCH

SW On Resistance RON I SW Leakage Current VSW = 12 V 0.01 20 µA Peak Current Limit

2

I

2.0 A

CLSET

OSCILLATOR

Oscillator Frequency f

RT = GND 0.49 0.7 0.885 MHz

OSC

RT = IN 0.89 1.23 1.6 MHz Maximum Duty Cycle D

COMP = open, VFB = 1 V, RT = GND 78 83 90 %

MAX

SHUTDOWN

Shutdown Input Voltage Low VIL Nonswitching state 0.6 V Shutdown Input Voltage High VIH Switching state 2.2 V Shutdown Input Bias Current I

SD

SOFT START

SS Charging Current VSS = 0 V 3 µA

UNDERVOLTAGE LOCKOUT

3

UVLO Threshold VIN rising 2.2 2.4 2.5 V UVLO Hysteresis 220 mV

1

This parameter specifies the average current while switching internally and with SW (Pin 5) floating.

2

Guaranteed by design and not fully production tested.

3

Guaranteed by characterization.

= 1.3 V, RT = VIN 390 600 A

FB

= 0 V 0.01 10 µA

SD

= 1.23 MHz, no load 1 2 mA

SW

12 V

IN

= 10 mA to 150 mA, V

LOAD

= 1.0 A 200 400 mΩ

SW

= 10 V 0.05 mV/mA

OUT

VSD = 3.3 V 0.01 1 µA

Rev. A | Page 3 of 20

ADP1610

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ABSOLUTE MAXIMUM RATINGS

Table 2.

Parameter Rating IN, COMP, SD, SS, RT, FB to GND

−0.3 V to +6 V SW to GND 14 V RMS SW Pin Current 1.2 A Operating Ambient Temperature Range −40°C to +85°C Operating Junction Temperature Range −40°C to +125°C Storage Temperature Range −65°C to +150°C θJA, Two Layers 206°C/W θJA, Four Layers 142°C/W Lead Temperature (Soldering, 60 sec) 300°C

Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

R

C

V

R1

1.2MHz

00kHz

C

SS

OUT

FB

R2

V

IN

SD

SS

REF

2

RAMP

GEN

RT

7

OSC

3

SOFT START

8

COMP

1 6

ERROR

AMP

g

m

COMPARATOR

Figure 2. Block Diagram and Typical Application Circuit

ESD CAUTION

IN

ADP1610

BIAS

F/F

QSR

DRIVER

CURRENT

SENSE

AMPLIFIER

C

IN

L1

D1

SW

5

4

GND

V

OUT

C

OUT

04472-002

Rev. A | Page 4 of 20

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PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

1

COMP

GND

ADP1610

FB

2

TOP VIEW

SD

3

(Not to Scale)

4

Figure 3. Pin Configuration

8

SS

RT

7

IN

6

SW

5

04472-003

Table 3. Pin Function Descriptions

Pin No. Mnemonic Description

1 COMP

Compensation Input. Connect a series resistor-capacitor network from COMP to GND to compensate the regulator.

2 FB

Output Voltage Feedback Input. Connect a resistive voltage divider from the output voltage to FB to set the regulator output voltage.

3

Shutdown Input. Drive SD low to shut down the regulator; drive SD high to turn it on.

SD 4 GND Ground. 5 SW

Switching Output. Connect the power inductor from the input voltage to SW and connect the external rectifier from SW to the output voltage to complete the step-up converter.

6 IN

Main Power Supply Input. IN powers the ADP1610 internal circuitry. Connect IN to the input source voltage. Bypass IN to GND with a 10 µF or greater capacitor as close to the ADP1610 as possible.

7 RT

Frequency Setting Input. RT controls the switching frequency. Connect RT to GND to program the oscillator to 700 kHz, or connect RT to IN to program the oscillator to 1.2 MHz.

8 SS Soft Start Timing Capacitor Input. A capacitor from SS to GND brings up the output slowly at power-up.

Rev. A | Page 5 of 20

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TYPICAL PERFORMANCE CHARACTERISTICS

100

V

= 10V

OUT

F

= 700kHz

SW

90

L = 10µH

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

110100

LOAD CURRENT (mA)

Figure 4. Output Efficiency vs. Load Current

VIN = 5.5V

VIN = 3.3V

VIN = 2.5V

04472-005

1k

100

V

= 7.5V

OUT

F

= 1.2MHz

SW

L = 4.7µH

90

80

70

60

EFFICIENCY (%)

50

40

30

110100

LOAD CURRENT (mA)

Figure 7. Output Efficiency vs. Load Current

VIN = 5.5V

VIN = 3.3V

VIN = 2.5V

04472-008

1k

100

V

= 10V

OUT

F

= 1.2MHz

90

SW

L = 4.7µH

80

70

60

50

40

EFFICIENCY (%)

30

20

10

0

1 10 100 1K

LOAD CURRENT (mA)

Figure 5. Output Efficiency vs. Load Current

100

V

= 7.5V

OUT

F

= 700kHz

SW

90

L = 10µH

80

70

60

EFFICIENCY (%)

50

40

30

110100

LOAD CURRENT (mA)

Figure 6. Output Efficiency vs. Load Current

VIN = 5.5V

VIN = 3.3V

VIN = 2.5V

VIN = 5.5V

VIN = 3.3V

VIN = 2.5V

04472-006

04472-007

1k

2.4

2.2

2.0

1.8

1.6

CURRENT LIMIT (A)

1.4

1.2 –40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

Figure 8. Current Limit vs. Ambient Temperature, V

1.4

1.2

1.0

0.8

0.6

0.4

OSCILLATORY FREQUENCY (MHz)

V

0.2

0

–40 –15 10 35 60 85

= 10V

OUT

V

= 3.3V

IN

AMBIENT TEMPERATURE (°C)

Figure 9. Oscillatory Frequency vs. Ambient Temperature

V

= 5.5V

IN

V

= 3.3V

IN

VIN = 2.5V

= 10 V

OUT

RT = V

RT = GND

04472-009

IN

04472-010

Rev. A | Page 6 of 20

ADP1610

A

R

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1.4

1.2

1.0

0.8

Y FREQUENCY (MHz)

0.6

TO

0.4

OSCILL

0.2 V

= 10V

OUT

0

2.5 3.0 3.5 4.0 4.5 5.0 5.5 SUPPLY VOLTAGE (V)

Figure 10. Oscillatory Frequency vs. Supply Voltage

350

300

250

200

150

100

SWITCH RESI STANCE (mΩ)

50

RT = V

RT = GND

= 2.5V

V

IN

V

= 3.3V

IN

VIN = 5.5V

0.50

FSW = 700kHz V

= 1.3V

FB

IN

04472-011

0.45

0.40

0.35

0.30

QUIESCENT CURRENT (mA)

0.25

0.20 –40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

VIN = 5.5V

V

= 3.3V

IN

VIN = 2.5V

04472-014

Figure 13. Quiescent Current vs. Ambient Temperature

0.60

FSW = 1.23kHz V

= 1.3V

FB

0.55

0.50

0.45

0.40

QUIESCENT CURRENT (mA)

0.35

= 5.5V

V

IN

V

= 3.3V

IN

VIN = 2.5V

0

–40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

Figure 11. Switch Resistance vs. Ambient Temperature

1.245

1.24

1.235

1.23

1.225

1.22

FB REGULATION VOLTAGE (V)

1.215

1.21 –40 –10–25 2055035 80 95 11065 125

AMBIENT TEMPERATURE (°C)

Figure 12. FB Regulation Voltage vs. Ambient Temperature

04472-012

04472-013

0.30 –40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

Figure 14. Quiescent Current vs. Ambient Temperature

2.0

FSW = 1.23kHz V

= 1V

FB

1.8

1.6

1.4

1.2

1.0

SUPPLY CURRENT (mA)

0.8

0.6 –40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

Figure 15. Supply Current vs. Ambient Temperature

V

= 5.5V

IN

VIN = 3.3V

VIN = 2.5V

04472-015

04472-016

Rev. A | Page 7 of 20

ADP1610

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1.4

FSW = 700kHz

1.3

V

= 1V

FB

1.2

1.1

1.0

0.9

0.8

0.7

SUPPLY CURRENT (mA)

0.6

0.5

0.4 –40 –15 10 35 60 85

AMBIENT TEMPERATURE (°C)

Figure 16. Supply Current vs. Ambient Temperature

3.5 VIN= 3.3V SD = 0.4V

3.0

2.5

2.0

VIN = 5.5V

VIN = 3.3V

VIN = 2.5V

04472-017

CH1 = IL 200mA/DIV CH2 = V

2

1

CH1 10.0mVΩ CH2 5.00V M400n s A CH2 10.0V

SW

5V/DIV

Figure 19. Switching Waveform in Discontinuous Conduction

VIN = 3.3V, V C

= 10μF, L = 10μH, RC= 130Ω

OUT

C

= 270pF, FSW = 700kHz

C

CH1 = V CH2 = I

1

OUT,

= 10V

OUT

200mV/DIV

200mA/DIV

VIN = 3.3V V

= 10V

OUT

I

= 20mA

LOAD

F

= 700kHz

SW

L = 10µH

T 136.000ns

04472-020

1.5

1.0

SUPPLY CURRENT (µA)

0.5

0

–40 15 70 125

AMBIENT TEMPERATURE (°C)

Figure 17. Supply Current in Shutdown vs. Ambient Temperature

CH1 = IL 500mA/DIV CH2 = V

2

1

CH1 10.0mVΩ CH2 5.00V M400ns A CH2 10.0V

SW

5V/DIV

VIN = 3.3V V

= 10V

OUT

I

= 200mA

LOAD

F

= 700kHz

SW

L = 10

μH

T 136.000ns

Figure 18. Switching Waveform in Continuous Conduction

04472-018

04472-019

2

CH1 200mV CH2 10.0mVΩ M200μs A CH2 7.60mV

Figure 20. Load Transient Response, 700 kHz , V

VIN = 3.3V, V C

= 10µF, L = 4.7µH, RC= 220kΩ

OUT

C

= 150pF, FSW = 1.2MHz

C

CH1 = V CH2 = I

1

2

CH1 200mV CH2 10.0mVΩ M200µs A CH2 7.60mV

OUT

, 200mV/DIV

OUT

, 200mA/DIV

OUT

= 10V

Figure 21. Load Transient Response, 1.2 MHz, V

OUT

04472-021

= 10 V

04472-022

= 10 V

Rev. A | Page 8 of 20

ADP1610

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2

4

CH1 = IL 1A/DIV CH2 = V

3

1

CH1 10.0mVΩ CH2 2.00V M100μs A CH2 680mV

CH3 10.0V CH4 10.00V

IN

CH3 = V

OUT

CH4 = SW,FSW= 700kHz

T 414.800μs

VIN = 3.3V V

OUT

I

= 0.2A

OUT

C

= 0nF

SS

= 10V

04472-023

2

4

CH1 = IL 1A/DIV CH2 = V

3

1

CH1 10.0mVΩ CH2 2.00V M100μs A CH2 1.72V

CH3 10.0V CH4 10.00V

IN

CH3 = V

OUT

CH4 = SW,FSW= 700kHz

T 405.600μs

Figure 22. Start-Up Response from VIN, SS = 0 nF Figure 24. Start-Up Response from Shutdown, SS = 0 nF

2

4

CH1 = IL 1A/DIV CH2 = V

3

IN

CH3 = V

OUT

CH4 = SW,FSW = 700kHz

VIN = 3.3V V

= 10V

OUT

I

= 0.2A

OUT

C

= 10nF

SS

2

4

CH1 = IL 1A/DIV

3

CH2 = SD CH3 = V

OUT

CH4 = SW,FSW= 700kHz

VIN = 3.3V V

= 10V

OUT

I

= 0.2A

OUT

C

= 0nF

SS

I

= 0.2A

OUT

V

= 3.3V

IN

V

= 10V

OUT

C

= 10nF

SS

04472-025

1

CH1 10.0mVΩ CH2 2.00V M100μs A CH2 680mV

CH3 10.0V CH4 10.00V

T 414.800μs

04472-024

1

CH1 10.0mVΩ CH2 2.00V M100μs A CH2 1.72V

CH3 10.0V CH4 10.00V

T 405.600μs

04472-026

Figure 23. Start-Up Response from VIN, SS = 10 nF Figure 25. Start-Up Response from Shutdown, SS = 10 nF

Rev. A | Page 9 of 20

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THEORY OF OPERATION

The ADP1610 current mode step-up switching converter converts a 2.5 V to 5.5 V input voltage up to an output voltage as high as 12 V. The 1.2 A internal switch allows a high output current, and the high 1.2 MHz switching frequency allows tiny external components. The switch current is monitored on a pulse-bypulse basis to limit it to 2 A.

CURRENT MODE PWM OPERATION

The ADP1610 uses current mode architecture to regulate the output voltage. The output voltage is monitored at FB through a resistive voltage divider. The voltage at FB is compared to the internal 1.23 V reference by the internal transconductance error amplifier to create an error current at COMP. A series resistor capacitor at COMP converts the error current to a voltage. The switch current is internally measured and added to the stabilizing ramp. The resulting sum is compared to the error voltage at COMP to control the PWM modulator. This current mode regulation system allows fast transient response, while maintaining a stable output voltage. By selecting the proper resistor capacitor network from COMP to GND, the regulator response is optimized for a wide range of input voltages, output voltages, and load conditions.

FREQUENCY SELECTION

The frequency of the ADP1610 is selectable to operate at either 700 kHz to optimize the regulator for high efficiency or at 1.2 MHz for small external components. Connect RT to IN for 1.2 MHz operation, or connect RT to GND for 700 kHz operation. To achieve the maximum duty cycle, which may be required for converting a low input voltage to a high output voltage, use the lower 700 kHz switching frequency.

SOFT START

To prevent input inrush current at startup, connect a capacitor from SS to GND to set the soft start period. When the ADP1610 is in shutdown (

2.4 V undervoltage lockout voltage, SS is internally shorted to GND to discharge the soft start capacitor. When the ADP1610 is turned on, SS sources 3 A to the soft start capacitor at startup. As the soft start capacitor charges, it limits the voltage at COMP. Because of the current mode regulator, the voltage at COMP is proportional to the switch peak current, and, therefore, the input current. By slowly charging the soft start capacitor, the input current ramps slowly to prevent the current from overshooting excessively at startup.

SD

is at GND) or the input voltage is below the

ON/OFF CONTROL

The SD input turns the ADP1610 regulator on or off. Drive SD low to turn off the regulator and reduce the input current to 10 nA.

SD

Drive

When the dc-to-dc step-up switching converter is turned off, there is a dc path from the input to the output through the inductor and output rectifier. This causes the output voltage to remain slightly below the input voltage by the forward voltage of the rectifier, preventing the output voltage from dropping to zero when the regulator is shut down. Figure 28 shows the application circuit to disconnect the output voltage from the input voltage at shutdown.

high to turn on the regulator.

SETTING THE OUTPUT VOLTAGE

The ADP1610 features an adjustable output voltage range of VIN to 12 V. The output voltage is set by the resistive voltage divider (R1 and R2 in Figure 2) from the output voltage (V

1.230 V feedback input at FB. To calculate the output voltage use the following equation:

V

= 1.23 × (1 + R1/R2) (1)

OUT

To prevent output voltage errors due to the 10 nA FB input bias current, an R2 resistance of 10 kΩ or less is to be used. Choose R1 based on the following equation:

V

−

23.1

R1 = R2 ×

⎛

OUT

⎜ ⎝

⎞

(2)

⎟

23.1

⎠

OUT

) to the

INDUCTOR SELECTION

The inductor is an essential part of the step-up switching converter. It stores energy during the on-time, and transfers that energy to the output through the output rectifier during the off-time. Use inductance in the range of 1 H to 22 µH. In general, lower inductance values have higher saturation current and lower series resistance for a given physical size. However, lower inductance results in higher peak current that can lead to reduced efficiency and greater input and/or output ripple and noise. A peak-to-peak inductor ripple current at approximately 30% of the maximum dc input current typically yields an optimal compromise.

For determining the inductor ripple current, the input (V output (V the following equation:

D =

) voltages determine the switch duty cycle (D) by

OUT

VV

IN

OUT

V

(3)

OUT

) and

IN

Rev. A | Page 10 of 20

ADP1610

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Table 4. Recommended Capacitor and Inductor Manufacturers

Maximum

Vendor Part L (μH) C (μF)/Rating (V)

Sumida CMD4D11-2R2MC 2.2 0.95 116 1.2

CMD4D11-4R7MC 4.7 0.75 216 1.2 CDRH4D28-100 10 1.00 128 3.0 CDRH5D18-220 22 0.80 290 2.0 CR43-4R7 4.7 1.15 109 3.5 CR43-100 10 1.04 182 3.5

Coilcraft DS1608-472 4.7 1.40 60 2.9

DS1608-103 10 1.00 75 2.9

Toko D52LC-4R7M 4.7 1.14 87 2.0

D52LC-100M 10 0.76 150 2.0

Murata GRM32ER61A226KE20L 22/10

GRM31CR61E106KA12L 10/25

GRM31CR61A106KA01L 10/10

GRM31CR71E475KA88L 4.7/25

DC Current

Using the duty cycle and switching frequency (F is calculated using the following equation:

D

t

=

ON

The inductor ripple current (I

(4)

f

SW

) in steady state is calculated

L

using the following equation:

V

×

IN



I

=

L

ON

(5)

L

The inductance value (L) is calculated using the following equation:

×

IN

L =

ON

(6)

ItVΔ

L

The peak inductor current (the maximum input current plus half the inductor ripple current) must be less than the rated saturation current of the inductor. Likewise, the maximum rated rms current of the inductor must be greater than the maximum dc input current to the regulator.

For duty cycles greater than 50%, which occur with input

), the on-time

SW

CHOOSING THE INPUT AND OUTPUT CAPACITORS

The ADP1610 requires input and output bypass capacitors to supply transient currents while maintaining constant input and output voltage. To prevent noise at the ADP1610 input, a low ESR (equivalent series resistance), 10 µF or greater input capacitor is to be used. Place the capacitor between IN and GND as close to the ADP1610 as possible. Ceramic capacitors are preferred because of their low ESR characteristics. Alternatively, use a high value, medium ESR capacitor in parallel with a 0.1 µF low ESR capacitor as close to the ADP1610 as possible.

The output capacitor maintains the output voltage and supplies current to the load while the ADP1610 switch is on. The value and characteristics of the output capacitor greatly affect the output voltage ripple and stability of the regulator. Use a low ESR output capacitor; ceramic dielectric capacitors are preferred.

The input bypass capacitor limits the high frequency impedance of the input voltage source. The value of this capacitor may need to be increased if the wire length from the input voltage source to the applications board is longer than 18” to optimize the transient performance.

voltages greater than one-half the output voltage, slope compensation is required to maintain stability of the current-mode regulator. For stable current-mode operation, ensure that the selected inductance is equal to or greater than L

using the following

MIN

equation:

−

VV

IN

OUT

=>

LL

MIN

A8.1

(7)

×

f

SW

Maximum DCR (mΩ) Height (mm) Comments

Used only for 3.3 V input and 5 V output

Used for 9 V, 10 V, and 12 V outputs

Used for inputs and outputs of 5 V

Rev. A | Page 11 of 20

ADP1610

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For very low ESR capacitors, such as ceramic capacitors, the ripple current due to the capacitance is calculated using the following equation. Because the capacitor discharges during the on-time

), the charge removed from the capacitor (QC) is the load

(t

ON

current multiplied by the on-time. The output voltage ripple

) is

(V

OUT

Q

V

OUT

C

OUT

t

×

L

ON

==Δ (8)

C

OUT

where:

is the output capacitance.

C

OUT

I

is the average inductor current.

L

D

t

= (9)

ON

F

SW

and

−

VV

IN

OUT

=

D

V

(10)

OUT

Choose the output capacitor based on the following equation:

OUT

IN

(11)

VVF

Δ××

OUTOUT

L

≥

C

OUT

SW

)(

VVI

The following are recommended capacitor manufacturers:

• AVX

• Kyo ce r a

• Murata

• Sanyo

• Tai y o –Yu d e n

• TDK

Rev. A | Page 12 of 20

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DIODE SELECTION

The output rectifier conducts the inductor current to the output capacitor and load while the switch is off. For high efficiency, minimize the forward voltage drop of the diode. For this reason, Schottky rectifiers are recommended. The following are recommended Schottky diode manufacturers:

• ON Semiconductor

• Diodes, Inc.

However, for high voltage, high temperature applications, where the Schottky rectifier reverse leakage current becomes significant and can degrade efficiency, use an ultrafast junction diode.

Make sure that the diode is rated to handle the average output load current. Many diode manufacturers derate the current capability of the diode as a function of the duty cycle. Verify that the output diode is rated to handle the average output load current with the minimum duty cycle. The minimum duty cycle of the ADP1610 is calculated by the following equations:

VV

−

D

where V

OUT

=

MIN

is the maximum input voltage.

IN-MAX

MAXIN

V

OUT

(12)

LOOP COMPENSATION

The ADP1610 uses external components to compensate the regulator loop, allowing optimization of the loop dynamics for a given application.

The step-up converter produces an undesirable right-half plane zero in the regulation feedback loop. This requires compensating the regulator such that the crossover frequency occurs well below the frequency of the right-half plane zero. The right-half plane zero is determined by the following equation:

Z

⎜ ⎝

⎛ ⎜

=2)(

RHPF

where:

F

(RHP) is the right-half plane zero.

Z

R

is the equivalent load resistance or the output voltage

LOAD

divided by the load current.

To stabilize the regulator, make sure that the regulator crossover frequency is less than or equal to one-fifth of the right-half plane zero and less than or equal to one-fifteenth of the switching frequency.

2

⎞

V

OUT

R

IN

LOAD

⎟

×

⎟ ⎠

(13)

L

×π

The regulator loop gain is calculated using the following equation:

A ×××××=

VL

V

OUT

IN

FB

V

OUT

CSCOMPMEA

(14)

ZGZG

OUT

V

where:

is the loop gain.

A

VL

V

is the feedback regulation voltage, 1.230 V.

FB

V

is the regulated output voltage.

OUT

V

is the input voltage.

IN

G

is the error amplifier transconductance gain.

MEA

Z

is the impedance of the series RC network from COMP

COMP

to GND.

G

is the current sense transconductance gain (the inductor

CS

current divided by the voltage at COMP), which is internally set by the ADP1610.

Z

is the impedance of the load and output capacitor.

OUT

To determine the crossover frequency, it is important to note that at that frequency the compensation impedance (Z dominated by the resistor, and the output impedance (Z

COMP

OUT

) is

) is dominated by the impedance of the output capacitor. Therefore, when solving for the crossover frequency, the equation (by definition of the crossover frequency) is simplified to

V

A (15)

VL

V

OUT

IN

FB

V

OUT

×××××=

GRG

CSCOMPMEA

1

2

C

=

××π

Cf

OUT

where:

f

is the crossover frequency.

C

R

is the compensation resistor.

COMP

is calculated using the following equation:

R

COMP

VVCf

MEA

××××π=2

OUTOUTOUT

GCSGVV

×××

(16)

= 1.23, G

V

FB

R

COMP

C

INFB

= 100 S, and GCS = 2 S is calculated using the

MEA

following equation:

4

R

COMP

1055.2

=

C

V

IN

VVCf

×××××

OUTOUTOUT

(17)

When the compensation resistor is known, the zero formed by the compensation capacitor and resistor is to be set to onefourth of the crossover frequency using the following equation:

2

(18)

RfC××π

COMPC

where C

=

COMP

is the compensation capacitor.

COMP

1

Rev. A | Page 13 of 20

ADP1610

www.BDTIC.com/ADI

Table 5. Recommended External Components for Popular Input/Output Voltage Conditions

VIN (V) V

3.3 5 0.700 4.7 22 22 30.9 10 68 150 600 5 1.23 2.7 10 22 30.9 10 100 100 600 9 0.700 10 10 10 63.4 10 20 680 9 1.23 4.7 4.7 10 63.4 10 47 150 10 0.700 10 10 10 71.5 10 33 470 10 1.23 4.7 4.7 10 71.5 10 39 220 12 0.700 10 10 10 88.7 10 33 470 12 1.23 4.7 10 10 88.7 10 75 150 5 9 0.700 10 10 10 63.4 10 24 470 9 1.23 4.7 4.7 10 63.4 10 47 150 10 0.700 10 10 10 71.5 10 33 330 10 1.23 4.7 4.7 10 71.5 10 51 220 12 0.700 10 10 10 88.7 10 33 330 12 1.23 4.7 4.7 10 88.7 10 75 150 350

The capacitor (C2) is chosen to cancel the zero introduced by output capacitance ESR. C2 is calculated using the following equation:

For low ESR output capacitance such as with a ceramic capacitor, C2 is optional. For optimal transient performance, the

and C

R

COMP

transient response of the ADP1610. For most applications, the compensation resistor is to be in the range of 30 kΩ to 400 kΩ, and the compensation capacitor is to be in the range of 100 pF to 1.2 nF. Tab le 5 shows external component values for several applications.

SOFT START CAPACITOR

The voltage at SS ramps up slowly by charging the soft start capacitor (C listed the values for the soft start period, based on maximum output current and maximum switching frequency.

(V) FSW (MHz) L (μH) C

OUT

ERROR AMP

REF

FB

2

Figure 26. Compensation Components

=

CESRC2×

R

COMP

may need to be adjusted by observing the load

COMP

) with an internal 3 µA current source. Table 6

SS

g

OUT

(19)

COMP

m

1

R

C

(μF) CIN (μF) R1 (kΩ) R2 (kΩ) R

OUT

The soft start capacitor limits the rate of voltage rise on the COMP pin. This limits the peak switch current at startup. Table 6 shows a typical soft start period (t

C2

for several conditions.

A 20 nF soft start capacitor results in negligible input current overshoot at startup and is therefore suitable for most applications.

04472-004

However, if an unusually large output capacitor is used, a longer soft start period is required to prevent input inrush current.

Table 6. Typical Soft Start Period

VIN (V) V

OUT

3.3 5 10 20 0.3 5 10 100 2 9 10 20 9 10 100 12 10 20 12 10 100 5 9 10 20 9 10 100 12 10 20

12 10 100 2

Conversely, if fast startup is a requirement, the soft start capacitor can be reduced or even removed, allowing the ADP1610 to start quickly, but allowing greater peak switch current (see Figure 22 to Figure 25).

COMP

SS

(V) C

(kΩ) C

(pF) I

comp

OUT_MAX

350 350 300 300 250 250 450 450 425 425 350

) at maximum output current (I

(μF) CSS (nF) tSS (ms)

OUT

(mA)

)

OUT_MAX

2.5

8.2

3.5 15

0.4

1.5

0.62

Rev. A | Page 14 of 20

ADP1610

3.3V

www.BDTIC.com/ADI

APPLICATION CIRCUITS

The circuit in Figure 27 shows the ADP1610 in a step-up configuration that can be used to generate a 10 V regulator with the following specifications: V

≤ 400 mA.

I

OUT

IN

ON

3

SD

7

C

10µF

IN

C

22nF

RT

8 1

SS

Figure 27. 3.3 V to 10 V Step-Up Regulator

= 2.5 V to 5.5 V, V

IN

4.7µH

L

ADP1610

GND

4

SW

FB

COMP

56

2

OUT

D1

71.3kΩ

10kΩ

R

COMP

220kΩ

C

COMP

150pF

R1

R2

= 10 V, and

10V

C

OUT

10µF

04472-030

The output is set to the desired voltage using Equation 1. To change the compensation network, see Equation 17 and Equation 18.

DC-TO-DC STEP-UP SWITCHING CONVERTER WITH TRUE SHUTDOWN

Some battery-powered applications require very low standby current. The ADP1610 typically consumes 10 nA from the input, which makes it suitable for these applications. However, the output is connected to the input through the inductor and the rectifying diode, allowing load current draw from the input while shut down. The circuit in Figure 28 enables the ADP1610 to achieve output load disconnect at shutdown. To shut down the ADP1610 and disconnect the output from the input, drive

SD

pin below 0.4 V.

the

4.7µH

L

Q1 FDC6331

3.3V 10V A

R3

10kΩ

Q1

B

C

IN

10µF

OFF

C

22nF

SS

ADP1610

IN

3

SD

7

RT

8 1

SS

SW

FB

COMP

GND

4

Figure 28. Step-Up Regulator with True Shutdown

D1

56

R1

71.3kΩ

2

R2

10kΩ

R

COMP

220kΩ

C

COMP

150pF

C

OUT

10mF

TFT LCD BIAS SUPPLY

Figure 29 shows a power supply circuit for TFT LCD module applications. This circuit has +10 V, −5 V, and +22 V outputs. The +10 V is generated in the step-up configuration. The −5 V and +22 V are generated by the charge-pump circuit. During the step-up operation, the SW node switches between 10 V and ground (neglecting forward drop of the diode and on resistance of the switch). When the SW node is high, C5 charges up to 10 V. C5 holds its charge and forward-biases D8 to charge C6 to −10 V. The Zener diode (D9) clamps and regulates the output to −5 V.

The VGH output is generated in a similar manner by the pump capacitors, C1, C2, and C4. The output voltage is tripled and regulated down to 22 V by the Zener diode, D5.

R4

BAV99

200Ω

C6

10µF

4.7mH

D8

D7

L

D9

ADP1610

IN

3

SD

7

RT

8 1

SS

SW

FB

COMP

GND

4

3.3V

C

10µF

VGL

–5V

BZT52C5V IS

ON

IN

C

SS

22nF

Figure 29. TFT LCD Bias Supply

04472-031

C5

10nF

56

2

C4

10nF

C1

10nF

D1

R

COMP

220kΩ

C

COMP

150pF

D5

D4

BAV99

D3

D2

BAV99

71.3kW

10kΩ

C3

10µF

C2 1µF

R1

R2

charge-

R3

200Ω

BZT52C22

10V

C

OUT

10µF

D5

VGH 22V

04472-033

Rev. A | Page 15 of 20

ADP1610

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SEPIC POWER SUPPLY

The circuit in Figure 30 shows the ADP1610 in a single-ended primary inductance converter (SEPIC) topology. This topology is useful for an unregulated input voltage, such as a batterypowered application in which the input voltage can vary between

2.7 V to 5 V, and the regulated output voltage falls within the input voltage range.

The input and the output are dc-isolated by a coupling capacitor (C1). In steady state, the average voltage of C1 is the input voltage. When the ADP1610 switch turns on and the diode turns off, the input voltage provides energy to L1, and C1 provides energy to L2. When the ADP1610 switch turns off and the diode turns on, the energy in L1 and L2 is released to charge the output capacitor (C

) and the coupling capacitor (C1) and to supply current to

OUT

the load.

4.7µH

L1

2.5V–5.5V 3.3V

ON

C

IN

10µF

C

SS

22nF

ADP1610

IN

3

SD

7

RT

8 1

SS

SW

FB

COMP

GND

4

Figure 30. 3.3 V DC-to-DC Converter

C1

10µF

56

R1

16.8kΩ

4.7µH

R

COMP

60kΩ

C 1nF

COMP

L2

C 10µF

R2

10kΩ

2

OUT

04472-032

Rev. A | Page 16 of 20

ADP1610

www.BDTIC.com/ADI

LAYOUT PROCEDURE

To get high efficiency, good regulation, and stability, a welldesigned PCB layout is required. Where possible, use the sample application board layout as a model.

When designing PCBs the following guidelines are to be used (see Figure 2):

The low ESR input capacitor (C

•

) is to be kept close to IN

IN

and GND.

•

The high current path from C

through the inductor L1 to

IN

SW and PGND is to be kept as short as possible.

•

The high current path from C

D1, and the output capacitor C

through L1, the rectifier

IN

is to be kept as short as

OUT

possible.

•

High current traces are to be kept as short and as wide as

possible.

•

The feedback resistors are to be placed as close to the FB

pin as possible to prevent noise pickup.

•

The compensation components are to be placed as close as

possible to COMP.

•

To prevent radiated noise injection, high impedance traces

are not to be routed near any node connected to SW or near the inductor.

04472-028

Figure 32. Sample Application Board (Top Layer)

04472-027

Figure 31. Sample Application Board (Bottom Layer)

Rev. A | Page 17 of 20

04472-029

Figure 33. Sample Application Board (Silkscreen Layer)

ADP1610

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OUTLINE DIMENSIONS

3.20

3.00

2.80

8

5

4

SEATING PLANE

5.15

4.90

4.65

1.10 MAX

0.23

0.08

8° 0°

0.80

0.60

0.40

3.20

3.00

2.80

PIN 1

0.95

0.85

0.75

0.15

0.00

COPLANARITY

1

0.65 BSC

0.38

0.22

0.10

COMPLIANT TO JEDEC STANDARDS MO-187-AA

Figure 34. 8-Lead Mini Small Outline Package [MSOP]

(RM-8)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option Branding

ADP1610ARMZ-R7 ADP1610-EVALZ

1

Z = RoHS Compliant Part.

1

−40°C to +85°C 8-Lead Mini Small Outline Package [MSOP] RM-8 P03

Evaluation Board

Rev. A | Page 18 of 20

ADP1610

www.BDTIC.com/ADI

NOTES

Rev. A | Page 19 of 20

ADP1610

www.BDTIC.com/ADI

NOTES

Rev. A | Page 20 of 20

Datasheet ADP1610 Datasheet (ANALOG DEVICES)

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES

APPLICATIONS

FUNCTIONAL BLOCK DIAGRAM

GENERAL DESCRIPTION

TABLE OF CONTENTS

REVISION HISTORY

SPECIFICATIONS

ABSOLUTE MAXIMUM RATINGS

ESD CAUTION

PIN CONFIGURATION AND FUNCTION DESCRIPTIONS

TYPICAL PERFORMANCE CHARACTERISTICS

THEORY OF OPERATION

CURRENT MODE PWM OPERATION

FREQUENCY SELECTION

SOFT START

ON/OFF CONTROL

SETTING THE OUTPUT VOLTAGE

INDUCTOR SELECTION

CHOOSING THE INPUT AND OUTPUT CAPACITORS

DIODE SELECTION

LOOP COMPENSATION

SOFT START CAPACITOR

APPLICATION CIRCUITS

DC-TO-DC STEP-UP SWITCHING CONVERTER WITH TRUE SHUTDOWN

TFT LCD BIAS SUPPLY

SEPIC POWER SUPPLY

LAYOUT PROCEDURE

OUTLINE DIMENSIONS

ORDERING GUIDE